JP2016111607A - Transmission device, reception device, communication system, transmission method, reception method and communication method - Google Patents

Abstract

PROBLEM TO BE SOLVED: To reduce the frequency with which in-phase synthesis is performed on phases between sub spectrums, and to reduce a peak-to-average power ratio in a communication system where a modulation signal is transmitted/received while being divided into the sub spectrums of a plurality of bands.SOLUTION: When transmitting a data signal to be transmitted while modulating and dividing it into a plurality of sub spectrums, a transmission device calculates a peak-to-average power ratio that is a ratio of maximum power and average power by giving a different phase rotation for each transmission system and each of sub spectrums included in the transmission system, selects and transmits a transmission signal of a transmission system with the smallest peak-to-average power ratio. When demodulating the received transmission signal after extracting the plurality of sub spectrums included therein, recovering and synthesizing the frequency bands, a reception device calculates a correlation between neighboring sub spectrums of frequency bands which are overlapped on a frequency axis, and demodulates a modulation signal for which sub spectrums compensating the phase rotation given to each of the sub spectrums are synthesized.SELECTED DRAWING: Figure 2

Description

  The present invention relates to a transmission device, a reception device, a communication system, a transmission method, a reception method, and a communication method.

  A communication apparatus system (communication system) that transmits and receives a modulated signal by dividing the modulated signal into a plurality of bands has been put into practical use. FIG. 9 is a block diagram showing an outline of a circuit configuration in a transmission device and a reception device that constitute a conventional communication system. FIG. 9A shows a schematic configuration of a transmission circuit that divides a modulated signal into a plurality of bands and transmits the transmission signal in a conventional communication system, and FIG. 9B shows a conventional communication. 1 shows a schematic configuration of a receiving circuit that receives a signal transmitted from a transmitting apparatus, combines a signal divided into bands, and restores / demodulates the signal into a modulated signal in a receiving apparatus constituting the system.

  As shown in FIG. 9A, a conventional transmission circuit includes a modulation circuit 501, a waveform shaping filter 502, a DFT (discrete Fourier transform) circuit 503, a plurality of band division filters 504, a plurality of frequency shifters 505, and an adder. 506, an IDFT (Inverse Discrete Fourier Transform) circuit 507 is provided. 9B, the conventional receiving circuit includes a DFT circuit 601, a plurality of extraction filters 602, a plurality of frequency shifters 603, an adder 604, an IDFT circuit 605, and a demodulation circuit 606. .

FIG _{9 (a), N D (} N D is an integer of 2 or more) with a number of band division filter 504-1～ band division filter _{504-N D, N D} number corresponding to each of the band division filter 504 It shows an example of a transmission circuit provided with a frequency shifter 505-1～ frequency shifter _{505-N D.} Further, in FIG. 9 (b), _{N D} number of extraction filter 602-1～ comprising an extraction filter _{602-N D, N D} number of frequency shifter 603-1～ frequency shifter 603 corresponding to each of the extraction filter 602 It shows an example of a transmission circuit having a -N _D.

  Here, communication processing in a conventional communication system including a transmission device including the transmission circuit as illustrated in FIG. 9A and a reception device including the reception circuit as illustrated in FIG. A flow of transmission processing by the transmission circuit and reception processing by the reception circuit) will be described. FIG. 10 is a diagram illustrating an example of a signal at each stage of transmission processing by a transmission circuit included in a transmission device constituting a conventional communication system. FIG. 11 is a diagram illustrating an example of signals at each stage of reception processing by a reception circuit provided in a reception device constituting a conventional communication system. FIG. 10 and FIG. 11 show examples of signal frequency bands at the respective stages of communication processing. In the following description, the flow of communication processing in the communication system will be described with reference to the diagrams showing the frequency bands of the respective signals shown in FIGS.

  In the conventional transmission circuit, the modulation circuit 501 modulates the data signal to be transmitted by a modulation scheme such as QPSK (Quadrature Phase Shift Keying), and the waveform shaping filter 502 modulates the data modulated by the modulation circuit 501. Band-limit the signal. In the conventional transmission circuit, the DFT circuit 503 converts the data signal band-limited by the waveform shaping filter 502 into a frequency domain modulation signal. FIG. 10A shows an example of the frequency band of the modulated signal converted by the DFT circuit 503.

In the conventional transmission circuit, the modulated signal converted by the DFT circuit 503 is input to each of the plurality of band division filters 504, and each band division filter 504 divides the signal into a plurality of spectra. In the following description, each divided spectrum is referred to as a “sub-spectrum SS”, and a code corresponding to the band-splitting filter 504 is assigned in order from the sub-spectrum SS having the lowest frequency, so that the sub-spectrum SS1, the sub-spectrum SS2, ..., sub-spectrum SSN _D. Incidentally, without distinction respective sub spectrum SS, to represent any one of the sub-spectrum SS represents a sub-spectrum _{SSk (1 ≦ k ≦ N D} ). In FIG. 10B, there are four band division filters 504 provided in the conventional transmission circuit, that is, k = 4, and four modulation signals converted by the DFT circuit 503, that is, the sub-spectrum SS1. ~ Shows an example of the frequency band of each sub-spectrum SSk when divided into sub-spectrum SS4.

  As can be seen from FIG. 10B, the frequency band of each divided sub-spectrum SSk has an overlapping region that overlaps the frequency band of the adjacent sub-spectrum SSk. Therefore, in the conventional transmission circuit, the frequency shifter 505 corresponding to each sub-spectrum SSk shifts each frequency band of the corresponding sub-spectrum SSk to a target frequency band. Thereby, each sub-spectrum SSk is distributed and arranged in a frequency band that does not overlap with the adjacent sub-spectrum SSk. FIG. 10C shows an example of each frequency band of sub-spectrum SS1 to sub-spectrum SS4 in which the frequency band of each sub-spectrum SSk is dispersed so as not to overlap with the frequency band of the adjacent sub-spectrum SSk. ing.

  In the conventional transmission circuit, the adder 506 adds each of the sub-spectrum SS1 to sub-spectrum SS4 in which the frequency bands are dispersed by the respective frequency shifters 505, and the IDFT circuit 507 outputs the output signal of the adder 506. It is converted into a time domain signal (transmission signal) and transmitted.

  On the other hand, in the conventional receiving circuit, the time domain transmission signal transmitted from the transmission circuit is received, and the DFT circuit 601 converts the received time domain transmission signal into a frequency domain signal. FIG. 11A shows an example of the frequency band of the signal converted by the DFT circuit 601. As can be seen from FIG. 11A, the signals converted by the DFT circuit 601 include the respective subspectra SSk (subspectra SS1 to subspectrum SS4 included in the transmission signal transmitted by the transmission circuit). Yes.

  In the conventional receiving circuit, the frequency domain signal converted by the DFT circuit 601 is input to each of the plurality of extraction filters 602, and each extraction filter 602 extracts each sub-spectrum SSk. In FIG. 11B, there are four extraction filters 602 provided in the conventional receiving circuit, that is, k = 4, and four spectrums SSk from the frequency domain signals converted by the DFT circuit 601, that is, sub-spectrums. An example of the frequency band of each sub-spectrum SSk when SS1 to sub-spectrum SS4 is extracted is shown.

  In the conventional receiving circuit, the frequency shifter 603 corresponding to the sub-spectrum SSk extracted by each extraction filter 602 shifts the frequency band of the sub-spectrum SSk. At this time, the frequency shifter 603 shifts to a frequency band before the frequency shifter 505 provided in the transmission circuit disperses. FIG. 11C shows an example of a state where the frequency band of each sub-spectrum SSk (sub-spectrum SS1 to sub-spectrum SS4) is shifted. As a result, the frequency band of each sub-spectrum SSk in the conventional receiving circuit overlaps with the frequency band of the adjacent sub-spectrum SSk. In other words, the frequency shifter 505 provided in the transmission circuit is in a state where there is an overlapping region before the respective subspectra SS1 to subspectrum SS4 are dispersed (see FIG. 10B).

  In the conventional receiving circuit, the adder 604 adds each of sub-spectrum SS1 to sub-spectrum SS4 whose frequency band is shifted by each frequency shifter 505 to obtain a modulated signal. FIG. 11D shows an example of the frequency band of the modulation signal obtained by the adder 604 adding each sub-spectrum SSk (sub-spectrum SS1 to sub-spectrum SS4). As a result, in the conventional receiving circuit, a modulated signal having the same frequency band as the modulated signal converted by the DFT circuit 503 provided in the transmitting circuit is obtained (see FIG. 10A).

  Thereafter, in the conventional receiving circuit, the IDFT circuit 605 converts the modulation signal obtained by the addition by the adder 604 into a time domain signal, and the demodulation circuit 606 converts the time domain signal converted by the IDFT circuit 605, The data signal transmitted by the transmission circuit is extracted by demodulating with a demodulation method corresponding to a modulation method such as QPSK in which the data signal is modulated.

  As described above, in the communication method in the conventional communication system, the transmission device divides and transmits the spectrum of the transmission signal, and the reception device transmits the data signal by combining the divided spectrum. For this reason, in the communication in the conventional communication system, a vacant frequency band can be used effectively.

Junichi Abe, Fumihiro Yamashita, Kiyoshi Kobayashi, “Proposal of spectrum-editing band-distributed transmission to achieve high frequency utilization efficiency”, IEICE Technical Report SAT 2009-48, IEICE, December 2009

  As described above, in the communication method in the conventional communication system, the frequency domain modulation signal converted by the DFT circuit 503 is divided to generate the sub-spectrum SSk, and the frequency shifter 505 shifts each sub-frequency to the desired frequency band. The spectrum SSk is added. For example, considering the case where the shift amount of the frequency band of the sub-spectrum SSm is the shift amount Sm, the frequency component F (k) included in the sub-spectrum SSm becomes the frequency component F (k + Sm) by the shift amount Sm. Then, after adding each sub-spectrum SSk, the IDFT circuit 507 converts it into a signal in the time domain. The conversion of the time domain signal in the IDFT circuit 507 at this time is expressed by the following equation (1).

In the above equation (1), N represents the size of the DFT, x represents a time domain signal, and F represents a frequency domain signal. When the frequency band of the frequency component F (k) included in the sub-spectrum SSm is not shifted, the IDFT circuit 507 calculates F (k) W ^nk from the above equation (1).

However, since the frequency shift of the shift amount Sm is performed in the conventional communication system, the IDFT circuit 507 calculates F (k + Sm) ^{Wn (k + Sm)} . That is, in the IDFT circuit 507, when the frequency component F (k + Sm) frequency-shifted by the shift amount Sm is subjected to inverse discrete Fourier transform, the inverse discrete Fourier transform is performed without shifting the frequency band of the frequency component F (k) included in the subspectrum SSm. Compared to the case of the conversion, the phase rotation is extra performed by W ^nSm . That is, in each sub-spectrum SSk transmitted in the conventional communication system, phase rotation corresponding to each shift amount Sm occurs.

  For this reason, in the conventional communication system, the phase relationship of each sub-spectrum SSk is changed by the phase rotation when the IDFT circuit 507 performs inverse discrete Fourier transform, and the adder 506 has a mutual phase between each sub-spectrum SSk. The in-phase synthesis timing also changes. This change in the in-phase combining timing causes an increase in peak-to-average power ratio (PAPR). For this reason, in the conventional communication system, it is necessary to use a large-sized and high-power amplifier for the transmission device as compared with the average power.

  The present invention has been made on the basis of the above problem recognition, and in a communication system that transmits and receives a modulated signal by dividing it into sub-spectrums of a plurality of bands, the phases of each sub-spectrum are in-phase combined. It is an object of the present invention to provide a transmission device, a reception device, a communication system, a transmission method, a reception method, and a communication method that can reduce the frequency of transmission and the peak-to-average power ratio.

  In order to solve the above problems, the transmitting apparatus of the present invention converts a modulated signal obtained by modulating a data signal to be transmitted into a frequency domain signal and divides it into N (N is an integer of 2 or more) sub-spectrums. A transmission apparatus for transmitting a transmission signal in which each of the sub-spectrums is distributed in a discontinuous frequency band on a frequency axis, wherein each of the divided sub-spectra A plurality of transmission sequences including a spectrum are generated, a different phase rotation is given to each of the transmission sequence and the sub-spectrum included in the transmission sequence, and the phase rotation corresponding to each of the divided sub-spectrums Transmission signal generating means for generating the transmission signal of each of the transmission sequences including the given sub-spectrum, and For each transmission signal of each of the transmission sequences, a peak-to-average power ratio that is a ratio of maximum power to average power is calculated, and the transmission signal of the transmission sequence having the smallest peak-to-average power ratio is selected. Transmission signal selection means for transmission.

  Further, in the transmission apparatus of the present invention, the transmission signal generating means corresponds to each of the transmission sequences, and the transmission sequence corresponding to each of the divided sub-spectrums included in the corresponding transmission sequence. A plurality of phase rotation means for applying the phase rotation to each of the sub-spectrums by adding different phase offsets, and corresponding to each of the phase rotation means, and the phase rotation is applied by the corresponding phase rotation means. A plurality of the sub-spectrums, and signal dispersion means for generating the transmission signal in the time domain in which each of the sub-spectrums is dispersed in the discontinuous frequency bands on the frequency axis. To do.

  The receiving apparatus of the present invention extracts N (N is an integer of 2 or more) sub-spectrums distributed in discontinuous frequency bands on the frequency axis included in the received transmission signal, and each extracted A receiving device that demodulates the modulated signal synthesized after returning the frequency band of the sub-spectrum to the frequency band before being dispersed on the frequency axis, and for each sub-spectrum after returning the frequency band, A phase added to calculate a correlation between the adjacent sub-spectrums whose frequency bands overlap on the frequency axis, and to add a phase rotation to each of the sub-spectrums based on the calculated correlation An offset is estimated, and the phase rotation applied to each sub-spectrum is compensated based on the estimated phase offset Phase estimation means for determining a phase compensation offset for the phase compensation means for compensating the phase rotation applied to each sub-spectrum by adding the phase compensation offset corresponding to each sub-spectrum, And demodulating a modulation signal obtained by synthesizing the sub-spectrum compensated for the phase rotation given by the phase compensation means.

  Further, in the receiving apparatus of the present invention, the phase estimation means includes a reference k-1th sub-spectrum of each of the sub-spectrums returned to the frequency band before being distributed on the frequency axis. (K is an integer of 2 ≦ k ≦ N) and the k−1th subspectrum and the frequency band overlap on the frequency axis, and the higher frequency side than the k−1th subspectrum Signal components in the transition region on the high frequency side in the k−1 subspectrum and signal components in the transition region on the low frequency side in the kth subspectrum in the kth subspectrum in the frequency band of The phase compensation offset corresponding to the k th sub-spectrum by multiplying by the complex conjugate of The frequency of the k-th sub-spectrum and the frequency band overlap on the frequency axis with the k-th sub-spectrum compensated for the phase rotation as a reference, and the frequency on the higher frequency side than the k-th sub-spectrum The phase compensation offset corresponding to the (k + 1) th sub-spectrum in the band is determined, and the phase compensation means gives each of the sub-spectrums by multiplying each of the phase compensation offsets corresponding to the sub-spectrum. The received phase rotation is sequentially compensated, and the receiving apparatus demodulates a modulation signal obtained by synthesizing the k−1th sub-spectrum to the Nth sub-spectrum.

  The communication system of the present invention converts a modulated signal obtained by modulating a data signal to be transmitted into a frequency domain signal and divides it into N (N is an integer of 2 or more) sub-spectrums, and then each of the sub-spectrums. A transmission device that transmits a transmission signal dispersed in a discontinuous frequency band on the frequency axis, and N (N is 2 or more) distributed in the discontinuous frequency band on the frequency axis included in the received transmission signal A receiver that extracts (integer) sub-spectrums and demodulates the combined modulated signal after returning the frequency bands of the extracted sub-spectrums to the frequency bands before being distributed on the frequency axis. In the system, the transmission device may include a plurality of divided sub-spectrums, each of which includes the divided sub-spectrums. Generating a transmission sequence, assigning a different phase rotation to each of the transmission sequence and the sub-spectrum included in the transmission sequence, and applying the phase rotation corresponding to each of the divided sub-spectrums A transmission signal generation means for generating the transmission signal of each of the transmission sequences, and a peak-to-average power ratio that is a ratio of maximum power to average power for the transmission signals of each of the transmission sequences; Transmission signal selection means for selecting and transmitting the transmission signal of the transmission sequence having the smallest peak-to-average power ratio, and the receiving apparatus, for each of the sub-spectrum after returning the frequency band, Calculate the correlation of each adjacent sub-spectrum where the frequency bands overlap on the frequency axis, Based on the calculated correlation, the phase offset added to give the phase rotation to each sub-spectrum is estimated, and the phase rotation given to each sub-spectrum based on the estimated phase offset Phase estimation means for determining a phase compensation offset for compensation, and phase compensation means for compensating for the phase rotation applied to each sub-spectrum by adding the phase compensation offset corresponding to each sub-spectrum And demodulating a modulation signal obtained by synthesizing the sub-spectrum compensated for the phase rotation given by the phase compensation means.

  In the transmission method of the present invention, a modulated signal obtained by modulating a data signal to be transmitted is converted into a frequency domain signal, divided into N (N is an integer of 2 or more) sub-spectrums, and then each of the sub-spectrums. Is a transmission method in a transmission apparatus for transmitting a transmission signal distributed in discontinuous frequency bands on the frequency axis, and includes each of the divided sub-spectrums by duplicating each of the divided sub-spectrums A plurality of transmission sequences generated, a different phase rotation is given to each of the transmission sequence and the sub-spectrum included in the transmission sequence, and the phase rotation corresponding to each of the divided sub-spectrums is given A transmission signal generation procedure for generating the transmission signal of each of the transmission sequences including the sub-spectrum; For the transmission signal of the transmission sequence, a peak-to-average power ratio that is a ratio of maximum power to average power is calculated, and the transmission signal of the transmission sequence with the smallest peak-to-average power ratio is selected and transmitted A transmission signal selection procedure.

  Further, the reception method of the present invention extracts N (N is an integer of 2 or more) sub-spectrums distributed in discontinuous frequency bands on the frequency axis included in the received transmission signal, and each extracted A receiving method for demodulating a synthesized modulated signal after returning the frequency band of the sub-spectrum to the frequency band before being dispersed on the frequency axis, wherein each sub-frequency after the frequency band is returned A spectrum is added to calculate the correlation of each adjacent sub-spectrum where the frequency bands overlap on the frequency axis, and to add phase rotation to each sub-spectrum based on the calculated correlation. Estimated phase offsets, and based on the estimated phase offset, A phase estimation procedure for determining a phase compensation offset for compensating for phase rotation and the phase compensation offset corresponding to each subspectrum are added to compensate for the phase rotation given to each subspectrum. A modulated signal obtained by synthesizing the sub-spectrum compensated for the phase rotation given by the phase compensation procedure.

  In the communication method of the present invention, a modulated signal obtained by modulating a data signal to be transmitted is converted into a frequency domain signal, divided into N (N is an integer of 2 or more) sub-spectrums, and then each of the sub-spectrums. And a transmission method for transmitting a transmission signal dispersed in a discontinuous frequency band on the frequency axis, and N (N is distributed in a discontinuous frequency band on the frequency axis included in the received transmission signal) Receiving in a receiving apparatus that demodulates a modulated signal synthesized after extracting sub-spectrums of integers of 2 or more and returning the frequency bands of the extracted sub-spectrums to the frequency bands before being distributed on the frequency axis A transmission method comprising: replicating each divided sub-spectrum to each of the divided sub-spectrums. A plurality of transmission sequences including a tram are generated, a different phase rotation is given to each of the transmission sequence and the sub-spectrum included in the transmission sequence, and the phase rotation corresponding to each of the divided sub-spectrums A transmission signal generation procedure for generating the transmission signal of each of the transmission sequences including the given sub-spectrum, and a peak-to-average that is a ratio of maximum power and average power for the transmission signals of each of the transmission sequences A transmission signal selection procedure for calculating a power ratio and selecting and transmitting the transmission signal of the transmission sequence having the smallest peak-to-average power ratio, and the reception method is configured to return the frequency band, respectively. For each of the adjacent sub-spectrums adjacent to each other in which the frequency bands overlap on the frequency axis. And calculating a correlation between the sub-spectrums based on the calculated correlation, estimating a phase offset added to add the phase rotation to each of the sub-spectrums, and calculating each of the sub-correspondences based on the estimated phase offset. A phase estimation procedure for determining a phase compensation offset for compensating for the phase rotation given to the spectrum, and the phase compensation offset corresponding to each of the sub-spectrums are added to each of the sub-spectrums. A phase compensation procedure for compensating for the phase rotation, and demodulating a modulation signal obtained by synthesizing the sub-spectrum compensated for the phase rotation provided by the phase compensation procedure.

  According to the present invention, in a communication system that performs transmission / reception by dividing a modulation signal into sub-spectrums of a plurality of bands, the frequency with which the phases of each sub-spectrum are combined in phase is reduced, and peak-to-average power The effect that the ratio can be reduced is obtained.

It is the block diagram which showed schematic structure of the communication system in embodiment of this invention. It is the block diagram which showed schematic structure of the transmission circuit with which the transmission apparatus which comprises the communication system of this embodiment was equipped. It is the flowchart which showed the process sequence of the transmission process in the transmission circuit with which the transmission apparatus which comprises the communication system of this embodiment was equipped. It is the figure which showed an example of the signal of each step of the transmission process by the transmission circuit with which the transmission apparatus which comprises the communication system of this embodiment was equipped. It is the block diagram which showed schematic structure of the receiving circuit with which the receiver which comprises the communication system of this embodiment was equipped. It is a figure explaining the calculation method of the correlation value of a signal in the receiver which comprises the communication system of this embodiment. It is the flowchart which showed the process sequence of the reception process in the receiver circuit with which the receiver which comprises the communication system of this embodiment was equipped. It is the figure which showed an example of the signal of each step | level of the reception process by the receiving circuit with which the receiver which comprises the communication system of this embodiment was equipped. It is the block diagram which showed the outline of the circuit structure in the transmitter which comprises the conventional communication system, and a receiver. It is the figure which showed an example of the signal of each step of the transmission process by the transmission circuit with which the transmission apparatus which comprises the conventional communication system was equipped. It is the figure which showed an example of the signal of each step of the reception process by the receiving circuit with which the receiver which comprises the conventional communication system was equipped.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings. FIG. 1 is a block diagram showing a schematic configuration of a communication system in the present embodiment. In FIG. 1, the communication system 1 includes a transmission device 10 and a reception device 20. The communication system 1 is a system that transmits a data signal from the transmission device 10 to the reception device 20 by wireless communication.

  The communication system 1 shown in FIG. 1 has a configuration including one transmission device 10 and one reception device 20, but in general, the transmission device 10 and the reception device 20 included in the communication system 1. It is considered that there are multiple units. In the communication system 1 shown in FIG. 1, a configuration in which a radio signal (transmission signal) is transmitted from the transmission device 10 to the reception device 20 is shown. It is considered that the apparatus has both functions of the transmission apparatus 10 and the reception apparatus 20, that is, the communication apparatus included in the communication system 1 is a transmission / reception apparatus. However, in the following description, for ease of explanation, a case will be described in which the communication system 1 includes one transmission device 10 that transmits a transmission signal and one reception device 20 that receives a transmission signal.

  When transmitting the data signal to the receiving device 20, the transmitting device 10 divides the modulated signal including the data signal into a plurality of spectrums, and transmits each of the divided spectra as transmission signals in different frequency bands from an antenna (not shown). A communication device for transmission. The transmission apparatus 10 includes a transmission circuit 100 that performs various processes on a data signal to be transmitted to generate a transmission signal. A detailed description of the configuration and operation of the transmission circuit 100 will be described later.

  The reception device 20 receives a transmission signal transmitted from the transmission device 10 in a plurality of frequency bands by an antenna (not shown), extracts a spectrum divided by the transmission device 10, and generates a modulated signal. It is a communication apparatus which takes out the data signal contained in. The receiving device 20 includes a receiving circuit 200 that performs various processes on the received transmission signal and extracts a data signal. A detailed description of the configuration and operation of the receiving circuit 200 will be described later.

  With this configuration, the communication system 1 transmits the data signal from the transmission device 10 to the reception device 20.

  Next, the configuration and operation of the transmission device 10 provided in the communication system 1 of the present embodiment will be described. FIG. 2 is a block diagram showing a schematic configuration of the transmission circuit 100 provided in the transmission apparatus 10 constituting the communication system 1 of the present embodiment. The transmission circuit 100 includes a modulation circuit 101, a waveform shaping filter 102, a DFT (discrete Fourier transform) circuit 103, a plurality of band division filters 104, a plurality of phase shifters 105, a plurality of frequency shifters 106, a plurality of adders 107, a plurality of adders 107, An IDFT (Inverse Discrete Fourier Transform) circuit 108, a plurality of PAPR (Peak to Average Power Ratio) calculation circuit 109, and a minimum PAPR selector 110 are provided.

  The modulation circuit 101 modulates the data signal transmitted by the transmission apparatus 10 using a modulation method such as QPSK and outputs the modulated data signal to the waveform shaping filter 102.

  The waveform shaping filter 102 performs band limitation on the modulated data signal input from the modulation circuit 101, and outputs the band-limited data signal to the DFT circuit 103.

  The DFT circuit 103 performs a discrete Fourier transform on the band-limited data signal input from the waveform shaping filter 102 to generate a frequency domain modulation signal. Then, the DFT circuit 103 outputs the generated modulation signal to each of the band division filters 104.

Each of the band division filters 104 generates a spectrum (hereinafter referred to as “sub-spectrum SS”) obtained by extracting a signal in a corresponding frequency band from the modulation signal input from the DFT circuit 103. FIG _{2, N D} (N D is an integer of 2 or more) shows the configuration of the transmitting circuit 100 having a number of band division filter 104-1～ band division filter _{104-N D.} Then, in the following description, the sub-spectrum SS that each band division filter 104-1～ band division filter _{104-N D} is generated, indicating to impart code corresponding to the band division filter 104.

In the configuration of the transmission circuit 100 shown in FIG. 2, the code following “−” after the code given to the band division filter 104 indicates the order of each band division filter 104 and the frequency of the sub-spectrum SS to be generated. Are shown in ascending order. Therefore, the sub-spectrum SS generated by each of the band division filter 104-1～ band division filter _{104-N D,} the sub-spectrum SS1, sub spectrum SS2, · · ·, a frequency in the order of sub-spectrum SSN _D increases. In addition, when expressing any one sub-spectrum SS without distinguishing each sub-spectrum SS, it is expressed as “sub-spectrum SSk” (1 ≦ k ≦ N _D ). Further, the band division filter 104 to generate the sub-spectrum SSk, referred to as "band-splitting filter 104-k" _{(1 ≦ k ≦ N D)} .

  In the transmission circuit 100, a plurality of frequency shifters 106 corresponding to the band division filter 104-k provided in the transmission circuit 100, one adder 107, one IDFT circuit 108, and one PAPR calculation circuit 109. A processing circuit group corresponding to one transmission sequence (hereinafter referred to as “transmission sequence circuit group”) is configured by a combination of processing circuits including The transmission circuit 100 includes a plurality of transmission sequence circuit groups in order to support a plurality of transmission sequences. A plurality of phase shifters 105 corresponding to the band division filter 104-k provided in the transmission circuit 100 other than the first transmission series circuit group among the plurality of transmission series circuit groups provided in the transmission circuit 100 are provided. include. In other words, the processing by the phase shifter 105 (more specifically, the processing for imparting phase rotation) is not performed in one transmission sequence among a plurality of types of transmission sequences.

FIG. 2 shows a configuration of the transmission circuit 100 in which C transmission sequence circuit groups are configured, that is, corresponding to C types of transmission sequences. That is, in the configuration of the transmission circuit 100 shown in FIG. 2, and the _{N D} frequency shifter 106 corresponding to each of the _{N D} band division filter 104-1～ band division filter _{104-N D,} 1 single addition The unit 107, one IDFT circuit 108, and one PAPR calculation circuit 109 constitute a first transmission series circuit group. In the configuration of the transmitting circuit 100 shown in FIG. 2, and the _{N D} phase 105 corresponding to each of the _{N D} band division filter 104-1～ band division filter _{104-N D,} the _{N D} The frequency shifter 106, one adder 107, one IDFT circuit 108, and one PAPR calculation circuit 109 constitute a second to Cth transmission sequence circuit group.

More specifically, in the configuration of the transmission circuit 100 illustrated in FIG. 2, the frequency shifter 106-1-1 to the frequency shifter 106-1 -N _D , the adder 107-1, the IDFT circuit 108-1, and the PAPR calculation. The circuit 109-1 constitutes the first transmission series circuit group. In the configuration of the transmission circuit 100 shown in FIG. 2, the phase shifter 105-2-1 to the phase shifter 105-2 -N _D , the frequency shifter 106-2-1 to the frequency shifter 106-2 -N _D , and the adder 107-2, IDFT circuit 108-2, and PAPR calculation circuit 109-2 constitute a second transmission series circuit group. In the configuration of the transmitting circuit 100 shown in FIG. 2, the phase shifter 105-C-1~ phaser _{105-C-N D,} the frequency shifter 106-C-1~ frequency shifter _{106-C-N D,} the adder 107-C, IDFT circuit 108-C, and PAPR calculation circuit 109-C constitute a C-th transmission series circuit group.

In the configuration of the transmission circuit 100 shown in FIG. 2, the components included in each transmission sequence circuit group (that is, the phase shifter 105, the frequency shifter 106, the adder 107, the IDFT circuit 108, and the PAPR calculation circuit 109). The code following “−” after the code given to the number represents the component included in the number of transmission sequence circuit groups. Further, in the phase shifter 105 and the frequency shifter 106 included in each transmission sequence circuit group, the code following “−” represents the band division filter 104-k to which the component corresponds. . In addition, when representing a component included in any one transmission sequence circuit group without distinguishing each transmission sequence circuit group, a code following “-” after a code assigned to each component Is represented by q (1 ≦ q ≦ C). More specifically, the adder 107, the IDFT circuit 108, and the PAPR calculation circuit 109 included in the q-th transmission series circuit group are referred to as “adder 107-q”, “IDFT circuit 108-q”, and It is expressed as “PAPR calculation circuit 109-q” (1 ≦ q ≦ C). The phase shifter 105 and the frequency shifter 106 included in the qth transmission sequence circuit group and corresponding to the band division filter 104-k are respectively referred to as “phase shifter 105-qk” and “frequency shifter 106-q. −k ”(1 ≦ q ≦ C, 1 ≦ k ≦ N _D ).

Each of the band division filters 104-k duplicates the generated sub-spectrum SSk by the number of transmission series circuit groups included in the transmission circuit 100, and the duplicated sub-spectrum SSk is included in each transmission series circuit group. Output to each of the frequency shifter 106-q-k and the corresponding phase shifter 105-q-k. More specifically, since the configuration of the transmission circuit 100 shown in FIG. 2 includes C transmission series circuit groups, the band division filter 104-1 converts the generated subspectrum SS1 into the frequency shifter 106- 1-1, phase shifter 105-2-1,..., And phase shifter 105-C-1. Further, the band division filter 104-2 outputs the generated sub-spectrum SS2 to each of the frequency shifter 106-1-2, the phase shifter 105-2-2, ..., the phase shifter 105-C-2. Further, the band division filter _{104-N D,} the sub-spectrum SSN _D which generated the frequency shifters _{106-1-N D,} phaser _105-2-N D, · · ·, phaser _{105-C-N D} Output to each.

Each of the phase shifters 105-q-k adds a predetermined offset phase (hereinafter referred to as “phase offset”) to the sub-spectrum SSk input from the corresponding band-splitting filter 104-k, so that the sub-spectrum Rotate the phase of SSk. At this time, in the phase shifter 105-q-k, by using the phase sequence Θ _q represented by the following equation (2) prepared in advance, a different phase offset is added to the input sub-spectrum SSk, respectively. A different phase rotation is given to the sub-spectrum SSk. As described above, the sub-spectrum SSk phaser 105-q-k to adding a phase offset is sub spectrum SS2~ sub spectrum SSN _D. Therefore, the phase sequence Θ _q (Θ ₂ , Θ ₃ ,..., Θ _q ,..., Θ _C ) is sub-spectrum SSq 2, sub-spectrum SSq 3,. phase offset theta _q2 which added to _{SSqN D, θ q3, ···,} θ qk, ···, comprised of equation (3).

In the above equation (2), the value of each phase offset θ _{qk is} a value determined randomly between 0 and 2π, that is, a random value. Each of the phase shifters 105-q-k outputs the sub-spectrum SSk to which the phase offset θ _qk is added to the corresponding frequency shifter 106-q-k. At this time, the output signal formula (4) output from each of the phase shifters 105-q-k, that is, the sub-spectrum SSk to which the phase offset θ _qk is added is expressed by the following formula (5).

Each of the frequency shifters 106-q-k is obtained after adding the sub-spectrum SSk input from the corresponding band-splitting filter 104-k or the phase offset θ _qk input from the corresponding phase shifter 105-q-k. The frequency band of the sub spectrum SSk is shifted to the target frequency band, and the sub spectrum SSk after the frequency band is shifted is output to the corresponding adder 107-q.

  Each of the adders 107-q adds (synthesizes) each of the sub-spectrums SSk input from the corresponding frequency shifter 106-q-k after shifting the frequency band, and adds a plurality of sub-spectrums SSk. The output signal is output to the corresponding IDFT circuit 108-q.

  Each of the IDFT circuits 108-q performs inverse discrete Fourier transform on the output signal input from the corresponding adder 107-q to generate a time domain transmission signal. Each of IDFT circuits 108-q then outputs the generated transmission signal in the time domain to the corresponding PAPR calculation circuit 109-q and minimum PAPR selector 110.

  Each of the PAPR calculation circuits 109-q has a peak-to-average power ratio (Peak-to-Average Power Ratio) that is a ratio between the maximum power and the average power for the time domain transmission signal input from the corresponding IDFT circuit 108-q. : PAPR). The calculation of the peak-to-average power ratio in the PAPR calculation circuit 109-q is, for example, squared with respect to the C time domain signals input to the PAPR calculation circuit 109-1 to PAPR calculation circuit 109-C. The maximum value and the average value are calculated, and the calculated maximum value is divided by the average value (maximum value / average value). Each of the PAPR calculation circuits 109-q outputs the calculated peak-to-average power ratio result to the minimum PAPR selector 110.

  Based on the calculation result of the peak-to-average power ratio input from each PAPR calculation circuit 109-q, the minimum PAPR selector 110 selects among the transmission signals in the time domain input from each IDFT circuit 108-q. The transmission signal in the time domain with the smallest peak-to-average power ratio is selected. Then, the minimum PAPR selector 110 transmits the selected time domain transmission signal to the reception device 20 as a final transmission signal.

  Next, operation | movement of the transmitter 10 with which the communication system 1 of this embodiment was equipped is demonstrated. FIG. 3 is a flowchart showing a processing procedure of transmission processing in the transmission circuit 100 provided in the transmission device 10 constituting the communication system 1 of the present embodiment. FIG. 4 is a diagram illustrating an example of signals at each stage of transmission processing by the transmission circuit 100 included in the transmission apparatus 10 included in the communication system 1 of the present embodiment. FIG. 4 shows an example of the frequency band of the signal of the component in the transmission circuit 100 at each stage of the transmission processing by the transmission apparatus 10. In the following description, the procedure of the transmission processing in the transmission apparatus 10 is four band division filters 104-k provided in the transmission circuit 100 shown in FIG. 2, that is, k = 4, and the transmission sequence circuit group is The case where there are three, that is, C = 3 will be described as an example with reference to the frequency band of each signal shown in FIG.

  First, in the transmission processing in the transmission apparatus 10, the modulation circuit 101 modulates the data signal to be transmitted by a modulation method such as QPSK, and outputs the modulated data signal to the waveform shaping filter 102 (step S100). Subsequently, the waveform shaping filter 102 multiplies the modulated data signal input from the modulation circuit 101 by a predetermined filter coefficient, thereby shaping the waveform of the modulated data signal and band limiting. Then, the waveform shaping filter 102 outputs the data signal after band limitation to the DFT circuit 103 (step S101). Thereafter, the DFT circuit 103 performs a discrete Fourier transform on the band-limited data signal input from the waveform shaping filter 102 to generate a frequency domain modulation signal. Then, the DFT circuit 103 outputs the generated frequency domain modulation signal to each of the band division filter 104-1 to band division filter 104-4 (step S102).

  The frequency domain modulation signal output from the DFT circuit 103 to each of the band division filter 104-1 to band division filter 104-4 by the process of step S102 is like the modulation signal A shown in FIG. It is a frequency band. FIG. 4A shows the phase θ in the modulation signal A.

  Subsequently, each of the band division filters 104-k multiplies the frequency domain modulation signal input from the DFT circuit 103 by a predetermined division coefficient, thereby obtaining a sub-spectrum SSk extracted from the corresponding frequency band signal. Generate (step S103). That is, the frequency domain modulation signal input from the DFT circuit 103 is divided into a plurality of sub-spectrums SSk.

  Here, since the transmission circuit 100 includes four band division filters 104-1 to 104-4, in the process of step S 103, the modulation signal A shown in FIG. Four division coefficients corresponding to each of the division filter 104-1 to the band division filter 104-4 are multiplied for each frequency to generate sub-spectrum SS1 to sub-spectrum SS4. FIG. 4A shows a state in which sub-spectrum SS1 to sub-spectrum SS4 are generated from modulated signal A by each of band-division filter 104-1 to band-division filter 104-4. Since FIG. 4 shows the frequency band of each signal, the amplitude shown on the vertical axis in FIG. 4A is the amplitude of the modulation signal A and each of the sub-spectrum SS1 to sub-spectrum SS4. It does not represent the relationship of amplitude. As can be seen from FIG. 4A, each divided sub-spectrum SSk has an overlapping region in which the frequency band overlaps with the adjacent sub-spectrum SSk. That is, the transition areas of the divided sub-spectrums SSk overlap each other. FIG. 4B shows the frequency band and phase θ for each sub-spectrum SSk output from each of the band division filter 104-1 to band division filter 104-4.

  Subsequently, each of the band division filters 104-k replicates the generated sub-spectrum SSk to the number of transmission series circuit groups provided in the transmission circuit 100. Then, each of the band division filters 104-k outputs each of the duplicated sub-spectrum SSk to the corresponding transmission sequence circuit group (step S104). Here, since the transmission circuit 100 includes three transmission series circuit groups, in the process of step S104, each of sub-spectrum SS1 to sub-spectrum SS4 shown in FIG. To the corresponding transmission sequence circuit group. In the following description, when each of the sub-spectrum SSk replicated by each of the band division filters 104-k is expressed together, it is referred to as a “replicated signal”. Here, since the transmission circuit 100 includes four band division filters 104-1 to 104-4, one replica signal includes each of sub-spectrum SS 1 to sub-spectrum SS 4. ing.

  Subsequently, each of the transmission sequence circuit groups performs the following processing on the duplicate signal input from each of the band division filters 104-k. As can be seen from the configuration of the transmission circuit 100 shown in FIG. 2, each transmission series circuit group can execute the processing for the input replica signal, that is, each sub-spectrum SSk in parallel. However, in the following description, for ease of explanation, it is assumed that each transmission series circuit group included in the transmission circuit 100 sequentially executes processing for each duplicate signal, and the transmission circuit included in the transmission device 10 The procedure of the transmission process by 100 will be described.

  First, each of the frequency shifter 106-q-k or the phase shifter 105-q-k included in the transmission sequence circuit group determines whether or not the input replica signal is the first replica signal ( Step S105). More specifically, each of the frequency shifters 106-qk or phase shifters 105-q-k included in each transmission sequence circuit group includes the input subspectrum SSk in the first duplicate signal. It is determined whether or not the sub-spectrum SSk.

  As a result of the determination in step S105, if the input duplicate signal is the first duplicate signal (“YES” in step S105), the frequency shifter 106- included in the transmission sequence circuit group corresponding to the first duplicate signal Each of 1-k shifts the frequency band of the corresponding sub-spectrum SSk included in the input duplicate signal to a predetermined target frequency band. Each of the frequency shifters 106-1-k includes the sub-spectrum SSk after the frequency band is shifted in the corresponding adder 107-q, that is, the transmission sequence circuit group corresponding to the first duplicate signal. Is output to the adder 107-1 (step S106).

  Here, the transmission circuit 100 includes four band division filters 104-1 to 104-4, and includes three transmission series circuit groups. For this reason, each of the frequency shifters 106-1-1 to 106-1-4 included in the transmission sequence circuit group corresponding to the first duplicated signal has a corresponding sub-spectrum SS1 to SS1 in the process of step S106. The frequency band of sub-spectrum SS4 is shifted and output to adder 107-1.

  Subsequently, the adder 107-q included in the transmission sequence circuit group adds (synthesizes) each of the sub-spectra SSk input from the corresponding frequency shifter 106-q-k after shifting the frequency band. . Then, the adder 107-q outputs an output signal obtained by adding the plurality of sub-spectrums SSk to the corresponding IDFT circuit 108-q (step S107). More specifically, the adder 107-1 included in the transmission sequence circuit group corresponding to the first replica signal is input from each of the frequency shifters 106-1-1-1 to 106-1-4. An output signal obtained by adding (synthesizing) each sub-spectrum SSk is output to the IDFT circuit 108-1 included in the transmission sequence circuit group corresponding to the first duplicate signal.

  Subsequently, the IDFT circuit 108-q included in the transmission sequence circuit group performs inverse discrete Fourier transform on the output signal input from the corresponding adder 107-q to generate a time-domain transmission signal. . Then, the IDFT circuit 108-q outputs the generated time domain transmission signal to the corresponding PAPR calculation circuit 109-q and the minimum PAPR selector 110 (step S108). More specifically, the IDFT circuit 108-1 included in the transmission sequence circuit group corresponding to the first duplicated signal performs inverse discrete Fourier transform on the output signal input from the adder 107-1. The generated time domain transmission signal is output to PAPR calculation circuit 109-1 and minimum PAPR selector 110 included in the transmission sequence circuit group corresponding to the first duplicate signal.

  Subsequently, the PAPR calculation circuit 109-q included in the transmission sequence circuit group calculates a peak-to-average power ratio (PAPR) for the time-domain transmission signal input from the corresponding IDFT circuit 108-q. Then, the PAPR calculation circuit 109-q outputs the calculated peak-to-average power ratio result to the minimum PAPR selector 110 (step S109). More specifically, the PAPR calculation circuit 109-1 included in the transmission sequence circuit group corresponding to the first duplicated signal calculates the peak-to-average calculated for the time-domain transmission signal input from the IDFT circuit 108-1. The result of the power ratio is output to the minimum PAPR selector 110.

  Subsequently, each of the transmission sequence circuit groups determines whether or not the processing has been completed for all the duplicate signals duplicated by the band division filter 104-k (step S110). As a result of the determination in step S110, if the processing has not been completed for all the duplicate signals (“NO” in step S110), the process returns to step S105, and the processes in steps S105 to S110 are repeated.

  On the other hand, if the input replica signal is not the first replica signal as a result of the determination in step S105 ("NO" in step S105), the phase shifter included in the transmission sequence circuit group corresponding to the replica signal other than the first Each of 105-q-k adds a predetermined phase offset to the corresponding sub-spectrum SSk included in the input duplicate signal. Then, each of the phase shifters 105-q-k outputs the sub-spectrum SSk to which the phase offset is added to the corresponding frequency shifter 106-q-k (step S111).

  Here, the transmission circuit 100 includes four band division filters 104-1 to 104-4, and includes three transmission series circuit groups. For this reason, for example, each of the phase shifters 105-2-1 to 105-2-4 included in the transmission sequence circuit group corresponding to the second duplicated signal corresponds to the corresponding sub-spectrum in the process of step S111. A phase offset is added to SS1 to sub-spectrum SS4 and output to each of the corresponding frequency shifters 106-2-1 to 106-2-4. Further, for example, each of the phase shifters 105-3-1 to 105-3-4 included in the transmission sequence circuit group corresponding to the third replica signal corresponds to the corresponding sub-spectrum SS 1 in the process of step S 111. ~ A phase offset is added to the sub-spectrum SS4 and output to each of the corresponding frequency shifters 106-3-1 to 106-3-4.

In FIG. 4 (c), each of the phase shifters 105-2-1 to 105-2-4 included in the transmission sequence circuit group corresponding to the second duplicated signal causes the band division filters 104-1 to 104-1 to be included. A state in which the phase offset θ _qk in the phase sequence Θ _q represented by the above equation (2) is added to the sub-spectrum SS1 to the sub-spectrum SS4 output by each of the band division filters 104-4 is shown. More specifically, the phase offset θ ₂₂ is added to the sub-spectrum SS 2, the phase offset θ ₂₃ is added to the sub-spectrum SS 3, and the phase offset θ ₂₄ is added to the sub-spectrum SS 4. As can be seen from FIG. 4C, no phase offset is added to the sub-spectrum SS1. This is because, as described above, the sub-spectrum SSk phaser 105-q-k to adding a phase offset is because a sub-spectrum SS2~ sub spectrum SSN _D.

  Thereafter, each of the frequency shifter 106-q-k, the adder 107-q, the IDFT circuit 108-q, and the PAPR calculation circuit 109-q included in the transmission sequence circuit group corresponding to the replica signal other than the first is The corresponding processing of step S106 to step S109 is executed. More specifically, for example, each of the frequency shifters 106-2-1 to 106-2-4 included in the transmission sequence circuit group corresponding to the second duplicated signal has the phase in the process of step S 106. The frequency bands of sub-spectrum SS1 to sub-spectrum SS4 input from each of the devices 105-2-1 to 105-2-4 are shifted and output to the adder 107-2. The adder 107-2 included in the transmission sequence circuit group corresponding to the second duplicated signal is input from each of the frequency shifters 106-2-1 to 106-2-4 in the process of step S107. An output signal obtained by adding (synthesizing) each of the sub-spectrums SSk is output to the IDFT circuit 108-2. Then, the IDFT circuit 108-2 included in the transmission sequence circuit group corresponding to the second duplicated signal performs inverse discrete Fourier transform on the output signal input from the adder 107-2 in the process of step S108. The generated time domain transmission signal is output to the PAPR calculation circuit 109-2 and the minimum PAPR selector 110. The PAPR calculation circuit 109-2 included in the transmission sequence circuit group corresponding to the second duplicated signal calculates the peak calculated for the time domain transmission signal input from the IDFT circuit 108-2 in the process of step S109. The result of the average power ratio is output to the minimum PAPR selector 110.

  Further, for example, each of the frequency shifters 106-3-1 to 106-3-4 included in the transmission sequence circuit group corresponding to the third replica signal is the phase shifter 105-3 in the process of step S 106. The frequency bands of sub-spectrum SS1 to sub-spectrum SS4 input from −1 to phase shifter 105-3-4 are shifted and output to adder 107-3. The adder 107-3 included in the transmission sequence circuit group corresponding to the third duplicated signal is input from each of the frequency shifters 106-3-1 to 106-3-4 in the process of step S107. An output signal obtained by adding (synthesizing) each of the sub-spectrums SSk is output to the IDFT circuit 108-3. Then, the IDFT circuit 108-3 included in the transmission sequence circuit group corresponding to the third duplicated signal performs inverse discrete Fourier transform on the output signal input from the adder 107-3 in the process of step S108. The generated time domain transmission signal is output to the PAPR calculation circuit 109-3 and the minimum PAPR selector 110. Then, the PAPR calculation circuit 109-3 included in the transmission sequence circuit group corresponding to the third duplicated signal calculates the peak calculated for the time domain transmission signal input from the IDFT circuit 108-3 in the process of step S109. The result of the average power ratio is output to the minimum PAPR selector 110.

  In FIG. 4D, the frequencies output by the frequency shifters 106-2-1 to 106-2-4 included in the transmission sequence circuit group corresponding to the second duplicated signal in the process of step S106. An output signal obtained by adding (synthesizing) the sub-spectrum SS1 to sub-spectrum SS4 after the band is shifted (dispersed) by the adder 107-2 in the process of step S107 is shown. The IDFT circuit 108-2 included in the transmission sequence circuit group corresponding to the second duplicate signal outputs an output signal including a plurality of sub-spectrums SSk in which frequency bands are dispersed as shown in FIG. Is subjected to inverse discrete Fourier transform to generate a transmission signal in the time domain, and the generated transmission signal in the time domain is output to the PAPR calculation circuit 109-2 and the minimum PAPR selector 110.

  In this way, each transmission series circuit group executes processing for the corresponding duplicate signal. As a result of the determination in step S110, when the processing is completed for all the duplicate signals (“YES” in step S110), the minimum PAPR selector 110 receives the peaks input from the respective PAPR calculation circuits 109-q. Based on the calculation result of the average power ratio, the transmission signal in the time domain having the smallest peak-to-average power ratio is selected and transmitted to the reception device 20 as the final transmission signal (step S112).

  In this way, in the transmission apparatus 10, the transmission circuit 100 divides the modulated signal including the data signal into a plurality of spectrums, and further generates a plurality of time domain transmission signals with different phase offsets. Then, the transmitter 10 calculates the peak-to-average power ratio of each of the generated transmission signals in the plurality of time domains, and the time with the smallest peak-to-average power ratio among the generated transmission signals in the plurality of time domains. The transmission signal of the area is transmitted as the final transmission signal. Thereby, in the transmission apparatus 10, it is possible to transmit a transmission signal having a small peak-to-average power ratio in which the frequency with which the phases of each sub-spectrum included in the transmission signal are in-phase combined is reduced. This eliminates the need for using a large, high-power amplifier in the transmission device 10, thereby realizing a reduction in the size of the transmission device 10.

  Next, the configuration and operation of the receiving device 20 provided in the communication system 1 of the present embodiment will be described. FIG. 5 is a block diagram showing a schematic configuration of the receiving circuit 200 provided in the receiving device 20 configuring the communication system 1 of the present embodiment. The receiving circuit 200 includes a DFT (Discrete Fourier Transform) circuit 201, a plurality of extraction filters 202, a plurality of frequency shifters 203, a plurality of phase estimators 204, a plurality of phase shifters 205, an adder 206, and an IDFT (Inverse Discrete Fourier Transform). A circuit 207 and a demodulation circuit 208 are provided.

In the reception circuit 200, the same number of extraction filters 202 and frequency shifters 203 as the band division filter 104 included in the transmission circuit 100, the number of the phase division filters 104 included in the transmission circuit 100 −1 phase estimators 204 and phases A container 205 is provided. Figure 5 is a band same the _{N D} extraction filter 202-1～ extraction filter _{202-N D} and the number of division filter 104, and frequency shifter 203-1～ frequency shifter _{203-N D} having the transmission circuit 100 shows the configuration of the receiving circuit 200 with the _(N D -1) pieces of phase estimator 204-2～ phase estimator _{204-N D,} and phase shifter 205-2～ phaser _{205-N D} . In the following description, a symbol representing the band division filter 104-k to which the component corresponds, following “-” after the symbol assigned to each component of the receiving circuit 200 shown in FIG. Is given. That is, the extraction filter 202, the frequency shifter 203, the phase estimator 204, and the phase shifter 205 corresponding to the band division filter 104-k are respectively referred to as “extraction filter 202-k”, “frequency shifter 203-k”, “phase”. It is expressed as “estimator 204-k” and “phaser 205-k” (1 ≦ k ≦ N _D ).

  The DFT circuit 201 performs a discrete Fourier transform on the time domain transmission signal transmitted from the transmission apparatus 10 to generate a frequency domain signal. Then, the DFT circuit 201 outputs the generated frequency domain signal to each of the extraction filters 202-k.

  Each of the extraction filters 202-k extracts a signal corresponding to the sub-spectrum SSk generated by the corresponding band division filter 104-k from the frequency domain signal input from the DFT circuit 201. Then, each of the extraction filters 202 outputs a signal corresponding to the extracted sub-spectrum SSk to the corresponding frequency shifter 203-k.

  Each of the frequency shifters 203-k shifts the frequency band of the sub-spectrum SSk input from the corresponding extraction filter 202-k to the target frequency band. Note that the shift of the frequency band of the sub-spectrum SSk by each of the frequency shifters 203-k is a process of returning to the frequency band before each of the sub-spectrum SSk is shifted by the frequency shifter 106-q-k provided in the transmission circuit 100. It is. By the frequency band shift processing by each of the frequency shifters 203-k, the frequency band of each sub-spectrum SSk in the receiving circuit 200 is overlapped with the frequency band of the adjacent sub-spectrum SSk. That is, each of the frequency shifters 203-k has a superimposed region before the frequency shifter 106-q-k provided in the transmission circuit 100 disperses the frequency band of each sub-spectrum SSk (FIG. 4C). Return to). Each of the frequency shifters 203-k shifts the frequency band, that is, the sub-spectrum SSk after returning the frequency band to the adder 206 or the corresponding phase estimator 204-k and phase shifter 205-k. Output.

In the configuration of the receiving circuit 200 shown in FIG. 5, the frequency shifter 203-1 is configured to output the sub-spectrum SS1 after shifting the frequency band to the adder 206 and the phase estimator 204-2. In the configuration of the receiving circuit 200 shown in FIG. 5, the frequency shifter 203-2 outputs the sub-spectrum SS2 after the frequency band shift to the phase estimator 204-2 and the phase shifter 205-2, and the frequency The shifter 203-3 is configured to output the sub-spectrum SS3 after shifting the frequency band to the phase estimator 204-3 and the phase shifter 205-3. Similarly, in the configuration of the receiving circuit 200 shown in FIG. 5, the frequency shifter _{203-N D} is, the sub-spectrum SSN _D phase estimator _{204-N D} and the phase shifter _{205-N D} after shifting the frequency band To output to.

Each of the phase estimators 204-k is operated by a phase shifter 105-q-k included in the transmission circuit 100 based on each sub-spectrum SSk in which the frequency bands input from the two frequency shifters 203-k are adjacent. The value of the added phase offset θ _qk is estimated and the sub-spectrum SSk is returned to the original phase, that is, the phase offset formula (6) for compensating the phase of the sub-spectrum SSk (hereinafter referred to as “phase compensation offset ^ θ _qk ")).

More specifically, each of the phase estimators 204-k calculates a correlation value R _{k_ab} in the transition region between the sub-spectrum SS (k−1) and the sub-spectrum SSk whose frequency bands are adjacent. Then, each of the phase estimators 204-k estimates the value of the phase offset θ _qk added by the phase shifter 105-q-k included in the transmission circuit 100 based on the calculated correlation value R _{k_ab} , determining the value of the phase compensation offset ^ theta _qk to be added to the spectrum SSk. Then, each of the phase estimators 204-k _outputs the determined value of the phase compensation offset ^ θ _qk to the corresponding phase shifter 205-k.

As described above, in the transmission circuit 100, no phase offset is added to the sub-spectrum SS1 (see FIG. 4C). Therefore, the sub-spectrum SS1 is used as a reference in determining the value of the phase compensation offset ^ theta _qk for the sub spectrum _{SSk (2 ≦ k ≦ N D} ). Further, when determining the value of the phase compensation offset ^ θ _qk for the subsequent sub-spectrum SSk (3 ≦ k ≦ N _D ), the phase compensated based on the determined value of the phase compensation offset ^ θ _qk The spectrum SSk (2 ≦ k ≦ N _D ) sequentially becomes the reference sub-spectrum SS (k−1). A detailed description of the method of estimating the value of the phase offset θ _{qk in} each of the phase estimators 204-k, that is, the method of determining the value of the phase compensation offset ^ θ _qk will be described later.

Each of the phase shifters 205-k adds the phase compensation offset ^ θ _qk to the sub-spectrum SSk input from the corresponding phase estimator 204-k, thereby changing the phase of the sub-spectrum SSk output by the frequency shifter 203-k. To compensate. That is, the phase is rotated so that the phase offset θ _qk added by the phase shifter 105-q-k provided in the transmission circuit 100 is subtracted from the phase of the sub-spectrum SSk output from the frequency shifter 203-k. The phase rotation given to SSk is eliminated, and the phase is returned to the same phase as that of the sub-spectrum SSk output from the band division filter 104-k. Then, each of the phase shifters 205-k outputs the sub-spectrum SSk whose phase has been compensated to the adder 206 and the corresponding phase estimator 204-k. A detailed description of the phase compensation method of the sub-spectrum SSk in each of the phase shifters 205-k will be described later.

  The adder 206 adds (synthesizes) each of the sub-spectrums SSk input from the corresponding frequency shifter 203-k or phase shifter 205-k after returning the shift of the frequency band. As a result, the reception circuit 200 obtains a modulation signal in the same frequency domain as the modulation signal generated by the DFT circuit 103 provided in the transmission circuit 100. Then, the adder 206 outputs a modulation signal obtained by adding the plurality of input sub-spectrums SSk to the IDFT circuit 207.

  The IDFT circuit 207 performs inverse discrete Fourier transform on the modulated signal input from the adder 206 to generate a time domain modulated signal. The IDFT circuit 207 then outputs the generated time domain modulation signal to the demodulation circuit 208.

  The demodulation circuit 208 demodulates the time-domain modulation signal input from the IDFT circuit 207 with a demodulation method corresponding to a modulation method such as QPSK that modulates the data signal transmitted from the modulation circuit 101 included in the transmission circuit 100. The data signal transmitted by the transmission circuit 100 is taken out.

Here, the method of estimating the value of the phase offset θ _{qk in} each of the phase estimators 204-k, that is, the method of determining the value of the phase compensation offset ^ θ _qk , and the phase of the sub-spectrum SSk in each of the phase shifters 205-k A compensation method for this will be described.

First, a method of calculating the correlation value R _{k — ab} calculated when the phase estimator 204-k estimates (determines) the value of the phase offset θ _qk will be described. Each sub-spectrum SSk included in the transmission signal transmitted from the transmission circuit 100 is distributed in frequency band by the frequency shifter 106-q-k included in the transmission circuit 100 as shown in FIG. ing. For this reason, in the receiving circuit 200, the frequency shifter 203-k returns the frequency band of each sub-spectrum SSk to the state before being shifted. Thereby, the sub-spectrum SS (k−1) and the sub-spectrum SSk input to the phase estimator 204-k are each sub-spectrum dispersed in the transmission circuit 100 as shown in FIG. The transition areas of SSk overlap each other. For this reason, adjacent sub-spectrum SSk transition regions have a common signal component. That is, each subspectrum SSk includes a common signal component that is discretized in the frequency band of each subspectrum SSk by the discrete Fourier transform by the DFT circuit 201. Therefore, the phase estimator 204-k calculates a correlation value R _{k_ab} from the overlapping signal components in the transition region.

FIG. 6 is a diagram illustrating a method for calculating the correlation value R _{k_ab} of the signal in the receiving device 20 configuring the communication system 1 of the present embodiment. FIG. 6 shows an example of the sub-spectrum SS (k−1) and the sub-spectrum SSk that are input to the phase estimator 204-k. In the following description, the signal components discretized in the transition region on the low frequency side in each sub-spectrum SSk are a _k1 , a _k2 ,..., A _{kp in} order from the low frequency side, The signal components discretized in the transition region on the high frequency side in the sub spectrum SSk are defined as b _k1 , b _k2 ,..., B _{kp in} order from the low frequency side. Note that p is the number of discretized signal components included in the frequency band of the transition region. At this time, for example, assuming that the frequency resolution in the DFT circuit 201 is r and the bandwidth of the frequency band in the transition region is B _t , the number p of signal components is p = [B _t / r]. However, [x] is the maximum integer not exceeding x.

Then, the correlation value R _{k_ab} is calculated based on the discretized signal component a and signal component b in the adjacent sub-spectrum SS (k−1) and sub-spectrum SSk. More specifically, the correlation value R k — _ab is expressed by the signal components b _{(k−1) 1} , b _(k− ) on the high frequency side of the sub-spectrum SS (k−1) as expressed in the following equation (7). _{1) 2, ···, b (} k1) and _p, the signal components of the low frequency side of the sub-spectrum _{SSk a k1, a k2, ···} , after multiplying the complex conjugate of _{a kp,} the frequency axis Calculated by smoothing in the direction.

FIG. 6 shows an example when the number of signal components is p = 3. The correlation value R _{k_ab in the} example shown in FIG. 6 is expressed by the following equation (8).

In the phase estimator 204-k, when the correlation value R _{k_ab} calculated in this way is equal to or _larger than a predetermined threshold value α, the value of the phase offset θ _qk is transmitted from the transmission circuit 100 by the phase shifter 105-q-k. It is determined that the added sub-spectrum SSk has been transmitted as a transmission signal. On the other hand, when the correlation value R _{k_ab} is less than the threshold value α, the phase estimator 204-k determines that the sub-spectrum SSk to which the value of the phase offset θ _qk is not added is transmitted as a transmission signal from the transmission circuit 100. To do.

When the phase estimator 204-k determines that the sub-spectrum SSk to which the value of the phase offset θ _qk is added by the phase shifter 105-q-k is transmitted from the transmission circuit 100 as a transmission signal, the calculated correlation value R _Based on _{k_ab} , the value of the phase compensation offset ^ θ _qk for returning each sub-spectrum SSk included in the transmission signal to the original phase is determined.

Subsequently, a method of determining the value of the phase compensation offset ^ θ _qk in each of the phase estimators 204-k and a method of compensating the phase of the sub-spectrum SSk in each of the phase shifters 205-k will be described. In the method of determining the value of the phase compensation offset ^ θ _qk by the phase estimator 204-k, the value of the phase compensation offset ^ θ _qk is sequentially _applied based on the correlation value R _{k_ab} calculated from the signal components in the overlapping transition region. decide.

First, in phase estimator _204-2 , based on correlation value _{R2_ab} calculated from the transition region between _subspectrum SS1 output from frequency shifter 203-1 and subspectrum SS2 output from frequency shifter 203-2. The value of the phase compensation offset ^ θ _q2 to be added to the sub-spectrum SS2 is determined by the following equation (9). Then, the phase estimator 204-2 outputs the determined value of the phase compensation offset ^ θ _q2 to the corresponding phase shifter 205-2.

In the above equation (9), k represents the corresponding band dividing filter 104-k. In the following description, for ease of explanation, a in the above equation (9) is a reference sub-spectrum SSk input to the phase estimator 204-k (sub-spectrum in the phase estimator 204-2). SS1, i.e. "1"), and b is the _{subspectrum SSk} that determines the value of the phase compensation offset ^ _{qq in} the phase estimator 204-k (subspectrum SS2 in the phase estimator 204-2, i.e. "2"")" Will be described. Therefore, the phase estimator 204-2 uses the value of the phase compensation offset ^ θ _q2 (= arctan (R 2 — ₁₂ )) determined by the above equation (9) from the calculated correlation value R 2 — ₁₂ and the corresponding phase shifter 205-2. Output to.

Then, the phase shifter 205-2 sets the value of the phase compensation offset ^ θ _q2 for the sub-spectrum SS2 input from the corresponding phase estimator 204-2 to the sub-spectrum SS2 input from the corresponding frequency shifter 203-2. By adding, the phase of the sub-spectrum SS2 output from the frequency shifter 203-2 is compensated. More specifically, the phase shifter 205-2 includes the value of Expression (10) based on the value of the phase compensation offset ^ θ _{q2 in} the sub-spectrum SS2 (in the phase shifter 205-2, “exp (−j ^ θ _q2 ) ") To compensate for the phase of the subspectrum SS2.

Subsequently, in the phase estimator 204-3, the correlation value R 3 — ₂₃ calculated from the transition region between the sub-spectrum SS2 whose phase is compensated by the phase shifter 205-2 and the sub-spectrum SS3 output from the frequency shifter _203-3. Based on the above, the value (= arctan (R 3 — ₂₃ )) of the phase compensation offset ^ θ _q3 to be added to the sub-spectrum SS3 is determined by the above equation (9), and is output to the corresponding phase shifter 205-3.

Then, the phase shifter 205-3 converts the sub-spectrum SS3 input from the corresponding frequency shifter 203-3 to the value of the phase compensation offset ^ θ _q3 for the sub-spectrum SS2 input from the corresponding phase estimator 204-3. The phase of the sub-spectrum SS3 is compensated by multiplying the value of the formula (10) based on the above (in the phase shifter 205-3, “exp (−j ^ θ _q3 )”).

Then, since the phase estimator 204-k (4 ≦ k ≦ N D) Similarly, on the basis of the sub-spectrum SS (k-1) and the correlation value _{R K_ab} calculated from the transition zone between the sub-spectrum SSk, The value of the phase compensation offset ^ θ _qk (= arctan (R _{k — ab} )) is determined by the above equation (9). Similarly, the phase shifter 205-k (4 ≦ k ≦ N _D ) corresponding to the phase estimator 204-k also receives the phase compensation offset input to the sub-spectrum SSk input from the corresponding frequency shifter 203-k. The phase of the sub-spectrum SSk is sequentially compensated by multiplying the value of the above formula (10) based on the value of _{{circumflex over} (θ) _{} qk} (in the phase shifter 205-k, “exp (−j ^ θ _qk )”).

  Next, operation | movement of the receiver 20 with which the communication system 1 of this embodiment was equipped is demonstrated. FIG. 7 is a flowchart showing a processing procedure of reception processing in the reception circuit 200 provided in the reception device 20 configuring the communication system 1 of the present embodiment. FIG. 8 is a diagram illustrating an example of signals at each stage of reception processing by the reception circuit 200 included in the reception device 20 included in the communication system 1 of the present embodiment. FIG. 8 shows an example of the frequency band of the signal of the component in the receiving circuit 200 at each stage of the receiving process by the receiving device 20. In the following description, the procedure of the reception process in the reception apparatus 20 is described in the case where the reception circuit 200 illustrated in FIG. 5 is the reception circuit 200 corresponding to the transmission circuit 100 including four band division filters 104-k. As an example, description will be made with reference to a diagram showing frequency bands of respective signals shown in FIG.

  First, in the reception process in the reception device 20, the DFT circuit 201 performs a discrete Fourier transform on the time-domain transmission signal transmitted from the transmission device 10 to generate a frequency-domain signal. Then, the DFT circuit 201 outputs the generated frequency domain signal to each of the extraction filters 202-1 to 202-4 (step S200).

  Subsequently, each of the extraction filter 202-1 to extraction filter 202-4 multiplies a frequency domain signal input from the DFT circuit 201 by a predetermined extraction coefficient, thereby corresponding bands provided in the transmission circuit 100. A signal corresponding to the sub-spectrum SSk generated by the division filter 104-k is extracted. Then, each of the extraction filters 202-1 to 202-4 outputs a signal corresponding to the extracted sub-spectrum SSk to each of the corresponding frequency shifters 203-1 to 203-4 (step S201). .

  Subsequently, each of the frequency shifter 203-1 to the frequency shifter 203-4 shifts the frequency band of the input sub-spectrum SSk to a target frequency band, so that the frequency shifter 106 -q provided in the transmission circuit 100 is shifted. -K returns to the frequency band before each sub-spectrum SSk is shifted. Each of the frequency shifter 203-1 to frequency shifter 203-4 converts the sub-spectrum SSk after shifting the frequency band into the corresponding adder 206 or the corresponding phase estimator 204-2 to phase estimator 204-. 4 and the phase shifters 205-2 to 205-4 (step S202).

The state of overlapping with the frequency band of the adjacent sub-spectrum SSk from each of the frequency shifter 203-1 to frequency shifter 203-4 by the processing of step S202, that is, each sub-spectrum SSk dispersed in the transmission circuit 100. Each of the sub-spectrums SSk similar to the state in which the transition regions overlap are output. For example, when the transmission signal received by the reception circuit 200 is a transmission signal including the sub-spectrum SSk to which the value of the phase offset θ _qk is added by the phase shifter 105-qk, the frequency shifter 203-2 to the frequency shifter The sub-spectrum SS2 to sub-spectrum SS4 in the frequency domain output from each of 203-4 are signals having characteristics different from the characteristics of the phase θ in the sub-spectrum SS1 output from the frequency shifter 203-1. FIG. 8A shows sub-spectrum SS2 to sub-spectrum SS4 having characteristics different from the characteristics of phase θ of sub-spectrum SS1 in sub-spectrum SSk output from each of frequency shifter 203-1 to frequency shifter 203-4. Show.

Subsequently, each of the phase estimator 204-2 to phase estimator 204-4 compares the phase θ between the subspectrum SS (k−1) and the subspectrum SSk with which the input frequency bands are adjacent, The difference of the phase θ in the transition region is calculated as the correlation value R _{k_ab} . Then, each of the phase estimator 204-2 to phase estimator 204-4 estimates (determines) the value of the phase compensation offset ^ θ _qk corresponding to the input sub-spectrum SSk based on the calculated correlation value R _{k_ab.} ) Then, each of the phase estimator 204-2 to phase estimator 204-4 _uses the value of the phase compensation offset ^ θ _qk corresponding to the estimated (determined) sub-spectrum SSk as the corresponding phase shifter 205-2 to phase shifter. It outputs to 205-4 (step S203).

Subsequently, each of the phase shifters 205-2 to 205-4 receives the phase compensation offsets input from the corresponding phase estimators 204-2 to 204-4 corresponding to the input sub-spectrum SSk. By adding the value of _{{circumflex over} (θ) _} qk, the phase of the input sub-spectrum SSk is compensated. Then, each of the phase shifters 205-k outputs the sub-spectrum SSk whose phase has been compensated to the adder 206 and the corresponding phase estimator 204-k.

FIG. 8B shows the phase θ of the subspectrum SS2 to the subspectrum SS4 output from the frequency shifter 203-2 to the frequency shifter 203-4 by the phase shifter 205-2 to the phase shifter 205-4, respectively. , _I.e. , returning to the phase before the phase offset θ _qk is added by the phase shifter 105 -q-k provided in the transmission circuit 100. More specifically, the value of the phase compensation offset ^ θ ₂₂ is added to the sub-spectrum SS2, the value of the phase compensation offset ^ θ ₂₃ is added to the sub-spectrum SS3, and the value of the phase compensation offset ^ θ ₂₄ is added to the sub-spectrum SS4. The state of adding is shown. Thereby, each phase θ of the sub-spectrum SS2 to the sub-spectrum SS4 has the same characteristics as the characteristics of the phase θ in the sub-spectrum SS1. As can be seen from FIG. 8B, the phase compensation offset ^ θ _qk is not added to the sub-spectrum SS1. This is because, as described above, in the transmission circuit 100, the phase offset θ _qk is not added to the sub-spectrum SS1.

  The compensation of phase θ for each of sub-spectrum SS2 to sub-spectrum SS4 in each of phase shifter 205-2 to phase shifter 205-4 is sequentially performed from the low-frequency side sub-spectrum SSk as described above. For this reason, the adder 206 completes the processing for all the sub-spectrums SSk depending on whether or not the sub-spectrum SSk is input from each of the frequency shifter 203-1 and the phase shifters 205-2 to 205-4. It is determined whether or not (step S205).

  If the result of determination in step S205 is that processing has not been completed for all sub-spectrums SSk (“NO” in step S205), the process returns to step S203, and the processes in steps S203 to S205 are repeated.

  On the other hand, if the result of determination in step S205 is that processing has been completed for all the sub-spectrums SSk (“YES” in step S205), the adder 206 includes the frequency shifter 203-1 and the phase shifter 205-2 to phase shifter. Each sub-spectrum SSk from 205-4, that is, each sub-spectrum SSk after returning the shift of the frequency band is added (synthesized). Then, the adder 206 outputs an output signal obtained by adding the plurality of sub-spectrums SSk to the IDFT circuit 207 (step S206). That is, the adder 206 receives the sub-spectrum SS1 to sub-spectrum SS4 after returning the frequency band shift from the frequency shifter 203-1 and the phase shifters 205-2 to 205-4. Then, an output signal obtained by adding (combining) all the input sub-spectrums SSk is output to the IDFT circuit 207.

  Subsequently, the IDFT circuit 207 performs inverse discrete Fourier transform on the output signal input from the adder 206 to generate a time domain modulated signal. Then, the IDFT circuit 207 outputs the generated time domain modulation signal to the demodulation circuit 208 (step S207).

  Subsequently, the demodulation circuit 208 uses a demodulation method corresponding to a modulation method such as QPSK obtained by modulating the data signal transmitted from the modulation circuit 101 included in the transmission circuit 100 by using the time domain modulation signal input from the IDFT circuit 207. Demodulate (step S208). Thereby, the receiving circuit 200 takes out the data signal transmitted by the transmitting circuit 100.

  As described above, in the reception device 20, the reception circuit 200 extracts each sub-spectrum SSk included in the time-domain transmission signal transmitted from the transmission device 10, and restores the frequency band shift to the original state. The receiving apparatus 20 further sequentially estimates the phase difference between adjacent sub-spectrums SSk, determines the value of the phase compensation offset for returning each sub-spectrum SSk to the original phase, and each sub-spectrum SSk. The data signal transmitted by the transmission circuit 100 is taken out by adding (synthesizing) the signal after compensating the phase (after returning to the original phase). As a result, the receiving apparatus 20 receives the transmission signal transmitted from the transmission circuit 100, which reduces the frequency of in-phase synthesis between the sub-spectrums and reduces the peak-to-average power ratio. can do.

  As described above, according to the mode for carrying out the present invention, the transmission apparatus converts the modulation signal including the data signal into a frequency domain signal and divides the signal into a plurality of sub-spectrums, and then converts each sub-spectrum. By shifting the frequency band of the spectrum, each subspectrum is arranged at distributed positions on the frequency axis. At this time, in the transmission apparatus according to the embodiment for carrying out the present invention, different phase rotations are given by adding different phase offsets to the respective sub-spectrums, and the phases of the sub-spectrums are combined in phase. Reduce frequency. Further, the transmission apparatus of the embodiment for carrying out the present invention calculates the peak-to-average power ratio after adding (synthesizing) each sub-spectrum and converting it into a time-domain modulation signal, and the highest peak-to-average power ratio. Is transmitted as a final transmission signal.

  Further, according to the embodiment for carrying out the present invention, the receiving apparatus converts the received transmission signal into a frequency domain signal, and then extracts each sub-spectrum included in the frequency domain signal to Undo the shift. Then, in the receiving apparatus of the embodiment for carrying out the present invention, after compensating the phase of each sub-spectrum based on the phase difference between adjacent sub-spectrums (after returning to the original phase), Addition (combination) is performed and converted into a time-domain modulation signal to extract a data signal.

  As a result, in the embodiment for implementing the present invention, the modulated signal obtained by modulating the data signal to be transmitted is converted into a frequency domain signal and divided into a plurality of sub-spectrums, and then each sub-spectrum is displayed on the frequency axis. In a communication system that disperses and transmits in a continuous frequency band, it is possible to reduce the frequency with which the phases of the sub-spectrums are combined in phase, and perform transmission with a reduced peak-to-average power ratio. As a result, in the communication system according to the embodiment for carrying out the present invention, it is not necessary to use a large-sized and high-power amplifier as the peak-to-average power ratio is reduced, and the transmission apparatus can be downsized. it can.

  In the present embodiment, the configuration in which the frequency shifter 106-q-k is arranged between the phase shifter 105-q-k and the adder 107-q has been described as the configuration of the transmission circuit 100. However, the position where the frequency shifter 106-qk is arranged in the transmission circuit 100 is not limited to the mode for carrying out the present invention. For example, the frequency shifter 106-qk is disposed between the band division filter 104-k and the phase shifter 105-qk, and is output from the band division filter 104-k by the frequency shifter 106-qk. After shifting the frequency band of the sub-spectrum SSk to the target frequency band, that is, distributing the sub-spectrum SSk to the target frequency band, the phase shifter 105-qk adds a phase offset to each sub-spectrum SSk. May be.

  In the present embodiment, the configuration in which the frequency shifter 203-k is disposed between the extraction filter 202-k and the phase estimator 204-k has been described as the configuration of the receiving circuit 200. However, the position where the frequency shifter 203-k is arranged in the receiving circuit 200 is not limited to the form for carrying out the present invention. For example, the frequency shifter 203-k is disposed between the phase shifter 205-k and the adder 206, and after the phase offset provided by the transmission circuit 100 is compensated by the phase shifter 205-k (phase rotation due to the phase offset). The frequency shifter 203-k shifts the frequency band of the sub-spectrum SSk to a target frequency band, that is, returns the frequency band of each sub-spectrum SSk to the frequency band before being shifted. Also good.

  In the present embodiment, the case where the configuration of the transmission circuit 100 is different in the configuration of the first transmission sequence circuit group and the transmission sequence circuit group other than the first has been described. More specifically, a configuration has been described in which the first transmission sequence circuit group does not include the phase shifter 105 and the transmission sequence circuit groups other than the first transmission sequence circuit group include the phase shifter 105. However, the configuration of the transmission sequence circuit group included in the transmission circuit 100 is not limited to the mode for carrying out the present invention. For example, all transmission series circuit groups may include the phase shifter 105. In the case of this configuration, the phase shifter 105 included in the first transmission series circuit group performs the process of giving the phase rotation with the rotation amount “0”, so that the phase shifter 105 described in the present embodiment is The same operation (processing) as that of the first transmission sequence circuit group not included can be performed. With this configuration, the circuit scale of the transmission circuit 100 may increase, but the operation (processing) for generating a transmission signal is the same in all transmission series circuit groups, and the processing in the transmission apparatus is performed. The effect that it becomes easy is acquired.

  In the present embodiment, the configuration of the transmission circuit 100 includes a plurality of transmission sequence circuit groups, and each transmission sequence circuit group is a phase offset with respect to each sub-spectrum SSk duplicated by the band division filter 104-k. The configuration has been described in which the processes of adding the frequency band, shifting the frequency band, adding (combining) the sub-spectrum SSk, inverse discrete Fourier transform, and calculating the peak-to-average power ratio are executed in parallel. However, the number of transmission series circuit groups provided in the transmission circuit 100 is not limited to the form for carrying out the present invention. For example, a configuration in which only one transmission sequence circuit group having the same configuration as the second transmission sequence circuit group shown in FIG. 2 is provided and the same processing for each sub-spectrum SSk is sequentially executed, that is, in FIG. A configuration may be adopted in which each processing is executed in the same manner as the processing procedure of the transmission processing in the transmission circuit 100 described with reference to the flowchart shown. With this configuration, there is a possibility that the speed at which the transmission signal is transmitted may be reduced, but there is an effect that the circuit scale of the transmission circuit 100 can be greatly reduced.

  In the present embodiment, the case where the modulation scheme is QPSK has been described. However, the modulation scheme is not limited to the mode for carrying out the present invention. For example, the idea of the present invention can be similarly applied to other modulation schemes such as 8PSK (8 Phase Shift Keying).

  The embodiment of the present invention has been described above with reference to the drawings. However, the specific configuration is not limited to this embodiment, and includes various modifications within the scope of the present invention. It is.

DESCRIPTION OF SYMBOLS 1 ... Communication system 10 ... Transmission apparatus 100 ... Transmission circuit (transmission apparatus)
101: Modulation circuit (transmission device)
102 ... Waveform shaping filter (transmitting device)
103 ... DFT (Discrete Fourier Transform) circuit (transmitting device)
_{104,104-1,104-2,104-N} D ··· band division filter (transmitting device)
_{105,105-2-1,105-2-2,105-2-N D, 105-C} -1,105-C-2,105-C-N D ··· phaser (transmission device, the transmission signal Generation means, phase rotation means)
_{106,106-1-1,106-1-2,106-1-N D, 106-2-1,106-2-2,106-2-N} D, 106-C-1,106-C- _2,106-C-N D ··· frequency shifter (transmitting device, transmitting signal generating unit, signal variance means)
107, 107-1, 107-2, 107 -C... Adder (transmission device, transmission signal generation means, signal dispersion means)
108, 108-1, 108-2, 108-C ... IDFT (Inverse Discrete Fourier Transform) circuit (transmitting device, transmission signal generating means, signal dispersion means)
109, 109-1, 109-2, 109-C... PAPR (peak-to-average power ratio) calculation circuit (transmission apparatus, transmission signal selection means)
110: Minimum PAPR selector (transmitter, transmission signal selection means)
20... Receiving device 200... Receiving circuit (receiving device)
201: DFT (Discrete Fourier Transform) circuit (receiving device)
_{202,202-1,202-2,202-3,202-N} D ··· extraction filter (reception apparatus)
_{203,203-1,203-2,203-3,203-N} D ··· frequency shifter (receiving apparatus)
_{204,204-2,204-3,204-N} D ··· phase estimator (receiving apparatus, the phase estimation means)
_{205,205-2,205-3,205-N} D ··· phaser (receiving apparatus, the phase compensation means)
206... Adder (receiving device)
207... IDFT (Inverse Discrete Fourier Transform) circuit (receiving device)
208: Demodulation circuit (receiving device)
501... Modulation circuit 502... Waveform shaping filter 503... DFT (discrete Fourier transform) circuit 504, 504-1, 504-2, 504-ND _D. Band division filters 505, 505-1 _505-2,505-N D ··· frequency shifter 506 ... adder 507 ... IDFT (inverse discrete Fourier transform) circuit 601 ... DFT circuit 602,602-1,602-2,602-N _D: Multiple extraction filters 603, 603-1, 603-2, 603-ND _D: Frequency shifter 604 ... Adder 605 ... IDFT circuit 606 ... Demodulation circuit

Claims (8)

A modulated signal obtained by modulating a data signal to be transmitted is converted into a frequency domain signal and divided into N (N is an integer of 2 or more) sub-spectrums, and then each of the sub-spectrums has a discontinuous frequency on the frequency axis. A transmission device for transmitting a transmission signal distributed in a band,
Each of the divided sub-spectrums is duplicated to generate a plurality of transmission sequences including each of the divided sub-spectrums, and the phase rotation different for each of the transmission sequences and the sub-spectrums included in the transmission sequences Transmission signal generating means for generating the transmission signal of each of the transmission sequences including the sub-spectrum to which the phase rotation corresponding to each of the divided sub-spectrums is provided,
For each transmission signal of each of the transmission sequences, a peak-to-average power ratio that is a ratio of maximum power to average power is calculated, and the transmission signal of the transmission sequence having the smallest peak-to-average power ratio is selected and transmitted. Transmission signal selection means for
A transmission device comprising: The transmission signal generating means includes
Corresponding to each of the transmission sequences, the phase is added to each sub-spectrum by adding a different phase offset in the corresponding transmission sequence to each of the divided sub-spectrums included in the corresponding transmission sequence. A plurality of phase rotation means for imparting rotation;
A plurality of the sub-spectrums corresponding to each of the phase rotation means and provided with the phase rotation by the corresponding phase rotation means, and each of the sub-spectrums including the discontinuous frequency bands on the frequency axis Signal dispersion means for generating the transmission signal in the time domain dispersed in
The transmission apparatus according to claim 1, further comprising: N (N is an integer of 2 or more) sub-spectrums distributed in discontinuous frequency bands on the frequency axis included in the received transmission signal are extracted, and the frequency bands of the extracted sub-spectrums are used as frequencies. A receiving device that demodulates a modulated signal synthesized after returning to a frequency band before being distributed on an axis,
For each sub-spectrum after returning the frequency band, calculate the correlation of each adjacent sub-spectrum where the frequency band overlaps on the frequency axis, and based on the calculated correlation Estimate the phase offset added to give the phase rotation to the sub-spectrum, and determine the phase compensation offset to compensate the phase rotation given to each of the sub-spectrum based on the estimated phase offset Phase estimation means for
Phase compensation means for compensating for the phase rotation applied to each sub-spectrum by adding the phase compensation offset corresponding to each sub-spectrum;
With
Demodulating a modulation signal obtained by synthesizing the sub-spectrum compensated for the phase rotation given by the phase compensation means;
A receiving apparatus. The phase estimation means includes
Of each of the sub-spectrums returned to the frequency band before being dispersed on the frequency axis, a reference k-1th sub-spectrum (k is an integer of 2 ≦ k ≦ N), The k-1 sub-spectrum on the frequency axis overlaps the frequency band, and the k-th sub-spectrum is in the frequency band on the higher frequency side than the k-1 sub-spectrum. Multiplying the signal component of the transition region on the high frequency side in the (k−1) -th sub-spectrum by the complex conjugate of the signal component of the transition region on the low-frequency side in the k-th sub-spectrum, the k th sub-spectrum. The phase compensation offset corresponding to a spectrum is determined, and the kth subspectrum whose phase rotation is compensated by the determined phase compensation offset is determined. As a quasi, the phase compensation corresponding to the (k + 1) th sub-spectrum in the frequency band higher in frequency than the k-th sub-spectrum overlaps the k-th sub-spectrum on the frequency axis. Determine the offset,
The phase compensation means includes
By sequentially multiplying each phase compensation offset corresponding to the sub-spectrum, the phase rotation imparted to each sub-spectrum is compensated,
The receiving device is
Demodulating a modulation signal obtained by synthesizing the k-1 th sub-spectrum to the N th sub-spectrum;
The receiving apparatus according to claim 3. A modulated signal obtained by modulating a data signal to be transmitted is converted into a frequency domain signal and divided into N (N is an integer of 2 or more) sub-spectrums, and then each of the sub-spectrums has a discontinuous frequency on the frequency axis. A transmission apparatus that transmits a transmission signal dispersed in a band, and N (N is an integer of 2 or more) sub-spectrums distributed in discontinuous frequency bands on the frequency axis included in the received transmission signal are extracted. A receiving device that demodulates the modulated signal synthesized after returning the extracted frequency band of each of the sub-spectrums to the frequency band before being distributed on the frequency axis,
The transmitter is
Each of the divided sub-spectrums is duplicated to generate a plurality of transmission sequences including each of the divided sub-spectrums, and the phase rotation different for each of the transmission sequences and the sub-spectrums included in the transmission sequences Transmission signal generating means for generating the transmission signal of each of the transmission sequences including the sub-spectrum to which the phase rotation corresponding to each of the divided sub-spectrums is provided,
For each transmission signal of each of the transmission sequences, a peak-to-average power ratio that is a ratio of maximum power to average power is calculated, and the transmission signal of the transmission sequence having the smallest peak-to-average power ratio is selected and transmitted. Transmission signal selection means for
With
The receiving device is:
For each sub-spectrum after returning the frequency band, calculate the correlation of each adjacent sub-spectrum where the frequency band overlaps on the frequency axis, and based on the calculated correlation The phase offset added to give the phase rotation to the sub-spectrum is estimated, and the phase compensation offset for compensating the phase rotation given to each sub-spectrum is determined based on the estimated phase offset Phase estimation means for
Phase compensation means for compensating for the phase rotation applied to each sub-spectrum by adding the phase compensation offset corresponding to each sub-spectrum;
With
Demodulating a modulation signal obtained by synthesizing the sub-spectrum compensated for the phase rotation given by the phase compensation means;
A communication system characterized by the above. A modulated signal obtained by modulating a data signal to be transmitted is converted into a frequency domain signal and divided into N (N is an integer of 2 or more) sub-spectrums, and then each of the sub-spectrums has a discontinuous frequency on the frequency axis. A transmission method in a transmission apparatus for transmitting a transmission signal distributed in a band,
Each of the divided sub-spectrums is duplicated to generate a plurality of transmission sequences including each of the divided sub-spectrums, and the phase rotation different for each of the transmission sequences and the sub-spectrums included in the transmission sequences And a transmission signal generation procedure for generating the transmission signal of each of the transmission sequences including the sub-spectrum to which the phase rotation corresponding to each of the divided sub-spectrums is provided,
For each transmission signal of each of the transmission sequences, a peak-to-average power ratio that is a ratio of maximum power to average power is calculated, and the transmission signal of the transmission sequence having the smallest peak-to-average power ratio is selected and transmitted. Transmission signal selection procedure to be
The transmission method characterized by including. N (N is an integer of 2 or more) sub-spectrums distributed in discontinuous frequency bands on the frequency axis included in the received transmission signal are extracted, and the frequency bands of the extracted sub-spectrums are used as frequencies. A receiving method in a receiving apparatus for demodulating a modulated signal synthesized after returning to a frequency band before being distributed on an axis,
For each sub-spectrum after returning the frequency band, calculate the correlation of each adjacent sub-spectrum where the frequency band overlaps on the frequency axis, and based on the calculated correlation Estimate the phase offset added to give the phase rotation to the sub-spectrum, and determine the phase compensation offset to compensate the phase rotation given to each of the sub-spectrum based on the estimated phase offset Phase estimation procedure to
A phase compensation procedure for compensating for the phase rotation imparted to each sub-spectrum by adding the phase compensation offset corresponding to each sub-spectrum;
Including
Demodulating a modulation signal obtained by synthesizing the sub-spectrum compensated for the phase rotation given by the phase compensation procedure;
And a receiving method. A modulated signal obtained by modulating a data signal to be transmitted is converted into a frequency domain signal and divided into N (N is an integer of 2 or more) sub-spectrums, and then each of the sub-spectrums has a discontinuous frequency on the frequency axis. Transmission method in transmission apparatus for transmitting transmission signal dispersed in band, and N (N is an integer of 2 or more) sub-spectrums distributed in discontinuous frequency bands on the frequency axis included in the received transmission signal And a receiving method in a receiving device that demodulates the modulated signal synthesized after returning the extracted frequency bands of the sub-spectrums to the frequency bands before being distributed on the frequency axis. And
The transmission method is:
Each of the divided sub-spectrums is duplicated to generate a plurality of transmission sequences including each of the divided sub-spectrums, and the phase rotation different for each of the transmission sequences and the sub-spectrums included in the transmission sequences And a transmission signal generation procedure for generating the transmission signal of each of the transmission sequences including the sub-spectrum to which the phase rotation corresponding to each of the divided sub-spectrums is provided,
For each transmission signal of each of the transmission sequences, a peak-to-average power ratio that is a ratio of maximum power to average power is calculated, and the transmission signal of the transmission sequence having the smallest peak-to-average power ratio is selected and transmitted. Transmission signal selection procedure to be
Including
The receiving method is:
For each sub-spectrum after returning the frequency band, calculate the correlation of each adjacent sub-spectrum where the frequency band overlaps on the frequency axis, and based on the calculated correlation A phase offset added to give the phase rotation to the sub-spectrum is estimated, and a phase compensation offset for compensating the phase rotation given to each of the sub-spectrums based on the estimated phase offset A phase estimation procedure to determine;
A phase compensation procedure for compensating for the phase rotation imparted to each sub-spectrum by adding the phase compensation offset corresponding to each sub-spectrum;
Including
Demodulating a modulation signal obtained by synthesizing the sub-spectrum compensated for the phase rotation given by the phase compensation procedure;
A communication method characterized by the above.

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