JP2015162928A - Ac motor controlling device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide an AC motor controlling device that reduces an impact of sharp change in a current detection value in an AC motor controlling device that detects phase current of one phase among multiple phases by a current sensor and calculates current estimation values of the other phases without using current command values in an α-β coordination system.SOLUTION: A current estimating unit 40 calculates α-axis current iα and β-axis current iβ in a fixed coordination system comprising an α-axis according with an axis of a W phase which is a sensor phase and a β-axis orthogonal to the α-axis. An α-axis current calculation unit 41 calculates the α-axis current iα on the basis of a current detection value iw_sns of the sensor phase. A β-axis current calculation unit 42 calculates the β-axis current iβ on the basis of shift current Ishift, which is an integrated value of a section from a phase as far back as 180° or 540° from any reference phase to the reference phase of the current detection value iw_sns of the sensor phase. A sensor phase reference current phase calculation unit 43 calculates sensor phase reference current phase θx. An other-phase current estimating unit 44 calculates a U-phase current estimation value iu_est.

Description

  The present invention relates to a control apparatus for an AC motor that detects a phase current of one phase among multiple phases by a current sensor and controls energization of the AC motor.

  In recent years, electric vehicles and hybrid vehicles equipped with AC motors have attracted attention as a power source for vehicles due to social demands for low fuel consumption and low exhaust emissions. For example, in a hybrid vehicle, a DC power source composed of a secondary battery or the like and an AC motor are connected via a power converter composed of an inverter or the like, and the DC voltage of the DC power source is converted into an AC voltage by the inverter. There is one that drives an AC motor.

  In such a control device for an AC motor mounted in a hybrid vehicle or an electric vehicle, a current sensor for detecting a phase current is provided in one “sensor phase”, and the “sensor phase” estimated based on the current detection value of the sensor phase is provided. A technique for controlling energization of an AC motor by feeding back an estimated current value other than “estimated phase” is known (see, for example, Patent Document 1). By providing the current sensor only in one phase, the number of current sensors is reduced, and the size near the output terminal of the inverter is reduced and the cost of the control system is reduced.

  In the control apparatus for an AC motor disclosed in Patent Document 1, an α-β coordinate system in which an α axis is defined in a direction that coincides with a sensor phase and a β axis is defined in a direction orthogonal to the α axis is used. The α-axis current iα is calculated based on iw_sns, and the β-axis current iβ is calculated based on the differential value Δiα of the α-axis current iα. Then, the sensor phase reference current phase θx is calculated from the α-axis current iα and the β-axis current iβ, and the estimated phase current estimation value iu_est is calculated based on the sensor phase reference current phase θx and the sensor phase current detection value iw_sns. calculate.

This control device pays attention to the fact that the α-axis current iα and the β-axis current iβ are in the relationship between the sin wave and the cosine wave, and the differential value of the α-axis current iα without using the current command value of the phase other than the sensor phase. A β-axis current iβ can be calculated based on Δiα. Therefore, the present invention can be applied to a control mode in which a current command value is not used for energization control of an AC motor, such as a torque feedback control rectangular wave control mode. The “rectangular wave” in the rectangular wave control mode refers to a waveform of one pulse in one cycle of current.
In the actual control, the “differential value Δiα” of the α-axis current iα is calculated as a difference value of the α-axis current iα with respect to the electrical angle movement amount during the current detection timing.

JP 2013-172594 A

In the control apparatus for an AC electric motor disclosed in Patent Document 1, as a method of calculating the β-axis current iβ without using the current command value, the differential calculation of the α-axis current iα is performed. However, the differential calculation has a problem that it is easily affected by a sudden change in the current detection value due to noise or the like.
The present invention has been made in view of the above-described problems, and the object of the present invention is to detect a phase current of one phase among the polyphases with a current sensor, without using a current command value in an α-β coordinate system. Another object of the present invention is to provide an AC motor control device that reduces the influence of a sudden change in a current detection value in an AC motor control device that calculates an estimated current value of another phase.

The present invention relates to an inverter that drives a multi-phase AC motor having three or more phases, a current sensor that detects a current flowing in one sensor phase among the multi-phases of the AC motor, and a plurality of switching elements that constitute the inverter. The present invention relates to an AC motor control device including control means for switching on / off to control energization of the AC motor.
Here, the “AC motor” includes an AC drive motor, a generator, and a motor generator. For example, a motor that is used as a main machine of a hybrid vehicle or an electric vehicle and generates torque for driving drive wheels. Applicable to generators. In addition, for example, an electric motor control device that drives a motor generator corresponds to an “AC electric motor control device”.

The control means of the present invention provides a current phase based on a sensor phase based on an α-axis current and a β-axis current in a fixed coordinate system comprising an α axis that coincides with the axis of the sensor phase and a β axis that is orthogonal to the α axis. Current estimation means for calculating a sensor phase reference current phase (θx), and calculating a current estimation value of a phase other than the sensor phase based on the sensor phase reference current phase and the detected current value (iw_sns) of the sensor phase Have.
The current estimation means calculates an α-axis current based on the current detection value of the sensor phase, and “a reference phase from a phase that is 180 [°] or 540 [°] backward from an arbitrary reference phase of the current detection value of the sensor phase. The β-axis current is calculated based on the “integrated value in the interval until”.

The current estimation means of the present invention calculates the β-axis current based on the integral value in the predetermined phase section of the detected current value of the sensor phase without using the current command value. Then, a sensor phase reference current phase is calculated based on the α-axis current and the β-axis current, and further, an estimated current value of another phase is calculated from the detected current value of the sensor phase and the sensor phase reference current phase. Therefore, the present invention is applicable to current estimation in a control mode that does not use a current command value, such as a rectangular wave control mode of a torque feedback control method.
Also, in the present invention, unlike the α-axis current differential calculation according to the prior art disclosed in Patent Document 1, the β-axis current is calculated based on an integral value in a predetermined phase section of the sensor phase current detection value. Compared to the differential calculation, the calculation of the present invention reflects the accumulated result of past current detection values, so that the influence of sudden change due to noise or the like can be reduced.

  The current estimation means of the present invention is a sensor phase current detection value in at least one of “switch timing” for switching on / off of a switching element or “intermediate timing” set at least once between successive switch timings. It is preferable to calculate the α-axis current based on the above and calculate the integral value of the α-axis current in at least one of the phase intervals between the switch timings or the phase interval between the intermediate timings. As a result, the β-axis current iβ can be accurately calculated without being substantially affected by the detected current value that irregularly increases or decreases between the switch timing and the intermediate timing.

It is a figure which shows the structure of the motor drive system to which the control apparatus of the alternating current motor by one Embodiment of this invention is applied. 1 is an overall configuration diagram of an AC motor control device according to an embodiment of the present invention. FIG. It is a block diagram which shows the structure of the control part of the control apparatus of the alternating current motor of FIG. It is a block diagram which shows the structure of the electric current estimation part of the control part of FIG. It is a figure explaining the fixed coordinate system (alpha-beta coordinate system) on the basis of a sensor phase. It is a figure explaining a W phase current waveform and a W phase orthogonal current waveform. It is a figure explaining the switch timing and intermediate | middle timing in a rectangular wave control mode. It is a flowchart of the electric current estimation process by one Embodiment of this invention.

Hereinafter, an AC motor control apparatus for controlling driving of an AC motor according to the present invention will be described with reference to the drawings. In the following embodiments, substantially the same components are denoted by the same reference numerals and description thereof is omitted.
(One embodiment)
As shown in FIG. 1, an electric motor control device 10 as an “AC electric motor control device” according to an embodiment of the present invention is applied to an electric motor drive system 1 that drives a hybrid vehicle.
The motor drive system 1 includes an AC motor 2, a DC power source 8, a motor control device 10, and the like.
The AC motor 2 is an electric motor that generates torque for driving the drive wheels 6 of the electric vehicle, for example. The AC motor 2 of the present embodiment is a permanent magnet type synchronous three-phase AC motor.

  The electric vehicle includes a vehicle that drives the drive wheels 6 with electric energy, such as a hybrid vehicle, an electric vehicle, and a fuel cell vehicle. The electric vehicle of the present embodiment is a hybrid vehicle including the engine 3, and the AC electric motor 2 is transmitted from the engine 3 and the drive wheel 6 as a function of an electric motor that generates torque for driving the drive wheel 6. It is a so-called motor generator (denoted as “MG” in the drawing) having a function as a generator that can be driven by the kinetic energy of the vehicle to generate electric power.

The AC motor 2 is connected to the axle 5 via a gear 4 such as a transmission. Thereby, the driving force of the AC motor 2 drives the drive wheels 6 by rotating the axle 5 via the gear 4.
The DC power supply 8 is a chargeable / dischargeable power storage device such as a secondary battery such as nickel metal hydride or lithium ion or an electric double layer capacitor. The DC power supply 8 is connected to an inverter 12 (see FIG. 2) of the motor control device 10 and is configured to be able to exchange power with the AC motor 2 via the inverter 12.

  The vehicle control circuit 9 is configured by a microcomputer or the like, and includes a CPU, a ROM, an I / O, a bus line that connects these, and the like, all of which are not shown. The vehicle control circuit 9 controls the entire electric vehicle by software processing by executing a program stored in advance by the CPU or hardware processing by a dedicated electronic circuit.

The vehicle control circuit 9 obtains signals from various sensors and switches such as an accelerator signal from an accelerator sensor (not shown), a brake signal from a brake switch, a shift signal from a shift switch, and a vehicle speed signal related to the vehicle speed. It is configured to be possible. The vehicle control circuit 9 detects the driving state of the vehicle based on these acquired signals and outputs a torque command value trq ^* corresponding to the driving state to the motor control device 10. The vehicle control circuit 9 outputs a command signal to an engine control circuit (not shown) that controls the operation of the engine 3.

As shown in FIG. 2, the motor control device 10 includes an inverter 12, a current sensor 13, and a control unit 15 as “control means”.
A boost voltage of a DC power source by a boost converter (not shown) is input to the inverter 12 as a system voltage VH. The inverter 12 has six switching elements (not shown) that are bridge-connected. As the switching element, for example, an insulated gate bipolar transistor (IGBT), a metal oxide semiconductor (MOS) transistor, a bipolar transistor, or the like can be used. On / off of the switching element is controlled based on the PWM signals UU, UL, VU, VL, WU, WL output from the PWM signal generation unit 25 of the control unit 15, so that the three applied to the AC motor 2 are controlled. The driving of the AC motor 2 is controlled based on the phase AC voltages vu, vv, vw.

The current sensor 13 is provided in any one phase of the AC motor 2. In the present embodiment, the current sensor 13 is provided in the W phase. Hereinafter, the W phase in which the current sensor 13 is provided is referred to as a “sensor phase”. The current sensor 13 detects the W-phase current as the sensor-phase current detection value iw_sns and outputs the detected current to the control unit 15.
Hereinafter, the description of the present embodiment will be made on the assumption that the sensor phase is the W phase. However, in other embodiments, the U phase or the V phase may be the sensor phase.

The rotation angle sensor 14 is provided in the vicinity of a rotor (not shown) of the AC motor 2, detects the electrical angle θe, and outputs it to the control unit 15. Further, based on the electrical angle θe detected by the rotation angle sensor 14, the rotational speed N of the rotor of the AC motor 2 is calculated. Hereinafter, “the rotational speed N of the rotor of the AC motor 2” is simply referred to as “the rotational speed N of the AC motor 2”.
The rotation angle sensor 14 of the present embodiment is a resolver, but in other embodiments, other types of sensors such as a rotary encoder may be used.

  The control unit 15 is configured by a microcomputer or the like, and includes a CPU, a ROM, an I / O, a bus line that connects these configurations, and the like, all of which are not shown. The control unit 15 controls the operation of the AC motor 2 by software processing by executing a program stored in advance by the CPU or hardware processing by a dedicated electronic circuit.

The motor control device 10 sets the AC motor 2 as “motor” according to the rotational speed N of the AC motor 2 based on the electrical angle θe detected by the rotation angle sensor 14 and the torque command value trq ^* from the vehicle control circuit 9. The power is consumed by “power running operation” or generated by “regenerative operation as a generator”. Specifically, the operation is switched in the following four patterns depending on whether the rotation speed N and the torque command value trq ^* are positive or negative.
<1. Forward rotation power consumption> Power consumption when the rotational speed N is positive and the torque command value trq ^* is positive.
<2. Forward rotation regeneration> When the rotational speed N is positive and the torque command value trq ^* is negative, power is generated.
<3. Reverse running power> Power consumption when the rotational speed N is negative and the torque command value trq ^* is negative.
<4. Reverse regeneration> Electricity is generated when the rotational speed N is negative and the torque command value trq ^* is positive.

When the rotational speed N> 0 (forward rotation) and the torque command value trq ^* > 0, or when the rotational speed N <0 (reverse rotation) and the torque command value trq ^* <0, the inverter 12 With this switching operation, the AC motor 2 is driven so that the DC power supplied from the DC power supply 8 side is converted into AC power and torque is output (powering operation).
On the other hand, when the rotational speed N> 0 (forward rotation) and the torque command value trq ^* <0, or when the rotational speed N <0 (reverse rotation) and the torque command value trq ^* > 0, the inverter 12 By the switching operation of the switching element, the AC power generated by the AC motor 2 is converted into DC power and supplied to the DC power supply 8 side to perform a regenerative operation.

Next, the configuration of the control unit 15 mainly applied to the rectangular wave control mode of the torque feedback control method will be described with reference to FIGS.
As shown in FIG. 3, the control unit 15 includes a torque subtractor 52, a PI calculation unit 53, a rectangular wave generator 54, a signal generator 55, a current estimation unit 40 as a “current estimation unit”, and a torque estimation unit 56. Have.

The torque subtractor 52 calculates a torque deviation Δtrq that is a difference between the estimated torque value trq_est fed back from the torque estimation unit 56 and the torque command value trq ^* .
The PI calculation unit 53 performs PI calculation on the “voltage phase command value VΨ” that is the phase command value of the voltage vector so that the torque deviation Δtrq converges to 0 so that the estimated torque value trq_est follows the torque command value trq ^*. Calculated by

The rectangular wave generator 54 generates a rectangular wave based on the voltage phase command value VΨ and the electrical angle θe, and generates a U phase voltage command value vu ^* , a V phase voltage command value vv ^* , and a W phase voltage command value vw ^*. Is output.
Based on the U-phase voltage command value vu ^* , the V-phase voltage command value vv ^* , and the W-phase voltage command value vw ^* , the signal generator 55 is a voltage command signal UU related to switching on / off of the switching element of the inverter 12. , UL, VU, VL, WU, WL are generated and output to the inverter 12.
Three-phase AC voltages vu, vv, vw are generated by controlling on / off of the switching elements of the inverter 12 based on the voltage command signals UU, UL, VU, VL, WU, WL. The three-phase AC voltages vu, vv, vw are applied to the AC motor 2, and the drive of the AC motor 2 is controlled so that torque according to the torque command value trq ^* is output.

The current estimation unit 40 calculates the current estimation value iu_est of the estimation phase by a characteristic calculation described later based on the current detection value iw_sns of the sensor phase detected by the current sensor 13 and the electrical angle θe acquired from the rotation angle sensor 14. Further, based on the detected current value iw_sns of the sensor phase and the estimated current value iu_est of the estimated phase, the d-axis current estimated value id_est and the q-axis current estimated value iq_est are calculated.
Based on the d-axis current estimated value id_est and the q-axis current estimated value iq_est estimated by the current estimating unit 40, the torque estimating unit 56 calculates the torque estimated value trq_est using a map or a mathematical formula and feeds it back to the torque subtractor 52. .

Next, a detailed configuration of the current estimation unit 40 will be described with reference to FIG.
The current estimation unit 40 includes an α-axis current calculation unit 41, a β-axis current calculation unit 42, a sensor phase reference current phase calculation unit 43, an other-phase current estimation unit 44, and a dq conversion unit 45. The detected current value iw_sns of the sensor phase is input to the α-axis current calculation unit 41, the β-axis current calculation unit 42, the other-phase current estimation unit 44, and the dq conversion unit 45 in a multiplexed manner, and calculation is executed step by step in each unit. Is done.
In addition, since the control unit 15 of the present embodiment is applied to the torque feedback control type rectangular wave control mode, basically, the d-axis current command value id ^* and the q-axis current command value iq ^* are not used. .

  After calculating the α-axis current iα and the β-axis current iβ using the α-β coordinate system which is a fixed coordinate system, the α-axis current calculation unit 41, the β-axis current calculation unit 42, and the sensor phase reference current phase calculation unit 43 The sensor phase reference current phase θx is calculated. Regarding the symbol of the sensor phase reference current phase, “xθ” described in Patent Document 1 (Japanese Patent Laid-Open No. 2013-172594) and “θx” used in this specification are synonymous.

As shown in FIG. 5, the α axis coincides with the axis of the W phase that is the sensor phase, and the β axis is orthogonal to the α axis. The sensor phase reference current phase θx is an angle synchronized with the detected current value iw_sns of the sensor phase formed by the α axis and the current vector (Ia∠θx) of the current amplitude Ia. The sensor phase current detection value iw_sns is expressed by equation (1) using the sensor phase reference current phase θx and the current amplitude Ia.
iw_sns = Ia × sin (θx) (1)
Since the current detection value iw_sns of the sensor phase is a continuous value that changes with a phase change, for example, when expressed as a continuous waveform with respect to the phase, it is also referred to as “W-phase detection current iw_sns” depending on the context. W-phase detection current iw_sns has a sine waveform shown in FIG.

The sensor phase reference current phase θx is 0 [°] when the waveform of the W-phase current iw crosses from negative to positive in the power running state of positive rotation and positive torque, and the waveform of the W-phase current iw changes from positive to negative. The sensor phase reference current phase θx at the time of zero crossing is 180 [°].
In addition, the “β axis orthogonal to the α axis” includes an axis (“β axis” in FIG. 5) having a sensor phase reference current phase θx of 90 [°] as a positive direction, and a sensor phase reference current. There are two types of axes (the “β ′ axis” in FIG. 5) whose phase θx is the positive direction in the direction of 270 °.
In the following, the angle unit is displayed as [°] in the text and without [] in the formula.

また、式（３）のキルヒホッフの法則より、三相電流ｉｕ、ｉｖ、ｉｗの瞬時値の和は０となる。
ｉｕ＋ｉｖ＋ｉｗ＝０ ・・・（３） Here, the α-axis current iα and the β-axis current iβ used to calculate the sensor phase reference current phase θx will be described. The α-axis current iα is expressed as in Expression (2) using the phase currents iu, iv, iw and the conversion coefficient K _αβ from the three-phase axis to the αβ axis.

Further, the sum of instantaneous values of the three-phase currents iu, iv, iw is 0 based on Kirchhoff's law of the equation (3).
iu + iv + iw = 0 (3)

When Expression (2) is transformed using Expression (3), Expression (4) is obtained. Then, when the current detection value iw_sns of the sensor phase is used as the W-phase current iw of Expression (4), the α-axis current detection value iα_sns is expressed by Expression (4 ′). The α-axis current calculation unit 41 calculates the α-axis current iα by the equation (4 ′).

Here, the current detection timing in the rectangular wave control mode will be described.
As shown in FIG. 7, the voltage waveform of each phase in the rectangular wave control mode is a current cycle in which 0 [V] in the off state and the system voltage VH in the on state are switched every 180 [°]. The waveform of one pulse. The phases of the three-phase voltage waveforms are shifted from each other by 120 [°], and a switching element (not shown) of any phase of the inverter 12 is turned on / off every electrical angle of 60 [°]. The waveform is switched on / off.

The on / off timing of the switching element is referred to as “switch timing”. The difference in electrical angle θe between successive switch timings is 60 [°]. The timing set between successive switch timings is called “intermediate timing”. The difference in electrical angle θe between the intermediate timings is also 60 [°].
In FIG. 7, the electrical angle difference between successive switch timings is divided into two equal parts, and the intermediate timing is set to an electrical angle θe that is shifted by 30 [°] from the switch timing. However, the present invention is not limited to this example, and the intermediate timing may be set to an electrical angle θe other than 30 [°] from the switch timing, or may be set twice or more between successive switch timings. .

  The sensor phase current detection value iw_sns is detected at least one of the switch timing and the intermediate timing. Therefore, a current detection value with an electrical angle of 60 [°] is obtained. The α-axis current iα is calculated in synchronization with the detection timing of the sensor phase current detection value iw_sns.

On the other hand, the β-axis current calculation unit 42 calculates the β-axis current based on the integration value of the current detection value iw_sns of the sensor phase in the “section from the phase that is 180 ° or 540 ° backward from the arbitrary reference phase to the reference phase”. iβ is calculated. Here, the “arbitrary reference phase” may be a current phase at the time of calculation or a phase before the time of calculation. Hereinafter, this integration calculation will be described.
First, “current (Ia × sin φ) as a function of phase φ is a section from phase (θ−ξ) to phase θ (where ξ is a phase where 0 ° <ξ <720 ° and ξ ≠ 360 °) Is defined as “shift current Ishift”. Assuming that the amplitude Ia is invariant with respect to the phase θ, the shift current Ishift is expressed by Expression (5), where “arbitrary reference phase” is θ. Equation (5) is not limited to the sensor phase reference current phase θx with respect to the W phase, but is an equation for a general phase θ.

When formula (5) is transformed, formula (6) is obtained. Ks defined by Equation (7) represents an amplitude ratio.

According to Expression (6), it can be seen that the shift current Ishift has an amplitude Ks times that of the current (Ia × sin θ) and the phase is shifted by (ξ / 2).
When applied to the present embodiment in which the W phase is the sensor phase and “θ” is replaced with “sensor phase reference current phase θx” in equation (5), the current (Ia × sin θ) in equations (5) and (6) is , “W-phase detection current iw_sns = Ia × sin (θx)”.

Substituting ξ = 180 ° into equation (6) yields equation (8.1).
Ishift = 2Ia × sin (θx−90 °) (8.1)
Further, when ξ = 540 ° is substituted into equation (6), equation (8.2) is obtained.
Ishift = 2Ia × sin (θx−270 °)
= 2Ia × sin (θx + 90 °) (8.2)
When formulas (8.1) and (8.2) are put together, they are expressed as formula (8.3).
Ishift = 2Ia × sin (θx ± 90 °) = ± 2Ia × cos (θx)
... (8.3)

As shown in FIG. 6, the shift current Ishift corresponding to ξ = 180 ° or ξ = 540 ° is orthogonal to the W-phase detection current iw_sns and has a double amplitude. The direction of the shift current Ishift corresponds to the β axis in FIG. 5 when ξ = 180 °, and corresponds to the β ′ axis in FIG. 5 when ξ = 540 °.
The β-axis current iβ is calculated from the shift current Ishift by the equation (9) using the conversion coefficient K _αβ of the equations (2) and (4).

  As described above, the β-axis current calculation unit 42 calculates the β-axis current iβ by the integral calculation based on the detected current value iw_sns of the sensor phase. This integration operation corresponds to the detection timing of the sensor phase current detection value iw_sns, and is preferably executed in at least one of the phase intervals between the switch timings or the phase interval between the intermediate timings.

The sensor phase reference current phase calculation unit 43 calculates the sensor phase reference current phase θx by Equation (10) based on the α axis current iα and the β axis current. Note that precautions such as inversion of the sign in equation (10) are in accordance with Patent Document 1.

The other-phase current estimation unit 44 is based on the sensor phase reference current phase θx calculated by the sensor phase reference current phase calculation unit 43 and the current detection value iw_sns of the sensor phase. iu_est is calculated.
As shown in FIG. 6, the U-phase current estimated value iu_est is expressed by Expression (11) in which the phase is shifted by 120 [°] with respect to the W-phase current detection value iw_sns of Expression (1).
iu_est = Ia × sin (θx−120 °) (11)

  When Expression (11) is transformed, Expression (12) using the sensor phase reference current phase θx and the sensor phase current detection value iw_sns is obtained.

The U-phase current estimated value iu_est may be calculated by Expression (14) using the estimation coefficient iu_kp defined by Expression (13). Here, the estimation coefficient iu_kp may be directly calculated from the equation (13), or a part or the whole of the equation (13) is mapped in advance based on the sensor phase reference current phase θx, and this map is referred to. You may calculate by doing.
Regarding the symbol of the estimation coefficient, “fu (xθ)” described in Patent Document 1 and “iu_kp” used in this specification are synonymous.

The dq conversion unit 45 performs dq conversion on the U-phase current estimation value iu_est and the sensor phase (W-phase) current detection value iw_sns using Equation (15), and calculates the d-axis current estimation value id_est and the q-axis current estimation value iq_est. To do.

Next, a current estimation processing routine executed by the current estimation unit 40 will be described with reference to the flowchart of FIG. In the following description of the flowchart, the symbol S indicates “step”.
This current estimation routine is repeatedly executed at a predetermined calculation period during the power-on period of the control unit 15. When this routine is started, in the first S10, the current detection value iw_sns of the sensor phase detected by the current sensor 13 is acquired, and the electrical angle θe of the AC motor 2 detected by the rotation angle sensor 14 is acquired.

S20 and S30 are executed in parallel.
In S20, the α-axis current iα is calculated based on the detected current value iw_sns of the sensor phase according to the equation (4 ′). In S30, the current detection value iw_sns (= Ia × sin (θx)) of the sensor phase is integrated with respect to a predetermined phase interval according to the equation (5) to calculate the β-axis current iβ.

In S40, the sensor phase reference current phase θx is calculated by the equation (10) using the α-axis current iα and the β-axis current iβ.
In S50, an estimated current value iu_est for the U phase is calculated from the detected current value iw_sns for the sensor phase and the sensor phase reference current phase θx according to the equation (12) or the equation (14) using the estimation coefficient iu_kp. This step includes determination as to whether or not the sensor phase current is at zero crossing, as disclosed in Patent Document 1 in detail.

In S60, the U-phase current estimation value iu_est and the sensor phase (W-phase) current detection value iw_sns are dq-transformed by the equation (15) to calculate the d-axis current estimation value id_est and the q-axis current estimation value iq_est. .
Above, the current estimation routine which the current estimation part 40 performs is complete | finished.

(effect)
(1) The motor control apparatus 10 of this embodiment detects the phase current of one phase among the three phases by the current sensor 13, and estimates the other two-phase phase currents. By providing the current sensor 13 only in the sensor phase, the number of current sensors 13 can be reduced. Thereby, the three-phase output terminal vicinity of the inverter 12 can be reduced in size, and the cost of the motor control apparatus 10 can be reduced.
Further, by making the number of the current sensors 13 one, the influence of the gain error of the current sensor which can be generated in the conventional AC motor control system using a plurality of current sensors is eliminated. Thereby, in the AC motor 2, the output torque fluctuation caused by the gain error of the plurality of current sensors can be eliminated. Can be removed.

(2) The current estimation unit 40 calculates the sensor phase reference current phase θx based on the α-axis current iα and the β-axis current iβ in the α-β coordinate system based on the sensor phase. The actual current phase θx can be calculated.
Further, since the estimated current value iu_est of the estimated phase is calculated based on the sensor phase reference current phase θx and the detected current value iw_sns of the sensor phase, the influence of the harmonic component of the actual current phase θx and the fluctuation that can normally occur is incorporated. Thus, the estimated current value iu_est of the estimated phase can be calculated with high accuracy.

(3) The current estimation unit 40 calculates the α-axis current iα based on the current detection value iw_sns of the sensor phase, and from the phase (θx−180 °) to the phase θx of the current detection value iw_sns of the sensor phase. Alternatively, the β-axis current iβ is calculated based on the shift current Ishift calculated as an integral value in the section from the phase (θx−540 °) to the phase θx.
Thereby, in the rectangular wave control mode of the torque feedback control system that does not use the current command value for the energization control of the AC motor 2, the β-axis current iβ can be calculated without using the current command value. In addition, the method of differentiating the α-axis current iα disclosed in Patent Document 1 is easily affected by a sudden change in the current detection value due to noise or the like. In the present embodiment, the accumulated result of past current detection values is obtained. Since it is reflected, the influence of sudden change can be reduced.

  (4) The current estimation unit 40 calculates the α-axis current iα based on the detected current value iw_sns of the sensor phase at least one of the switch timing and the intermediate timing, and the phase interval between the switch timings or the phase between the intermediate timings The β-axis current iβ is calculated based on the integral value (that is, the shift current Ishift) of the current detection value iw_sns of the sensor phase in at least one of the sections.

  While the waveform of the current detection value detected at each switching timing is affected by the switching operation, the waveform is distorted, while the waveform of the current detection value detected at each intermediate timing is not affected by the switching operation. Is hardly distorted. Therefore, the current waveform consisting of both the current detection value at each switch timing and the current detection value at each intermediate timing does not increase or decrease regularly like a sine wave, but tends to increase or decrease irregularly.

  On the other hand, both of the waveform of the current detection value detected at each switching timing and the waveform of the current detection value detected at each intermediate timing increase or decrease almost regularly in each waveform. Therefore, if the integral calculation of the shift current Ishift is executed in the phase interval between the switch timings or in the phase interval between the intermediate timings, the influence of the current detection value that irregularly increases and decreases between the switch timing and the intermediate timing is almost eliminated. Without being received, the β-axis current iβ can be accurately calculated. As a result, the calculation accuracy of the sensor phase reference current phase θx can be improved.

(Other embodiments)
(A) The sensor phase for detecting the phase current by the current sensor is not limited to the W phase in the above embodiment, and may be the U phase or the V phase. In addition, the estimation phase for calculating the current estimation value from the current detection value of the sensor phase and the sensor phase reference current phase θx is not limited to the U phase of the above embodiment, and may be the V phase or the W phase.

  (A) The AC motor control device according to the present invention is effective when applied mainly to a rectangular wave control mode of a torque feedback control system that does not use a current command value for energization control. However, the β-axis current iβ may be calculated by the integral operation of the present invention even in the current feedback control type sine wave PWM control mode or overmodulation PWM control mode in which the current command value is used for energization control.

  (C) The AC motor of the above embodiment is a permanent magnet type synchronous three-phase AC motor. However, in other embodiments, an induction motor or other synchronous motor may be used. In addition, the AC motor of the above embodiment is a so-called motor generator having both a function as a motor and a function as a generator. However, in other embodiments, the AC motor may not have a function as a generator. Furthermore, the present invention is not limited to a three-phase AC motor, and can be widely applied to a multi-phase AC motor having three or more phases.

  (D) The control device for an AC motor according to the present invention is not limited to a system in which only one set of an inverter and an AC motor is provided as in the above embodiment, but may be applied to a system in which two or more sets of inverters and AC motors are provided. Good. Further, the present invention may be applied to a system such as a train in which a plurality of AC motors are connected in parallel to one inverter.

(E) The control device for an AC motor according to the present invention is not limited to the AC motor of the hybrid vehicle having the configuration shown in FIG. 1, and may be applied to an AC motor of an electric vehicle having any configuration. Moreover, you may apply to AC motors other than an electric vehicle.
As mentioned above, this invention is not limited to the said embodiment at all, In the range which does not deviate from the meaning of invention, it can implement with a various form.

2 ... AC motor,
10: Electric motor control device (AC motor control device),
12 ... Inverter,
13 ... Current sensor,
15 ... control unit (control means),
40: Current estimation unit (current estimation means).

Claims (2)

An inverter (12) for driving a three-phase or more multi-phase AC motor (2);
A current sensor (13) for detecting a current flowing in a single sensor phase among the multiphases of the AC motor;
Control means (15) for controlling energization of the AC motor by switching on / off of a plurality of switching elements constituting the inverter;
With
The control means has a current phase based on the sensor phase based on an α-axis current and a β-axis current in a fixed coordinate system including an α axis that coincides with the axis of the sensor phase and a β axis that is orthogonal to the α axis. Current estimation for calculating a current estimation value of a phase other than the sensor phase based on the sensor phase reference current phase (θx) and a current detection value (iw_sns) of the sensor phase Means (40),
The current estimation means calculates the α-axis current based on the detected current value of the sensor phase, and goes back 180 [°] or 540 [°] from an arbitrary reference phase of the detected current value of the sensor phase. The control apparatus (10) for an AC motor, wherein the β-axis current is calculated based on an integral value in a section from the phase to the reference phase. The current estimation means includes
The α-axis current is calculated based on the detected current value of the sensor phase at at least one of a switch timing for switching on / off of the switching element or an intermediate timing set at least once between successive switch timings. And
2. The β-axis current is calculated based on an integrated value of current detection values of the sensor phase in at least one of a phase interval between the switch timings or a phase interval between the intermediate timings. The control apparatus of the alternating current motor described in 1.

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