JP2015047004A - Step-up converter - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a step-up converter which does not generate a surge voltage, and allows for low noise and high frequency.SOLUTION: A first switching element and a first reactor are connected in series with a DC power supply, a series connection of a second reactor, a first capacitor and a second capacitor is connected in parallel with the first reactor, a second switching element is connected in parallel with the second capacitor, a third switching element and a third capacitor are connected in parallel with the second switching element, and an output is taken out from the third capacitor by performing on/off control of the first switching element, second switching element and third switching element on the basis of the opposite ends of the third capacitor.

Description

  The present invention relates to a boost converter that boosts an input DC voltage.

  Conventionally, a boost converter that converts a low DC voltage to a high DC voltage is known. In particular, non-insulated boost converters are widely used because of their simple circuit configuration. A conventional non-insulated boost converter is shown in FIG. In a conventional boost converter, an inductor L and a switching element Q1 are connected in series to a DC power source Vi, and a series circuit of a diode D and a smoothing capacitor Co is connected in parallel to both ends of the switching element Q1, so that the switching element Q1 is turned on / off. A control circuit Cont3 for controlling the turning-off is provided. A snubber circuit composed of a series circuit of a resistor R and a capacitor C is connected in parallel to both ends of the switching element. An output voltage Vo is obtained from both ends of the smoothing capacitor Co.

A circuit diagram of the control circuit Cont3 is shown in FIG. The control circuit Cont3 includes a triangular wave oscillator (triangular wave OSC) serving as a reference frequency, a PWM comparator (PWMCOMP), an operational amplifier (OPAMP) as an error amplifier, a reference voltage (ES), a resistor (Rs), a resistor (Rf), and a capacitor (Cf ), And a buffer circuit (BUFF1).
The control circuit Cont3 compares the output voltage Vo detected at the FB terminal and the reference voltage ES by the operational amplifier OPAMP. The output of the operational amplifier OPAMP is sent to the PWM comparator PWMCOMP as an error signal. The PWM comparator PWMCOMP generates a PWM signal by comparing the error signal and the triangular wave signal. The switching element Q1 is on / off controlled by this PWM signal.

  However, a boost converter generally has a leakage (floating) inductance not shown in FIG. For example, it exists in series with the switching element Q1, the diode D, and the like. For this reason, a surge voltage is generated by a steep current change of the switching element Q1 and the diode D. A large voltage is applied to the semiconductor element by the surge voltage, and the element may be destroyed. In order to prevent this, a snubber circuit composed of R and C is inserted into the switching element Q1 in the example of FIG. The surge voltage is absorbed as heat by the snubber circuit composed of R and C, and the surge voltage is reduced. As described above, since the conventional boost converter shown in FIG. 8 is so-called hard switching, a surge voltage is generated due to the leakage inductance. Therefore, the energy stored in the leakage inductance is converted into heat by some method to reduce the surge voltage. The occurrence was prevented. For this reason, the conversion efficiency was reduced. In addition, sharp switching has caused high frequency noise. Furthermore, since the generation of surge voltage is proportional to the switching frequency, it has also hindered high frequency operation.

  In order to improve the above problems, a chopper reactor connected in series to a DC power source, a first switching element connected in series between the chopper reactor and ground, and a series between the chopper reactor and the output point are connected. A rectifier connected to the rectifier, a smoothing capacitor charged by receiving the output of the rectifier, a resonance reactor connected in series with the first switching element, and one end point of the chopper reactor connected in series. A resonance capacitor, a second switching element connected in series between the resonance capacitor and the ground, a connection point between the resonance reactor and the first switching element, an output point, and a resonance capacitor And a second switching element are connected between the connection point and the output point, respectively, and a back electromotive voltage generated by the resonance reactor is a predetermined voltage. Boost converter that includes a bypass means for suppressing has been proposed (Patent Document 1). In the boost converter of Patent Document 1, the back electromotive force generated by the resonance reactor when the first and second switching elements are turned off is output by the bypass means provided between the input terminal and the output point of each switching element. By clamping with voltage, it is possible to suppress the vibration of the voltage waveform applied to the switching element to prevent the generation of noise, improve the conversion efficiency, and further prevent the switching element from being damaged.

JP 2006-042443 A

  However, in Patent Document 1, since the resonance current flows by controlling on / off of the second switching element every time the first switching element is turned on / off, the control becomes complicated. In addition, there is a problem that a bypass means for suppressing the back electromotive voltage of the resonance reactor to a predetermined voltage is required. The present invention seeks to provide a boost converter that does not generate surge voltage, and that can achieve low noise and high frequency.

  In the present invention, a first switching element and a first reactor are connected in series to a DC power source, and a second reactor, a first capacitor, and a second capacitor are connected in series to the first reactor. A second switching element connected in parallel to the second capacitor; a third switching element and a third capacitor connected in series to the second switching element; and This is a boost converter characterized by taking output from both ends of a capacitor.

Furthermore, the first switching element and the second switching element are turned on / off in synchronization, and the second switching element and the third switching element have a dead time period in which they are turned off simultaneously. The inverting boost converter is characterized by being exclusively turned on / off.

In addition, the first switching element and the third switching element are turned on and off in synchronization, and the second switching element and the third switching element have a dead time period in which they are turned off simultaneously. Alternatively, it may be turned on and off exclusively. Operates as a boost converter.

  According to the present invention, the boost converter can be reduced in noise and frequency without generating a surge voltage.

It is a circuit diagram of Example 1 of the present invention. It is a block diagram of the control circuit used for FIG. 3 is a timing chart of the control circuit of FIG. It is an operation | movement waveform of each part of Example 1 shown in FIG. It is a circuit diagram which shows Example 2 of this invention. FIG. 6 is a block diagram of a control circuit used in FIG. 5. It is an operation | movement waveform of each part of Example 2 shown in FIG. It is a circuit diagram of the conventional boost converter. FIG. 9 is a block diagram of a control circuit used in FIG. 8.

  FIG. 1 is a circuit diagram of Embodiment 1 of the present invention. The first switching element Q1 and the first reactor Ll are connected in series to the DC power source Vi, and further in parallel with the first reactor Ll, the second reactor Ls, the first capacitor Cri, and the second capacitor Crv. Are connected directly, and the second switching element Q2 is connected in parallel to the second capacitor Crv, and the third switching element Q3 and the third capacitor Co are connected in series in parallel to the second switching element Q2. Connect to. A parasitic diode exists in parallel with each of the first switching element Q1, the second switching element Q2, and the third switching element Q3.

The boost converter control circuit Cont1 detects the output voltage Vo, and based on the detected output voltage Vo, simultaneously turns on and off the first switching element Q1 and the third switching element Q3, The first switching element Q1 and the third switching element Q3 are alternately turned on / off with a dead time. The output is taken from both ends of the third capacitor C _O.

  FIG. 2 shows the boost converter control circuit Cont1 according to the first embodiment of the present invention. The output voltage Vo is input to the feedback terminal FB. An inverting amplifier circuit composed of an operational amplifier OPAMP, resistors Rs and Rf, and a capacitor Cf compares the output voltage input to the feedback terminal FB with the reference voltage Es, and the voltage of the difference between the output voltage Vo and the reference voltage Es is output from the output. Output as an error signal. The output of the operational amplifier OPAMP is input to an oscillator VCO which is a voltage controlled variable frequency oscillator, and the output of the oscillator VCO outputs a pulse signal VCO output whose frequency changes based on an error signal. The pulse signal VCO output is input to the one-shot multivibrator circuit ONE-SHOT. The one-shot multivibrator circuit ONE-SHOT outputs a pulse ONE-SHOT output which is a signal having a predetermined pulse width when the pulse signal VCO output is input. The pulse ONE-SHOT output is input to the second dead time generation circuit DT2, and the dead time required by the first switching element Q1 and the third switching element Q3 is given from the second dead time generation circuit DT2. The signal DT2 output is output. The output of the signal DT2 is converted into a drive signal Q3D of the third switching element Q3 via the level shift circuit LEVELSHIFT1 and the buffer circuit BUFF2, and drives the third switching element Q3. To the drive signal Q1D of the first switching element Q1 to drive the first switching element Q1. Further, the pulse ONE-SHOT output is connected to the input of the first dead time generation circuit DT1 through the inverter INV. The first dead time generation circuit DT1 outputs a signal DT1 output to which a dead time required by the second switching element Q2 is given. The output of the signal DT1 is converted to the drive signal Q2D of the second switching element Q2 via the buffer circuit BUFF1, and drives the second switching element Q2.

  The operation waveforms of the boost converter control circuit Cont1 are shown in FIG. According to FIG. 3, the pulse signal VCO output is output at time t0, the one-shot multivibrator circuit ONE-SHOT is activated, and the pulse ONE-SHOT output is generated. When a predetermined time elapses and time t1 is reached, the signal DT2 output is output from the second dead time generation circuit DT2. The signal DT2 output generates a signal Q1D via the level shift circuit LEVELSHIFT2 and the buffer circuit BUFF3, and turns on the first switching element Q1. Further, Q3D is generated via the level shift circuit LEVELSHIFT1 and the buffer circuit BUFF2, and the third switching element Q3 is turned on. At time t2, the pulse ONE-SHOT output disappears, and the first switching element Q1 and the third switching element Q3 are turned off. At the same time, a signal INV output is output from the inverter INV, and a signal DT1 output is output at time t3 after a predetermined time elapses in the first dead time generation circuit DT1. The signal DT1 output generates a signal Q2D output via the buffer circuit BUFF1 and turns on the second switching element Q2. Next, when the VCO signal is output, the one-shot multivibrator circuit ONE-SHOT is activated and the pulse ONE-SHOT output is generated, so that the signal DT1 disappears and the second switching element Q2 is turned off. The above operation is repeated, and the switching elements Q1, Q2, and Q3 are repeatedly turned on and off. For this reason, the first switching element Q1 and the third switching element Q3 are turned on only by subtracting the dead time generated by the second dead time generation circuit DT2 from the pulse ONE-SHOT output, and the second switching element Q2 is turned on. The first switching element Q1 and the third switching element Q3 are turned on simultaneously by subtracting the pulse ONE-SHOT output and the predetermined dead time generated by the first dead time generation circuit DT1 from the period of the pulse signal VCO output. The second switching element Q2 is turned on, and the first switching element Q1 and the third switching element Q3 are alternately turned on.

  Here, when the output voltage Vo increases, the error signal output from the operational amplifier OPAMP decreases. Since the output of the oscillator VCO generates a pulse signal having a longer period when the error signal becomes smaller based on the error signal, the signal VCO output has a longer period. In FIG. 3, the cycle of the VCO output was from time t0 to t4 at time t0, whereas at time t4, the error signal is smaller than at time t0, so the VCO output has a longer cycle from time t4 to t8. At this time, the ON period of the first switching element Q1 and the third switching element Q3 is the period t1 to t2 or the period t5 to t6, and there is no change, but the ON period of the second switching element Q2 is the period from the period t3 to t4 From t7 to t8, the signal VCO output is turned on for a longer period.

  The operation of voltage conversion will be described with reference to FIG. When the first switching element Q1 and the third switching element Q3 are turned on and the second switching element Q2 is turned off, the exciting current of the first reactor Ll flows through the first path of Vi-Q1-L1-Vi. At the same time, a resonance current having a resonance frequency determined by the second reactor Ls, the first capacitor Cri, and the second capacitor Crv flows through the path of Vi-Q1-Ls-Cri-Crv-Vi, and the second capacitor Crv is substantially omitted. It is charged until it reaches Vo. When the second capacitor Crv becomes approximately Vo, a current flows through the second path of Vi-Q1-Ls-Cri-Q3-Co-Vi and charges the third capacitor Co with the output voltage Vo. At this time, the resonance frequency of the second reactor Ls, the first capacitor Cri, the second capacitor Crv, and the third capacitor Co has the highest resonance frequency of the second reactor Ls and the first capacitor Cri. Become. The output pulse ONE-SHOT output of the one-shot multivibrator circuit ONE-SHOT determines the ON period of the first switching element Q1 and the third switching element Q3, and is approximately ½ of the period of the resonance frequency of Ls and Cri. Set.

  Next, when the first switching element Q1 and the third switching element Q3 are turned off, a current flows through the path of L1-Crv-Cri-Ls, and the first reactor Ll, the second reactor Ls, and the first capacitor. When the voltage of the second capacitor Crv charged to approximately Vo voltage gradually decreases and reaches zero volts by the resonance circuit of Cri and the second capacitor Crv, the current flowing through the third switching element Q3 is changed to the first voltage. 2 commutates to the parasitic diode of the switching element Q2. At this time, a voltage quasi-resonant waveform appears at both ends of the second switching element Q2 and the third switching element Q3. When the drive signal is supplied to the second switching element Q2 and turned on during this commutation period, the second switching element Q2 performs zero volt switching. When the second switching element Q2 is turned on, a current flows through the path of L1-Q2-Cri-Ls-Ll. Since the relationship of L1 >> Ls is established, the resonance frequency at this time is dominated by the resonance frequency of L1 and Cri, which is a period much larger than the switching period, and the resonance current is a sine whose frequency is much lower than the switching frequency. A part of the wave is observed linearly as indicated by IQ2 in FIG. While the second switching element Q2 is on, the resonance current is inverted and flows through the path of L1-Ls-Cri-Q2-Ll.

  Next, when the second switching element Q2 is turned off, the current of the second switching element Q2 is commutated to the second capacitor Crv, and the current flows through the path of L1-Ls-Cri-Crv. When the voltage of the second capacitor Crv is gradually charged by the resonance circuit of Ll, the second reactor Ls, the first capacitor Cri, and the second capacitor Crv and is charged to approximately Vo voltage, The current flowing in the capacitor Crv is commutated through a parasitic diode connected in parallel to the third switching element Q3. At this time, a voltage quasi-resonant waveform appears at both ends of the second switching element Q2 and the third switching element Q3. When a drive signal is supplied to the third switching element Q3 during this commutation period to turn on the third switching element Q3, the third switching element Q3 is switched to zero volts. At the same time, the first switching element Q1 is also turned on.

The on-time of the first switching element Q1 and the third switching element Q3 is set to approximately ½ of the period of the resonance frequency of Ls and Cri. The on-time ratio Don (Q2) of Q2 is approximately the following expression when dead time << switching period T, and is determined by the input / output voltage.
Don (Q2) = Vi / Vo Equation (1)
Similarly, the on-time ratio of Q1 and Q3 is
Don (Q1, Q3) = 1-Vi / Vo ... Formula (2)
It can be expressed. That is, in the output voltage control, the output voltage Vo is detected at the FB terminal of the boost converter control circuit Cont1, compared with the reference voltage Es, and the voltage control type variable frequency oscillator VCO is controlled by the error signal. Since the ON time of the first switching element Q1 and the third switching element Q3 is controlled to be constant by the one-shot multivibrator circuit ONE-SHOT, the first switching element Q1 and the third switching element Q3 are turned off. The time is varied by the VCO.

  FIG. 4 shows waveforms at various parts of the boost converter according to the first embodiment. In FIG. 4, the voltage VQ1 and current IQ1 of the first switching element Q1, the voltage VQ2 and current IQ2 of the second switching element Q2, the voltage VQ3 and current IQ3 of the third switching element Q3, the first The change of current ILl of reactor Ll and current ILs of second reactor Ls is shown. As can be seen from the figure, the on-time ratio of each switching element varies depending on the input / output voltage ratio according to equations (1) and (2). For load fluctuations, only the resonance current changes and the on-time ratio does not change. In practice, the on-time ratio varies to compensate for stray impedance such as line impedance.

  As described above, the second switching element Q2 and the third switching element Q3 operate by voltage pseudo resonance and current resonance, and perform zero volt switching and soft switching. Further, although the first switching element Q1 is hard switching, it is turned off after the current at the time of turning off is reduced, so that a surge voltage is hardly generated. That is, surge voltage countermeasures can be achieved with a slight snubber circuit. FIG. 4 shows a waveform when a snubber (not shown) of R and C is connected. It is settled with a slight surge voltage. In the boost converter according to the present invention, the on-time ratio of each switching element does not change due to load fluctuations, and therefore the response speed of the control circuit is not related to the transient response speed. Therefore, there is theoretically no response delay.

  FIG. 5 shows a second embodiment of the present invention, which operates as an inverting boost converter. The polarity of the direct current power source Vi is reversely connected to the first embodiment shown in FIG. The first switching element Q1, the second switching element Q2, and the third switching element Q3 are on / off controlled by the boost converter control circuit Cont2, respectively, and output is taken out from both ends of the capacitor Co.

The operation of the second embodiment shown in FIG. 5 is as follows. The first embodiment of FIG. 1 simultaneously turns on and off the first switching element Q1 and the third switching element Q3, and sets the second switching element Q2, the first switching element Q1, and the third switching element Q3 to dead. In the second embodiment shown in FIG. 5, the first switching element Q1 and the second switching element Q2 are simultaneously turned on / off, and the second switching element Q2 is turned on and off alternately. The third switching element Q3 is alternately turned on / off with a dead time.
In order to perform the control as described above, the boost converter control circuit Cont2 is configured as shown in FIG. That is, the boost converter control circuit Cont1 in FIG. 2 inverts the pulse ONE-SHOT output from the one-shot multivibrator circuit ONE-SHOT by the inverter INV in order to generate the drive signal Q2D of the second switching element Q2. However, in the step-up converter control circuit Cont2 of the second embodiment shown in FIG. 6, the pulse ONE-SHOT output from the one-shot multivibrator circuit ONE-SHOT is generated to generate the drive signal Q3D of the third switching element Q3. The output is inverted by the inverter INV. Note that the first switching element Q1 and the second switching element Q2 are not inverted and are driven as they are with the pulse ONE-SHOT output. However, the first switching element Q1 changes the signal level via the level shift circuit LEVELSHIFT2 similarly to the boost converter control circuit Cont1.

  The voltage conversion operation of the second embodiment will be described with reference to FIG. Q1 and Q2 are turned on simultaneously. A current flows through the first path of Vi-Ll-Q1-Vi and the second path of Vi-Q2-Cri-Ls-Q1-Vi. That is, in the first path, the exciting current of the first reactor Ll flows. In the second path, a resonance current having a resonance frequency flows through the second reactor Ls and the first capacitor Cri. The ON period output from ONE-SHOT is set to approximately ½ of the period of this resonance frequency.

  When the first switching element Q1 and the second switching element Q2 are turned off, a current flows through the path L1-Ls-Cri-Crv, and the first reactor Ll, the second reactor Ls, and the first capacitor Cri When the voltage of the second capacitor Crv discharged to zero volt gradually rises and reaches substantially the output voltage Vo by the resonance circuit of the second capacitor Crv, the current flowing through the second capacitor Crv becomes the third value. The commutation is performed via a parasitic diode connected in parallel to the switching element Q3. At this time, a voltage quasi-resonant waveform appears at both ends of the second switching element Q2 and the third switching element Q3. When the third switching element Q3 is turned on during the commutation period, the third switching element Q3 is switched to zero volts. When the third switching element Q3 is turned on, a current flows through the path of L1-Ls-Cri-Q3-Co-Ll. Since there is a relationship of Ll >> Ls, Co >> Cri, the resonance frequency at this time is dominated by the resonance frequency of the first reactor Ll and the first capacitor Cri, and becomes a period much larger than the switching period. As the resonance current, a part of a sine wave having a frequency much lower than the switching frequency is linearly observed.

  When the third switching element Q3 is turned off, the current of the third switching element Q3 is commutated to the second capacitor Crv, the current flows through the path of Crv-Cri-Ls-Ll-Crv, and the first reactor Ll. When the voltage of the second capacitor Crv is gradually discharged by the resonance circuit of the first and second reactors Ls, the first capacitor Cri, and the second capacitor Crv to zero volts, the second capacitor Crv is The flowing current is commutated through a parasitic diode connected in parallel to the second switching element Q2. At this time, a voltage quasi-resonant waveform appears at both ends of the second switching element Q2 and the third switching element Q3. During the commutation period, a drive signal is supplied to the second switching element Q2 to turn it on. At the same time, the first switching element Q1 is also turned on.

The on-time of the first switching element Q1 and the second switching element Q2 is set to approximately ½ of the period of the resonance frequency of Ls and Cri. The on-time ratio Don (Q3) of Q3 is determined by the input / output voltage as shown in the following equation, assuming that dead time << switching period T.
Don (Q3) = Vi / Vo Equation (3)
Similarly, the on-time ratio of Q1 and Q2 is
Don (Q1, Q2) = 1-Vi / Vo ... Formula (4)
It can be expressed.

    As described above, the second embodiment operates as an inverting boost converter. FIG. 7 shows waveforms at various parts of the boost converter according to the second embodiment. FIG. 7 shows that the voltage VQ1 and current IQ1 of the first switching element Q1, the voltage VQ2 and current IQ2 of the second switching element Q2, the voltage VQ3 and current IQ3 of the third switching element Q3, the first The change of current ILl of reactor Ll and current ILs of second reactor Ls is shown. As shown in FIG. 7, the second embodiment also operates by voltage pseudo resonance current resonance, and operates by zero volt switching and soft switching, as in the first embodiment. As in the first embodiment, the first switching element Q1 is hard-switched, but it is turned off after the off-state current is reduced, so that a surge voltage is hardly generated. Since the on-time ratio does not change due to load fluctuation, the response speed of the control circuit is not related to the transient response speed. Therefore, there is theoretically no response delay.

  The present invention can be applied to an apparatus and a power supply apparatus that require a boosted power supply.

Q1, Q2, Q3 Switching element L1, Ls Reactor Cri, Crv Capacitor Co Capacitor Cont1, Cont2, Cont3 Control circuit

Claims (4)

The first switching element and the first reactor are connected in series to the DC power source,
In parallel with the first reactor, a second reactor, a first capacitor, and a second capacitor are connected in series;
A second switching element is connected in parallel to the second capacitor;
In parallel with the second switching element, a third switching element and a third capacitor are connected in series;
A step-up converter characterized in that outputs are taken from both ends of the third capacitor. The first switching element and the second switching element are turned on and off in synchronization,
2. The boost converter according to claim 1, wherein the second switching element and the third switching element are exclusively turned on / off with a dead time period in which they are simultaneously turned off. The first switching element and the third switching element are turned on and off in synchronization,
2. The boost converter according to claim 1, wherein the second switching element and the third switching element are exclusively turned on / off with a dead time period in which they are simultaneously turned off. 4. The boost converter according to claim 1, wherein an ON period of the first switching element is approximately ½ of a resonance period of the second reactor and the first capacitor. 5.

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