JP2015027018A - Impedance adjustment element and high frequency module - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide an impedance adjustment element having a wider frequency range capable of impedance matching as compared to the prior configuration.SOLUTION: An impedance adjustment element 10 includes a rectangular waveguide 100, metal meshes 21-28, and ferrite 31-38. The metal meshes 21-28 and the ferrite 31-38 are mutually disposed with gaps along the high frequency signal transmission direction of the rectangular waveguide 100. In other words, the impedance adjustment element 10 includes a structure in which each unit structure including the metal meshes and the ferrite juxtaposed to the high frequency signal transmission direction is periodically disposed along the high frequency signal transmission direction.

Description

  The present invention relates to an impedance adjustment element that adjusts the impedance of a high-frequency transmission line.

  In wireless communication using the megahertz band or gigahertz band, transmission impedance of a high-frequency signal occurs unless the impedance of the high-frequency signal used in the wireless communication is matched between two connected circuit elements.

  For this reason, an impedance matching circuit is connected to the input end and output end of the circuit element. For example, in Patent Document 1, an input matching circuit is connected to an input terminal of a power amplifier, and an additional matching circuit is connected to an output terminal.

  Such a normal matching circuit includes an inductor, a capacitor, and the like, and is realized by a mounting circuit component mounted on a substrate and a conductor pattern formed on the substrate.

  On the other hand, currently, in wireless communication devices represented by mobile phones, the number of communication bands (BAND) used by one wireless communication device is increasing. That is, the frequency band transmitted and received by one wireless communication device is widened. Therefore, the antenna and the power amplifier are also required to have a wide band, and various broadband antennas and broadband power amplifiers have been put into practical use.

JP 2005-328194 A

  However, the matching circuit having the conventional configuration described above has a limit in the frequency range in which impedance matching is possible. This is due to the parasitic components (parasitic capacitance) generated by realizing the matching circuit with the mounted circuit components mounted on the substrate and the conductor pattern formed on the substrate, and the parasitic components generated in the mounted circuit components and mounted circuit patterns to be matched. This is due to the component (parasitic capacitance).

  Accordingly, an object of the present invention is to provide an impedance adjusting element that enables impedance matching in a wider band than in the conventional configuration.

  The present invention is an impedance adjusting element that is connected between a first port and a second port and adjusts impedance for a high-frequency signal transmitted between the first port and the second port. It is characterized by having a structure in which dθ / dω, which is a frequency characteristic of the change amount dθ of the phase θ, is a positive value.

  In this configuration, since the negative impedance characteristic can be realized by the impedance adjustment element, the parasitic component of the impedance of the circuit element connected to the impedance adjustment element can be canceled.

  The impedance adjusting element of the present invention may include a negative dielectric part having a negative dielectric constant with respect to a high-frequency signal and a negative magnetic part having a negative permeability with respect to the high-frequency signal. preferable. This configuration shows the conditions of the magnetic permeability and dielectric constant of the impedance adjusting element for dθ / dω to be a negative value.

  Moreover, it is preferable that the impedance adjusting element of this invention is the following structures. The negative dielectric portion is composed of a plurality of metal meshes arranged at intervals along the transmission direction of the high-frequency signal in the rectangular waveguide. The negative magnetic part is made of a plurality of ferrites arranged at intervals along the transmission direction of the high-frequency signal in the rectangular waveguide. In this configuration, a structural example for realizing a negative dielectric constant and a negative magnetic permeability is shown.

  In the impedance adjusting element of the present invention, it is preferable that a plurality of metal meshes and a plurality of ferrites are alternately arranged along the transmission direction of the high-frequency signal. This configuration shows a more specific structural example for realizing a negative dielectric constant and a negative magnetic permeability.

  According to another aspect of the present invention, there is provided a high-frequency module according to any one of the above-described impedance adjusting elements, a first high-frequency circuit unit connected to the first port, and a second high-frequency circuit unit connected to the second port. It is characterized by providing.

  In this configuration, by using the above-described impedance matching element, impedance matching between the first high-frequency circuit unit and the second high-frequency circuit unit can be realized in a wide frequency band.

  The high-frequency module according to the present invention includes any one of the impedance adjusting elements described above, and a variable trap filter connected between a transmission conductor connecting the first port and the second port and the ground. The impedance adjusting element is connected between the transmission conductor and the ground.

  In this configuration, a trap filter having a wide variable range of the attenuation pole frequency can be realized by using the above-described impedance matching element.

  According to the present invention, it is possible to realize an impedance that could not be realized conventionally. Thereby, impedance matching can be performed in a wider band than in the conventional configuration.

It is a graph which shows the frequency characteristic of phase (theta) of a transmission line. It is a conceptual diagram which shows the change of the passage characteristic (S21) by the aspect of impedance matching. 1 is an external perspective view of an impedance adjustment element according to an embodiment of the present invention. It is a three-view figure (a front view, a top view, and a side view) of the impedance adjustment element which concerns on embodiment of this invention. It is a graph showing the frequency characteristic of the impedance adjustment element which consists of a structure of FIG. 3, FIG. It is a functional block diagram of the high frequency module concerning the embodiment of the present invention. It is a functional block diagram of the high frequency module of another mode concerning the embodiment of the present invention. It is a graph which shows the reflection characteristic of the trap filter of the structure shown in FIG. 7, and the trap filter of a conventional structure. It is a figure which shows the input-output structure with respect to the impedance adjustment element which concerns on embodiment of this invention.

  An impedance adjustment element according to an embodiment of the present invention will be described with reference to the drawings. First, the principle for realizing the impedance adjusting element according to the embodiment of the present invention will be described.

When a high-frequency signal having a frequency f (angular frequency ω) is transmitted through a transmission line having a dielectric constant ε and a magnetic permeability μ, the wave number k [m ⁻¹ ] per unit length is expressed by the following (Expression 1). be able to.

ただし、誘電率εは、真空の誘電率ε_０と伝送線路の比誘電率ε_ｒから次式となる。

However, the dielectric constant ε is _expressed by the following equation from the dielectric constant ε _{0 of} vacuum and the relative dielectric constant ε _{r of the} transmission line.

ε = ε ₀ ε _r
Further, the magnetic permeability μ is _expressed by the following expression from the magnetic permeability μ ₀ in vacuum and the relative magnetic permeability μ _{r of the} transmission line.

μ = μ ₀ μ _r
Here, assuming a state where a high-frequency signal is transmitted in a TEM mode through a transmission line composed of parallel flat plates extending infinitely in the transmission direction, an electric field component E _{z1 in} a direction (z direction) perpendicular to the flat plate surface of the parallel flat plates. (X, t) [A / m] is expressed by the following (Formula 2). Note that x [m] represents a position from the reference point along the signal transmission direction, and t [sec. ] Represents the time.

ここで、Ｅ_Ｚ０１ ［Ａ／ｍ］はｚ方向電界成分の最大値である。

Here, E _Z01 [A / m] is the maximum value of the z-direction electric field component.

  At this time, when the first point x1 [m] having a different transmission direction is the port 1 and the second point x2 [m] is the port 2, the S parameter between the ports 1-2 is expressed by the following (Equation 3): It is obtained by (Expression 4) and (Expression 5). Port 1 is an incident port, and port 2 is an emission port.

ここで、θはポート１−２間の通過位相を表し、Ｌはポート１−２間の伝送線路長を表す。

Here, θ represents the passing phase between the ports 1-2, and L represents the transmission line length between the ports 1-2.

Here, since the measurement by the network analyzer is assumed, the electric field E _Z1 (x, t) is expressed by (Equation 2).

  Considering that the dielectric constant ε and the magnetic permeability μ have frequency dependency, the passing phase θ is expressed by the following (Expression 6) from (Expression 5) and (Expression 1).

（式６）を微分すると、すなわち、位相θの各周波数ω特性を算出すると、次の（式７）になる。

When (Equation 6) is differentiated, that is, when each frequency ω characteristic of the phase θ is calculated, the following (Equation 7) is obtained.

（式７）の微分項を計算すると、次の（式８）になる。

When the differential term of (Expression 7) is calculated, the following (Expression 8) is obtained.

従来ある通常の伝送線路（正の電気長を有する伝送線路）では、図１（Ａ）に示すように、ｄθ／ｄωは負の特性を有することが知られている。図１は伝送線路の位相θの周波数特性を示すグラフである。図１（Ａ）が従来ある通常の伝送線路の特性を示し、図１（Ｂ）が後述する本願構成の伝送線路の特性を示す。図１（Ａ）に示すように、従来ある通常の伝送線路では、周波数の増加に伴って位相が負の方向に回転する。

In a conventional normal transmission line (a transmission line having a positive electrical length), as shown in FIG. 1A, dθ / dω is known to have a negative characteristic. FIG. 1 is a graph showing frequency characteristics of the phase θ of the transmission line. FIG. 1A shows the characteristics of a conventional normal transmission line, and FIG. 1B shows the characteristics of the transmission line of the present application configuration described later. As shown in FIG. 1A, in a conventional normal transmission line, the phase rotates in the negative direction as the frequency increases.

Here, from (Equation 8)
dθ / dω <0 − (Formula 9)
Consider the conditions that become.

In a conventional normal transmission line, the relative dielectric constant ε _r (assumed to be ε _rn ) and the relative permeability μ _r (tentatively assumed to be μ _rn ) are positive constants. Therefore, dε / dω = 0 and dμ / dω = 0. Further, by inputting this condition into (Expression 8), the following expression is obtained.

比誘電率ε_ｒｎ、および、比透磁率μ_ｒｎは正の定数であるので、（式１０）からｄθ／ｄω＜０となる。このように、図１（Ａ）に示す従来ある通常の伝送線路（正の電気長を有する伝送線路）の特徴と一致する。したがって、上述の（式８）は現実の伝送線路の特性と一致するものと判断できる。

Since the relative dielectric constant ε _rn and the relative magnetic permeability μ _rn are positive constants, dθ / dω <0 from (Equation 10). Thus, it matches the characteristics of the conventional normal transmission line (transmission line having a positive electrical length) shown in FIG. Therefore, it can be determined that the above (Equation 8) matches the actual characteristics of the transmission line.

For such a transmission line having a positive electrical length, a transmission line or an impedance adjusting element having a negative electrical length is considered. That is, consider a transmission line or an impedance adjustment element (hereinafter simply referred to as an impedance adjustment element) having the characteristics shown in FIG. When an impedance element having such characteristics is used, the characteristic of increasing the phase θ as the frequency increases, that is,
dθ / dω> 0 − (formula 11)
The characteristics can be realized. By realizing such a characteristic of dθ / dω> 0, a negative impedance can be realized.

  As a result, the parasitic component (parasitic capacitance) generated by realizing the matching circuit by the mounting circuit component mounted on the substrate or the conductor pattern formed on the substrate, or the parasitic component generated in the mounting circuit component or mounting circuit pattern to be matched A phase characteristic (phase frequency characteristic) opposite to that of the component (parasitic capacitance) can be obtained. Therefore, parasitic components can be canceled in a wide frequency band. Thereby, impedance matching is possible in a wide frequency band.

  FIG. 2 is a conceptual diagram showing a change in pass characteristic (S21) according to an impedance matching mode. FIG. 2 is based on the simulation result calculated based on the above equation (8). FIG. 2 shows pass characteristics (S21) when impedance matching is not performed, when conventional impedance matching is performed, and when impedance matching of the present application is performed.

  As shown in FIG. 2, when conventional impedance matching (impedance matching (dθ / dω <0) using a transmission line with a positive electrical length) is performed, the transmission loss of the specific frequency fs is reduced, but the transmission loss is low. The bandwidth of the frequency band becomes narrower.

  On the other hand, as shown in FIG. 2, when the impedance matching of the present application (impedance matching (dθ / dω> 0) using a transmission line having a negative electrical length) is performed, transmission loss can be reduced in a wide frequency band.

  As described above, by using an impedance matching element that realizes dθ / dω> 0, impedance matching can be performed in a wider frequency band than the conventional configuration.

  Next, a specific configuration example of the impedance matching element that realizes the above-described dθ / dω> 0 will be described with reference to the drawings. FIG. 3 is an external perspective view of the impedance adjusting element according to the embodiment of the present invention. FIG. 4 is a three-side view (a front view, a plan view, and a side view) of the impedance adjusting element according to the embodiment of the present invention. FIGS. 3 and 4 show a perspective view of the waveguide. FIG. 5 is a graph showing the frequency characteristics of the impedance adjusting element having the structure shown in FIGS. FIG. 5A shows frequency characteristics of relative permittivity, FIG. 5B shows frequency characteristics of relative permeability, and FIG. 5C shows frequency characteristics of dθ / dω.

  As shown in FIGS. 3 and 4, the impedance adjusting element 10 includes a rectangular waveguide 100, metal meshes 21, 22, 23, 24, 25, 26, 27, 28, and ferrites 31, 32, 33, 34, 35. , 36, 37, 38. In addition, in the impedance adjusting element 10 of this embodiment, although the example which uses a metal mesh and a ferrite 8 each was shown, if the number of the metal mesh and a ferrite to be used is suitably set based on the desired impedance characteristic Good. The portion made of the metal mesh 21-28 corresponds to the “negative dielectric portion” of the present invention, and the portion made of the ferrite 31-38 corresponds to the “negative magnetic body portion” of the present invention.

  The rectangular waveguide 100 has a shape extending in the x direction with the z direction as the height direction and the y direction as the width direction. One end of the waveguide 100 in the high-frequency signal transmission direction corresponds to the first port, and the other end corresponds to the second port.

  The metal mesh 21-28 and the ferrite 31-38 are disposed in the rectangular waveguide 100. The metal mesh 21-28 is formed on the surface of a flat insulator, for example, and the insulator is disposed in the rectangular waveguide 100 so that the flat surface of the insulator is orthogonal to the high-frequency signal transmission direction. By doing so, the metal mesh 21-28 can be arranged in a standing state in the rectangular waveguide 100. The metal meshes 21-28 and philite 31-38 are alternately arranged along the x direction of the rectangular waveguide 100. In other words, the metal meshes 21-28 and the philites 31-38 are alternately arranged along the high-frequency signal transmission direction of the rectangular waveguide 100. Specifically, the metal mesh 21, ferrite 31, metal mesh 22, ferrite 32, metal mesh 23, ferrite 33, metal mesh 24, ferrite 34, metal mesh 25, ferrite along the x direction (high-frequency signal transmission direction). 35, metal mesh 26, ferrite 36, metal mesh 27, ferrite 37, metal mesh 28, and ferrite 38 are arranged in this order. At this time, a region between adjacent ferrites may be filled with air or a dielectric having a relative dielectric constant εr (≠ 1).

  As described above, the impedance adjusting element 10 has a structure in which the unit structures made of the metal mesh and the ferrite arranged in the high-frequency signal transmission direction are periodically arranged along the high-frequency signal transmission direction.

  In addition, the space | interval of an adjacent metal mesh and a ferrite is Pc, and the space | interval of a metal mesh is Pd. These intervals Pc and Pd are appropriately set based on desired impedance characteristics. Moreover, although the example which provided the space | interval between a ferrite and a metal mesh is shown in the above-mentioned structure, you may use the structure which forms a metal mesh in one main surface of a ferrite as a unit structure.

  The metal mesh 21-28 is made of a material such as silver Ag, aluminum Al, or nickel Ni. The metal mesh 21-28 has the same structure, and the configuration of the metal mesh 21 will be specifically described below as a representative.

  The metal mesh 21 includes vertical line conductors 201, 202, 203, 204, 205, 206, and 207 extending along the height direction (z direction) and horizontal line conductors extending along the width direction (y direction). 211, 212, 213, 214. The vertical line conductors 201-207 and the horizontal line conductors 211-214 are connected to the waveguide 100.

  The vertical linear conductors 201 to 207 have a width (length in the y direction) of Ww, and are arranged at intervals Pw along the width direction (y direction). The horizontal linear conductors 211-214 have a width (length in the z direction) of Wh, and are arranged at an interval Ph along the height direction (z direction). In addition, the width | variety and space | interval of the vertical linear conductor 201-207 and the horizontal linear conductor 211-214 are set based on the desired impedance characteristic.

Here, when the width Wh of the vertical linear conductor Ww and the horizontal linear conductor in each metal mesh 21-28 is Wr, and the intervals Pw, Ph, and Pd are Pa, the relative dielectric constant of the periodic structure of the metal mesh 21-28. ε _rcy can be expressed by the following formula (formula 12). Note that ε _rM is the dielectric constant between the linear conductors.

（式１２）より、比誘電率ε_ｒｃｙは、周波数分散特性を有する。すなわち、ｄε_ｒｃｙ／ｄω≠０である。したがって、各金属メッシュ２１−２８における線状導体の幅Ｗｒと、線状導体の間隔Ｐａを設定することで、所望とする周波数で比誘電率ε_ｒｃｙを負値にすることができる。例えば、比誘電率ε_ｒｃｙを、図５（Ａ）に示すような特性にすることができる。

From (Equation 12), the relative dielectric constant ε _rcy has frequency dispersion characteristics. That is, dε _rcy / dω ≠ 0. Therefore, by setting the width Wr of the linear conductor in each metal mesh 21-28 and the interval Pa between the linear conductors, the relative dielectric constant ε _rcy can be made negative at a desired frequency. For example, the relative dielectric constant ε _rcy can have a characteristic as shown in FIG.

  The ferrite 31-38 is a flat plate extending in the width direction (y direction) and the height direction (z direction) of the rectangular waveguide 100. Each side surface orthogonal to the flat plate surface of the ferrite 31-38 is in contact with the inner wall surface of the rectangular waveguide 100.

The ferrite 31-38 has a magnetic resonance point, and the relative permeability μ _rcy that the periodic structure of the ferrite 31-38 has can be expressed by the following equation (13). H ₀ [Oe] is the internal magnetic field of the ferrite, M _s [G] is the saturation magnetization, γ is the gyromagnetic ratio, and ΔH is the loss term.

なお、

である。

In addition,

It is.

From (Equation 13), the relative permeability μ _rcy has frequency dispersion characteristics. That is, dμ _rcy / dω ≠ 0. Therefore, by setting the material and shape of the ferrite 31-38, the relative permeability μ _rcy can be made negative at a desired frequency. For example, the relative magnetic permeability μ _rcy can have characteristics as shown in FIG.

As described above, by _setting both the relative permittivity ε _rcy and the relative permeability μ _rcy to negative values (ε _rcy <0, μ _rcy <0), one condition in which the right side of the above (formula 8) becomes a positive value. Can be obtained. That is, one condition that satisfies dθ / dω> 0 expressed by (Equation 11) can be obtained.

Furthermore, when the equation (8) is modified by applying the conditions of ε _rcy <0, μ _rcy <0, dε _rcy / dω, dμ _rcy / dω, and dθ / dω> 0, the following equation (14) ) Is obtained.

この場合、ｄθ／ｄωの特性図は、図５（Ｃ）となる。

In this case, the characteristic diagram of dθ / dω is as shown in FIG.

Thus, by configuring the periodic structure of the metal mesh 21-28 and the ferrite 31-38 so as to satisfy the condition of (Equation 14) and the condition of ε _rcy <0, μ _rcy <0, It is possible to realize the impedance matching element 10 having an electrical length of. As described above, by using the impedance matching element 10 having this configuration, impedance matching can be performed in a wide frequency band that cannot be realized by the conventional configuration.

  Note that the frequency bandwidth capable of impedance matching can be adjusted by the loss term ΔH of the ferrite 31-38. Specifically, the greater the loss term ΔH of the ferrite 31-38, the wider the frequency band in which impedance matching is possible. However, the transmission loss increases as the loss term ΔH increases. Therefore, the loss term ΔH may be determined based on the frequency bandwidth for impedance matching and the transmission loss.

  The impedance matching element 10 described above can be applied to the following high frequency module. FIG. 6 is a functional block diagram of the high-frequency module according to the embodiment of the present invention.

  As shown in FIG. 6, the high-frequency module 90 includes an RF front end circuit 91 and an antenna 92, and also includes the impedance matching element 10 having the above-described configuration. The impedance matching element 10 is connected between the RF front end circuit 91 and the antenna 92.

  The transmission signal generated by the RF front end circuit 91 is supplied to the antenna 92 via the impedance matching element 10. The antenna 92 radiates a transmission signal to the outside. The reception signal received by the antenna 92 is output to the RF front end circuit 91 via the impedance matching element 10. The RF front end circuit 91 demodulates the received signal.

  In the high-frequency module 90, the impedance matching between the RF front end circuit 91 and the antenna 92 is performed by the impedance matching element 10 described above. Therefore, the RF front end circuit 91, the antenna 92, the RF front end circuit 91, and the impedance Parasitic components (parasitic capacitance) such as a transmission line between the matching element 10 and a transmission line between the antenna 92 and the impedance matching element 10 can be canceled out. Therefore, impedance matching between the RF front-end circuit 91 and the antenna 92 can be realized in a wide frequency band, and a high-frequency signal can be transmitted between the RF front-end circuit 91 and the antenna 92 with low loss. it can.

  At this time, even if the broadband antenna and the RF front-end circuit 91 corresponding to the broadband are connected, only one impedance matching element 10 described above is used without using a plurality of impedance matching circuits or impedance matching elements. It is possible to transmit a plurality of high-frequency signals spread over a wide band with low loss. That is, a high-frequency module that transmits and receives high-frequency signals of a plurality of bands can be formed in a small size.

  Moreover, the above-described impedance matching element 10 can be applied to the following high-frequency module. FIG. 7 is a functional block diagram of a high-frequency module of another aspect according to the embodiment of the present invention. FIG. 8 is a graph showing the reflection characteristics of the trap filter having the configuration shown in FIG. 7 and the trap filter having the conventional configuration. The broken line in FIG. 8 shows the reflection characteristic of the trap filter having the conventional configuration, and the solid line shows the reflection characteristic of the trap filter having the configuration shown in FIG.

  As shown in FIG. 7, the high-frequency module 90A includes a SAW resonator 93 and a variable capacitance element 94, and also includes the impedance matching element 10 having the above-described configuration. The high frequency module 90A includes a first port P1 and a second port P2. The SAW resonator 93 and the variable capacitance element 94 are connected in parallel, and the parallel circuit of the SAW resonator 93 and the variable capacitance element 94 is between the transmission conductor connecting the first and second ports P1 and P2 and the ground. It is connected to the. Thereby, the high frequency module 90A functions as a trap filter whose attenuation pole frequency is variable.

  Further, in the high frequency module 90A, the impedance matching element 10 is connected between the transmission conductor connecting the first and second ports P1 and P2 and the ground. With such a configuration, the impedance matching element 10 can cancel the parasitic component (parasitic capacitance) included in the parallel circuit of the SAW resonator 93 and the variable capacitance element 94. Therefore, the variable frequency range of the attenuation pole frequency can be widened. For example, as shown in FIG. 8, the variable frequency range Δfoi of the high-frequency module 90A can be made wider than the variable frequency range Δfop of the conventional configuration. Furthermore, with this configuration, a trap filter with a variable attenuation pole frequency in a wide band can be realized by adding one impedance matching element 10, so that the trap filter can be formed in a small size.

  In addition, what is necessary is just to input / output a high frequency signal with the following structure with respect to the impedance adjustment element which consists of the above-mentioned structure, for example. FIG. 9 is a diagram showing an input / output structure for the impedance adjustment element according to the embodiment of the present invention. FIG. 9A is a perspective view showing an input / output structure, and FIG. 9B is a schematic circuit diagram.

  Antenna conductors 40 are respectively disposed at both ends of the rectangular waveguide 100 in the high-frequency signal transmission direction (x direction). The antenna conductor 40 includes a first antenna conductor 41 and a second antenna conductor 42 that extend in two directions orthogonal to the high-frequency signal transmission direction. The first antenna conductor 41 has a shape extending in the y direction, and the second antenna conductor 42 has a shape extending in the z direction. The second antenna conductor 42 is divided into two. With this configuration, the first antenna conductor 41 and the second antenna conductor 42 are arranged in an orthogonal manner in a state of being separated from each other.

  Connectors 50 are respectively disposed on the outer walls at both ends of the rectangular waveguide 100 in the high-frequency signal transmission direction. The connector 50 is a two-conductor connector, for example, a coaxial connector. One conductor of the connector 50 is connected to the first antenna conductor 41, and the other conductor of the connector 50 is connected to the second antenna conductor 42. At this time, a 90 ° phase shifter 60 (not shown in FIG. 9A) is connected between the other conductor of the connector 50 and the second antenna conductor 42.

  With such a configuration, when a high-frequency signal is input through one connector 50, the high-frequency signal is propagated from the one antenna conductor 40 into the rectangular waveguide 100 as a helically polarized wave. Further, the high-frequency signal propagated through the impedance adjustment element 10 is received by the other antenna conductor 40 and output from the other connector 50 to the outside.

  In the above description, the impedance adjusting element having a configuration in which unit structures made of a metal mesh and ferrite are periodically arranged in a rectangular waveguide has been described as an example. However, the relative permittivity and relative permeability are negative values. There may be other configurations as long as the structure satisfies the condition of (Expression 14) described above, that is, a structure satisfying dθ / dω> 0.

10: Impedance adjusting element 21-28: Metal mesh 31-38: Ferrite 90, 90A: High frequency module 91: RF front end circuit 92: Antenna 93: SAW resonator 94: Variable capacitance element 100: Rectangular waveguide 201-207 : Vertical linear conductor 211-214: Horizontal linear conductor

Claims (6)

An impedance adjusting element that is connected between the first port and the second port and adjusts an impedance for a high-frequency signal transmitted between the first port and the second port;
An impedance adjusting element having a structure in which dθ / dω, which is a frequency characteristic of a change amount dθ of a phase θ of a high frequency signal to be transmitted, is a negative value. A negative dielectric portion having a negative dielectric constant with respect to the high-frequency signal;
A negative magnetic part having a negative permeability with respect to the high-frequency signal;
The impedance adjusting element according to claim 1, comprising: The negative dielectric part is composed of a plurality of metal meshes arranged at intervals along the transmission direction of the high-frequency signal in the rectangular waveguide,
The negative magnetic body portion is composed of a plurality of ferrites arranged at intervals along the transmission direction of the high-frequency signal in the rectangular waveguide.
The impedance adjusting element according to claim 2. The plurality of metal meshes and the plurality of ferrites are alternately arranged along a transmission direction of the high-frequency signal.
The impedance adjusting element according to claim 3. The impedance adjusting element according to any one of claims 1 to 4,
A first high-frequency circuit unit connected to the first port;
A second high-frequency circuit unit connected to the second port;
High frequency module with The impedance adjusting element according to any one of claims 1 to 4,
A variable trap filter connected between a transmission conductor connecting the first port and the second port and a ground;
The impedance adjusting element is a high frequency module connected between the transmission conductor and a ground.

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