JP2014212390A - Oscillator circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To obtain an oscillation signal having a stable oscillation frequency in a wide frequency range.SOLUTION: An oscillator circuit includes: a first amplifier 1 for amplifying an inputted signal; a second amplifier 21 for invert-amplifying an output signal of the first amplifier 1 and inputting the resultant signal to the first amplifier 1; a resistor 26 connected between an input and an output of the second amplifier 21; a first oscillator 5 and a third amplifier 3 serially connected to each other in parallel to the resistor 26; and a second oscillator 6 and a fourth amplifier 4 serially connected to each other in parallel to the resistor 26. The first oscillator 5 has a different antiresonance frequency from the second oscillator 6, and at least one of the third amplifier 3 and the fourth amplifier 4 has variable gain.

Description

  The present invention relates to an oscillation circuit capable of adjusting an oscillation frequency.

  Conventionally, by using a plurality of crystal resonators having different resonance frequencies, an anti-resonance circuit that can adjust an oscillation frequency in a frequency range wider than a frequency range that can be adjusted by a single crystal resonator is known (for example, (See Patent Document 1).

  FIG. 15 shows a configuration example of a conventional anti-resonance circuit 400. In FIG. 15, the anti-resonance circuit 400 is connected to the output resistor 440 and the load resistor 450 of the AC signal source 430.

  The anti-resonance circuit 400 includes a crystal resonator 411 and a crystal resonator 421 connected to different paths between the output resistor 440 and the load resistor 450. In the first path to which the crystal resonator 411 is connected, an attenuator 412, an inductor 413, and a capacitor 414 are provided in series. The crystal resonator 411 is connected to a connection point between the inductor 413 and the capacitor 414 and the ground. Similarly, an attenuator 422, an inductor 423, and a capacitor 224 are provided in series on the second path to which the crystal resonator 421 is connected. The crystal resonator 421 is connected to a connection point between the inductor 423 and the capacitor 424 and the ground.

  The crystal resonator 411 and the crystal resonator 421 have different resonance frequencies, and are connected to each other via a capacitor 414 and a capacitor 424. As a result, the anti-resonance circuit 400 resonates at a frequency between the resonance frequency of the crystal resonator 411 and the resonance frequency of the crystal resonator 421. By changing the attenuation rates of the attenuator 412 and the attenuator 422, the anti-resonance frequency of the anti-resonance circuit 400 changes.

JP 2007-295256 A

By the way, the resonance frequency of a resonator having a high Q such as a crystal resonator or a MEMS (Micro-Electro-Mechanical Systems) resonator is f _L = (1 / 2π) √ {(C ₁ + C _L ) / L ₁ C. ₁ C _L }. Here, C ₁ is the motional capacitance of the equivalent circuit of the resonator, the C _L load capacitor, L ₁ is a series inductance of the transducer.

In the oscillation circuit C ₁ is used a relatively small MEMS resonator, the frequency adjustment is performed by adjusting the bias voltage applied to the vibrator. However, if the C ₁ with respect to the capacitance value of a few pF orders can be realized in integrated circuits and discrete components are used very small vibrator oscillator, f based on the relationship between C L _>> C _{1 L} = Since it can be approximated as (1 / 2π) √ (1 / L ₁ C ₁ ), the resonance frequency is determined based on L ₁ and C ₁ of the vibrator. Therefore, when the above vibrator is used in an oscillation circuit, the temperature characteristic of the resonance frequency of the vibrator is directly reflected in the temperature characteristic of the oscillation frequency.

  In particular, the temperature characteristic of the resonance frequency of the MEMS vibrator is about −30 ppm / ° C., and the frequency change range with respect to the temperature change is relatively large. Therefore, in an oscillation circuit using a MEMS vibrator, it is difficult to obtain a stable oscillation frequency by canceling the temperature change only by adjusting the bias voltage.

  In the anti-resonance circuit 400 shown in FIG. 15, the anti-resonance frequency can be changed in a wider frequency range than when the bias voltage of a single MEMS vibrator is adjusted. However, in the anti-resonance circuit 400, in order to make the Q value at the anti-resonance frequency large enough to be used in the oscillation circuit, the inductance values of the inductor 413 and the inductor 423 must be sufficiently large. It was. Specifically, in Patent Document 1, 27 μH is exemplified as the inductance values of the inductor 413 and the inductor 423.

  However, the inductance value of the inductor changes greatly according to the temperature change. It is also difficult to adjust the inductance value according to the variation of the vibrator. Therefore, an oscillation signal having a stable oscillation frequency cannot be obtained in a resonance circuit using an inductor. Furthermore, it has been difficult to incorporate an inductor having an inductance value on the order of μH in an integrated circuit. Therefore, an oscillation circuit that can obtain an oscillation signal having a stable oscillation frequency using the conventional anti-resonance circuit 400 cannot be mass-produced at a low cost.

  Accordingly, the present invention has been made in view of these points, and an oscillation circuit that generates an oscillation signal that can be incorporated in an integrated circuit and that can change an oscillation frequency in a wide frequency range is provided. The purpose is to provide.

  In an embodiment of the present invention, a first amplifier that amplifies an input signal, a second amplifier that inverts and amplifies an output signal of the first amplifier, and inputs the first amplifier, and an input / output of the second amplifier A connected resistor; a first oscillator and a third amplifier connected in series with each other in parallel; and a second oscillator and a fourth amplifier connected in series with each other in parallel with the resistor; The first vibrator has an anti-resonance frequency different from that of the second vibrator, and at least one of the third amplifier and the fourth amplifier provides an oscillation circuit having a variable gain. For example, the resonance frequency of the first vibrator is lower than the resonance frequency of the second vibrator, and the anti-resonance frequency of the first vibrator is higher than the resonance frequency of the second vibrator.

  In the oscillation circuit described above, the resonance frequency of the first vibrator is lower than the resonance frequency of the second vibrator, is provided in parallel with the first vibrator, and is opposite to the equivalent parallel capacitance of the first vibrator. A first negative capacitance circuit having a capacitance of the following polarity may be further provided. In this case, the resonance frequency and antiresonance frequency of the first vibrator are lower than, for example, the resonance frequency and antiresonance frequency of the second vibrator.

  The first negative capacitance circuit includes, for example, an inverting amplifier connected to one terminal of the first vibrator, and a first capacitor connected to the other terminal of the first vibrator and the inverting amplifier. .

  The oscillation circuit may further include a third negative capacitance circuit provided between the input terminal of the second amplifier and the ground.

  According to the oscillation circuit of the present invention, it is possible to generate an oscillation signal that can be built in an integrated circuit and can change the oscillation frequency in a wide frequency range.

2 shows a configuration example of an oscillation circuit according to the first embodiment. The frequency characteristic of the impedance of a 1st vibrator and a 2nd vibrator is shown. The simulation result of the transfer characteristic from Vi to Vo of the oscillation circuit shown in FIG. 1 is shown. 3 shows a configuration example of an oscillation circuit according to a second embodiment. The frequency characteristic of the impedance of the 1st vibrator and the 2nd vibrator in a 2nd embodiment is shown. The equivalent circuit of a 1st vibrator is shown. The equivalent circuit of a 2nd vibrator is shown. The simulation result of the transfer characteristic from Vi to Vo of the oscillation circuit shown in FIG. 4 is shown. The simulation result of the transfer characteristic from Vi to Vo of the oscillation circuit shown in FIG. 4 is shown. The structural example of a 1st negative capacitance circuit is shown. 10 shows simulation results of transfer characteristics when the first negative capacitance circuit shown in FIG. 9 is connected and when it is not connected. 4 shows a configuration example of an oscillation circuit according to a third embodiment. The structural example of a 3rd negative capacitance circuit is shown. The result of having obtained | required by simulation the imaginary number component of the input admittance of the 3rd negative capacitance circuit shown in FIG. 12 shows the open loop characteristics of the feedback path alone of the oscillation circuit shown in FIG. The structural example of the conventional antiresonance circuit is shown.

<First Embodiment>
[Circuit Configuration of Oscillation Circuit 100]
FIG. 1 shows a configuration example of the oscillation circuit 100 according to the first embodiment. The oscillation circuit 100 includes a first amplifier 1, an inverting amplifier 2 including a second amplifier 21, a third amplifier 3, a fourth amplifier 4, a first oscillator 5, and a second oscillator 6.

The first amplifier 1 amplifies an input signal and generates an oscillation signal. The inverting amplifier 2 has an input side connected to the output side of the first amplifier 1 via the capacitor 14, and an output side connected to the input side of the first amplifier 1 via the capacitor 15. The third amplifier 3 and the first vibrator 5 are connected in series with each other in parallel with the second amplifier 21 and the resistor 26. The fourth amplifier 4 and the second vibrator 6 are connected in series with each other in parallel with the second amplifier 21 and the resistor 26. At least one of the third amplifier 3 and the fourth amplifier 4 has a variable gain, and can change the gain according to a control signal input from the outside of the oscillation circuit 100, for example. The first vibrator 5 and the second vibrator 6 have different antiresonance frequencies.
Hereinafter, details of the circuit of the oscillation circuit 100 will be described.

  The first amplifier 1 includes a transistor 11, a resistor 12, and a resistor 13. The transistor 11 is an NPN transistor. The emitter of the transistor 11 is connected to the ground. The collector of the transistor 11 is connected to the power supply via the resistor 12. The collector of the transistor 11 is connected to the input side of the inverting amplifier 2 via the capacitor 14, and the base of the transistor 11 is connected to the output side of the inverting amplifier 2 via the capacitor 15. A resistor 13 is provided between the base and collector of the transistor 11.

The inverting amplifier 2 includes a second amplifier 21, a resistor 22, a resistor 23, a capacitor 24, a resistor 25, and a resistor 26.
The second amplifier 21 inverts and amplifies the output signal of the first amplifier 1 and inputs it to the first amplifier 1. Specifically, the second amplifier 21 is, for example, an operational amplifier, and one end of a resistor 22, a resistor 23, and a capacitor 24 is connected to the non-inverting input terminal. The other end of the resistor 22 is connected to the power source, and the other ends of the resistor 23 and the capacitor 24 are connected to the ground. A voltage obtained by dividing the power supply voltage according to the resistance ratio between the resistor 22 and the resistor 23 is applied to the non-inverting input terminal of the second amplifier 21. The inverting input terminal of the second amplifier 21 is connected to the resistor 25, the resistor 26, the first vibrator 5, and the second vibrator 6.

  The resistor 26 is connected between the input and output of the second amplifier 21. That is, one end of the resistor 26 is connected to the inverting input terminal of the second amplifier 21 and the resistor 25, and the other end of the resistor 26 is connected to the output terminal of the second amplifier 21.

  The third amplifier 3 includes a transistor 31, a variable resistor 32, a variable resistor 33, and a resistor 34. The collector of the transistor 31 is connected to the power source. The base of the transistor 31 is connected to the variable resistor 32 and the variable resistor 33. The emitter of the transistor 31 is connected to the ground via the resistor 34 and is connected to one end of the first vibrator 5.

  The transistor 31, the variable resistor 32, the variable resistor 33, and the resistor 34 constitute a collector ground circuit. That is, the transistor 31 functions as an emitter follower that outputs from the emitter a voltage that changes according to the resistance ratio between the variable resistor 32 and the variable resistor 33. The variable resistor 32 and the variable resistor 33 are digital potentiometers that can be adjusted to a resistance value according to a control signal from the outside, for example.

  The fourth amplifier 4 includes a transistor 41, a variable resistor 42, a variable resistor 43, and a resistor 44. The collector of the transistor 41 is connected to the power source. The base of the transistor 41 is connected to the variable resistor 42 and the variable resistor 43. The emitter of the transistor 41 is connected to the ground via the resistor 44 and is connected to one end of the second vibrator 6.

  The transistor 41, the variable resistor 42, the variable resistor 43, and the resistor 44 constitute a collector ground circuit. That is, the transistor 41 functions as an emitter follower that outputs a voltage that changes according to the resistance ratio between the variable resistor 42 and the variable resistor 43 from the emitter. The variable resistor 42 and the variable resistor 43 are digital potentiometers that can be adjusted to a resistance value according to a control signal from the outside, for example.

  In FIG. 1, the variable resistor 32, the variable resistor 33, the variable resistor 42, and the variable resistor 43 are shown as variable resistors, but at least one of these variable resistors may be a variable resistor. Further, as a method of changing the gains of the third amplifier 3 and the fourth amplifier 4, a method other than the method of changing the resistance values of the variable resistor 32, the variable resistor 33, the variable resistor 42, and the variable resistor 43 may be used.

The first vibrator 5 and the second vibrator 6 are, for example, an AT cut crystal vibrator, an SC cut crystal vibrator, and a MEMS vibrator. The first vibrator 5 and the second vibrator 6 have different resonance frequencies (series resonance frequencies) and antiresonance frequencies (parallel resonance frequencies), respectively. The resonance frequency of the first vibrator 5 is fr ₁ and the anti-resonance frequency is fa ₁ . The resonance frequency of the second vibrator 6 is fr ₂ and the anti-resonance frequency is fa ₂ .

In the present embodiment, the resonance frequency of the second vibrator 6 is larger than the resonance frequency of the first vibrator 5, the anti-resonance frequency of the first vibrator 5 is larger than the resonance frequency of the second vibrator 6, and the first The anti-resonance frequency of the second vibrator 6 is larger than the anti-resonance frequency of the vibrator 5. That is, the relationship between the resonance frequency and antiresonance frequency of the first vibrator 5 and the second vibrator 6 is fr ₁ <fr ₂ <fa ₁ <fa ₂ .

FIG. 2 shows the frequency characteristics of the impedances of the first vibrator 5 and the second vibrator 6. The dotted line indicates the impedance of the first vibrator 5, and the chain line indicates the impedance of the second vibrator 6. The impedance of the first vibrator 5 is minimum at the resonance frequency fr _{1 and} maximum at the anti-resonance frequency fa ₁ . The impedance of the second vibrator 6 is minimum at the resonance frequency fr _{2 and} maximum at the anti-resonance frequency fa ₂ .

ここで、Ｇ_１は、第３増幅器３の入力から出力までの利得、Ｇ_２は、第４増幅器４の入力から出力までの利得、Ｚ_１は第１振動子５のインピーダンス、Ｚ_２は第２振動子６のインピーダンスを示す。Ｒ_１は抵抗２５の抵抗値、Ｒ_２は抵抗２６の抵抗値を示す。 [Gain characteristics of oscillation circuit 100]
Estimate the transfer characteristics from the input Vi to the output Vo when the oscillation circuit 100 is an open loop at the input terminal of the inverting amplifier 2, the voltage input to the inverting amplifier 2 is Vi, and the voltage output from the inverting amplifier 2 is Vo. Then, it is expressed by the following formula.

Here, G ₁ is the gain from the input to the output of the third amplifier 3, G ₂ is the gain from the input to the output of the fourth amplifier 4, Z ₁ is the impedance of the first vibrator 5, and Z ₂ is the _first gain. The impedance of the two vibrators 6 is shown. R ₁ represents the resistance value of the resistor 25, and R ₂ represents the resistance value of the resistor 26.

When the gain G ₁ of the third amplifier 3 is sufficiently larger than the gain G ₂ of the fourth amplifier 4, the transfer characteristic is mainly determined by the ratio of R ₁ and Z _1, and the impedance Z of the first vibrator 5 is determined. The denominator of the above transfer characteristic equation is minimized at the anti-resonance frequency of the first vibrator 5 where ₁ is the maximum. Therefore, a peak in which the gain of the transfer characteristic is greater than 1 appears at the antiresonance frequency of the first vibrator 5, and the oscillation condition of the oscillation circuit 100 is satisfied.

Similarly, if the gain G ₂ of the fourth amplifier 4 is sufficiently larger than the gain G ₁ of the third amplifier 3, the transfer characteristic is mainly determined by the ratio between R ₂ and Z _2, the second oscillator 6 impedance Z ₂ is the denominator of the equation above the transfer characteristic is minimized by anti-resonance frequency of the second oscillator 6 is a frequency at which the maximum. Therefore, a peak at which the gain of the transfer characteristic is greater than 1 appears at the antiresonance frequency of the second vibrator 6, and the oscillation condition of the oscillation circuit 100 is satisfied.

As understood from the above description, in the oscillation circuit 100, by adjusting the ratio of the gain G ₁ of the third amplifier 3 and the gain G ₂ of the fourth amplifier 4, the gain of the transfer characteristic is greater than 1 The frequency at which the peak appears can be changed. Specifically, in the third amplifier 3 shown in FIG. 1, the gain G ₁ of the third amplifier 3 can be changed by adjusting at least one of the variable resistor 32 and the variable resistor 33. Further, in the fourth amplifier 4 shown in FIG. 1, by adjusting at least one of the variable resistor 42 and variable resistor 43, it is possible to vary the gain G ₂ of the fourth amplifier 4. The gains of the third amplifier 3 and the fourth amplifier 4 may be changed by other methods.

FIG. 3 shows a simulation result of the transfer characteristic from Vi to Vo of the oscillation circuit 100 shown in FIG. In the example shown in FIG. 3, the resonance frequency of the first vibrator 5 is 3.816 MHz, the anti-resonance frequency is 4.104 MHz, the resonance frequency of the second vibrator 6 is 3.823 MHz, and the anti-resonance frequency is 4.115 MHz. It is said. The solid line shows the characteristics when adjusted to G ₁ = G ₂ = 0.5 times. Dotted lines _indicate the characteristics when the adjusted G 1 = 0.95 _times, the G 2 = 0.05 times. The chain line indicates the characteristics when G ₁ = 0.05 times and G ₂ = 0.95 times. In the gain characteristics, it can be seen that different peak frequencies are obtained in the vicinity of 4.1 MHz, and the phase changes 180 degrees in conjunction with the peak frequencies.

  Here, the first amplifier 1 changes the phase of the input signal by 180 degrees and outputs it. Further, the inverting amplifier 2 changes the phase of the output signal of the first amplifier 1 by 180 degrees and inputs it to the first amplifier 1. Therefore, by combining the gain of the first amplifier 1 and the peak gain of the inverting amplifier 2 to 0 dB or more, both the gain and phase oscillation conditions can be satisfied.

  As described above, according to the oscillation circuit 100 according to the first embodiment, the first amplifier 1, the second amplifier 21, the resistor 26, and the resistor 26 are connected in series with each other in series. 5 and the third amplifier 3, the second vibrator 6 and the fourth amplifier 4 connected in series with each other in parallel with the resistor 26, and the first vibrator 5 is different from the second vibrator 6 in antiresonance. An oscillation frequency between the anti-resonance frequency of the first vibrator 5 and the anti-resonance frequency of the second vibrator 6 because at least one of the third amplifier 3 and the fourth amplifier 4 has a variable gain. Can be changed.

<Second Embodiment>
[Equipped with floating negative capacitance circuit]
FIG. 4 shows a configuration example of the oscillation circuit 200 according to the second embodiment. The oscillation circuit 200 differs from the oscillation circuit 100 shown in FIG. 1 in that it further includes a first negative capacitance circuit 7 and a second negative capacitance circuit 8, and is the same in other respects. The first negative capacitance circuit 7 is provided in parallel with the first vibrator 5, and has a capacitance opposite to that of the equivalent parallel capacitance of the first vibrator 5 and a size approximately equal to the equivalent parallel capacitance. The second negative capacitance circuit 8 is provided in parallel with the second vibrator 6, and has a capacitance opposite to that of the equivalent parallel capacitance of the second vibrator 6 and a size that is approximately equal to the equivalent parallel capacitance.

  Note that the oscillation circuit 200 only needs to be provided with a negative capacitance circuit on the vibrator having a lower antiresonance frequency. For example, when the anti-resonance frequency of the first vibrator 5 is lower than the anti-resonance frequency of the second vibrator 6, the oscillation circuit 200 only needs to include at least the first negative capacitance circuit 7.

FIG. 5 shows the frequency characteristics of the impedances of the first vibrator 5 and the second vibrator 6 in the second embodiment. The relationship between the resonance frequency and the antiresonance frequency of the first vibrator 5 and the second vibrator 6 shown in FIG. 5 is fr ₁ <fa ₁ <fr ₂ <fa ₂ . That is, the resonance frequency and antiresonance frequency of the first vibrator 5 are smaller than the resonance frequency and antiresonance frequency of the second vibrator 6. In this case, the resonance frequency fr ₂ of the second vibrator 6 is between the anti-resonance frequency fa ₁ of the first vibrator 5 and the anti-resonance frequency fa ₂ of the second vibrator 6. In the configuration of the oscillation circuit 100 that is assumed to change the frequency between the anti-resonance frequency fa ₁ and the anti-resonance frequency fa ₂ of the second vibrator 6, the anti-resonance frequency fa ₁ of the first vibrator and the first resonance frequency fa The oscillation condition is not satisfied in the entire frequency range between the anti-resonance frequency fa ₂ of the two vibrators 6, and the oscillation condition is satisfied only in a narrow frequency range.

  In contrast, in the oscillation circuit 200, the first negative capacitance circuit 7 and the second negative capacitance circuit 8 cancel out the equivalent parallel capacitance of the first vibrator 5 and the second vibrator 6, thereby reducing the anti-resonance frequency. The influence is suppressed, and oscillation occurs at a frequency from the resonance frequency of the first vibrator 5 to the resonance frequency of the second vibrator 6.

  FIG. 6A shows an equivalent circuit of the first vibrator 5. In the first vibrator 5, an equivalent parallel capacitor 51 and an equivalent series resistor 52, an equivalent series inductor 53, and an equivalent series capacitor 54 connected in series with each other are connected in parallel. FIG. 6B shows an equivalent circuit of the second vibrator 6. In the second vibrator 6, an equivalent parallel capacitor 61 and an equivalent series resistor 62, an equivalent series inductor 63, and an equivalent series capacitor 64 connected in series with each other are connected in parallel.

  The first negative capacitance circuit 7 is provided in parallel with the first vibrator 5, and has a capacitance opposite to the capacitance of the equivalent parallel capacitor 51 and substantially the same size. Can be countered. Similarly, the second negative capacitance circuit 8 is provided in parallel with the second vibrator 6 and has a capacitance opposite to that of the equivalent parallel capacitor 61 and substantially the same size. The equivalent parallel capacitor 61 of the two vibrators 6 can be canceled out.

  FIG. 7 shows a simulation result of the transfer characteristic from Vi to Vo of the oscillation circuit 200. In this embodiment, the resonance frequency of the first vibrator 5 is 11.0000 MHz, and the resonance frequency of the second vibrator 6 is 11.0592 MHz. A dotted line indicates characteristics when the oscillation circuit 200 includes the first vibrator 5 and the first negative capacitance circuit 7 and does not include the second vibrator 6 and the second negative capacitance circuit 8. A chain line indicates characteristics when the oscillation circuit 200 includes the second vibrator 6 and the second negative capacitance circuit 8 and does not include the first vibrator 5 and the first negative capacitance circuit 7. A solid line indicates characteristics when the oscillation circuit 200 includes the first vibrator 5 and the first negative capacitance circuit 7, and the second vibrator 6 and the second negative capacitance circuit 8.

As indicated by the dotted line, when the oscillation circuit 200 includes the first vibrator 5 and the first negative capacitance circuit 7 and does not include the second vibrator 6 and the second negative capacitance circuit 8, impedance Z ₁ is minimized at the resonant frequency of the first oscillator 5. Further, as indicated by a chain line, when the oscillation circuit 200 includes the second vibrator 6 and the second negative capacitance circuit 8 and does not include the first vibrator 5 and the first negative capacitance circuit 7. the impedance Z ₂ is minimized at the resonant frequency of the second oscillator 6. Therefore, in these cases, since the negative feedback of the inverting amplifier 2 is increased, the gain of the transfer function Vo / Vi is decreased, and the oscillation condition of the oscillation circuit 200 is not satisfied.

  On the other hand, when the oscillation circuit 200 includes the first vibrator 5 and the first negative capacitance circuit 7, the second vibrator 6 and the second negative capacitance circuit 8, the resonance frequency of the first vibrator 5. 1 to a resonance frequency of the second vibrator 6, a gain peak of approximately 1 occurs, and the phase changes by 180 degrees at the frequency. As a result, the oscillation condition of the oscillation circuit 200 is satisfied.

FIG. 8 shows a simulation result of the transfer characteristics from Vi to Vo of the oscillation circuit 200 when the gains of the third amplifier 3 and the fourth amplifier 4 are changed. The solid line in FIG. 8 shows the characteristics in the case where the gain G ₁ of the third amplifier 3 and the gain G ₂ of the fourth amplifier 4 is adjusted to 0.5 times. The dotted lines show the characteristics when adjusted to G ₁ = 0.9 times and G ₂ = 0.1 times. The chain line indicates the characteristics when adjusted to G ₁ = 0.1 times and G ₂ = 0.9 times. It can be seen from FIG. 8 that by adjusting the gain of the third amplifier 3 and the gain of the fourth amplifier 4, it is possible to control the frequency at which the gain of the transfer characteristic becomes maximum and the frequency at which the phase changes by 180 degrees.

  As described above, according to the oscillation circuit 200, since the influence of the anti-resonance frequency is eliminated by canceling out the equivalent parallel capacitance of the first vibrator 5 and the second vibrator 6, the resonance frequency of the first vibrator 5 and the first It oscillates in a wide frequency range between the resonance frequencies of the two vibrators 6. For example, in the example shown in FIG. 8, a frequency variable range of ± 2000 ppm or more is secured.

  FIG. 9 shows a configuration example of the first negative capacitance circuit 7. The first negative capacitance circuit 7 includes an operational amplifier 71, a resistor 72, a resistor 73, and a capacitor 74. The operational amplifier 71 functions as an inverting amplifier, and a non-inverting input terminal is connected to the ground. The inverting input terminal of the operational amplifier 71 is connected to the first terminal of the first vibrator 5 through the resistor 72 and is connected to the output terminal of the operational amplifier 71 through the resistor 73. The output terminal of the operational amplifier 71 is connected to the second terminal of the first vibrator 5 via the capacitor 74. The operational amplifier 71 generates a signal having a phase opposite to that of the signal input from the inverting input terminal and inputs the signal to the capacitor 74.

  The capacitor 74 is connected to the second terminal of the first vibrator 5 and the operational amplifier 71. The capacitor 74 has a capacity equivalent to that of the equivalent parallel capacitor of the first vibrator 5. Since the signal output from the operational amplifier 71 is the same magnitude and opposite phase signal as the signal passing through the equivalent parallel capacitor of the first vibrator 5, the equivalent negative capacitor of the first vibrator 5 is obtained by the first negative capacitance circuit 7. Can offset the effects of

  FIG. 10 shows simulation results of the transfer characteristics of the first vibrator 5 when the first negative capacitance circuit 7 shown in FIG. 9 is connected and when it is not connected. A dotted line indicates the transfer characteristic of the first vibrator 5 when the first negative capacitance circuit 7 is not connected. The solid line indicates the transfer characteristic of the first vibrator 5 when the first negative capacitance circuit 7 is connected in parallel with the first vibrator 5. The first vibrator 5 has the minimum impedance at the resonance frequency of the first vibrator 5 and the maximum impedance at the anti-resonance frequency. Therefore, in the transfer characteristic of the single unit of the first vibrator 5 in the state where the first negative capacitance circuit 7 is not connected, as shown by the dotted line in FIG. There is a peak where the gain is minimized at the anti-resonance frequency.

  On the other hand, in the state where the first negative capacitance circuit 7 is connected, as shown by the solid line in FIG. 10, the decrease in gain that occurs at the antiresonance frequency of the first vibrator 5 disappears, and the resonance frequency Only the peaks that appear in As a result, the oscillation circuit 200 is not affected by the anti-resonance frequency of the first vibrator 5 and has a wide frequency range between the resonance frequency of the first vibrator 5 and the resonance frequency of the second vibrator 6. The oscillation frequency can be changed.

The configuration of the first negative capacitance circuit 7 is not limited to the configuration shown in FIG. For example, A. Antoniou, “Floating Negative-Impedance Converters,” IEEE
A floating negative capacitance circuit having another configuration as shown in Transactions on Circuit Theory, vol. 19, pp.209-212, March 1972 may be used.

  As described above, according to the oscillation circuit 200 according to the second embodiment, since the first negative capacitance circuit 7 is provided in parallel with the first vibrator 5, the anti-resonance frequency of the first vibrator 5 is achieved. The oscillation frequency can be changed in a wide frequency range between the resonance frequency of the first vibrator 5 and the resonance frequency of the second vibrator 6 without being affected by the above.

<Third Embodiment>
FIG. 11 shows a configuration example of an oscillation circuit 300 according to the third embodiment. In the oscillation circuit 100 illustrated in FIG. 1 and the oscillation circuit 200 illustrated in FIG. 4, the output signal of the first amplifier 1 includes a path in which the inverting amplifier 2 is provided, a path in which the first vibrator 5 is provided, The oscillation condition is satisfied by returning to the input side of the first amplifier 1 through the path in which the two vibrators 6 are provided. However, in an actual circuit, unnecessary oscillation may occur at a frequency other than the desired oscillation frequency due to the influence of a path not shown in the circuit diagram due to the influence of the stray capacitance not shown in FIGS. is there.

  Therefore, in order to prevent unnecessary oscillation, the oscillation circuit 300 further includes a third negative capacitance circuit 9 provided between the inverting input terminal of the second amplifier 21 and the ground. In other respects, the oscillation circuit 300 is the same as the oscillation circuit 200 shown in FIG.

  The magnitude of the negative capacitance of the third negative capacitance circuit 9 is C = −2 (C01 + C02). Here, C01 indicates the magnitude of the parallel capacitance of the first vibrator 5, and C02 indicates the magnitude of the parallel capacity of the second vibrator 6. By setting the negative capacitance of the third negative capacitance circuit 9 as described above, it is possible to reduce the influence of the parallel capacitance of the first vibrator 5 and the second vibrator 6 on the transfer characteristics. As a result, it is possible to eliminate the gain peak and phase rotation caused by the resonance between the parallel capacitance of the first vibrator 5 and the second vibrator 6 and the equivalent series resistance, the equivalent series inductor, and the equivalent series capacitor. Can be prevented.

  FIG. 12 shows a configuration example of the third negative capacitance circuit 9. The third negative capacitance circuit 9 includes an operational amplifier 91, a capacitor 92, a resistor 93, and a resistor 94. The non-inverting input terminal of the operational amplifier 91 is connected to the first vibrator 5 and the second vibrator 6. The capacitor 92 is provided between the non-inverting input terminal and the output terminal of the operational amplifier 91. The resistor 93 is provided between the inverting input terminal and the output terminal of the operational amplifier 91. The resistor 94 is provided between the inverting input terminal of the operational amplifier 91 and the ground.

ここで、Ｉ_ｉｎはオペアンプ９１の非反転入力端子への入力電流、Ｖ_ｉｎはオペアンプ９１の非反転入力端子への入力電圧、Ａはオペアンプ９１のオープンループ利得である。
上記の式においてＡが十分に大きければ、第３負性容量回路９の構成によって、Ｃ_ｆにほぼ等しい容量値の負性容量を得ることができる。 The capacitance value of the capacitor 92 with a C _f, at equal resistance value of the resistor 93 and the resistance value of the resistor 94, when the gain of the operational amplifier 91 functioning as a non-inverting amplifier to double, the third negative capacitance The admittance Y _{m when the} circuit 9 is viewed from the input is expressed by the following equation.

Here, I _in is an input current to the non-inverting input terminal of the operational amplifier 91, V _in is an input voltage to the non-inverting input terminal of the operational amplifier 91, and A is an open loop gain of the operational amplifier 91.
If A is sufficiently large in the above formula, a negative capacitance having a capacitance value substantially equal to C _f can be obtained by the configuration of the third negative capacitance circuit 9.

FIG. 13 shows the result of obtaining the imaginary component of the input admittance of the third negative capacitance circuit 9 shown in FIG. 12 by simulation. The vertical axis in FIG. 13 indicates the susceptance component of the third negative capacitance circuit 9. FIG. 13 shows how the imaginary number component of the third negative capacitance circuit 9 decreases as the frequency increases. The dotted line shows an asymptotic line of −jω × 24 pF, and it can be seen that substantially −C _f = −24 pF is realized at a frequency smaller than 20 MHz. Since the oscillation circuit 300 includes the third negative capacitance circuit 9, unnecessary oscillation in the feedback path of the third amplifier 3, the first oscillator 5, and the fourth amplifier 4, the second oscillator 6 can be prevented.

  FIG. 14 shows the open loop characteristics of the feedback path alone of the oscillation circuit 300. The dotted line indicates the characteristic when the third negative capacitance circuit 9 is not provided, and the solid line indicates the characteristic when the third negative capacitance circuit 9 is provided. As indicated by the dotted line, when the third negative capacitance circuit 9 is not provided, the steep phase intersects with the phase 0 near the resonance frequency of the first vibrator 5 and the second vibrator 6. There is a case where a change has occurred and a condition for causing unnecessary oscillation is satisfied. On the other hand, as shown by the solid line, when the third negative capacitance circuit 9 is provided, there is no steep phase change in the vicinity of the resonance frequency of the first vibrator 5 and the second vibrator 6. It can be seen that the condition for generating unnecessary oscillation is not satisfied. As a result, the oscillation circuit 300 can change the frequency over a wide frequency range without causing unnecessary oscillation.

Note that the configuration of the third negative capacitance circuit 9 is not limited to the configuration shown in FIG. 12, and the same effect can be obtained by a negative capacitance circuit having another configuration. For example, as the third negative capacitance circuit 9, RL Brennan, et al., “The CMOS Negative Impedance Converter,” IEEE
Circuits described in J. of Solid-State Circuits, vol. 23, No. 5, Oct. 1988, or H.
Mostafa, et al., “Novel Timing Yield Improvement Circuits for High-Performance
Low-Power Wide Fan-In Dynamic OR Gates, ”IEEE Transactions on Circuits and
Systems- I: Circuits described in Regular Papers, vol. 58, No. 8, Aug. 2011 can be applied.

  As described above, the oscillation circuit 300 according to the third embodiment further includes the third negative capacitance circuit 9 provided between the connection point of the first vibrator 5 and the second amplifier 21 and the ground. Thus, unnecessary oscillation can be prevented and the frequency can be changed in a wide frequency range.

  As mentioned above, although this invention was demonstrated using embodiment, the technical scope of this invention is not limited to the range as described in the said embodiment. It will be apparent to those skilled in the art that various modifications or improvements can be added to the above embodiment. It is apparent from the scope of the claims that the embodiments added with such changes or improvements can be included in the technical scope of the present invention.

  For example, in the above embodiment, the configuration including the first vibrator 5 and the second vibrator 6 has been described, but three or more vibrators may be provided. Further, as the first vibrator 5 and the second vibrator 6, not only a MEMS oscillator but also a crystal vibrator, a ceramic vibrator, a SAW (Surface Acoustic Wave) filter, or an LC oscillator composed of an inductor and a capacitor, etc. It can be applied to various oscillation circuits.

  In the above embodiment, an example in which a bipolar transistor is used has been described. However, a field effect transistor may be used instead of the bipolar transistor. In this case, the collector of the bipolar transistor is replaced by the drain of the field effect transistor, the emitter of the bipolar transistor is replaced by the source of the field effect transistor, and the base of the bipolar transistor is replaced by the gate of the field effect transistor.

DESCRIPTION OF SYMBOLS 1 ... 1st amplifier, 2 ... Inverting amplifier, 3 ... 3rd amplifier, 4 ... 4th amplifier, 5 ... 1st vibrator, 6 ... 2nd vibrator, 7 ... 1st negative capacitance circuit, 8 ... 2nd negative capacitance circuit, 9 ... 3rd negative capacitance circuit, 11 ... Transistor, 12 ... Resistance, 13 ... Resistance, 14 ... capacitor, 15 ... capacitor, 21 ... second amplifier, 22 ... resistor, 23 ... resistor, 24 ... capacitor, 25 ... resistor, 26 ... resistor, 31 ... transistor 32 ... variable resistor 33 ... variable resistor 34 ... resistor 41 ... transistor 42 ... variable resistor 43 ... variable resistor 44 ... Resistance 51 equivalent parallel capacitor 52 equivalent series resistance 53 equivalent series inductor 54 equivalent series Capacitor 61 ... Equivalent parallel capacitor 62 ... Equivalent series resistance 63 ... Equivalent series inductor 64 ... Equivalent series capacitor 71 ... Operational amplifier 72 ... Resistance 73 ... Resistor, 74 ... capacitor, 91 ... operational amplifier, 92 ... capacitor, 93 ... resistor, 94 ... resistor, 100 ... oscillator circuit, 200 ... oscillator circuit, 224 ... Capacitor 300 ... Oscillator circuit 400 ... Anti-resonant circuit 411 ... Vibrator 412 ... Attenuator 413 ... Inductor 414 ... Capacitor 421 ... Vibrator 422 ... Attenuator, 423 ... Inductor, 424 ... Capacitor, 430 ... AC signal source, 440 ... Output resistance, 450 ... Load resistance

Claims (6)

A first amplifier for amplifying an input signal;
A second amplifier that inverts and amplifies the output signal of the first amplifier and inputs it to the first amplifier;
A resistor connected between the input and output of the second amplifier;
A first vibrator and a third amplifier connected in series with each other in parallel with the resistor;
A second vibrator and a fourth amplifier connected in series with each other in parallel with the resistor;
With
The first vibrator has an anti-resonance frequency different from that of the second vibrator;
At least one of the third amplifier and the fourth amplifier has a variable gain;
Oscillator circuit. The resonance frequency of the first vibrator is smaller than the resonance frequency of the second vibrator,
The anti-resonance frequency of the first vibrator is greater than the resonance frequency of the second vibrator;
The oscillation circuit according to claim 1. The resonance frequency of the first vibrator is smaller than the resonance frequency of the second vibrator,
A first negative capacitance circuit provided in parallel with the first vibrator and having a capacitance opposite to the equivalent parallel capacitance of the first vibrator;
The oscillation circuit according to claim 1. The resonance frequency and anti-resonance frequency of the first vibrator are smaller than the resonance frequency and anti-resonance frequency of the second vibrator,
The oscillation circuit according to claim 3. The first negative capacitance circuit includes an inverting amplifier connected to one terminal of the first vibrator, and a first capacitor connected to the other terminal of the first vibrator and the inverting amplifier. Have
The oscillation circuit according to claim 3 or 4. A third negative capacitance circuit provided between the input terminal of the second amplifier and the ground;
The oscillation circuit according to any one of claims 3 to 5.

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