JP2013192436A - Power transmission device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To solve the problem that increase in the number of components or the like may result from providing a snubber circuit to absorb a surge voltage caused by a recovery current.SOLUTION: A voltage of an input capacitor 10 is converted into AC voltage through a full bridge circuit FB and applied to a primary-side coil w1. A voltage induced in a secondary-side coil w2 in accordance with the application is applied through a full-wave rectifier circuit RC to an output capacitor Co. An inductor 12 is provided between the secondary-side coil w2 and the full-wave rectifier circuit RC. Furthermore, the full-wave rectifier circuit RC and the output capacitor Co are short-circuited. The timing to invert a polarity of the voltage to be applied to the primary-side coil w1 is after turning a current flowing in the inductor 12 to zero.

Description

  The present invention relates to a power transmission device that transmits power using a transformer.

  As this type of power transmission device, for example, the one shown in Patent Document 1 below has been proposed. This is because a series connection body of a pair of diodes is used as an output capacitor in order to apply a voltage of a DC voltage source to a primary coil of a transformer via a bridge circuit and convert an AC voltage generated in the secondary coil into a DC voltage. Are provided with a snubber capacitor for absorbing a surge caused by the recovery current of the diode. The charging energy of the snubber capacitor is discharged to the output capacitor via a snubber inductor that stores the discharging energy of the snubber capacitor.

JP-A-1-295675

  In the above apparatus, a snubber circuit is provided to absorb the surge voltage caused by the recovery current, which increases the number of components.

  The present invention has been made in the process of solving the above-described problems, and an object thereof is to provide a new power transmission apparatus that transmits power using a transformer.

  Hereinafter, means for solving the above-described problems and the operation and effect thereof will be described.

  The invention according to claim 1 is an AC voltage applying means (FB) for applying an AC voltage to the primary side coil of the transformer (T), and a rectifying means (RC) for rectifying the output of the secondary side coil of the transformer. Control that is connected to the secondary side of the transformer and variably controls the output voltage by operating an output capacitor (Co) that determines the magnitude of the output voltage by a charging voltage and the AC voltage applying means Means (20), and the voltage is applied by electronic operation of the AC voltage applying means from a state in which a voltage that gradually increases the current flowing between the AC voltage applying means and the rectifying means is applied by the AC voltage applying means. When the application of current is stopped, an inductor is provided between the AC voltage applying means and the rectifying means so that a closed loop path for applying a voltage for gradually decreasing the current is configured. When the control setting is performed and the control by the control means is performed, the inductance between the output capacitor and the rectifying means is equal to the inductance between the AC voltage applying means and the rectifying means. It is less than or equal to a value obtained by multiplying a value obtained by dividing the voltage of the output capacitor by the voltage induced in the secondary coil.

  In the above invention, the current can be kept flowing gradually through the rectifying means by the inductance setting in a period in which the application of the voltage for gradually increasing the current is stopped before the polarity of the voltage for gradually increasing the current is reversed. . As a result, if the current flowing through the rectifying unit becomes zero before the reverse polarity voltage is applied to the rectifying unit, a situation in which a recovery current is generated can be avoided.

1 is a system configuration diagram according to a first embodiment. FIG. The figure which shows the structure of the trans | transformer concerning the embodiment. The time chart which shows the switching operation method concerning the embodiment. The circuit diagram which shows the power conversion process concerning the embodiment. The circuit diagram which shows the power conversion process concerning the embodiment. The time chart which shows the switching operation method concerning 2nd Embodiment. The figure which shows the relationship between the output electric power and duty ratio concerning the embodiment. The figure which shows the relationship between the inductance of an inductor, and a duty ratio in the same output voltage. The time chart used in order to explain quantitatively the process concerning the said embodiment. The figure which shows formulas, such as an output voltage concerning each said embodiment.

<First Embodiment>
Hereinafter, a first embodiment in which a power transmission device according to the present invention is applied to an in-vehicle charging device will be described with reference to the drawings.

  The converter CNV shown in FIG. 1 is mounted on an in-vehicle charging device. That is, for the input side capacitor 10 serving as an output unit of a device (not shown) that uses electric power from a commercial power source supplied from the outside of the vehicle as a predetermined DC voltage, the voltage is converted into the voltage of the output capacitor Co. It is. Specifically, the converter CNV is an insulating converter, and is configured to include a transformer T.

  The primary side of the transformer T includes a series connection body of the switching element S1 and the switching element S2 connected in parallel between both electrodes of the input side capacitor 10, and a series connection body of the switching element S3 and the switching element S4. Yes. The switching elements S1 to S4 are AC voltage application means (full bridge circuit FB) that applies an AC voltage to the primary coil w1 of the transformer T. In the present embodiment, N-channel MOS field effect transistors are illustrated as the switching elements S1 to S4. In the figure, each of the diodes D1 to D4 is a body diode of each of the switching elements S1 to S4.

  A primary coil w1 of the transformer T is connected between a connection point between the switching element S1 and the switching element S2 and a connection point between the switching element S3 and the switching element S4.

  In parallel to the output capacitor Co, a series connection body of a diode Da and a diode Db and a series connection body of a diode Dc and a diode Dd are short-circuited. These diodes Da to Dd are rectification means (full-wave rectification circuit RC) for converting the AC voltage output from the secondary coil w2 into a DC voltage and outputting it.

  A secondary coil w2 of the transformer T is connected between a connection point between the diode Da and the diode Db and a connection point between the diode Dc and the diode Dd. In the figure, in addition to the secondary coil w2, an inductor l2 is shown between the pair of connection points. As shown in FIG. 2, this is an equivalent circuit representation of a leakage inductor generated by increasing the gap between the magnetic core 11 of the transformer T and the primary side coil w1 and the secondary side coil w2.

  A load 14 is connected in parallel to the output capacitor Co. The load 14 includes a battery as energy storage means of a rotating machine as a vehicle-mounted main machine.

  The control device 20 converts the voltage of the input side capacitor 10 and applies it to the output capacitor Co by outputting the operation signals ms1 to ms4 to the switching elements S1 to S4. Here, as shown in FIG. 3, the operation signals m1 and m4 and the operation signals m2 and m3 are the same as each other, and the operation signals m1 and m4 and the operation signals m2 and m3 are turned on. The phase is shifted by π.

  Hereinafter, with reference to FIG. 4 and FIG. 5, a qualitative explanation will be given regarding the power conversion operation by the operation of the converter CNV by the operation signals ms1 to ms4. In the following description, the definition shown in FIG. 1 is used for the polarity of the voltage of the primary coil w1, the polarity of the voltage of the secondary coil w2, and the like.

  FIG. 4A shows a case where the switching elements S1 and S4 are switched from the off state to the on state. As illustrated, in this case, the applied voltage Vw1 of the primary coil w1 has an absolute value that is the input voltage Vi and is negative. Then, a current flows through a closed loop path including the input side capacitor 10, the switching element S1, the primary side coil w1, and the switching element S4. On the other hand, the applied voltage Vw2 of the secondary side coil w2 is a voltage “−n” having a magnitude corresponding to the ratio (turn ratio n) between the number of turns N1 of the primary side coil w1 and the number of turns N2 of the secondary side coil w2.・ Vi ”. Thereby, a current flows through a path including the secondary coil w2, the diode Dc, the output capacitor Co, and the diode Db. Here, in this embodiment, since it is assumed that the voltage (n · Vin) induced in the secondary coil w2 is larger than the output voltage Vo, this current gradually increases.

  FIG. 4B shows a case where the switching elements S1 and S4 are switched to the off state. In this case, in order to release the energy stored in the inductor 12, a current flows through a closed loop path including the inductor 12, the secondary coil w2, the diode Dc, the output capacitor Co, and the diode Db. Since a current flows through the secondary coil w2, a current also flows through a closed loop path including the primary coil w1, the diode D3, the input capacitor 10, and the diode D2 in the primary coil w1. However, this current gradually decreases. When the current flows through the path, the applied voltage Vw1 of the primary coil w1 becomes “−Vi” having a polarity opposite to that of FIG. 4A. As a result, the applied voltage Vw2 of the secondary coil w2 (voltage induced in the secondary coil w2) becomes “n · Vi” having the opposite polarity to that of FIG. For this reason, the voltage VL applied to the inductor 12 becomes “−n · Vi−Vo”, and the current flowing through the inductor 12 gradually decreases according to the absolute value of the value obtained by dividing the voltage VL by the inductance of the inductor 12. Incidentally, at this time, the magnitude of the current flowing through the inductor 12, the magnitude of the current flowing through the electrical path connecting the full-wave rectifier circuit RC and the output capacitor Co, and the magnitude of the forward current flowing through the diodes Db and Dc are as follows. Always match. This is because the electrical path connecting the full-wave rectifier circuit RC and the output capacitor Co does not have a branch path, and the inductance of this electrical path is ideally zero. Incidentally, when the inductance of the electric path is large, the potential on the cathode side of the diodes Da and Dc tends to be lower than the potential on the anode side of the diodes Db and Dd.

  FIG. 4C shows a state in which the current flowing through the primary side coil w1, the secondary side coil w2, and the inductor 12 is gradually reduced to zero.

  FIG. 5A shows a state in which the switching elements S2 and S3 are switched to the on state. By switching the switching elements S2 and S3 to the on state, the applied voltage Vw1 of the primary coil w1 has an absolute value that is the input voltage Vi and a positive voltage. Then, a current flows through a closed loop path including the input side capacitor 10, the switching element S3, the primary side coil w1, and the switching element S2. On the other hand, the applied voltage Vw2 of the secondary coil w2 is a voltage “n · Vi” having a magnitude corresponding to the turn ratio n. Thereby, a current flows through a path including the secondary coil w2, the diode Da, the output capacitor Co, and the diode Dd.

  In the present embodiment, the recovery current does not flow through the diodes Dc and Db when the switching elements S2 and S3 are turned on. This is because, as shown in FIG. 4C, after the currents of the diodes Dc and Db become zero, the current flows in the path including the secondary coil w2, the diode Da, the output capacitor Co, and the diode Dd. This is because it flows. For this reason, the reverse bias is not applied while the forward current flows through the diodes Dc and Db.

  FIG. 5B shows a case where the switching elements S2 and S3 are switched to the off state. In this case, in order to release the energy stored in the inductor 12, a current flows through a closed loop path including the secondary coil w2, the inductor 12, the diode Da, the output capacitor Co, and the diode Dd. Since a current flows through the secondary coil w2, a current also flows through a closed loop path including the primary coil w1, the diode D1, the input capacitor 10, and the diode D4 in the primary coil w1. However, this current gradually decreases. As a current flows through the path, the applied voltage Vw1 of the primary coil w1 becomes “−Vi” having a polarity opposite to that of FIG. As a result, the applied voltage Vw2 of the secondary coil w2 becomes “−n · Vi” having a polarity opposite to that of FIG. For this reason, the voltage VL applied to the inductor 12 becomes “n · Vi + Vo”, and the current flowing through the inductor 12 gradually decreases according to a value obtained by dividing the voltage VL by the inductance of the inductor 12.

  FIG. 5C shows a state in which the current flowing through the primary side coil w1, the secondary side coil w2, and the inductor 12 is gradually reduced to zero. After this state is reached, the state shifts to the state shown in FIG.

  According to this embodiment in which the processing of the above aspect is performed, the surge voltage can be suitably reduced when switching the switching states of the switching elements S1 to S4. This is made possible by the following settings. The first setting is when the inductor 12 is provided between the full-bridge circuit FB and the full-wave rectifier circuit RC, and the application of the voltage that gradually increases the current flowing through the inductor 12 is stopped by the electronic operation of the full-bridge circuit FB. A closed loop path (FIGS. 4B and 5B) for applying a gradually decreasing voltage is configured. The second setting is to short-circuit between the full-wave rectifier circuit RC and the output capacitor Co.

  That is, the forward current flowing through the diodes Db and Dc (Da and Dd) can be gradually reduced by the first setting. And, in the process of gradually decreasing and before the forward current becomes zero yet, the forward current does not start to flow through the diodes Da, Dd (Db, Dc). This is because of the second setting. That is, in this case, no voltage is induced between the full-wave rectifier circuit RC and the output capacitor Co when the forward current flowing through the diodes Db, Dc (Da, Dd) is gradually decreasing. For this reason, the differential pressure between the potential of the terminal of the full-wave rectifier circuit RC and the potential of the terminal of the output capacitor Co connected thereto can be sufficiently reduced. As a result, the potential on the cathode side of the diodes Da and Dc (positive terminal side of the full-wave rectifier circuit RC) is lower than the potential on the anode side of the diodes Db and Dd (negative terminal side of the full-wave rectifier circuit RC). Never become. For this reason, the forward current does not start to flow through the diodes Da and Dd (Db and Dc) in the process in which the forward current flowing through the diodes Db and Dc (Da and Dd) gradually decreases. Therefore, when a forward current flows through the diodes Db and Dc (Da and Dd), a reverse bias is not applied to them, and hence no recovery current flows through them.

  Further, when switching elements S1, S4 (S2, S3) are switched to the OFF state, since inductor 12 is provided, a situation in which the current flowing through primary side coil w1 and secondary side coil w2 suddenly becomes zero is avoided. be able to. That is, after the switching elements S1, S4 (S2, S3) are switched to the OFF state, the decreasing rate of the current flowing through the primary side coil w1, the secondary side coil w2, and the diodes Db, Dc (Da, Dd) The absolute value of the value obtained by subtracting the voltage of the capacitor Co and the voltage of the secondary coil w2 by the inductance of the inductor 12 is limited.

  Since the inductor 12 is provided, when the switching elements S1, S4 (S2, S3) are switched to the on state, the increasing speed of the current flowing through them can be limited. Switching loss can also be reduced.

  As shown in FIG. 1, in this embodiment, the time ratio D of the ON time with respect to the ON / OFF cycle T of the switching elements S1, S4 (S2, S3) is set as the first output voltage Vo of the converter CNV. The operation amount. Here, the duty ratio D is variably set according to the output current Io of the converter CNV in addition to the charging voltage (input voltage Vi) of the input side capacitor 10 and the output voltage Vo. Specifically, even when the input voltage Vi and the output voltage Vo are the same, the duty ratio D is set to a larger value as the output current Io increases. Thus, when the input voltage Vi and the output voltage Vo are the same, the duty ratio D depends on the output current Io is one of the characteristics of the converter CNV according to the present embodiment.

  Furthermore, in this embodiment, the switching frequency f is the second manipulated variable of the output voltage Vo. A quantitative explanation that the output voltage Vo depends on the switching frequency f and the output current Io is given in the remarks column at the end of this specification.

  According to the embodiment described above, the following effects can be obtained.

  (1) The inductor 12 is provided between the full bridge circuit FB and the full-wave rectifier circuit RC, and the output capacitor Co is short-circuited to the full-wave rectifier circuit RC. Thereby, the magnitude of the current flowing between the full-wave rectifier circuit RC and the output capacitor Co, the magnitude of the current flowing through the inductor 12, and the magnitude of the current flowing through the diodes Db and Dc (Da, Dd) are matched. Can do. For this reason, the situation where the recovery current flows through the diodes Db and Dc (Da, Dd) can be preferably avoided, and the surge voltage caused by the recovery current can be avoided.

  (2) By short-circuiting the output capacitor Co to the full-wave rectifier circuit RC, the potential of the cathodes of the diodes Da and Dc (the positive side terminal of the full-wave rectifier circuit RC) becomes the anode of the diodes Db and Dc (full-wave rectifier). The situation where the potential becomes lower than the potential of the negative terminal of the circuit RC) can be surely avoided. As a result, the situation where the reverse bias is applied during the period in which the forward current flows through the diodes Db and Dc (Da, Dd). It can be suitably avoided.

  (3) A voltage is applied to the primary coil w1 by turning on the switching elements S1, S4 (S2, S3) after the time when the current flowing through the secondary coil w2 gradually decreases to zero. Thereby, it becomes easy to improve the controllability of the output voltage Vout. As can be seen from the derivation of the relational expression between the output voltage Vout and the duty ratio D described in the “Remarks” column at the end of the present specification, the current is caused by the ON operation of the switching elements S1, S4 (S2, S3). This is because it is possible to easily derive a relational expression between the time ratio D and the output voltage Vout by assuming that the voltage gradually increases immediately. On the other hand, when the switching elements S1, S4 (S2, S3) are turned on before the time point when the current flowing through the secondary coil w2 gradually decreases to zero, FIG. 4 (b) and FIG. During the period until the current shown in b) becomes zero, the current gradually decreases.

  (4) When setting the duty ratio D, the output current Io was referred to. Thereby, the controllability of the output voltage Vo can be improved.

  (5) The switching frequency f was manipulated to control the output voltage Vout. Thereby, the output voltage Vout can be controlled without depending on the duty ratio D.

  (6) Both the duty ratio D and the switching frequency f are employed as the manipulated variable of the output voltage Vout. Thereby, the freedom degree of control can be improved.

(7) The inductor 12 is constituted by a leakage inductor of the transformer T. Thereby, the increase in the circuit scale accompanying the inductance setting between full bridge circuit FB and full wave rectifier circuit RC can be suppressed suitably.
<Second Embodiment>
Hereinafter, the second embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 6 shows operation signals m1 to m4 according to the present embodiment. As illustrated, in the present embodiment, the switching timing to the ON state and the switching timing to the OFF state are shifted from each other by the switching element S1 (S2) and the switching element S4 (S3). This is for performing phase shift processing. By using the phase shift process, the switching loss at the time of switching the switching elements S1 to S4 to the ON state can be further reduced.

  Here, the phase shift processing according to the present embodiment is realized using the inductor 12. That is, in the phase shift process, for example, instead of the state shown in FIG. 4B, the switching element S1 is maintained in the on state, so that the loop including the primary coil w1, the diode D3, and the switching element S1. Current flows through the path. Here, the current flows in this loop path because the inductor 12 is provided so that the current continues to flow in the secondary coil w2.

FIG. 7 shows the relationship between the output power and the duty ratio D when the output voltage Vo is “260 V” and “410 V” in this embodiment. FIG. 8 shows the relationship between the inductance of the inductor 12 and the duty ratio D when the output voltage Vo is “410 V” and the output power is “3.3 kW”. As shown in the figure, when the inductance of the inductor 12 is about 15 μH and the actual inductance has an error of “± 30%”, the duty ratio D varies by “± 10%”. For this reason, even if an error of “± 30%” occurs in the inductance by setting the assumed usage region of the duty ratio D in the case where there is no error in the inductance to “20 to 80%”, for example, the output voltage Vo Controllability can be maintained.
<Other embodiments>
Each of the above embodiments may be modified as follows.

"About control means"
In the above embodiment, the switching elements S1, S4 (S2, S3) are switched to the ON state in order to perform the process of gradually increasing again after the absolute value of the current iL flowing through the inductor 12 gradually decreases to zero. However, it is not limited to this. For example, it may be switched to the on state in synchronization with the timing when it becomes zero.

  However, it is also possible to switch to the on state before it becomes zero. In this case, zero volt switching can be performed when switching the switching state to the on state. At this time, in order to easily improve the controllability of the output voltage Vout, it is desirable to switch to the ON state after it becomes zero in most of the variable region of the duty ratio D (for example, 80%). .

About AC voltage application means
It is not limited to the full bridge circuit FB. For example, the series connection body of the switching elements S3 and S4 may be changed to a series connection body of a pair of capacitors.

"About rectification means"
For example, a half-wave rectifier circuit or the like may be used instead of the full-wave rectifier circuit RC.

“Inductance setting”
In the above embodiment, the inductor 12 is configured as a leakage inductor of the transformer T. However, the present invention is not limited to this, and an inductance element may be used. At this time, the inductance element may be disposed between the secondary coil w2 and the full-wave rectifier circuit RC. Alternatively, it may be arranged between the full bridge circuit FB and the primary coil w1.

About duty ratio operation means
In the above embodiment, the manipulated variable of the output voltage Vo is both the duty ratio D and the switching frequency f. However, the present invention is not limited to this, and for example, only the switching frequency f may be used. That is, the duty ratio operation means is not essential.

“Frequency control means”
In the above embodiment, the manipulated variable of the output voltage Vo is both the duty ratio D and the switching frequency f. However, the present invention is not limited to this. For example, only the duty ratio D may be used. That is, the frequency operation means is not essential.

"Connection between full-wave rectifier circuit RC and output capacitor Co"
A smoothing coil may be provided between the positive terminal (the cathodes of the diodes Da and Dc) of the full-wave rectifier circuit RC and the output capacitor Co. Even in this case, depending on the setting of the inductance of the smoothing coil, the currents of the inductor 12 and the smoothing coil can be the same amount and the gradual reduction speed can be made the same. Can be suitably avoided, and as a result, the surge voltage can be suitably reduced. Here, it is desirable that the smoothing coil inductance Lo is set to satisfy “Vo−VLo ≧ 0” using the smoothing coil voltage VLo (the output capacitor side is positive). If this condition is not satisfied, a forward current may flow through the diode Da (Dc) when the current flowing through the diode Dc (Da) is not yet zero. This condition can be regarded as a condition that “Lo ≦ (Ls · Vo) / (n · Vin)” is satisfied in the range of values that the voltage Vi of the input capacitor Ci and the voltage Vo of the output capacitor Co can take. .

"Application of converter CNV"
The power conversion circuit for charging the on-vehicle high voltage battery with the electric power of the commercial power source outside the vehicle is not limited to the configuration. In particular, it is effective to apply the converter according to the present invention that can reduce a surge or the like with a simple configuration in a situation where it is desirable to use an insulating converter.
<Remarks>
Here, first, for the case of the first embodiment (in the case of hard switching), the relationship between the output voltage Vo and the duty ratio D is derived.

  FIG. 9 shows transitions of the operation signals ms1, ms2, ms3, ms4, the applied voltage Vw1 of the primary coil w1, the current iL of the inductor 12, and the voltage VL applied to the inductor 12 during hard switching. The applied voltage Vw1 of the primary coil w1, the current iL of the inductor 12, and the polarity of the voltage VL applied to the inductor 12 are in accordance with the definitions in FIG.

  In a period in which the operation signals ms2 and ms3 are on-operation commands (period of FIG. 5A), if the turn ratio n between the primary side coil w1 and the secondary side coil w2 is used, the secondary side coil w2 The applied voltage Vw2 becomes “n · Vi”. Therefore, the voltage VL applied to the inductor 12 is “−n · Vi + Vo” due to the relationship “n · Vi = −VL + Vo”. Therefore, the current iL (> 0) flowing through the inductor 12 gradually increases according to the speed of a value obtained by dividing “n · Vi−Vo (> 0)” by the inductance L of the inductor 12. Here, the peak value iPeak of the current iL flowing through the inductor 12 is expressed by the following equation (c1) using the duty ratio D and the period T.

iPeak = (n · Vi−Vo) · D · T / L (c1)
On the other hand, the sign of the voltage Vw2 of the secondary coil w2 is inverted during the period when the current flows through the inductor 12 in the period in which the operation signals ms1 to ms4 are the off operation command (period of FIG. 5B). Therefore, the voltage VL applied to the inductor 12 becomes “n · Vi + Vo” from “−n · Vi = −VL + Vo”. Here, the time a required until the current iL flowing through the inductor 12 becomes zero satisfies the following expression (c2).

n · Vi + Vo = L · iPeak / a (c2)
When the above equation (c2) is solved for time a, the following equation (c3) is obtained.

a = (n * Vi-Vo) * DT * (n * Vi + Vo) (c3)
Here, in the steady state, since the average value of the current output to the output capacitor Co during the period T matches the output current Io of the converter CNV, the following equation (c4) is established.

iPeak · (D · T + a) = Io · T (c4)
By substituting the above equation (c1) into the above equation (c4) and solving for the time ratio D or solving for the output voltage Vo, the equation relating to hard switching shown in FIG. 10 can be derived. As shown in FIG. 10, the output voltage Vo depends not only on the input voltage Vi and the duty ratio D but also on the output current Io and the switching frequency f.

  In the case of the phase shift, since the applied voltage Vw2 of the secondary coil w2 becomes zero during the period in which the absolute value of the current iL of the inductor 12 gradually decreases, the voltage VL applied to the inductor 12 is the output voltage. It can be calculated in the same manner as described above except that it becomes Vo. This is also described in FIG.

  DESCRIPTION OF SYMBOLS 10 ... Input capacitor, 12 ... Inductor, T ... Transformer, Co ... Output capacitor, RC ... Full wave rectifier circuit, FB ... Full bridge circuit.

Claims (7)

AC voltage application means (FB) for applying an AC voltage to the primary coil of the transformer (T);
Rectifying means (RC) for rectifying the output of the secondary coil of the transformer;
An output capacitor (Co) connected to the secondary side of the transformer and determining the magnitude of the output voltage by a charging voltage;
Control means (20) for variably controlling the output voltage by operating the AC voltage applying means;
With
When the application of the voltage is stopped by the electronic operation of the AC voltage application means from the state where the voltage gradually increasing the current flowing between the AC voltage application means and the rectification means is applied by the AC voltage application means, Inductance setting (12) is performed between the AC voltage applying means and the rectifying means so that a closed loop path for applying a voltage for gradually decreasing the current is configured.
When controlled by the control means, the inductance between the output capacitor and the rectifying means is the inductance between the AC voltage applying means and the rectifying means due to the voltage induced in the secondary coil. A power transmission device having a value equal to or less than a value obtained by multiplying a value obtained by dividing a voltage of an output capacitor.   The power transmission device according to claim 1, wherein the output capacitor is short-circuited to the rectifier.   The control means sets the voltage for gradually increasing the current flowing through the secondary coil after the time when the current flowing through the secondary coil gradually decreases to zero due to the formation of the closed loop path. The power transmission device according to claim 1 or 2, wherein the AC voltage applying means is operated so as to be applied to the secondary coil.   The gradual decrease rate of the current when the closed loop path for applying the gradually decreasing voltage is formed is inversely proportional to the inductance between the AC voltage applying unit and the rectifying unit set by the inductance setting. The power transmission device according to claim 1. The control means operates a time ratio of a period in which a voltage for gradually increasing the current by the AC voltage application means is applied to one period between a period in which the voltage is applied to the primary coil and a period in which the voltage is not applied. Comprising a time ratio operation means for controlling the voltage of the output capacitor;
5. The electric power according to claim 1, wherein the duty ratio operation unit increases the duty ratio as the output current increases even when the output voltage is the same. 6. Transmission equipment.   The control means outputs an output as a charging voltage of the output capacitor when the voltage for gradually increasing the current by the AC voltage applying means is periodically switched between a period in which the voltage is applied to the primary coil and a period in which the voltage is not applied. The power transmission according to any one of claims 1 to 5, further comprising frequency operation means for operating a total length of the applied period and the non-applied period in order to control a voltage. apparatus.   The power transmission device according to claim 1, wherein the inductance setting is realized by using a leakage inductance of the transformer.

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