JP2012223059A - Motor controller and electric power steering device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a motor controller capable of effectively suppressing a motor current without degrading stability of motor control.SOLUTION: A current command value restriction section restricts a γ-axis current command value to be not more than a current command upper limit value Ilim calculated by a current command upper limit value calculation section. A switching control section provided at the current command upper limit value calculation section estimates ("sinθL" that is a sine component) of a load angle (error angle) θL indicating a deviation between a control angle as a virtual motor rotational angle in its control and an actual rotational angle and determines a motor control state based on the load angle θL. In addition, the switching control section, when its control state is determined to be unstable, determines that the current command upper limit value Ilim shall be changed to a higher value (Ilim_b) than a value (Ilim_a) when the control state is stable.

Description

  The present invention relates to a motor control device and an electric power steering device.

  2. Description of the Related Art Conventionally, some electric power steering devices (EPS) that apply assist force to a steering system using a motor as a drive source include a motor control device that can control a brushless motor without detecting the motor rotation angle. As an aspect of sensorless (resolverless) drive control that does not use such a rotation angle sensor (motor resolver), an addition angle corresponding to the motor rotation angle change amount for each calculation cycle is calculated, and the addition angles are integrated. A method of executing current feedback control in a rotating coordinate system according to the motor rotation angle on the control obtained by the above is proposed.

  For example, the motor control device described in Patent Document 1 calculates an addition angle corresponding to the motor rotation angle change amount for each calculation cycle based on the deviation between the target torque to be generated by the motor and the actual torque. . Further, the motor control device described in Patent Document 2 estimates the motor rotation angular velocity based on the motor current and the motor voltage. Then, the addition angle is calculated using the motor rotation angular velocity as a change component for each calculation cycle.

  In other words, even if the actual motor rotation angle (actual rotation angle) and the control motor rotation angle (control angle) do not exactly match, the brushless motor can be controlled as long as the deviation remains within a certain range. is there. Then, the difference between the actual rotation angle and the control angle is calculated by executing the current feedback control using the control angle obtained by calculating the addition angle by the method described in each of the above patent documents and integrating the addition angles. Can be kept within the range in which the motor can be controlled.

JP 2010-11709 A JP 2010-29031 A

  In the resolverless control using the virtual control angle as described above, as long as the stator can generate the magnetomotive force necessary to maintain the rotational position of the rotor, the control angle and the actual rotation angle are It is possible to keep the deviation in a range that can be stably controlled. That is, by generating a larger motor current, the stability of the motor control can be improved.

  However, by continuously energizing a motor with a large current, its energy efficiency decreases. And since the reliability of the motor may decrease due to the remarkable heat generation of the motor, the improvement has been left as a problem to be solved.

  The present invention has been made to solve the above-described problems, and an object of the present invention is to provide a motor control device and an electric power that can effectively suppress the motor current without impairing the stability of the motor control. The object is to provide a steering device.

  In order to solve the above problems, the invention according to claim 1 is a motor control signal output means for outputting a motor control signal, and a drive circuit for supplying three-phase drive power to the motor based on the motor control signal. And the motor control signal output means performs a torque feedback control based on a torque deviation between a target torque to be generated by the motor and an actual torque, thereby obtaining a motor rotation angle change amount for each calculation cycle. The corresponding addition angle is calculated, and the addition angle is integrated to calculate the motor rotation angle for control. At each calculation cycle, the increase / decrease value based on the torque deviation is calculated and the increase / decrease value is integrated. By executing the current feedback control in the rotating coordinate system according to the motor rotation angle on the control while calculating the current command value, the motor control signal In the motor control device that outputs, the motor control signal output means limits the current command value to an upper limit value or less, and sets a load angle indicating a deviation between the motor rotation angle and the actual rotation angle on the control. And estimating the control state of the motor based on the estimated load angle, and when the control state is in an unstable state, the upper limit value is set more than when the control state is stable. To make it higher.

  That is, the motor (brushless motor) can be stably controlled as long as the deviation between the virtual motor rotation angle (control angle) and the actual rotation angle in control remains within a certain range. Therefore, when the load angle indicating the deviation is in the boundary region of the range in which the motor can be stably controlled, it can be determined that the control state is in an unstable state that tends to be unstable. And when it is determined that the control state is in an unstable state as in the above configuration, the upper limit value of the current command value is changed to a high value that allows energization of a large motor current. The stability of motor control can be improved. As a result, the upper limit value of the current command value when the control state is stable can be set to a low value in advance, and as a result, the motor current can be effectively reduced without impairing the stability of the motor control. It becomes possible to suppress.

The gist of the invention described in claim 2 is an electric power steering apparatus provided with the motor control device described in claim 1.
According to the above configuration, it is possible to effectively suppress the motor current while avoiding instability of the motor control and maintaining a good steering feeling.

  ADVANTAGE OF THE INVENTION According to this invention, the motor control apparatus and electric power steering apparatus which can suppress a motor current effectively, without impairing the stability of motor control can be provided.

The schematic block diagram of an electric power steering device (EPS). The block diagram which shows the electric constitution of EPS. The schematic block diagram of a 1st control part. The schematic block diagram of a 2nd control part. The block diagram which shows schematic structure of a disturbance observer. The flowchart which shows the process sequence of rotation angular velocity estimation. The flowchart which shows the process sequence of addition angle adjustment calculation. The schematic block diagram of the electric current command value calculating part by the side of a 2nd control part. The schematic block diagram of a current command upper limit calculating part. Explanatory drawing which shows the difference (load angle) of a control angle and an actual motor rotation angle, and the relationship of each axis current value. The flowchart which shows the process sequence at the time of determining an electric current command upper limit. The flowchart which shows the process sequence at the time of determining the electric current command upper limit of another example.

DESCRIPTION OF EXEMPLARY EMBODIMENTS Hereinafter, an embodiment of the invention will be described with reference to the drawings.
As shown in FIG. 1, in the electric power steering apparatus (EPS) 1 of this embodiment, a steering shaft 3 to which a steering 2 is fixed is connected to a rack shaft 5 via a rack and pinion mechanism 4. The rotation of the steering shaft 3 accompanying the steering operation is converted into a reciprocating linear motion of the rack shaft 5 by the rack and pinion mechanism 4. The steering shaft 3 of this embodiment is formed by connecting a column shaft 3a, an intermediate shaft 3b, and a pinion shaft 3c. Then, the reciprocating linear motion of the rack shaft 5 accompanying the rotation of the steering shaft 3 is transmitted to a knuckle (not shown) via tie rods 6 connected to both ends of the rack shaft 5, whereby the steering angle of the steered wheels 7. That is, the traveling direction of the vehicle is changed.

  Further, the EPS 1 includes an EPS actuator 10 as a steering force assisting device that applies an assist force for assisting a steering operation to the steering system, and an ECU 11 as a control unit that controls the operation of the EPS actuator 10. .

  The EPS actuator 10 of the present embodiment is configured as a so-called column-type EPS actuator in which a motor 12 that is a drive source is drivingly connected to a column shaft 3 a via a speed reduction mechanism 13. In the present embodiment, the motor 12 employs a brushless motor that rotates based on three-phase (U, V, W) driving power. The EPS actuator 10 is configured to apply an assist force based on the motor torque to the steering system by decelerating the rotation of the motor 12 and transmitting it to the column shaft 3a.

  On the other hand, a torque sensor 14 is connected to the ECU 11, and the ECU 11 detects a steering torque τ transmitted to the steering shaft 3 based on an output signal of the torque sensor 14. Further, the vehicle speed V detected by the vehicle speed sensor 15 and the steering angle θs detected by the steering sensor (steering angle sensor) 16 are input to the ECU 11 of the present embodiment. The ECU 11 calculates a target assist force to be applied to the steering system based on each of these state quantities, and drives the motor 12 by supplying drive power to generate a corresponding motor torque. The operation of the EPS actuator 10 as a source, that is, the assist force applied to the steering system is controlled (power assist control).

Next, the electrical configuration of the EPS of this embodiment will be described.
FIG. 2 is a control block diagram of the EPS of this embodiment. As shown in the figure, the ECU 11 supplies three-phase drive power to the motor 12 based on the microcomputer 17 serving as motor control signal output means for outputting a motor control signal and the motor control signal output from the microcomputer 17. And a drive circuit 18.

  Each control block shown below is realized by a computer program executed by the microcomputer 17. The microcomputer 17 detects each state quantity at a predetermined sampling period, and generates a motor control signal by executing each arithmetic processing shown in the following control blocks at every predetermined period.

  Specifically, in the drive circuit 18 of the present embodiment, three switching arms corresponding to the respective phase motor coils 12u, 12v, 12w are arranged in parallel with a pair of switching elements connected in series as a basic unit (switching arm). A well-known PWM inverter connected to is used. That is, the motor control signal output from the microcomputer 17 defines the on / off state (duty of each phase switching arm) of each phase switching element constituting this drive circuit. The drive circuit 18 is activated by the input of the motor control signal and supplies three-phase drive power based on the applied power supply voltage V_pig to the motor.

  More specifically, the ECU 11 is provided with a current sensor 21 for detecting each phase current value Iu, Iv, Iw of the motor 12. Note that the current sensor 21 of the present embodiment has a well-known configuration in which a shunt resistor is connected to the low potential side (ground side) of each switching arm constituting the drive circuit 18. The microcomputer 17 of the present embodiment detects the phase current values Iu, Iv, and Iw flowing through the phase motor coils 12u, 12v, and 12w based on the output signal of the current sensor 21 (the voltage across the terminals of the shunt resistor). To do.

  Further, the microcomputer 17 of the present embodiment detects the rotation angle (electrical angle) θm of the motor 12 based on the output signal of the motor resolver 23. In the present embodiment, the motor resolver 23 has a two-phase sinusoidal signal (a sine signal S_sin and a cosine signal S_cos) whose amplitude changes according to the actual rotation angle (electrical angle) of the motor 12 as the sensor signal. A winding type resolver that outputs is used. The microcomputer 17 of the present embodiment executes a current feedback control based on the phase current values Iu, Iv, Iw and the rotation angle θm of the motor 12 to output a motor control signal to the drive circuit 18. Is generated.

  More specifically, in the present embodiment, the motor controller 24 of the microcomputer 17 provides the phase voltage command values Vu *, Vv *, Vw * to be applied to each phase of the motor 12 by executing current control in the rotating coordinate system. A first control unit 25 and a second control unit 26 that calculate (Vu **, Vv **, Vw **), and a PWM conversion unit 27 that converts the phase voltage command value into a motor control signal are provided. Yes. The microcomputer 17 according to the present embodiment is configured to output the motor control signal generated by the motor control unit 24 to the drive circuit 18.

  As shown in FIG. 3, the first control unit 25 includes a current command value calculation unit 31 that calculates a current command value corresponding to the target assist force based on the steering torque τ and the vehicle speed V detected as described above. ing. In addition, the first control unit 25 includes a d / q conversion unit 32, and the d / q conversion unit 32 is based on the rotation angle θm detected by the motor resolver 23, and each phase current value Iu, By mapping Iv and Iw on the d / q coordinate, the d-axis current value Id and the q-axis current value Iq are calculated. And the 1st control part 25 performs the current feedback control in the rotation coordinate system (d / q coordinate system) according to the real rotation angle ((theta) m) of this motor 12, The voltage which should be applied to each phase of the motor 12 The phase voltage command values Vu *, Vv *, and Vw * are calculated.

  That is, the current command value calculation unit 31 calculates the q-axis current command value Iq * as the current command value. Specifically, the current command value calculation unit 31 calculates a q-axis current command value Iq * that generates a larger assist force as the input steering torque τ is larger and the vehicle speed V is smaller. . The d-axis current command value Id * is fixed to “0” (Id * = 0). The d-axis current command value Id * and the q-axis current command value Iq *, together with the d-axis current value Id and the q-axis current value Iq output from the d / q conversion unit 32, the corresponding subtractors 33d and 33q. Is input.

  Next, the current deviations ΔId and ΔIq of the respective axes calculated by the subtracters 33d and 33q are input to the corresponding F / B control units (feedback control units) 34d and 34q, respectively. Each of the F / B control units 34d and 34q executes d / q by executing a feedback control calculation based on the input current deviations ΔId and ΔIq and a predetermined feedback gain (proportional: P, integral: I). A d-axis voltage command value Vd * and a q-axis voltage command value Vq *, which are voltage command values in the coordinate system, are calculated.

  Specifically, each of the F / B control units 34d and 34q sets a proportional component obtained by multiplying the input current deviations ΔId and ΔIq by a proportional gain, and an integral value of the current deviations ΔId and ΔIq, respectively. The integral component obtained by multiplying the integral gain is calculated. Then, the d-axis voltage command value Vd * and the q-axis voltage command value Vq * are generated by adding the proportional component and the integral component.

  Next, the d-axis voltage command value Vd * and the q-axis voltage command value Vq * are mapped onto three-phase (U, V, W) AC coordinates in the d / q inverse conversion unit 35. The first control unit 25 is configured to output the phase voltage command values Vu *, Vv *, Vw * obtained by the reverse conversion executed by the d / q reverse conversion unit 35 to the PWM conversion unit 27. ing.

  On the other hand, as shown in FIG. 4, the second control unit 26 includes an addition angle calculation unit 41 that calculates an addition angle θa (θa ′) corresponding to the motor rotation angle change amount for each calculation cycle, and the addition angle θa ( and a control angle calculation unit 42 for calculating a control angle θc as a virtual motor rotation angle for control by integrating θa ′) for each calculation cycle. And the 2nd control part 26 performs phase feedback command value Vu **, Vv **, Vw ** by performing electric current feedback control in the rotation coordinate system ((gamma) / (delta) coordinate system) according to the control angle (theta) c. It is configured to calculate.

  Specifically, the steering angle τ, the vehicle speed V, and the steering angle θs detected as described above are input to the addition angle calculation unit 41 of the present embodiment. The addition angle calculation unit 41 includes a target steering torque calculation unit 45 that calculates a target steering torque τ * corresponding to the target value of the steering torque τ based on the steering angle θs generated in the steering 2 and the vehicle speed V. The target steering torque τ * calculated by the target steering torque calculation unit 45 is input to the subtractor 46 together with the steering torque τ. The addition angle calculation unit 41 of the present embodiment calculates the addition angle θa based on the torque deviation Δτ obtained by subtracting the target steering torque τ * from the actual steering torque τ detected by the torque sensor 14. To do.

  That is, in EPS that applies assist force based on motor torque to the steering system, the target steering torque τ * is a parameter corresponding to the motor torque (target torque) to be generated by the motor 12, and the steering torque τ is the motor 12. This parameter corresponds to the actual torque. That is, the difference (torque deviation Δτ) between the target steering torque τ * and the actual steering torque τ is a state quantity indicating the excess or deficiency of the actual torque with respect to the target torque. Then, the addition angle calculation unit 41 of this embodiment calculates the addition angle θa by executing torque feedback control so that the actual steering torque τ follows the target steering torque τ *.

  Specifically, the torque deviation Δτ calculated by the subtractor 46 is input to the F / B control unit 47. Then, the F / B control unit 47 calculates a proportional component obtained by multiplying the torque deviation Δτ by a proportional gain, and an addition value of an integral component obtained by multiplying the integral value of the torque deviation Δτ by an integral gain. Calculation is performed as the first change component dθτ of the motor rotation angle in each calculation cycle.

  In the present embodiment, the second control unit 26 is provided with a rotation angular velocity estimation calculation unit 50 as motor rotation angular velocity estimation means for estimating the motor rotation angular velocity, and the addition angle calculation unit 41 includes The motor rotation angular velocity ωm_e estimated by the rotation angular velocity estimation calculation unit 50 is input as the second change component dθω of the motor rotation angle in each calculation cycle. Then, the addition angle calculation unit 41 of the present embodiment calculates the addition angle θa using the first change component dθτ based on the torque deviation Δτ and the second change component dθω based on the motor rotation angular velocity ωm_e.

  More specifically, the second controller 26 includes phase voltage command values Vu *, Vv *, Vw * (Vu **, Vv **, Vw *) used when the PWM converter 27 generates a motor control signal. An internal command value corresponding to *), that is, Duty is input. Moreover, ECU11 of this embodiment detects the power supply voltage V_pig applied to the drive circuit 18 with the voltage sensor 51 (refer FIG. 2). The second control unit 26 is provided with a phase voltage calculation unit 52 that calculates the phase voltage values Vu, Vv, and Vw of the motor 12 based on the detected power supply voltage V_pig and the duty.

  Further, these phase voltage values Vu, Vv, Vw and the phase current values Iu, Iv, Iw of the motor 12 detected by the current sensor 21 are fixed in two phases in the α / β converter 53, respectively. They are converted into an α-axis voltage value Vα and β-axis voltage value Vβ, an α-axis current value Iα, and a β-axis current value Iβ in the coordinate system (α / β coordinate system). Then, the rotational angular velocity estimation calculation unit 50 of the present embodiment is based on the motor voltage and motor current indicated by the α-axis voltage value Vα and β-axis voltage value Vβ, and the α-axis current value Iα and β-axis current value Iβ. Estimate the motor rotation angular velocity ωm_e.

  More specifically, the rotational angular velocity estimation calculation unit 50 of this embodiment includes a disturbance observer 54 that estimates an induced voltage generated in the motor 12 as a disturbance based on the motor model.

  That is, in the block diagram shown in FIG. 5, the motor 12 is represented by the motor model M1 that generates the motor current (Iα, Iβ) based on the motor voltage (Vα, Vβ) and the induced voltage (Eα, Eβ). . Accordingly, the induced voltage as described above is obtained by the reverse motor model M2 having the motor current (Iα, Iβ) as an input, and the subtractor 55 having the output of the reverse motor model M2 and the motor voltage (Vα, Vβ) as inputs. A disturbance observer 54 that outputs estimated values (Eα_e, Eβ_e) can be formed. For example, if the motor model M1 is “1 / (R + pL)”, the reverse motor model M2 is “R + pL” (where R: armature winding resistance, L: inductance, p: differential operator). Then, the rotational angular velocity estimation calculation unit 50 of the present embodiment estimates the motor rotational angular velocity ωm_e based on the induced voltage estimated values (Eα_e, Eβ_e) output from the disturbance observer 54.

That is, the induced voltages (Eα, Eβ) in the α / β coordinate system are expressed by the following equations (1) and (2), respectively. In each equation, “Ke” is an induced voltage constant, and “ωm” is a motor rotation angular velocity.
Eα = −Ke × ωm × sinθ (1)
Eβ = Ke × ωm × cosθ (2)
Further, the following equation (3) is obtained by solving these equations (1) and (2) with respect to the angle “θ”. In the formula, “arctan” is “arc tangent”.

θ = arctan (−Eα / Eβ) (3)
Therefore, the motor rotation angle (θm_e) can be estimated from the estimated voltage estimated values (Eα_e, Eβ_e) output from the disturbance observer 54. Then, the rotational angular velocity estimation calculation unit 50 of the present embodiment calculates the motor rotational angular velocity (estimated value) ωm_e by differentiating the estimated value (θm_e) of the motor rotational angle.

  Specifically, as shown in the flowchart in FIG. 6, when the rotational angular velocity estimation calculation unit 50 estimates the induced voltage of the motor 12 by the disturbance observer 54 (Eα_e, Eβ_e, step 101), first, the induced voltage estimation is performed. Filter processing is performed on the values (Eα_e, Eβ_e) (LPF: low-pass filter, step 102). Next, the rotational angular velocity estimation calculation unit 50 estimates the motor rotational angle (θm_e) from the induced voltage estimated values (Eα_e, Eβ_e) by using the above formula (3) (rotational angle estimation, step 103). . Then, the motor rotation angular velocity (estimated value) ωm_e is calculated by differentiating the motor rotation angle (θm_e) (estimation of rotation angle, step 104).

  The rotational angular velocity estimation calculation unit 50 of the present embodiment is configured to output the motor rotation angular velocity ωm_e to the addition angle calculation unit 41 as the second change component dθω of the motor rotation angle in each calculation cycle. (Step 105).

  As shown in FIG. 4, in the addition angle calculation unit 41 of the present embodiment, the first change component dθτ of the motor rotation angle based on the torque deviation Δτ calculated by the F / B control unit 47, and the rotation angular velocity estimation calculation unit The second change component dθω of the motor rotation angle based on the motor rotation angular velocity ωm_e calculated by 50 is input to the addition angle adjustment calculation unit 58. In this embodiment, the rotational angular velocity estimation calculation unit 50 calculates the sum of squares of the induced voltage estimation values (Eα_e, Eβ_e) output from the disturbance observer 54 (Esq_αβ = (Eα_e) ^ 2 + (Eβ_e) ^ 2, where “^ 2” indicates a square), and the induced voltage square sum Esq_αβ is output to the addition angle adjustment calculation unit 58. Then, the addition angle calculation unit 41 of the present embodiment changes the calculation mode of the addition angle θa based on the value of the induced voltage square sum Esq_αβ.

  More specifically, the addition angle adjustment calculation unit 58 of the present embodiment compares the input induced voltage square sum Esq_αβ with a predetermined threshold value (E0). When the induced voltage square sum Esq_αβ exceeds the threshold value (E0), the addition value of the first change component dθτ based on the torque deviation Δτ and the second change component dθω based on the motor rotation angular velocity ωm_e is defined as an addition angle θa. When the value is equal to or less than the threshold value (E0), the first change component dθτ based on the torque deviation Δτ is set as the addition angle θa.

  That is, the motor rotation angular velocity ωm_e having one calculation cycle as a basic unit has an equivalent meaning to the motor rotation angle change amount per one calculation cycle. The estimation of the induced voltage based on the motor current and the motor voltage using the disturbance observer 54 as described above ensures higher accuracy in the high-speed rotation region where the induced voltage increases.

  In consideration of this point, the addition angle adjustment calculation unit 58 of the present embodiment compares the induced voltage square sum Esq_αβ with the threshold value (E0) to determine the rotation state (rotation speed) of the motor 12 from the estimated motor rotation. It is determined whether or not the angular velocity ωm_e is in a high-speed rotation region in which estimation accuracy that can be used as the second change component dθω of the motor rotation angle is ensured. The second variation component dθω based on the motor rotation angular velocity ωm_e is used only when the required estimation accuracy is in a high-speed rotation region.

  Specifically, as shown in the flowchart of FIG. 7, the addition angle adjustment calculation unit 58 firstly includes a first change component dθτ based on the torque deviation Δτ and a second change component dθω based on the motor rotation angular velocity ωm_e, The induced voltage square sum Esq_αβ is acquired (step 201 to step 203).

  Next, the addition angle adjustment calculation unit 58 determines whether or not the induced voltage square sum Esq_αβ exceeds the threshold value E0 (step 204), and if it exceeds the threshold value E0 (step 204: YES), It is determined whether or not an excess flag indicating that the induced voltage square sum Esq_αβ is in a state exceeding the threshold value E0 is set (step 205). If the excess flag is not set (step 205: NO), the excess flag is set (step 206), and the value of the first change component dθτ acquired in step 201 is cleared (dθτ = 0, step 207).

  If the excess flag has already been set in step 205 (step 205: YES), the processing in step 206 and step 207 is not executed. If it is determined in step 204 that the induced voltage square sum Esq_αβ exceeds the threshold value E0 (step 204: YES), the first change component based on the torque deviation Δτ regardless of the excess flag. The addition angle θa is calculated by adding the second change component dθω based on dθτ and the motor rotational angular velocity ωm_e (step 208).

  On the other hand, when it is determined in step 204 that the induced voltage square sum Esq_αβ is equal to or less than the threshold value E0 (step 204: NO), the addition angle adjustment calculation unit 58 also determines whether or not the excess flag is set. (Step 209). If the excess flag is set (step 209: YES), the excess flag is reset (step 210). If the excess flag is not set (step 209: NO), the process of step 210 is not executed. Then, the first change component dθτ acquired in step 201 is calculated as the addition angle θa (step 211).

  The addition angle adjustment calculation unit 58 of the present embodiment is configured to output the addition angle θa calculated in step 208 or step 211 as described above (step 212).

  That is, the first change component dθτ based on the torque deviation Δτ is a value corresponding to the magnitude of the deviation between the actual rotation angle of the motor 12 and the virtual motor rotation angle in control. Therefore, the value is less affected by the motor rotation state than the second change component dθω based on the motor rotation angular velocity ωm_e. In consideration of this point, in the present embodiment, when the motor rotation state is in the low speed region as described above, the first change component dθτ is set as the addition angle θa. In the first calculation cycle (step 204: YES and step 205: NO) in which the addition angle θa is calculated using the second change component dθω based on the motor rotation angular velocity ωm_e, the first change component dθτ is cleared. This is because (Step 207) the first change component dθτ reflects the state of the previous calculation cycle in which the second change component dθω was not used. In the present embodiment, this makes it possible to calculate the addition angle with high accuracy regardless of the motor rotation state.

  As shown in FIG. 4, in the addition angle calculation unit 41, the addition angle θa output from the addition angle adjustment calculation unit 58 is input to the addition angle restriction unit 59. Then, the addition angle calculation unit 41 of the present embodiment outputs the addition angle θa ′ after the addition angle restriction processing is performed in the addition angle restriction unit 59 to the control angle calculation unit 42.

  On the other hand, the control angle calculation unit 42 holds the previous value of the control angle θc calculated in the previous calculation cycle in a storage area (not shown), and adds a new control by adding the addition angle θa to the previous value. The angle θc is calculated. Then, by updating the previous value stored in the storage area with the new control angle θc, the control angle θc is calculated by adding the addition angle θa for each calculation cycle. Yes.

  In the second control unit 26, the control angle θc as the virtual motor rotation angle for control calculated in this way is the two-phase fixed coordinate system (α / β coordinate) output by the α / β conversion unit 53. System) and α-axis current value Iα and β-axis current value Iβ. In the present embodiment, the γ / δ converter 60 receives the α-axis voltage value Vα and the β-axis voltage value Vβ. Then, the γ / δ converter 60 converts the α-axis current value Iα and β-axis current value Iβ, and the α-axis voltage value Vα and β-axis voltage value Vβ into a rotational coordinate system according to the control angle θc, that is, γ / δ. By mapping onto the orthogonal coordinates of the coordinate system, the γ-axis current value Iγ and the δ-axis current value Iδ as the actual current value of the γ / δ coordinate system, and the γ-axis voltage value Vγ and the δ-axis voltage as the actual voltage value The value Vδ is calculated.

  The second control unit 26 includes a current command value calculation unit 61 that calculates a γ-axis current command value Iγ * and a δ-axis current command value Iδ * as the current command value of the γ / δ coordinate system. Then, the current command value calculation unit 61 calculates the γ-axis current command value Iγ * and the δ-axis current command value Iδ * based on the torque deviation Δτ calculated by the addition angle calculation unit 41 and the target steering torque τ *. Calculate.

  The γ-axis current command value Iγ * calculated by the current command value calculation unit 61 is input to the corresponding subtracter 64a together with the γ-axis current value Iγ. Similarly, the δ-axis current command value Iδ * is also input to the corresponding subtracter 64b together with the δ-axis current value Iδ. In this embodiment, the δ-axis current command value Iδ * is fixed to “0” (Iδ * = 0). The current deviations ΔIγ and ΔIδ calculated by the subtracters 64a and 64b are input to the corresponding F / B control units 65a and 65b, respectively.

  Next, each F / B control unit 65a, 65b executes a feedback control calculation based on the current deviations ΔIγ, ΔIδ and a predetermined feedback gain (proportional: P, integral: I), thereby obtaining a γ / δ coordinate system. The γ-axis voltage command value Vγ * and the δ-axis voltage command value Vδ * which are the voltage command values are calculated. The feedback control calculation performed by each of the F / B controllers 65a and 65b is the same as that of the F / B controllers 34d and 34q on the first controller 25 side. Is omitted.

  Further, the γ-axis voltage command value Vγ * and the δ-axis voltage command value Vδ * are mapped onto the three-phase (U, V, W) AC coordinates in the two-phase / three-phase converter 66. The second control unit 26 is configured to output the phase voltage command values Vu **, Vv **, and Vw ** generated by the two-phase / three-phase conversion unit 66 to the PWM conversion unit 27. ing. For details of the principle of resolverless control executed by the second control unit 26 as described above, refer to, for example, the descriptions in Patent Document 1 and Patent Document 2 above.

  As shown in FIG. 2, the microcomputer 17 according to the present embodiment includes a rotation angle abnormality detection unit 68 that detects an abnormality in the rotation angle θm detected by the motor resolver 23. Specifically, the rotation angle abnormality detection unit 68 of this embodiment determines whether or not the sum of squares of the sine signal S_sin and the cosine signal S_cos output from the motor resolver 23 is within an appropriate range. Then, based on the determination result, an abnormality in the rotation angle θm is detected as the actual rotation angle of the motor 12. For details of such rotation angle abnormality detection, refer to, for example, the description of JP-A-2006-177750.

  Further, in the present embodiment, the result of abnormality detection by the rotation angle abnormality detection unit 68 is input to the motor control unit 24 as a rotation angle abnormality detection signal S_rsf. Then, when there is no abnormality in the rotation angle θm, the motor control unit 24 of the present embodiment performs a motor control signal based on the phase voltage command values Vu *, Vv *, Vw * calculated by the first control unit 25. When the rotation angle θm is abnormal, the motor control signal is output based on the phase voltage command values Vu **, Vv **, Vw ** calculated by the second control unit 26. Execute.

  That is, the second control unit 26 uses the control angle θc, which is a virtual motor rotation angle on control, without using the rotation angle θm detected by the motor resolver 23, which is the actual rotation angle of the motor 12. The phase voltage command values Vu **, Vv **, and Vw ** are calculated. Then, the ECU 11 of this embodiment generates a motor control signal based on the phase voltage command values Vu **, Vv **, and Vw ** calculated by the second control unit 26, thereby causing an abnormal rotation angle θm. Even after this is detected, the motor control can be continued stably.

(Current command value calculation)
Next, a mode of current command value calculation by the current command value calculation unit 61 of the present embodiment will be described.

  As shown in FIG. 8, the current command value calculation unit 61 of the present embodiment uses the γ-axis current command value Iγ in each calculation cycle based on the torque deviation Δτ between the target steering torque τ * and the actual steering torque τ. A γ-axis current increase / decrease value calculation unit 71 that calculates an increase / decrease value of γ (γ-axis current increase / decrease value η) and an integration control unit 72 that integrates the input γ-axis current increase / decrease value η for each calculation cycle are provided. .

  The integration control unit 72 of the present embodiment holds the control output in the previous calculation cycle, that is, the previous value of the γ-axis current command value Iγ * in a storage area (not shown). Then, the integration control unit 72 calculates a new γ-axis current command value Iγ * by adding the input γ-axis current increase / decrease value η to the previous value, and the new γ-axis current command value Iγ *. To update the previous value held in the storage area.

The current command value calculation unit 61 of the present embodiment is configured such that the control output of the integration control unit 72, that is, the integrated value of the γ-axis current increase / decrease value η is the γ-axis current command value Iγ *.
More specifically, the γ-axis current increase / decrease value computing unit 71 of this embodiment includes two maps (71a, 71b) in which the torque deviation Δτ and the γ-axis current increase / decrease value η are associated. Specifically, the first map 71a is formed corresponding to the case where the sign (direction) of the target steering torque τ * is “positive (τ *> 0)”, while the second map 71b is the target map The sign of the steering torque τ * is formed corresponding to “when it is negative (τ * <0)”. When the target steering torque τ * is “0”, the code immediately before is used. Then, the γ-axis current increase / decrease value calculation unit 71 switches the map to be referred to according to the sign of the input target steering torque τ *, and based on the torque deviation Δτ, the γ-axis current increase / decrease value η in each calculation cycle. Is calculated.

  That is, when the target steering torque τ * is “positive value”, the torque deviation Δτ is “positive value”, or when the sign of the target steering torque τ * is “negative value”, the torque deviation Δτ is “ The state of “negative value” indicates that the actual torque is “insufficient” with respect to the target torque that should be generated by the motor 12. On the other hand, when the target steering torque τ * is “positive value”, the torque deviation Δτ is “negative value”, or when the sign of the target steering torque τ * is “negative value”, the torque deviation Δτ is “ The state of “positive value” indicates that the actual torque is “excessive” with respect to the target torque that should be generated by the motor 12. Then, the γ-axis current increase / decrease value calculation unit 71 of the present embodiment calculates the γ-axis current increase / decrease value in each calculation cycle based on the excess or deficiency of the actual torque with respect to the target torque to be generated by the motor 12 indicated by the torque deviation Δτ. Calculate η.

  Specifically, in the first map 71a, the γ-axis current increase / decrease value η is greater than or equal to the predetermined value A1 having the “positive value” and the torque deviation Δτ is smaller than the predetermined value A2 having the same “positive value”. (A1 ≦ Δτ <A2) is set such that the larger the torque deviation Δτ is, the more positive the value is. Further, when the torque deviation Δτ is smaller than the predetermined value A1 and equal to or larger than the predetermined value A3 having the same “positive value” (A3 ≦ Δτ <A1), the torque deviation Δτ becomes larger as the value becomes smaller. It is set to be a “negative value” having an absolute value. When the torque deviation Δτ is equal to or greater than the predetermined value A2 (A2 ≦ Δτ), the γ-axis current increase / decrease value η is a constant “positive value (maximum increase value γ1)”, and the torque deviation Δτ is a predetermined value. When smaller than A3 (Δτ <A3), the γ-axis current increase / decrease value η is set to be a constant “negative value (maximum decrease value γ2)”.

  On the other hand, in the second map 71b, the γ-axis current increase / decrease value η is in a range where the torque deviation Δτ is equal to or less than the predetermined value A4 having the “negative value” and larger than the predetermined value A5 having the same “negative value”. (A5 <Δτ ≦ A4) is set so that the smaller the torque deviation Δτ, the “positive value” having a larger absolute value. When the torque deviation Δτ is greater than the predetermined value A4 and equal to or less than the predetermined value A6 having the same “negative value” (A4 <Δτ ≦ A6), the torque deviation Δτ is a large value (small absolute value). Is set to be a “negative value” having a larger absolute value. When the torque deviation Δτ is equal to or less than the predetermined value A5 (Δτ ≦ A5), the γ-axis current increase / decrease value η is a constant “positive value (maximum increase value γ1)”, and the torque deviation Δτ is a predetermined value. When larger than A6 (A6 <Δτ), the γ-axis current increase / decrease value η is set to be a constant “negative value (maximum decrease value γ2)”.

  The γ-axis current increase / decrease value calculation unit 71 of the present embodiment refers to these two maps (71a, 71b), and the actual torque is “excess” with respect to the target torque that should be generated by the motor 12 ( When τ *> 0, Δτ <0, or when τ * <0, Δτ> 0), the γ-axis current increase / decrease value η having a “negative value” that reduces the γ-axis current command value Iγ * is calculated. To do.

  Furthermore, in the present embodiment, a range in which the “shortage of actual torque” is allowed is set for the region indicating that the real torque is “insufficient” with respect to the target torque to be generated by the motor 12 (τ). *> 0, 0 ≦ Δτ <A1, or τ * <0, A4 <Δτ ≦ 0). Then, the γ-axis current increase / decrease value calculating unit 71 reduces the γ-axis current command value Iγ * so as to reduce the γ-axis current command value Iγ * even when “insufficient actual torque” indicated by the torque deviation Δτ is within the allowable range. Γ-axis current increase / decrease value η having “value” is calculated.

  Then, the γ-axis current increase / decrease value calculation unit 71 of the present embodiment, when “insufficient actual torque” indicated by the torque deviation Δτ exceeds the allowable range (Δτ ≧ A1 or τ * < At 0, Δτ ≦ A4) is configured to calculate a γ-axis current increase / decrease value η having a “positive value” that increases the γ-axis current command value Iγ *.

(Current command value limit)
The current command value calculation unit 61 of the present embodiment includes a current command upper limit value calculation unit 73 that calculates the upper limit value (current command upper limit value Ilim) of the γ-axis current command value Iγ * calculated as described above. There is provided a current command value limiting unit 74 for limiting the γ-axis current command value Iγ * to be equal to or less than the current command upper limit value Ilim, in other words, correcting so as not to exceed the current command upper limit value Ilim. The current command value calculation unit 61 is configured to output the γ-axis current command value Iγ ** after the limit process is performed in the current command value limit unit 74 to the subtractor 64a (see FIG. 4).

  More specifically, as shown in FIG. 9, the current command upper limit value calculation unit 73 of the present embodiment displays two maps (73a, 73b) in which the vehicle speed V and the current command upper limit values (Ilim_a, Ilim_b) are associated with each other. I have. The current command upper limit value calculation unit 73 calculates the current command upper limit value Ilim by referring to these maps.

  In the first map 73a of the present embodiment, the current command upper limit value Ilim_a is set to be a constant value (predetermined value I1) regardless of the vehicle speed V. On the other hand, in the second map 73b, the current command upper limit value Ilim_b is set to be lower as the vehicle speed V is higher. The current command upper limit value Ilim_b in the second map 73b is set such that the minimum value I2 is higher than the predetermined value I1 in the first map 73a.

  In addition, the current command upper limit value calculation unit 73 of the present embodiment includes a switching control unit 75 that determines which of the first map 73a and the second map 73b is used when calculating the current command upper limit value Ilim. The current command upper limit value calculation unit 73 of the present embodiment switches the first map 73a and the second map 73b based on the determination of the switching control unit 75, and the current command upper limit value Ilim according to the vehicle speed V. Is calculated.

  That is, as described above, in the resolverless control using the virtual control angle (θc) in the control, a large current is supplied to the motor 12 to increase the magnetomotive force of the stator, thereby stabilizing the motor control. Can be increased. However, if a motor is continuously energized with a large current, its energy efficiency is lowered, and there is a risk of lowering reliability due to heat generation of the motor.

  Considering this point, in the present embodiment, a value (predetermined value I1) that suppresses the motor current is set as the current command upper limit value Ilim_a in the first map 73a, while the second map 73b includes As the current command upper limit value Ilim_b, a high value (I2 or more, I2> I1) that allows energization of a large motor current is set. Then, the current command upper limit calculation unit 73 of the present embodiment switches the map to be referred to when calculating the current command upper limit value Ilim, thereby effectively reducing the motor current without impairing the stability of the motor control. It is possible to suppress.

  More specifically, the switching control unit 75 of the present embodiment includes each axis current value and voltage value (Iγ, Iδ, Vγ, Vδ) of the γ / δ coordinate system calculated by the γ / δ conversion unit 60, In addition, the motor rotational angular velocity ωm_e and the induced voltage E that are calculated (estimated) by the rotational angular velocity estimation calculation unit 50 are input. The induced voltage E is calculated by obtaining the square root (√) of the induced voltage square sum Esq_αβ. Then, the switching control unit 75 determines whether or not the control state of the motor 12 is stable based on each of these state quantities (control state determination), and based on the determination result, the current command The map used for the calculation of the upper limit value Ilim is switched.

  Specifically, the switching control unit 75 according to the present embodiment substitutes each of the above state quantities into a voltage equation represented by the following equation (4), thereby controlling as a virtual motor rotation angle in the control. A load angle (error angle) θL indicating a deviation between the angle θc and the actual rotation angle (θm) is estimated. Then, the control state of the motor 12 is determined based on the load angle θL.

sin θL = (− Vγ + R · Iγ + L · (d / dt) · Iγ−ωm · L · Iδ) / E
... (4)
However, “R” is motor resistance, and “L” is motor inductance. The load angle θL can be obtained from the inverse function (arcsin) of the above equation (4).

  As shown in FIG. 10, in this embodiment, the angle formed by the “d-axis” of the actual rotation coordinate (d / q coordinate) with respect to the “U-axis” of the three-phase fixed coordinate is defined as the actual rotation angle (θm). In this case, when the actual rotation angle (θm) coincides with the control angle θc, γ / as a virtual coordinate according to the control angle θc so that the “γ axis” and the “d axis” coincide with each other. A δ coordinate is defined. Note that the above equation (4) is an equation corresponding to the case where the “δ axis” is defined at a position advanced by the “90 °” phase with respect to the “γ axis”, as shown in FIG. In the ECU 11 (second control unit 26) of the present embodiment, the magnitude (positive value) of the load angle θL indicating the deviation between the control angle θc and the actual rotation angle (θm) is “0 to 90 °”. As long as it stays within the range, the motor control can be executed stably.

  That is, as shown in FIG. 4, the second control unit 26 executes the torque feedback control, thereby controlling the control angle θc based on the torque deviation Δτ indicating the excess or deficiency of the actual torque with respect to the target torque that should be generated by the motor 12. Is calculated. As shown in FIG. 10, if the γ-axis current value Iγ is constant, a larger q-axis current flows as the load angle θL increases within the range in which the motor can be stably controlled. A motor torque corresponding to the q-axis current value Iq is generated (Iq = Iγ · sinθL).

  That is, in this embodiment, the larger the torque deviation Δτ is due to the lack of actual torque with respect to the target torque, the larger the added angle θa is calculated according to the torque deviation Δτ. Then, the load angle θL is enlarged by the change of the control angle θc according to the addition angle θa, and the motor torque is increased by the change of the q-axis current value Iq according to the load angle θL, so that the actual torque becomes the target torque. Follow.

  However, when the load angle θL exceeds the range in which the motor can be stably controlled, that is, in a region where the magnitude exceeds “90 °”, the change in the load angle θL and the q-axis current value Iq The relationship with changes is reversed. That is, as the torque deviation Δτ increases, the motor torque decreases and the load angle θL is increased. Then, at the moment when the magnitude of the load angle θL exceeds “180 °”, a motor torque opposite to the motor rotation direction is generated. Therefore, when the load angle θL is in the boundary region of the range where the motor can be stably controlled, the control state of the motor 12 is likely to become unstable.

  Focusing on this point, the switching control unit 75 of the present embodiment acquires each of the state quantities (Iγ, Iδ, Vγ, ωm_e, E) as shown in the flowchart of FIG. 4) When the sine component “sin θL” of the load angle θL is estimated based on the equation (step 302), the sine component (absolute value “| sin θL |”) thereof is compared with a predetermined threshold B0 (step 303). In this case, the threshold value B0 is set to a value corresponding to the boundary region of the range in which the motor can be stably controlled, for example, about “0.8”. When the sine component (absolute value) of the load angle θL is equal to or less than the threshold value B0 (| sin θL | ≦ B0, step 303: NO), it is determined that the control state of the motor 12 is stable. (Stable state), it is determined that the value (Ilim_a) based on the first map 73a should be calculated as the current command upper limit value Ilim (step 304).

  In step 303, if the sine component (absolute value) of the load angle θL exceeds the threshold value B0 (| sin θL |> B0, step 303: YES), the control state is likely to become unstable (see FIG. It is determined that the state is in an unstable state, and it is determined that the value (Ilim_b) based on the second map 73b should be calculated as the current command upper limit value Ilim (step 305).

  As described above, when it is determined that the control state is in an unstable state, the switching control unit 75 of the present embodiment has a value when the control state is stable, that is, in the first map 73a. It is determined that the current command upper limit value Ilim should be calculated using the second map 73b in which a value (Ilim_b) higher than the value (Ilim_a) is set.

  That is, by switching the current command upper limit value Ilim to a high value (Ilim_b) and allowing energization of a large motor current, the magnetomotive force of the stator is strengthened and the actual rotation angle with respect to the virtual control angle θc in the control The followability of (θm) can be improved. That is, according to the control angle θc, the rotor can be stably rotated (the rotor can be held at the rotational position corresponding to the changing control angle θc). In this embodiment, the motor control is thereby stabilized, and at the same time, the current command upper limit value Ilim when the control state is stable is set to a low value (Ilim_a) in advance. Thus, the motor current can be effectively suppressed.

As described above, according to the present embodiment, the following effects can be obtained.
(1) The current command value limiter 74 limits the γ-axis current command value Iγ * below the current command upper limit value Ilim calculated by the current command upper limit value calculation unit 73. Further, the switching control unit 75 provided in the current command upper limit value calculation unit 73 is a load angle (error) indicating a deviation between the control angle θc as a virtual motor rotation angle in the control and the actual rotation angle (θm). (Angle) θL (“sin θL” which is a sine component) is estimated, and the control state of the motor 12 is determined based on the load angle θL. When the switching control unit 75 determines that the control state is in an unstable state, the current command upper limit value Ilim is higher than the value (Ilim_a) when the control state is stable. It is determined that the value (Ilim_b) should be changed.

  That is, the brushless motor can be stably controlled as long as the deviation between the virtual control angle θc and the actual rotation angle (θm) is within a certain range. Therefore, when the load angle θL indicating the deviation is in the boundary region of the range in which the motor can be stably controlled, it can be determined that the control state is in an unstable state that tends to be unstable. When it is determined that the control state is in an unstable state as in the above configuration, the current command upper limit value Ilim is changed to a high value (Ilim_b) that allows energization of a large motor current. As a result, the stability of the motor control can be improved. As a result, the current command upper limit value Ilim when the control state is stable can be set to a low value (Ilim_a) in advance, and as a result, the motor control can be effectively performed without impairing the stability of the motor control. The current can be suppressed.

  (2) The current command upper limit calculation unit 73 has a first map 73a in which a low value (Ilim_a) for suppressing the motor current is set low and a high value (Ilim_b) in which a large motor current is allowed to be supplied. Two maps 73b are provided (Ilim_a <Ilim_b). Then, the current command upper limit value calculation unit 73 switches a map to be referred to when calculating the current command upper limit value Ilim based on the determination of the switching control unit 75.

  According to the above configuration, the value of the current command upper limit value Ilim can be changed according to the stability of the motor control, and an arbitrary state quantity and current command upper limit value can be changed in each map (73a, 73b). Ilim can be associated. As a result, it is possible to flexibly change the value of the current command upper limit value Ilim, thereby further optimizing the limiting process.

  (3) The current command upper limit value Ilim and the vehicle speed V are associated with each map (73a, 73b) provided in the current command upper limit value calculation unit 73. In the second map 73b, the current command upper limit value Ilim_b is set to be a lower value as the vehicle speed V is higher.

  That is, normally, as the vehicle speed V increases, the steering angle θs generated in the steering decreases, and as a result, the load torque applied to the motor 12 also decreases. Therefore, according to the above configuration, the motor current can be more appropriately suppressed without impairing the stability of the motor control.

In addition, you may change the said embodiment as follows.
In the above embodiment, the present invention is embodied in a so-called column type electric power steering device (EPS) 1. However, the present invention is not limited to this, and the present invention may be applied to a so-called pinion type or rack assist type EPS.

  -Moreover, you may apply this invention to the motor control apparatus used for uses other than EPS. In the above embodiment, the torque deviation Δτ between the target steering torque τ * and the actual steering torque τ is used as the “torque deviation between the target torque to be generated by the motor and the actual torque”. However, when applied to applications other than EPS, the actual “torque deviation between target torque and actual torque” may be used.

  In the above embodiment, the “δ axis” is defined at a position advanced by “90 °” phase with respect to the “γ axis”, and the load angle (error angle) θL (with a sine component) A certain “sin θL”) was estimated. However, the present invention is not limited to this, and when the δ-axis current command value Iδ * is fixed to “0” (Iδ * = 0), the “δ-axis” is positioned at a position delayed by “90 °” from the “γ-axis”. And the load angle θL may be estimated using a voltage equation matched to this.

  In addition to estimating the load angle θL using the voltage equation such as the above equation (4), the actual rotation angle (θm) is estimated and the actual rotation angle (θm) is subtracted from the control angle θc. Accordingly, the load angle θL may be estimated. Even if it does in this way, the effect similar to the said embodiment can be acquired.

  For example, as shown in the flowchart in FIG. 12, the switching control unit 75 estimates the control angle θc and the motor rotation angular velocity ωm_e (see FIG. 6, step 103), and the motor rotation as the actual rotation angle. The angle (estimated value) θm_e is acquired (step 401), and the load angle θL is estimated by subtracting the motor rotation angle θm_e from the control angle θc (step 402). Then, the control state of the motor 12 is determined by comparing the load angle θL (absolute value thereof) with a predetermined threshold value θ0 (step 403). In this case, the threshold value θ0 is set to a value corresponding to the boundary of the range where the motor can be stably controlled, for example, about “60 °”.

  That is, when the load angle θL (the absolute value thereof) is equal to or smaller than the threshold θ0 (| θL | ≦ θ0, Step 403: NO), the switching control unit 75 has a stable control state of the motor 12. And determines that the value (Ilim_a) based on the first map 73a should be calculated as the current command upper limit value Ilim (step 404). When the load angle θL exceeds the threshold θ0 (| θL |> θ0, step 403: YES), it is determined that the control state is in an unstable state, and a value based on the second map 73b ( It is determined that Ilim_b) should be calculated as the current command upper limit value Ilim (step 405).

  -Furthermore, in the above embodiment, although not particularly mentioned, not only when the load angle θL is in the boundary region of the range in which the motor can be stably controlled (when the control state is easily destabilized), Even when the load angle θL is outside the range in which stable motor control is possible, the current command upper limit value Ilim may be increased by including the destabilized state.

  In other words, the current command upper limit value Ilim is set to a high value (Ilim_b) to allow energization of a large motor current, and by increasing the magnetomotive force of the stator, the load angle θL can be quickly brought within a range where the motor can be stably controlled. Can be restored. As a result, the stability of the motor control can be improved.

  In the above embodiment, the current command upper limit value calculation unit 73 sets the first map 73a in which a low value (Ilim_a) that suppresses the motor current is set low and a high value (Ilim_b) that allows energization of a large motor current. The second map 73b is provided (Ilim_a <Ilim_b). The current command upper limit value calculation unit 73 changes the current command upper limit value Ilim by switching a map to be referred to when calculating the current command upper limit value Ilim based on the determination of the switching control unit 75. However, the present invention is not limited to this, and a configuration may be adopted in which preset two values (low value and high value) are switched according to the control state of the motor 12.

  In the above embodiment, in the second map 73b, the current command upper limit value Ilim_b is set to be lower as the vehicle speed V is higher. However, the present invention is not limited to this, and also in the first map 73a, the current command upper limit value Ilim_a is set to be lower as the vehicle speed V is higher. Alternatively, only in the first map 73a, the current command upper limit value Ilim_a may be configured to be set to a lower value as the vehicle speed V is higher.

  In the above embodiment, the current command upper limit value Ilim is associated with the vehicle speed V in each map (73a, 73b) provided in the current command upper limit value calculation unit 73. However, the present invention is not limited to this, and if it is determined that the control state of the motor 12 is in an unstable state, if the current command upper limit value Ilim can be changed to a high value, a state quantity other than the vehicle speed V And a map in which the current command upper limit value Ilim is associated with each other.

  In the above embodiment, the addition angle calculation unit 41 calculates the addition angle θa by executing torque feedback control based on the torque deviation Δτ. The γ-axis current increase / decrease value calculation unit 71 calculates the γ-axis current increase / decrease value η based on the torque deviation Δτ. However, when the target steering torque τ * is fixed to “0” and controlled, it goes without saying that the configuration using the steering torque τ instead of the torque deviation Δτ is completely equivalent (Δτ = τ−τ *). ).

In the above embodiment, the steering angle θs is detected using the steering sensor 16, but a configuration in which the steering angle θs is estimated from the wheel speed difference may be employed.
Next, technical ideas that can be grasped from the above embodiments will be described together with effects.

  (A) The motor control signal output means includes a first map in which a low value for suppressing the motor current is set low, and a second map in which a high value for allowing energization of a large motor current is set, The map to be referred to when calculating the upper limit value is switched based on the determination of the control state.

  According to the above configuration, the upper limit value can be changed according to the stability of the motor control, and an arbitrary state quantity and the upper limit value can be associated with each map. As a result, the value of the upper limit value can be flexibly changed, thereby further optimizing the limit process.

  (B) The upper limit value and the vehicle speed are associated with at least one of the maps, and in the map, the upper limit value is set to be lower as the vehicle speed is higher.

  That is, normally, the higher the vehicle speed, the smaller the steering angle generated in the steering, and as a result, the load torque applied to the motor also decreases. Therefore, according to the above configuration, the motor current can be more appropriately suppressed without impairing the stability of the motor control.

  (C) Even if the torque deviation is in a state indicating a shortage of the actual torque with respect to the target torque to be generated by the motor, the motor control signal output means The gist is to calculate the increase / decrease value so as to reduce the current command value.

  According to the above configuration, generation of an excessive motor current exceeding the amount required for maintaining the stability of motor control can be suppressed, and the heat generation of the motor can be effectively suppressed.

  DESCRIPTION OF SYMBOLS 1 ... Electric power steering apparatus (EPS), 10 ... EPS actuator, 11 ... ECU, 12 ... Motor, 12u, 12v, 12w ... Motor coil, 14 ... Torque sensor, 15 ... Vehicle speed sensor, 16 ... Steering sensor, 17 ... Microcomputer , 18 ... Drive circuit, 21 ... Current sensor, 23 ... Motor resolver, 24 ... Motor controller, 25 ... First controller, 26 ... Second controller, 27 ... PWM converter, 41 ... Addition angle calculator, 42 ... control angle calculation unit, 45 ... target steering torque calculation unit, 46 ... subtractor, 47 ... F / B control unit, 50 ... rotational angular velocity estimation calculation unit, 52 ... phase voltage calculation unit, 53 ... α / β conversion unit, 54 ... Disturbance observer, 58 ... Addition angle adjustment calculation unit, 59 ... Addition angle restriction unit, 60 ... γ / δ conversion unit, 61 ... Current command value calculation unit, 65a, 65b ... F / B control unit, 66 ... Two-phase Three-phase conversion unit, 68 ... rotation angle abnormality detection unit, 71 ... γ-axis current increase / decrease value calculation unit, 71a ... first map, 71b ... second map, 72 ... integration control unit, 73 ... current command upper limit value calculation unit, 73a ... 1st map, 73b ... 2nd map, 74 ... Current command value limiting part, 75 ... Switching control part, Iu, Iv, Iw ... Phase current value, θm ... Rotation angle, Id ... d-axis current value, Iq ... q-axis current value, Id * ... d-axis current command value, Iq * ... q-axis current command value, ΔId, ΔIq ... current deviation, Vu *, Vv *, Vw * ... phase voltage command value, τ ... steering torque, τ * ... target steering torque, Δτ ... torque deviation, A1-A6 ... predetermined value, η ... γ-axis current increase / decrease value, dθτ ... first change component, Iα ... α-axis current value, Iβ ... β-axis current value, Vα ... α Axis voltage value, Vβ: β axis voltage value, Eα, Eβ: Induced voltage, Eα_e, Eβ_e: Estimated induced voltage, Esq_αβ: Induced voltage square sum, E0: Threshold, ωm_e: Mode Data rotation angular velocity, dθω, second change component, θa, θa ′, addition angle, θc, control angle, Iγ, γ-axis current value, Iδ, δ-axis current value, Iγ *, Iγ **, γ-axis current command Value, Iδ *: δ-axis current command value, ΔIγ, ΔIδ: current deviation, Vu **, Vv **, Vw ** ... phase voltage command value, θs: steering angle, V: vehicle speed, Ilim: current command upper limit value Ilim_a ... current command upper limit value, I1 ... predetermined value, Ilim_b ... current command upper limit value, I2 ... minimum value, E ... induced voltage, θL ... load angle, B0 ... threshold, θm_e ... motor rotation angle, θ0 ... threshold, S_rsf ... Rotation angle abnormality detection signal.

Claims (2)

A motor control signal output means for outputting a motor control signal; and a drive circuit for supplying three-phase drive power to the motor based on the motor control signal. The motor control signal output means should be generated by the motor By executing torque feedback control based on the torque deviation between the target torque and the actual torque, an addition angle corresponding to the amount of change in the motor rotation angle for each calculation cycle is calculated, and the addition angle is integrated to achieve control. In accordance with the control motor rotation angle while calculating the current command value by calculating the increase / decrease value based on the torque deviation and integrating the increase / decrease value for each calculation cycle. In the motor control device that outputs the motor control signal by executing current feedback control in the rotating coordinate system,
The motor control signal output means limits the current command value to an upper limit value or less,
A load angle indicating a deviation between the motor rotation angle and the actual rotation angle on the control is estimated, the control state of the motor is determined based on the estimated load angle, and the control state is in an unstable state. In some cases, the upper limit value is made higher than when the control state is stable,
A motor control device.   An electric power steering device comprising the motor control device according to claim 1.

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