JP2011217898A - Ultrasonic diagnostic apparatus - Google Patents

Abstract

PROBLEM TO BE SOLVED: To improve selectivity of a target position in technique for selectively extracting information on the inside of a living body from the target position utilizing a continuous wave.SOLUTION: A PSK modulating part 22A conducts PSK modulation for a carrier wave A using a code A. A PSK modulating part 22B conducts PSK modulation for a carrier wave B using a code B. The code A and the code B have mutual complementary relationship. A multiprocessing part 24 multiplexes two PSK continuous waves by OFDM to form an OFDM continuous wave. Receiving mixers 30A, 30B respectively conduct demodulation for received signals using a reference signal whose correlation to the target position in a living body is adjusted, thereby obtaining a demodulation signal corresponding to the target position. A synthesizing part 52I adds signals obtained from addition parts 46A, 46B, and a synthesizing part 52Q adds signals obtained from addition parts 48A, 48B. Thus, the received signals from the outside of the target position are decreased to heighten the selectivity of the target position.

Description

  The present invention relates to an ultrasonic diagnostic apparatus, and more particularly to an ultrasonic diagnostic apparatus using a continuous wave.

  A continuous wave Doppler is known as a technique using a continuous wave of an ultrasonic diagnostic apparatus. In continuous wave Doppler, for example, a transmission wave that is a sine wave of several MHz is continuously emitted into the living body, and a reflected wave from the living body is continuously received. The reflected wave includes Doppler shift information by a moving body (for example, blood flow) in the living body. Therefore, by extracting the Doppler shift information and performing frequency analysis, a Doppler waveform reflecting the velocity information of the moving body can be formed.

  Continuous wave Doppler using a continuous wave is generally superior in speed measurement at a higher speed than pulse Doppler using a pulse wave. Under such circumstances, the inventor of the present application has conducted research on continuous wave Doppler. As one of the results, Patent Document 1 proposes a technique related to continuous wave Doppler (FMCW Doppler) subjected to frequency modulation processing.

  On the other hand, with continuous wave Doppler, position measurement is difficult due to the use of continuous waves. For example, a conventional general continuous wave Doppler device (a device that does not use FMCW Doppler) cannot perform position measurement. On the other hand, the inventor of this application has proposed a very epoch-making technique in Patent Document 2 that can selectively extract Doppler information from a desired position in a living tissue by FMCW Doppler.

JP 2005-253949 A JP 2008-289851 A

  The technology of FMCW Doppler described in Patent Literature 1 and Patent Literature 2 is an epoch-making technology with the possibility of ultrasonic diagnosis that has never existed before. The inventor of the present application has conducted further research and development on this revolutionary technology improvement. In particular, research and development have been repeated with a focus on techniques for selectively extracting in vivo information from target positions using continuous waves.

  The present invention has been made in the course of its research and development, and its purpose is to improve the selectivity of the target position in a technique for selectively extracting in-vivo information from the target position using continuous waves. It is in.

  A preferred ultrasonic diagnostic apparatus for the above object is a transmission signal processing for outputting a continuous wave transmission signal obtained by multiplexing two carrier signals digitally modulated on the basis of two periodic signal sequences complementary to each other. Correlation between the target position in the living body and the ultrasonic wave transmitting / receiving section that receives the ultrasonic wave corresponding to the transmission signal and receives the ultrasonic wave from the living body The received signal processing unit that obtains a demodulated signal corresponding to the target position by performing demodulation processing on the received signal while adjusting the in-vivo, and in vivo information that extracts in-vivo information from the demodulated signal corresponding to the target position And an information extraction unit.

In a preferred embodiment, the transmission signal processing unit includes a first carrier signal digitally modulated based on a periodic signal sequence that repeats a first code sequence _N having a code length N, and a second code sequence having a code length N. A continuous wave transmission signal in which the second carrier signal digitally modulated based on a periodic signal sequence of repeating _N is multiplexed is output.

In a preferred embodiment, the code length N is the power of 2, each of the first sign row _N and the second tally column _N is based first sign row _{N / 2} to the second code sequence _{N / 2} It is characterized by being formed.

In a preferred embodiment, the first code sequence _N is a code sequence A _N , the second code sequence _N is a code sequence B _N , and the code sequence A _N is a code sequence A _{N / 2} and a code sequence. B _{N / 2} is a code string in which the code string B _{N / 2} is connected in series, the code string B _N is a code string in which the code string A _{N / 2} and the code string -B _{N / 2} are connected in series, and the code string -B _{N / 2} is , A code string obtained by inverting the code value of the code string B _{N / 2} .

In a preferred embodiment, each of the code string A _N and the code string B _N is obtained from the code string A ₂ = (1, 1) and the code string B ₂ = (1, −1) having a code length of 2 in series. The connection is formed repeatedly.

In a preferred embodiment, the first code sequence _N is a code sequence A _N , the second code sequence _N is a code sequence B _N ′, and the code sequence A _N is encoded with a code sequence A _{N / 2.} The code sequence B _{N / 2} is a code sequence in which the code sequence B _{N / 2} is connected in series, and the code sequence B _N ′ is a code sequence in which the code sequence B _{N / 2} ^-1 and the code sequence -A _{N / 2} ^-1 are connected in series. B _{N / 2} is a code string in which a code string A _{N / 4} and a code string -B _{N / 4} are connected in series, and the code string B _{N / 2} ^-1 is a sequence inversion of codes of the code string B _{N / 2} The code string -A _{N / 2} ^-1 is a code string obtained by inverting both the code value and the sequence of the code string A _{N / 2} , and the code string -B _{N / 4} is a code string. It is a code string obtained by inverting the value of the code of the sequence B _{N / 4} .

In a preferred embodiment, each of the code string A _N and the code string B _N ′ is obtained from the code string A ₂ = (1, 1) and the code string B ₂ = (1, −1) having a code length of 2, It is formed by repeating series connection.

In a preferred embodiment, the transmission signal processing unit outputs a first carrier signal whose phase is changed by phase shift keying based on a periodic signal sequence that repeats the first code sequence _N , and the second code sequence _N. A continuous wave transmission signal in which a second carrier signal whose phase is changed by phase shift keying based on a repeated periodic signal sequence is multiplexed is output.

  In a preferred embodiment, the transmission signal processing unit outputs a continuous wave transmission signal formed by multiplexing the first carrier signal and the second carrier signal by orthogonal frequency division multiplexing (OFDM). Features.

  According to the present invention, the selectivity of a target position is improved in a technique for selectively extracting in-vivo information from a target position using a continuous wave.

1 is a diagram illustrating an overall configuration of an ultrasonic diagnostic apparatus that is preferable in the practice of the present invention. Is a graph showing the calculation results of the autocorrelation relates to code A _8. Is a graph showing the calculation results of the autocorrelation relates to code B _8. It is a graph showing the calculation results of the autocorrelation relates to code B _{8 '.} It is a figure which shows the time change of the carrier wave utilized for OFDM. It is a figure which shows the frequency spectrum of the PSK continuous wave utilized for OFDM. It is a figure for demonstrating the transmission / reception processing process of two PSK continuous waves. 6 is a diagram for explaining PSK modulation processing and position selectivity for carrier wave A. FIG. 6 is a diagram for explaining PSK modulation processing and position selectivity for carrier wave B. FIG. It is a figure which shows the characteristic of a phase detector. It is a figure for demonstrating extraction of a Doppler signal.

  FIG. 1 is a diagram showing an overall configuration of an ultrasonic diagnostic apparatus suitable for implementing the present invention. The transmitting vibrator 10 continuously transmits ultrasonic waves into the living body, and the receiving vibrator 12 continuously receives reflected ultrasonic waves from the living body. In this way, transmission and reception are performed by different vibrators, and transmission / reception is performed by a so-called continuous wave Doppler method. The transmitting transducer 10 includes a plurality of vibration elements, and the plurality of vibration elements are controlled to form an ultrasonic transmission beam. The receiving vibrator 12 also includes a plurality of vibration elements, and signals obtained by the plurality of vibration elements are processed to form a reception beam.

  The transmission beamformer (transmission BF) 14 outputs transmission signals to a plurality of vibration elements included in the transmission transducer 10. The transmission beamformer 14 is supplied with a continuous wave transmission signal via the multiplex processing unit 24, and the transmission beamformer 14 performs a delay process on the transmission signal in accordance with each vibration element, and each vibration element. A transmission signal corresponding to is formed. Note that power amplification processing may be performed on the transmission signal corresponding to each vibration element formed in the transmission beam former 14 as necessary. Thus, an ultrasonic transmission beam is formed.

  The continuous wave transmission signal in this embodiment is formed by the PSK modulation processing units 22A and 22B and the multiplex processing unit 24.

  The PSK modulation processing unit 22A generates a PSK continuous wave by performing digital modulation processing by phase shift keying (PSK) on the carrier A (RF wave) obtained from the carrier A generator 20A. On the other hand, the PSK modulation processing unit 22B generates a PSK continuous wave by performing digital modulation processing by phase shift keying (PSK) on the carrier B (RF wave) obtained from the carrier B generator 20B.

  Then, the two PSK continuous waves output from the PSK modulation processing units 22 </ b> A and 22 </ b> B are multiplexed by the multiplex processing unit 24. The multiplexing processing unit 24 multiplexes two PSK continuous waves by orthogonal frequency division multiplexing (OFDM), thereby forming an OFDM continuous wave as a continuous wave transmission signal. The continuous wave transmission signal formed by the PSK modulation processing units 22A and 22B and the multiplex processing unit 24 will be described in detail later.

  The reception beam former (reception BF) 16 forms a reception beam by phasing and adding a plurality of reception signals obtained from a plurality of vibration elements included in the reception transducer 12. That is, the reception beamformer 16 performs a delay process corresponding to the vibration signal obtained from each vibration element and adds a plurality of reception signals obtained from the plurality of vibration elements. Form a beam. Note that a plurality of received signals may be supplied to the reception beam former 16 after processing such as low noise amplification is performed on the received signals obtained from the respective vibration elements. In this way, a reception RF signal along the reception beam is formed and sent to BPF (band pass filters) 18A and 18B.

  The BPF 18A is a filter that extracts a signal in a frequency band corresponding to the carrier A, and the BPF 18B is a filter that extracts a signal in a frequency band corresponding to the carrier B. That is, the received RF signals corresponding to each of the two PSK continuous waves before being multiplexed are extracted by the BPFs 18A and 18B.

  The receiving mixer 30A is a circuit that generates a complex baseband signal by performing quadrature detection on the received RF signal output from the BPF 18A, that is, the received RF signal corresponding to the PSK continuous wave of the carrier A. 32A and 34A. Each mixer is a circuit that mixes the received RF signal with a predetermined reference signal.

  The reference signal supplied to each mixer of the reception mixer 30A is generated based on the PSK continuous wave of the carrier A output from the PSK modulation processing unit 22A. That is, the PSK continuous wave of the carrier wave A output from the PSK modulation processing unit 22A is subjected to delay processing in the delay circuit 25A, and the delayed PSK continuous wave is directly supplied to the mixer 32A as a reference signal, while being supplied to the mixer 34A. The delayed PSK continuous wave is supplied as a reference signal via the π / 2 shift circuit 26A.

  The π / 2 shift circuit 26A is a circuit that shifts the phase of the delayed reference signal by π / 2. As a result, an in-phase signal component (I signal component) is output from one of the two mixers 32A and 34A, and a quadrature signal component (Q signal component) is output from the other. The high-frequency components of the in-phase signal component and the quadrature signal component are cut by LPFs (low-pass filters) 36A and 38A provided at the subsequent stage of the reception mixer 30A, and a demodulated signal of only a necessary band after detection is obtained. Extracted.

  Adders 46A and 48A add the demodulated signals obtained from LPFs 36A and 38A over a predetermined period. As a result, the code pattern included in the PSK continuous wave is subjected to addition processing, and a demodulated signal from a target position that matches the code pattern of the reference signal is selectively extracted. This position selectivity will be described in detail later.

  On the other hand, the reception mixer 30B is a circuit that performs quadrature detection on the reception RF signal output from the BPF 18B, that is, the reception RF signal corresponding to the PSK continuous wave of the carrier B, and generates a complex baseband signal. It comprises two mixers 32B and 34B. The PSK continuous wave of the carrier wave B output from the PSK modulation processing unit 22B is subjected to delay processing in the delay circuit 25B, and the delayed PSK continuous wave is directly supplied to the mixer 32B as a reference signal, while being supplied to the mixer 34B. The delayed PSK continuous wave is supplied as a reference signal via the π / 2 shift circuit 26B.

  As a result, an in-phase signal component (I signal component) is output from one of the two mixers 32B and 34B, and a quadrature signal component (Q signal component) is output from the other, and detection is performed by LPFs (low-pass filters) 36B and 38B. A demodulated signal of only the necessary band later is extracted. Adders 46B and 48B add the demodulated signals obtained from LPFs 36B and 38B over a predetermined period. As a result, the code pattern included in the PSK continuous wave is subjected to addition processing, and a demodulated signal from a target position that matches the code pattern of the reference signal is selectively extracted. This position selectivity will be described in detail later.

  The combining unit 52I adds the signal obtained from the adding unit 46A and the signal obtained from the adding unit 46B to obtain a combined demodulated signal (in-phase signal component). On the other hand, combining section 52Q adds the signal obtained from adding section 48A and the signal obtained from adding section 48B to obtain a combined demodulated signal (orthogonal signal component). The selectivity of the target position is enhanced by the synthesis process in the synthesis units 52I and 52Q. This improvement in position selectivity will be described in detail later.

  The FFT processing unit (fast Fourier transform processing unit) 54 performs an FFT operation on each demodulated signal (in-phase signal component and quadrature signal component) obtained from the combining units 52I and 52Q. As a result, the demodulated signal is converted into a frequency spectrum in the FFT processing unit 54. The frequency spectrum output from the FFT processing unit 54 is output as frequency spectrum data having a frequency resolution δf depending on circuit setting conditions and the like.

  The Doppler information analysis unit 56 extracts a Doppler signal from the demodulated signal converted into a frequency spectrum. As will be described in detail later, in the ultrasonic diagnostic apparatus of FIG. 1, the target position is set by the delay processing in the delay circuits 25 </ b> A and 25 </ b> B, and the Doppler information analysis unit 56 selectively extracts the Doppler signal from the target position. For example, the Doppler information analysis unit 56 forms a display waveform of a Doppler signal that changes with time. In addition, Doppler signals are extracted for each depth (each position) in the living body, and for example, the velocity of the tissue in the living body is calculated for each depth on the ultrasonic beam (sound ray) and output in real time. May be. Alternatively, the velocity of each position of the in-vivo tissue may be calculated two-dimensionally or three-dimensionally by scanning an ultrasonic beam.

  The display unit 58 displays the waveform of the Doppler signal formed in the Doppler information analysis unit 56. Each unit in the ultrasonic diagnostic apparatus shown in FIG. 1 is controlled by the system control unit 60. That is, the system control unit 60 performs transmission control, reception control, display control, and the like.

  As described above, in the ultrasonic diagnostic apparatus of FIG. 1, a reception signal is obtained by transmitting and receiving ultrasonic waves using a continuous wave transmission signal (OFDM continuous wave) obtained by multiplexing two PSK continuous waves. The delay relationship between the reference signal and the received signal is adjusted according to the depth of the target position in the living body, and the demodulation process is performed by strengthening the correlation between the received signal from the target position and the reference signal. As a result, Doppler information from the target position is selectively extracted. Therefore, the principle of selectively extracting the PSK modulation processing and the Doppler information from the target position in the ultrasonic diagnostic apparatus of FIG. In addition, about the part (structure) shown in FIG. 1, the code | symbol of FIG. 1 is utilized also in the following description.

  In the ultrasonic diagnostic apparatus of FIG. 1, phase shift keying (PSK) is performed using two codes (code strings) that are complementary to each other. That is, the code A is used in the PSK modulation processing unit 22A, the code B is used in the PSK modulation processing unit 22B, and the code A and the code B are complementary to each other.

  First, the autocorrelation between code A and code B is defined as shown in Equation 1, and a specific code satisfying the condition of Equation 2 is examined.

For example, when n = 2, A ₂ = (1,1) and B ₂ = (1, −1) satisfy the above-described conditions. Specific calculation results based on Equation 1 and Equation 2 are shown in Equation 3.

  The code of n = 4 (4 bits) obtained by extending the code of n = 2 (2 bits) can be obtained from Equation 4.

A code of n = 4 obtained based on A ₂ = (1, 1) and B ₂ = (1, −1) is as shown in Formula 5.

The autocorrelation of the code A ₄ shown in Equation 5 (see Equation 1) is as shown in Equation 6.

Further, the autocorrelation of the code B ₄ shown in the equation 5 is as shown in the equation 7.

As shown in Equation 6 and Equation 7, the autocorrelation value of both the code A ₄ and the code B ₄ is a maximum value 4 when the code deviation is 0 (i = 0). The sum of the correlation values of both codes also has a maximum value of 8 when the code deviation is 0 (i = 0).

  Further, a code of n = 8 (8 bits) obtained by extending a code of n = 4 (4 bits) can be obtained from Equation 8.

A code of n = 8 obtained from A ₄ and B _{4 in} Expression 5 is as shown in Expression 9.

The autocorrelation of the code A ₈ shown in Equation 9 (see Equation 1) is as shown in Equation 10.

Figure 2 is a graph showing the calculation results of the autocorrelation relates to code A _8. When τ (= i) indicating code deviation is 1, 2, 4, 6, 7, the correlation value is 0, when τ is 3, 5, the correlation value is 4, and when τ is 0, 8 The correlation value is 8. Although it is smaller than the maximum value of 8, when τ is 3 or 5, the correlation value is 4 and is not 0.

On the other hand, the autocorrelation of the code B ₈ shown in Equation 9 is as shown in Equation 11.

Figure 3 is a graph showing the calculation results of the autocorrelation relates to code B _8. When τ (= i) indicating code deviation is 1, 2, 4, 6, 7, the correlation value is 0, and when τ is 3, 5, the correlation value is −4, and τ is 0,8. In this case, the correlation value is 8. When τ is 3 or 5, the correlation value is −4, and the absolute value is equal and the polarity is reversed when compared with the calculation result of the autocorrelation for the code A ₈ (FIG. 2).

Therefore, when the sum of the correlation values of both the codes A ₈ and B ₈ is calculated, the sum of the correlation values is the maximum value when τ indicating the code deviation is 0, 8, that is, when there is no code deviation. When τ is another value, that is, when there is a code shift, the sum of correlation values is always 0. As described above, the code A ₈ and the code B ₈ are in a complementary relationship in which the correlation values cancel each other when the codes are shifted.

Note that codes of n = 8 that are complementary to each other can also be obtained from the following equation (12). In Equation 12, code B ₄ ^-1 is a code obtained by inverting the arrangement of codes of code B ₄ , and code -A ₄ ^-1 is a code obtained by inverting both the code value and arrangement of code A _4. is there.

From A ₄ and B _{4 in} Formula 5, the code A ₈ obtained based on Formula 12 is the same as that in Formula 9, and the code B ₈ ′ obtained based on Formula 12 is expressed by Formula 13 As shown in the equation.

FIG. 4 is a diagram showing the calculation result of the autocorrelation for the code B ₈ ′. When τ (= i) indicating code deviation is 1, 2, 4, 6, 7, the correlation value is 0, and when τ is 3, 5, the correlation value is −4, and τ is 0,8. In this case, the correlation value is 8. That is, the same result as the calculation result of the autocorrelation for the code B ₈ (FIG. 3) is obtained, and it can be seen that the combination of the code A ₈ and the code B ₈ ′ is also complementary to each other.

  Further, the code of n = 16 (16 bits) obtained by extending the code of n = 8 (8 bits) can be obtained from Expression 14 or Expression 15.

  Furthermore, codes of 32 bits, 64 bits, 128 bits, 256 bits, etc. can be extended by applying the same rule. A general expression corresponding to Expression 14 is expressed as Expression 16, and a general expression corresponding to Expression 15 is expressed as Expression 17. N in Equation 16 and Equation 17 is the number of bits (code length) included in the code and is a power of 2.

In the ultrasonic diagnostic apparatus of FIG. 1, a code A _N and a code B _N (obtained by using Formula 16 or Formula 17 from A ₂ = (1,1) and B ₂ = (1, −1). Or the code B _N ′) is used. That is, the code A _N is used in the PSK modulation processing unit 22A, and the code B _N (or code B _N ′) is used in the PSK modulation processing unit 22B. Then, in order to transmit and receive the two PSK continuous waves output from the PSK modulation processing units 22A and 22B simultaneously (in parallel), the multiplexing processing unit 24 performs multiplexing processing by orthogonal frequency division multiplexing (OFDM). . In the OFDM (Orthogonal Frequency Division Multiplex) method, a plurality of digital modulation signals are transmitted and received using a plurality of carriers having orthogonal frequency spectra.

  FIG. 5 is a diagram showing a time change of a carrier wave used for OFDM. Carrier A in FIG. 5 is an example of a signal output from carrier A generator 20A, and carrier B in FIG. 5 is an example of a signal output from carrier B generator 20B.

The 1-bit length τ is a 1-bit period of a code (code A _N or the like) used for PSK modulation processing. The frequencies of the carrier A and the carrier B are set to an integer multiple of the bit rate 1 / τ (an integer multiple of the frequency f ₀ ).

FIG. 6 is a diagram showing a frequency spectrum of a PSK continuous wave used for OFDM. The PSK continuous wave (solid line) of carrier A shown in FIG. 6 is a continuous wave obtained by PSK modulation processing of carrier A using a code (for example, code A _N ) with a bit rate of 1 / τ, and is shown in FIG. The PSK continuous wave of carrier B (broken line) is a continuous wave obtained by performing PSK modulation processing on carrier B using a code (for example, code B _N ) with a bit rate of 1 / τ.

The center frequency 2f ₀ related to the PSK continuous wave of the carrier A and the center frequency 3f ₀ related to the PSK continuous wave of the carrier B are separated from each other by 1 / τ (= f ₀ ). Then, at the position of the center frequency 2f ₀ about PSK continuous wave carrier A, next the frequency spectrum 0 about PSK continuous wave carrier B, also at the position of the center frequency 3f ₀ about PSK continuous wave carrier B, the carrier A The frequency spectrum for the PSK continuous wave is zero.

  In OFDM, for example, as shown in FIG. 6, the frequency spectrum of other carriers becomes 0 at the position of the center frequency of each carrier, and it is possible to multiplex a plurality of carriers with extremely small interference between carriers. Become. That is, the two PSK continuous waves output from the PSK modulation processing units 22A and 22B are multiplexed by the OFDM method in the multiplexing processing unit 24, thereby suppressing the mutual interference between the two PSK continuous waves (preferably complete). Two PSK continuous waves can be transmitted and received in parallel.

FIG. 7 is a diagram for explaining a transmission / reception process process of two PSK continuous waves. Figure 7 shows an example using 4-bit code A _4. PSK modulation processing section 22A performs a two-phase phase-shift keying (PSK) modulation processing on the carrier A by using the code _{A 4.} That is, the PSK modulation processing unit 22A inverts the phase of the carrier A in the bit period of the code “−1” while keeping the phase of the carrier A in the bit period of the code A ₄ of “1” (only π). Shift). Thereby, the PSK continuous wave of the carrier A is obtained. The received signal A shown in FIG. 7 is a received signal from the target position obtained by transmitting and receiving the PSK continuous wave of the carrier wave A. The reception signal A is included in the reception signal that has passed through the BPF 18A.

The reference signal A shown in FIG. 7 is a signal obtained by delaying the PSK continuous wave of the carrier wave A output from the PSK modulation processing unit 22A in the delay circuit 25A. 7, the reference signal A, the phase pattern (pattern of code A ₄₎ is, to match the phase pattern of the received signal A from the target position, are delay processing by the delay circuit 25A.

  The multiplier output A shown in FIG. 7 corresponds to the multiplication result of the reference signal A and the reception signal A, and the signal component from the target position included in the signal obtained in the reception mixer 30A (for example, the signal output from the mixer 32A). It is. Since there is no delay time difference between the reference signal A and the received signal A, the multiplier output A includes a frequency component that is twice the carrier wave and a baseband signal A in the vicinity of direct current indicated by a broken line. For example, the LPF 36A removes the double frequency component of the carrier wave and extracts the baseband signal A near the direct current. The Doppler signal is detected as a low frequency signal near DC.

On the other hand, the PSK modulation processing unit 22B performs two-phase phase shift keying (PSK) modulation processing on the carrier B using, for example, the code B ₄ that is complementary to the code A ₄ . Thereby, the PSK continuous wave of the carrier wave B is obtained. The received signal B shown in FIG. 7 is a received signal obtained by transmitting and receiving the PSK continuous wave of the carrier B.

  The multiplier output B shown in FIG. 7 corresponds to the multiplication result of the reference signal A and the reception signal B, and the baseband signal B, which is the difference in frequency between the reference signal A and the reception signal B, and the reference signal A and the reception signal B. The sum component of the frequency is included. However, since the reference signal A corresponding to the carrier A and the reception signal B corresponding to the carrier B are different in center frequency by 1 / τ, the multiplier output B includes a baseband signal near DC. Absent. That is, in the multiplication process using the reference signal A, the baseband signal A near the direct current is extracted from only the received signal A corresponding to the carrier wave A. Therefore, the BPF 18A may be omitted as necessary.

  In this way, a signal obtained by multiplexing the PSK continuous wave of the carrier A and the PSK continuous wave of the carrier B, which are orthogonal to each other, is transmitted / received by the OFDM method, and only the received signal A corresponding to the carrier A is processed by the reception mixer 30A. A baseband signal A near the direct current can be extracted. Similarly to the processing in the reception mixer 30A, the baseband signal near the direct current is extracted from only the reception signal corresponding to the carrier wave B (the reception signal from the target position) by the processing in the reception mixer 30B. That is, the two PSK continuous waves output from the PSK modulation processing units 22A and 22B are transmitted simultaneously (in parallel), and the two received signals associated with the two PSK continuous waves are processed individually and in parallel. can do. Therefore, the position selectivity regarding each of the two PSK continuous waves will be described.

FIG. 8 is a diagram for explaining PSK modulation processing and position selectivity for carrier wave A. Figure 8 is example that a two-phase phase-shift keying (PSK) process is illustrated for carrier A with a 8-bit code A _8.

PSK modulation processing section 22A performs a two-phase phase-shift keying (PSK) modulation processing on the carrier A by using the code _{A 8.} As a result, the phase of the PSK continuous wave is adjusted as shown in FIG. FIG. 8 also shows the phase of the received signal obtained using this PSK continuous wave.

  In the present embodiment, the reference signal obtained by delaying the phase of the PSK continuous wave in the delay circuit 25A is multiplied by the reception signal in the reception mixer 30A. FIG. 8 shows the phase difference between the reference signal and the received signal and the multiplication result (multiplier voltage) of the reference signal and the received signal when the phase of the reference signal is changed from φ1 to φ9.

  For example, the phase (φ2) of the reference signal is a reference signal obtained by delaying the phase of the PSK continuous wave by one bit period. The multiplier voltage is a voltage obtained from the phase difference between the reference signal and the received signal, for example, based on the characteristics of the general-purpose phase detector shown in FIG.

  The sum shown in FIG. 8 is a total value of the multiplier voltages within the 8-bit period, and is obtained, for example, in each of the adders 46A and 48A. As shown in FIG. 8, in the case of the phase of the reference signal (φ1, φ9) that completely matches the phase of the received signal, the multiplier voltage is always 1 and the total value is the maximum value 8. On the other hand, in the case of the phase of the reference signal (φ2 to φ8) that does not match the phase of the received signal, the multiplier voltage changes randomly and the total value becomes 0 or 4.

  Therefore, the received signal from the target position can be selectively extracted by matching the phases of the received signal obtained from the target position and the reference signal. For example, by setting the delay time in the delay circuit 25A as the propagation time of the ultrasonic wave to and from the target position, the phase of the received signal obtained from the target position and the reference signal are matched, and the received signal from the target position is selectively selected. Can be extracted. Instead of the adders 46A and 48A, a random change in the multiplier voltage may be eliminated by using, for example, a low-pass filter, and the multiplier voltage that is always 1 shown in FIG. 8 may be extracted.

  As shown in FIG. 8, in the case of the phase (φ4, φ6) of the reference signal, the total value of the multiplier voltages is 4. In other words, a received signal from a position other than the target position is selected, although it is smaller than the selectivity (maximum value 8) regarding the received signal from the target position. In order to suppress the selectivity of the received signal from other than the target position, the PSK continuous wave of the carrier wave B is used together.

FIG. 9 is a diagram for explaining PSK modulation processing and position selectivity for carrier wave B. FIG. 9 shows an example in which a two-phase phase shift keying (PSK) modulation process is performed on a carrier B using an 8-bit code B ₈ that is complementary to an 8-bit code A _8. Yes.

The PSK modulation processing unit 22B adjusts the phase of the PSK continuous wave by performing two-phase phase shift keying (PSK) modulation processing on the carrier wave B using the code B ₈ as shown in FIG. . FIG. 9 also shows the phase of the received signal obtained using this PSK continuous wave.

  Then, the reference signal obtained by delaying the phase of the PSK continuous wave in the delay circuit 25B is multiplied by the reception signal in the reception mixer 30B. FIG. 9 shows the phase difference between the reference signal and the received signal and the multiplication result (multiplier voltage) of the reference signal and the received signal when the phase of the reference signal is changed from φ1 to φ9. The multiplier voltage is a voltage obtained from the phase difference between the reference signal and the received signal, for example, based on the characteristics of the general-purpose phase detector shown in FIG.

  The total shown in FIG. 9 is a total value of the multiplier voltages in the 8-bit period, and is obtained, for example, in each of the addition units 46B and 48B. Instead of the adders 46B and 48B, a random change in the multiplier voltage may be eliminated by using, for example, a low-pass filter.

  As shown in FIG. 9, in the case of the phase (φ1, φ9) of the reference signal that completely matches the phase of the received signal, the multiplier voltage is always 1 and the total value is the maximum value 8. In the case of the phase (φ4, φ6) of the reference signal, the total value of the multiplier voltages is −4, and compared with the total value obtained from the carrier A (FIG. 8), the absolute values are equal and the polarity is reversed. It has become.

  Therefore, when the sum of the total value obtained from the carrier wave A and the total value obtained from the carrier wave B is calculated, the sum of the total value becomes the maximum value 16 in the case of the phase (φ1, φ9) of the reference signal, and the phase of the reference signal ( In the case of φ2 to φ8), the sum of the total values is always 0. The addition of the total value is executed in each of the combining units 52I and 52Q.

The addition of the total value shown in FIG. 8 and the total value shown in FIG. 9 corresponds to the sum of the correlation values of both the codes A ₈ and B ₈ described with reference to FIGS. That is, by using code A ₈ and code B ₈ that are complementary to each other to form two PSK continuous waves, and using a continuous wave transmission signal obtained by multiplexing these two PSK continuous waves by the OFDM method, The component of the received signal obtained from other than the target position can be made extremely small, preferably completely zero, and the range side lobe is significantly reduced.

8 and 9, the PSK modulation processing and the position selectivity have been described using the 8-bit code A ₈ and the code B _8. However, in the realization of the device, the number of bits included in the code (code) It is desirable that the (long) is extended to, for example, 256 bits. The position resolution can be increased by increasing the number of bits and reducing the 1-bit period.

FIG. 11 is a diagram for explaining extraction of a Doppler signal. FIG. 11A shows a frequency spectrum of a PSK continuous wave including a transmission signal. The frequency f ₀ is the frequency of the RF signal (carrier signal). The frequency interval of the sideband spreading around the frequency f _{0 of the} RF signal is the repetition frequency f _p of the code A _N and the code B _N. In addition, there is a so-called null point where the power of the sideband spreading around the frequency f ₀ is 0 (zero). The frequency interval from the frequency f ₀ to the null point is the reciprocal of the 1-bit time interval T between the code A _N and the code B _N.

  FIG. 11B shows the frequency spectrum of the PSK continuous wave included in the received signal. The received signal has the same waveform as the transmitted signal when attenuation in the living body is ignored. Therefore, the frequency spectrum of the reception signal shown in FIG. 11B is almost the same as the frequency spectrum of the transmission signal shown in FIG. However, the phase differs between the transmission signal and the reception signal according to the propagation time of the ultrasonic wave in the living body.

  In this embodiment, the PSK continuous wave is subjected to delay processing to form a reference signal, and the reception mixer (reference numerals 30A and 30B in FIG. 1) uses the reference signal to perform mixer processing (reference signal) on the received signal. And the received signal).

  FIG. 11C shows the frequency spectrum of the demodulated signal obtained by the mixer processing. The demodulated signal in FIG. 11C corresponds to the multiplication result of the reference signal and the received signal when the correlation is maximum. That is, the multiplication result between the received signal from the target position and the reference signal whose phase is matched to the depth of the target position is the demodulated signal in FIG.

The demodulated signal shown in FIG. 11C includes a DC signal component and a harmonic component that is twice the frequency f ₀ of the RF signal. The Doppler signal appears in a form attached to these components. In the LPF (reference numerals 36A, 38A, 36B, and 38B in FIG. 1), the harmonic component is cut and only the DC signal component is extracted. Of the DC signal component shown and the harmonic component twice the frequency f ₀ , only the frequency spectrum of the DC signal component is formed.

  Then, the Doppler information analysis unit 56 extracts the Doppler signal from the frequency spectrum of the DC signal component shown in FIG. 11C, and calculates the flow velocity of the blood flow existing at the target position based on the Doppler shift amount and the like. . Since quadrature detection is performed in the receiving mixer (reference numerals 30A and 30B in FIG. 1), the polarity of the flow velocity can also be determined. Alternatively, the clutter signal may be extracted from the frequency spectrum of the DC signal component, and the position of the blood vessel wall existing at the target position may be calculated.

  In the above-described embodiment, phase shift keying (PSK) is used when digitally modulating a continuous wave. Instead of this PSK, frequency shift keying (FSK) may be used as a digital modulation method. Alternatively, digitally modulated continuous wave data may be stored in a memory or the like, and the continuous wave may be generated based on data read from the memory.

  Further, in FIG. 1, the delay circuits 25A and 25B, the π / 2 shift circuits 26A and 26B, the reception mixers 30A and 30B, the low-pass filters 36A to 38B, the addition units 46A to 48B, and the synthesis units 52I and 52Q are independent. However, the Doppler information may be extracted in parallel from a plurality of target positions along the ultrasonic beam by providing a plurality of receiving system circuits according to the plurality of target positions.

  The preferred embodiments of the present invention have been described above, but the above-described preferred embodiments of the present invention are merely examples in all respects, and do not limit the scope of the present invention. The present invention includes various modifications without departing from the essence thereof.

  22A, 22B PSK modulation processing unit, 24 multiplexing processing unit, 25A, 25B delay circuit, 30A, 30B reception mixer, 54 FFT processing unit, 56 Doppler information analysis unit.

Claims (9)

A transmission signal processing unit that outputs a continuous wave transmission signal obtained by multiplexing two carrier signals digitally modulated based on two periodic signal sequences that are complementary to each other;
An ultrasonic transmission / reception unit that obtains a reception signal by transmitting an ultrasonic wave corresponding to the transmission signal to a living body and receiving the ultrasonic wave from the living body;
A received signal processing unit that obtains a demodulated signal corresponding to the target position by performing demodulation processing on the received signal while adjusting the correlation between the target position in the living body;
An in-vivo information extraction unit for extracting in-vivo information from the demodulated signal corresponding to the target position;
Having
An ultrasonic diagnostic apparatus. The ultrasonic diagnostic apparatus according to claim 1,
The transmission signal processing unit periodically repeats a first carrier signal digitally modulated based on a periodic signal sequence that repeats a first code sequence _N having a code length N and a second code sequence _N having a code length N A continuous wave transmission signal that is multiplexed with a second carrier signal that has been digitally modulated based on a simple signal sequence;
An ultrasonic diagnostic apparatus. The ultrasonic diagnostic apparatus according to claim 2,
The code length N is a power of 2;
Each of the first sign row _N and the second tally row _N is formed based on the first sign row _{N / 2} to the second code sequence _{N / 2,}
An ultrasonic diagnostic apparatus. The ultrasonic diagnostic apparatus according to claim 3.
The first code sequence _N is a code sequence A _N.
The second sign column _N is a code sequence _{B N,}
The code string A _N is a code string obtained by connecting the code string A _{N / 2} and the code string B _{N / 2} in series.
The code string B _N is a code string obtained by connecting the code string A _{N / 2} and the code string -B _{N / 2} in series.
The code string -B _{N / 2} is a code string obtained by inverting the code value of the code string B _{N / 2} .
An ultrasonic diagnostic apparatus. The ultrasonic diagnostic apparatus according to claim 4,
Each of the code string A _N and the code string B _N is formed by repeating the serial connection from a code string A ₂ = (1, 1) and a code string B ₂ = (1, −1) having a code length of 2. To be
An ultrasonic diagnostic apparatus. The ultrasonic diagnostic apparatus according to claim 3.
The first code sequence _N is a code sequence A _N.
The second code sequence _N is a code sequence B _N ′,
The code string A _N is a code string obtained by connecting the code string A _{N / 2} and the code string B _{N / 2} in series.
The code string B _N ′ is a code string obtained by connecting the code string B _{N / 2} ⁻¹ and the code string −A _{N / 2} ⁻¹ in series.
The code string B _{N / 2} is a code string obtained by connecting the code string A _{N / 4} and the code string -B _{N / 4} in series.
The code string B _{N / 2} ⁻¹ is a code string obtained by inverting the arrangement of the codes of the code string B _{N / 2} .
The code string -A _{N / 2} ^-1 is a code string obtained by inverting both the code value and the sequence of the code string A _{N / 2} .
The code string -B _{N / 4} is a code string obtained by inverting the code value of the code string B _{N / 4} .
An ultrasonic diagnostic apparatus. The ultrasonic diagnostic apparatus according to claim 6,
Each of the code string A _N and the code string B _N ′ is obtained by repeating the series connection from the code string A ₂ = (1, 1) and the code string B ₂ = (1, −1) having the code length 2. It is formed,
An ultrasonic diagnostic apparatus. The ultrasonic diagnostic apparatus according to any one of claims 2 to 7,
The transmission signal processing unit includes a first carrier signal whose phase is changed by phase shift keying based on a periodic signal sequence that repeats the first code sequence _N, and a periodic signal that repeats the second code sequence _N Outputting a continuous wave transmission signal multiplexed with a second carrier signal whose phase has been changed by phase shift keying based on the sequence;
An ultrasonic diagnostic apparatus. The ultrasonic diagnostic apparatus according to any one of claims 2 to 8,
The transmission signal processing unit outputs a continuous wave transmission signal formed by multiplexing the first carrier signal and the second carrier signal by orthogonal frequency division multiplexing (OFDM).
An ultrasonic diagnostic apparatus.

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