JP2011198795A - Led driving device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide an LED driving device that improves accuracy of balance of current flowing in an LED.SOLUTION: The LED driving device has: a resonance-type power supply unit that has a series connection including a first transformer T2, first and second switching elements QH, QL, a primary winding Np, and a resonance capacitor Cri; a series circuit including first rectifying and smoothing circuits C1, D1 connected to the output of the power supply unit, a first LED group LD1 connected to the first rectifying and smoothing circuits and in which one or more LEDs are connected in series, and a first winding N1 connected between the power supply unit and the first rectifying and smoothing circuits; a series circuit including second rectifying and smoothing circuits C2, D2 connected to the output of the power supply unit, a second LED group LD2 connected to the second rectifying and smoothing circuits and in which one or more LEDs are connected in series, and a second winding S1 connected between the power supply unit and the second rectifying and smoothing circuits; and a control unit 1 for controlling switching frequencies of the QH and QL to a frequency higher than the resonance frequency of the power supply unit. First and second currents flowing in each series circuit are balanced on the basis of electromagnetic force generated in the first/second windings.

Description

  The present invention relates to an LED driving device that drives a plurality of LEDs connected in series.

  Conventionally, for example, Patent Document 1 is known as an LED driving device that lights a plurality of LEDs (Light Emitting Diodes) connected in series.

  Patent Document 1 discloses a light source circuit that supplies current using a plurality of transformers and drives a plurality of LEDs connected to a plurality of current paths, as shown in FIG. Each transformer consists of one current supply coil and an induction coil, which is used to provide the output current to different current paths, and the induction coil is connected to the induction coil of another transformer to form a current loop. Form. The relationship between the output currents of each transformer and another transformer is determined by the turns ratio of the coils of the transformers connected to each other. That is, when the coil turns ratio is 1: 1, the current flowing through each current path can be balanced.

JP 2006-352116 A

  However, the LED driving device disclosed in Patent Document 1 cannot accurately balance a plurality of LED currents connected in parallel. That is, current balance accuracy was not sufficient.

  The subject of this invention is providing the LED drive device which can improve the precision of the balance of the electric current which flows into several LED connected in parallel.

  In order to solve the above-described problem, an LED drive device according to the present invention is an LED drive device that rectifies and smoothes an alternating current and supplies it to an LED, and includes a first transformer having a primary winding and a secondary winding, First and second switching elements connected in series to both ends of a DC power source, and having a series connection portion connected in parallel to one of the first and second switching elements and having the primary winding and a resonant capacitor A resonance type power supply means for outputting the alternating current from the secondary winding; a first rectification smoothing circuit having a first rectifying element and a first smoothing element connected to the output of the power supply means; A first LED group connected to a rectifying / smoothing circuit and having at least one LED connected in series; a first series circuit having a first winding connected between the power supply means and the first rectifying / smoothing circuit; , Output of the power supply means A second rectifying / smoothing circuit having a second rectifying element and a second smoothing element connected; a second LED group connected to the second rectifying / smoothing circuit and connected in series with one or more LEDs; the power supply means; A second series circuit having a second winding connected between the two rectifying and smoothing circuits, and a control for controlling a switching frequency of the first and second switching elements to a frequency higher than a resonance frequency of the power supply means; And a first current and a second current flowing through each of the first and second series circuits are balanced based on electromagnetic force generated in the first and second windings.

  The first and second windings have inductance components connected to each of the first and second windings, and the first and second windings constitute a second transformer for balancing the first and second currents. The component is the leakage inductance of the second transformer.

  According to the present invention, since the control means controls the switching frequency of the first and second switching elements to be higher than the resonance frequency of the power supply means, the first and second currents flowing through the first and second series circuits, respectively. Two currents are balanced based on the electromagnetic force generated in the first and second windings. Therefore, it is possible to improve the balance accuracy of the currents flowing through the plurality of LEDs.

  In addition, since the inductance component connected to each of the first and second windings is the leakage inductance of the second transformer, the variation in current is suppressed by this leakage inductance, and the accuracy of balance of the current flowing through the plurality of LEDs is suppressed. Can be improved.

It is a block diagram of the LED drive device of Example 1 of this invention. It is a timing chart for demonstrating operation | movement of the LED drive device of Example 1 of this invention. It is a timing chart for demonstrating operation | movement of the LED drive device of Example 1 of this invention. FIG. 6 is a configuration diagram of an LED drive device according to a modification of Example 1. It is a figure which shows the relationship between the leakage inductance and the dispersion | variation in an electric current. It is a figure which shows the half wave voltage doubler rectification system of Example 1, and another full wave rectification system. It is a figure which shows the comparison result of the dispersion | variation in the electric current of each rectification system. It is a principal block diagram of the LED drive device of Example 2 of this invention. It is a figure which shows the result of having compared the dispersion | variation in the electric current at the time of adding the coil of the LED drive device of Example 2, and utilizing the leakage inductance of a transformer. It is a figure which shows the specific example of the conventional LED drive device.

  Hereinafter, an LED drive device according to an embodiment of the present invention will be described in detail with reference to the drawings. The present invention uses a resonant converter as the converter, controls the switching frequency of the switching element to be higher than the resonant frequency of the resonant converter, and configures the current balance transformer to include a large leakage inductance, thereby rectifying As a method, a half-wave voltage doubler rectification method is adopted.

  FIG. 1 is a configuration diagram of an LED driving apparatus according to Embodiment 1 of the present invention. In the power supply means 10, in order to supply a sinusoidal alternating current, a series circuit of a switching element QH made of a MOSFET and a switching element QL made of a MOSFET is connected to both ends of the DC power supply Vin.

  A series resonance circuit of the primary winding Np of the transformer T2 and the current resonance capacitor Cri is connected to a connection point between the switching element QH and the switching element QL. The transformer T2 has leakage inductances Lr1 and Lr2. Lp is the exciting inductance of the transformer T2. By alternately turning on and off the switching element QL and the switching element QH, a sinusoidal alternating current resonated by the leakage inductances Lr1 and Lr2 and the current resonance capacitor Cri or Lr1, Lp and Cri is supplied from the winding Ns of the transformer T2. be able to.

  One end of the secondary winding Ns is connected to one end of the winding N1, and the other end of the winding N1 is connected to the anode of the diode D1 for half-wave rectification of the alternating current and the cathode of the diode D11 via the capacitor C11. Is done. A capacitor C1 is connected between the cathode of the diode D1 and the other end of the secondary winding Ns, and a load LD1 (LED1a to LED1e) is connected in parallel to one end of the capacitor C1. The anode of the diode D11 is connected to the other end of the secondary winding Ns and the other end of the capacitor C1. Capacitors C1 and C11 and diodes D1 and D11 constitute a first half-wave double voltage rectifier circuit. The secondary winding Ns and the first half-wave voltage doubler rectifier circuit form a series circuit, and the load LD1 (LED1a to LED1e) is connected to the output of the first half-wave voltage doubler rectifier circuit.

  One end of the winding S1 is connected to one end of the secondary winding Ns of the transformer T2, and the other end of the winding S1 is connected to the anode of the diode D2 and the diode D12 that rectifies the alternating current through the capacitor C12. To the cathode. A capacitor C2 is connected between the cathode of the diode D2 and the other end of the secondary winding Ns, and a load LD2 (LED2a to LED2e) is connected in parallel to one end of the capacitor C2. The anode of the diode D12 is connected to the other end of the secondary winding Ns and the other end of the capacitor C2. Capacitors C2 and C12 and diodes D2 and D12 constitute a second half-wave double voltage rectifier circuit. The secondary winding Ns and the second half-wave voltage doubler rectifier circuit form a series circuit, and the load LD2 (LED2a to LED2e) is connected to the output of the second half-wave voltage doubler rectifier circuit.

  Winding N1 and winding S1 are electromagnetically coupled to each other to form transformer T1. The transformer T1 has an exciting inductance L1. Further, the impedance of the load LD1 and the impedance of the load LD2 in the first embodiment are different from each other due to variations in the forward voltage (Vf) of the LED.

  Further, one end of the resistor Rs and one end of the input terminal of the PFM circuit 1 are connected to the load LD1 (LD2), and the other end of the resistor Rs and the other end of the input terminal of the PFM circuit are grounded. The resistor Rs collectively detects the current flowing through the load LD1 (and LD2) and outputs a current detection value to the PFM circuit 1. The PFM circuit 1 compares the current detection value with the internal reference voltage, and controls the on / off frequency of the switching element QH and the switching element QL based on the error output so that the current flowing through the load is constant.

(Relationship between switching frequency and resonance frequency)
Next, the operation of the LED driving device of Example 1 will be described. First, as a comparative example, the operation when the switching frequency of the switching elements QH and QL is lower than the resonance frequency of the power supply means 10 will be described with reference to the timing chart shown in FIG.

  The resonance frequency of the power supply means 10 is determined by the excitation inductance, leakage inductance, and current resonance capacitor Cri of the transformer T2.

  Here, the value of the leakage inductance Lr2 of the transformer T2 is set to 100 μH, for example. That is, by reducing the value of the leakage inductance Lr2 of the transformer T2, the switching frequency is set to about 0.7 times the resonance frequency due to the leakage inductance (Lr1 + Lr2) and the current resonance capacitor Cri.

  Here, Vf (LD1)> Vf (LD2), where Vf (LD1) is the sum of forward voltages Vf of LEDs 1a to LED1e, and Vf (LD2) is the sum of forward voltages Vf of LEDs 2a to LED2e. It is assumed that

  In FIG. 2, V (QH) and V (QL) are voltages between the drain and source of the switching elements QH and QL, and I (QH) and I (QL) are currents flowing between the drain and source of the switching elements QH and QL. , V (Ns) is the voltage of the secondary winding Ns of the transformer T2, I (D1) and I (D11) are currents flowing through the diodes D1 and D11, and I (D1) -I (D2) are the diodes D1 and D2. I (L1) is a current flowing through the exciting inductance L1 of the transformer T1.

  The operating state is divided into six periods T11 to T16 depending on the on / off state of the switching elements QH and QL and the state of the voltage applied to the transformers T1 and T2.

  First, in the period T11, the switching element QH is off and the switching element QL is on, and the current on the primary side of the transformer T2 is Cri → Lp (Np → Lr2) → Lr1 → QL → Cri or Cri → QL. It flows through the route of (DL) → Lr1 → Lp (Np → Lr2) → Cri. The secondary current flows through a path of Ns → D11 → C11 → N1 (L1) → Ns and a path of Ns → D12 → C12 → S1 → Ns.

For the voltages of the two paths,
V (Ns) -V (N1) + V (C11) = 0
V (Ns) + V (S1) + V (C12) = 0
Is established. Further, the voltages V (N1) and V (S1) of the windings N1 and S1 of the transformer T1 are V (N1) = V (S1) due to the characteristics of the transformer. Therefore, when the voltage V (L1) = V (N1) = V (S1) of the excitation inductance L1 of the transformer T1,
V (Ns) =-(V (C11) + V (C12)) / 2
V (L1) = (V (C11) -V (C12)) / 2
Thus, a voltage half the difference between the voltage of the capacitor C11 and the voltage of the capacitor C12 is applied to the excitation inductance L1 of the transformer T1. In the voltage doubler rectification method, when the output voltage V (Ns) of the power supply means is 0.5 and the polarity is reversed symmetrically as shown in FIG.
V (L1) = (V (C1) -V (C2)) / 4
It becomes.

  When Vf (LD1)> Vf (LD2), since V (C1)> V (C2), V (L1) has a positive voltage at the beginning of winding. For this reason, the excitation current I (L1) flowing through the excitation inductance L1 increases with a slope of V (L1) / L1.

  Next, in the period T12, the switching element QH is turned off and the switching element QL is turned on as in the period T11. However, the resonance frequency changes compared to the period T11, and the inductance (Lr1 + Lp), the current resonance capacitor Cri, It becomes the resonance frequency by. The current on the primary side of the transformer T2 is the same as the current in the period T11 and flows through a path of Cri → Lp (Np → Lr2) → Lr1 → QL → Cri. The secondary current flows through a path of Ns → D12 → C12 → S1 → Ns and a path of L1 → N1 → L1. The current flowing on the secondary side during this period is the current that is stored in the exciting inductance L1, and when the exciting current becomes zero, this period T12 ends.

  Next, in the period T13, as in the periods T11 and T12, the switching element QH is off and the switching element QL is on. On the primary side of the transformer T2, current flows through a path of Cri → Lp → Lr1 → QL → Cri, and no current flows through Np and Ns of the transformer T2. On the secondary side, no current flows through any of the diodes D1, D2, D11, and D12, and no current flows through the exciting inductance L1.

  Next, in the period T14, the switching element QH is on and the switching element QL is off, and the current on the primary side of the transformer T2 is Vin → Cri → Lp (Lr2 → Np) → Lr1 → QH (DH). → Vin or Vin → QH → Lr1 → Lp (Lr2 → Np) → Cri → Vin The secondary-side current flows through a path of Ns → N1 (L1) → C11 → D1 → C1 (LD1) → Ns and a path of Ns → S1 → C12 → D2 → C2 (LD2) → Ns.

The voltage of the two paths is
V (Ns) -V (N1) + V (C11) -V (C1) = 0
V (Ns) + V (S1) + V (C12) -V (C2) = 0
It is. Further, the voltages V (N1) and V (S1) of the windings N1 and S1 of the transformer T1 are V (N1) = V (S1) due to the characteristics of the transformer. Therefore, when the voltage V (L1) = V (N1) = V (S1) of the excitation inductance L1 of the transformer T1,
V (Ns) = (V (C1) + V (C2) -V (C11) -V (C12)) / 2
V (L1) = (V (C2) -V (C1)) / 2+ (V (C11) -V (C12)) / 2
It becomes. Now, when the output voltage V (Ns) of the power supply means 10 is reversed with a time ratio of 0.5 as shown in FIG.
V (C11) = V (C1) / 2
V (C12) = V (C2) / 2
Because
V (L1) =-(V (C1) -V (C2)) / 4
It becomes.

  When Vf (LD1)> Vf (LD2), since V (C1)> V (C2), V (L1) has a negative voltage at the beginning. For this reason, the exciting current I (L1) flowing through the exciting inductance L1 decreases with a slope of | V (L1) / L1 |.

  Next, in the period T15, as in the period T14, the switching element QH is on and the switching element QL is off. However, the resonance frequency is changed to a resonance frequency by the inductance (Lr1 + Lp) and the current resonance capacitor Cri. The current on the primary side of the transformer T2 flows through a route of Vin → QH → Lr1 → Lp (Lr2 → Lp) → Cri → Vin. On the secondary side, a route of Ns → S1 → C12 → D2 → C2 (LD2) → Ns and a route of L1 → N1 → L1 flow. The current flowing on the secondary side during this period is the discharge of the current stored in the excitation inductance L1, and this period T15 ends when the excitation current becomes zero.

  Next, in the period T16, as in the periods T14 and T15, the switching element QH is on and the switching element QL is off. On the primary side of the transformer T2, a current flows through a path of Vin → QH → Lr1 → Lp → Cri → Vin, and no current flows through Np and Ns of the transformer T2. On the secondary side, as in the period T13, no current flows through any of the diodes D1, D2, D11, and D12, and no current flows through the exciting inductance L1.

  The LED driving device according to the comparative example repeats the above periodic operation. Among the six periods T11 to T16, the diodes D1 and D2 do not conduct in the periods T11, T12, T13, and T16. During the periods T14 and T15, the diodes D1 and D2 are turned on, and the excitation current flowing through the excitation inductance L1 is superimposed on the current I (D1), so that a difference between the current I (D1) and the current I (D2) occurs. Arise.

  Next, as an operation of the LED driving device according to the present invention, an operation when the switching frequency of the switching elements QH and QL is higher than the resonance frequency of the power supply means 10 will be described with reference to a timing chart shown in FIG.

  The PFM circuit 1 (corresponding to the control unit) controls the switching frequency of the switching elements QH and QL to a frequency higher than the resonance frequency of the power supply unit 10. In this case, the value of the leakage inductance (Lr1 + Lr2) of the transformer T2 is increased by loosely coupling the transformer T2. Lr2 is 3 mH, for example. As a result, the switching frequency is about twice the resonance frequency due to the leakage inductance (Lr1 + Lr2) and the resonance capacitor Cri.

  This Lr2 includes the inductance on the secondary side of the transformer T2. That is, the resonance frequency is determined by the inductance of the transformer T2, the leakage inductance (not shown) of the transformer T1, and the current resonance capacitor Cri.

  The operating state is divided into four periods T1 to T4 depending on the on / off state of the switching elements QH and QL and the state of the voltage applied to the transformers T1 and T2.

  First, in the period T1, the switching element QH is off and the switching element QL is on, and the current on the primary side of the transformer T2 is Cri → Lp (Np → Lr2) → Lr1 → QL → Cri or Cri → QL. It flows through the route of (DL) → Lr → Lp (Np → Lr2) → Cri. The resonance frequency is determined by the leakage inductance (Lr1 + Lr2) of the transformer T2 and the capacitor Cri. The windings Np and Ns of the transformer T2 both start with a negative voltage. For this reason, the secondary current flows through a path of Ns → D11 → C11 → N1 (L1) → Ns and a path of Ns → D12 → C12 → S1 → Ns.

The voltage V (L1) applied to the excitation inductance L1 is also similar to the period T11 in the example shown in FIG.
V (L1) = (V (C1) -V (C2)) / 4
Therefore, V (L1) has a positive voltage at the start of winding. For this reason, the excitation current I (L1) flowing through the excitation inductance L1 increases with a slope of V (L1) / L1.

  Since the primary winding N1 and the secondary winding S1 of the transformer T1 are magnetically coupled so that the currents flowing through them are balanced, the same current is charged in the capacitor C11 and the capacitor C12.

  Next, in the period T2, the switching element QH is turned on and the switching element QL is turned off, and the current on the primary side of the transformer T2 is Vin → Cri → Lp (Np → Lr2) → Lr1 → QH (DH ) → Vin path. Similarly to the period T1, the windings Np and Ns of the transformer T2 have a negative voltage at the beginning, so the secondary current is Ns → D11 → C11 → N1 (L1) → Ns and Ns → It flows along a route of D12 → C12 → S1 → Ns.

Similarly to the period T1, the voltage V (L1) applied to the excitation inductance is
V (L1) = (V (C1) -V (C2)) / 4
Therefore, V (L1) has a positive voltage at the beginning of winding. For this reason, the excitation current I (L1) flowing through the excitation inductance L1 increases with a slope of V (L1) / L1.

  Next, in the period T3, the switching element QH is turned on and the switching element QL is turned off. The current on the primary side of the transformer T2 flows through a path of Vin → Cri → Lp (Lr2 → Np) → Lr1 → QH (DH) → Vin or Vin → QH → Lr1 → Lp (Lr2 → Np) → Cri → Vin. . Since the windings Np and Ns of the transformer T2 both start with a positive voltage, the current on the secondary side is Ns → N1 (L1) → C11 → D1 → C1 (LD1) → Ns → S1 → C12 → D2 → C2 (LD2) → Ns.

The voltage V (L1) applied to the excitation inductance L1 is also similar to the period T14 in the example shown in FIG.
V (L1) =-(V (C1) -V (C2)) / 4
Therefore, V (L1) has a negative voltage at the start of winding. For this reason, the exciting current I (L1) flowing through the exciting inductance L1 decreases with a slope of | V (L1) / L1 |.

  Since the primary winding N1 and the secondary winding S1 of the transformer T1 are magnetically coupled so that the currents flowing therethrough are balanced, the same current is charged in the capacitor C1 and the capacitor C2. Accordingly, a balanced current flows through the load LD1 connected to the capacitor C1 and the load LD2 connected to the capacitor C2 even when the impedances are different.

  Next, in the period T4, the switching element QH is turned off and the switching element QL is turned on. The current on the primary side of the transformer T2 flows through a path of Cri → QL (DL) → Lr1 → Lp (Lr2 → Np) → Cri. The windings Np and Ns of the transformer T2 start with a positive voltage. For this reason, the secondary-side current has a path of Ns → N1 (L1) → C11 → D11 → D1 → C1 (LD1) → Ns and a path of Ns → S1 → C12 → D2 → C2 (LD2) → Ns. It flows in.

Similarly to the period T3, the voltage V (L1) applied to the excitation inductance L1 is
V (L1) =-(V (C1) -V (C2)) / 4
V (L1) is a negative voltage at the beginning of winding. For this reason, the exciting current I (L1) flowing through the exciting inductance L1 decreases with a slope of | V (L1) / L1 |.

  The LED drive device according to the present embodiment repeats the above periodic operation. Of the four periods T1 to T4, the diodes D1 and D2 do not conduct in the periods T1 and T2. In the periods T3 and T4, the diodes D1 and D2 are turned on, and the excitation current flowing through the excitation inductance L1 is superimposed on the current I (D1), so that a difference between the current I (D1) and the current I (D2) occurs. Arise. However, when I (D1) -I (D2) is integrated in one cycle (T1 to T4), the integrated value becomes substantially zero. Thus, when the switching frequency is higher (higher) than the resonance frequency, variations in the current flowing through each LED string can be reduced.

  In addition to increasing the leakage inductance of the transformer T2 as described above, the resonance frequency of the power supply means 10 can be lowered by increasing the leakage inductance of the transformer T1. FIG. 4 shows the configuration of an LED drive device according to a modification of the first embodiment. The LED drive device according to this modification is configured such that the primary winding N1 and the secondary winding S1 of the transformer T1 are loosely coupled and have large leakage inductances (Lrn1, Lrs1). The leakage inductance Lr2 of the transformer T2 includes an inductance on the secondary side of the transformer T2. That is, the resonance frequency of the power supply means 10 is determined by the excitation inductance, leakage inductance, current resonance capacitor Cri, and leakage inductance of the transformer T1. Therefore, even with this configuration, the resonance frequency of the power supply means 10 can be adjusted, and variations in the current I (D1) and the current I (D2) can be reduced. Furthermore, since the leakage inductance of the transformer T2 itself can be made relatively small, heat generation due to the leakage magnetic flux of the transformer T2 can be reduced.

  FIG. 5 is a diagram showing the relationship between leakage inductance and current variation. As shown in FIG. 5, the current variation ΔI decreases as the leakage inductance increases. In general, a transformer having a large leakage inductance is relatively inexpensive, and therefore an LED driving circuit that is inexpensive and has high current accuracy can be configured by using a transformer having a large leakage inductance.

  FIG. 6A shows the half-wave voltage doubler rectification method of the first embodiment, and FIG. 6B shows the full-wave rectification method. In addition, DD1 and DD2 shown in FIG.6 (b) show a full wave rectifier circuit.

  FIG. 7 is a diagram showing a comparison result of current variation of each rectification method. As shown in FIG. 7, even in the case of the full-wave rectification method, the current variation ΔI decreases as the leakage inductance of the transformer T2 increases. Furthermore, the half-wave voltage doubler rectification method can suppress the variation in current to half or less of the full-wave rectification method.

  FIG. 8 is a main configuration diagram of the LED drive device according to the second embodiment. As shown in FIG. 8A, the LED drive device of the second embodiment is different from the circuit using the leakage inductance of the transformer T1 shown in FIG. 8B (same as the configuration shown in FIG. 1). In addition to the leakage inductance, a coil Lr is connected to the transformer T1 outside.

  Even when this coil Lr is used, the same effect as when the transformer T1 has a leakage inductance can be obtained. FIG. 9 shows a result of comparison of current variations when the coil Lr is added and when the leakage inductance of the transformer T1 is used. As can be seen from FIG. 9, even when the coil Lr is added to the outside, the same effect as when the transformer T1 has a leakage inductance can be obtained. As an example, when a 5 μH coil Lr is connected to the transformer T1, the current accuracy is the same as when the transformer T1 has a leakage inductance of 10 μH.

  In the first and second embodiments, the case where the number of LEDs in series is 5 (LED1a to LED1e) and the number of parallel loads is 2 (LD1, LD2) is described, but the number is not limited to these. Further, in order to obtain a desired resonance frequency, the capacity of the current resonance capacitor Cri can be changed.

  The present invention is applicable to LED lighting devices and LED lighting for lighting LEDs.

DESCRIPTION OF SYMBOLS 1 PFM circuit 3 Comparison circuit 10 Power supply means Vin DC power supply QL, QH Switching element D1, D2, D11, D12, DL, DH Diode Cri Current resonance capacitor C1, C2, C11, C12 Capacitor T1, T2 Transformer Np, N1 1 Secondary windings Ns, S1 Secondary winding Lp Excitation inductances Lr1, Lr2, Lrn1, Lrs1 Leakage inductances Vref, Vref1 Reference power sources LD1, LD2 Load

Claims (5)

An LED driving device that rectifies and smoothes an alternating current and supplies the alternating current to the LED,
A first transformer having a primary winding and a secondary winding; first and second switching elements connected in series to both ends of a DC power supply; connected in parallel to one of the first and second switching elements; Resonant type power supply means having a series connection having the primary winding and a resonant capacitor, and outputting the alternating current from the secondary winding;
A first rectifying / smoothing circuit connected to the output of the power supply means and having a first rectifying element and a first smoothing element; a first LED group connected to the first rectifying / smoothing circuit and having one or more LEDs connected in series; A first series circuit having a first winding connected between the power supply means and the first rectifying and smoothing circuit;
A second rectifying / smoothing circuit having a second rectifying element and a second smoothing element connected to the output of the power supply means; a second LED group connected to the second rectifying / smoothing circuit and having one or more LEDs connected in series; A second series circuit having a second winding connected between the power supply means and the second rectifying and smoothing circuit;
Control means for controlling the switching frequency of the first and second switching elements to a frequency higher than the resonance frequency of the power supply means,
The LED driving device according to claim 1, wherein the first and second currents flowing through each of the first and second series circuits are balanced based on electromagnetic force generated in the first and second windings. The first transformer has a leakage inductance;
The LED driving device according to claim 1, wherein the resonance frequency is determined based on the leakage inductance. An inductance component connected to each of the first and second windings;
3. The LED driving device according to claim 1, wherein the resonance frequency is determined based on the resonance capacitor and the inductance component.   The first and second windings constitute a second transformer for balancing the first and second currents, and the inductance component is a leakage inductance of the second transformer. The LED driving device according to claim 2 or claim 3.   The LED drive device according to claim 2, wherein the inductance component is a coil connected to the first and second windings.

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