JP2011176493A - System for feedback of transmission route information - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a transmission route information feedback system, capable of estimating transmission route information of a channel impulse response, or the like, with high accuracy from a feedback signal, and also capable of transmitting information of an SNR (Signal to Noise Ratio). <P>SOLUTION: The channel impulse response which is the transmission route information of a down-link, is estimated using a known pilot signal for the down-link output from a down-link pilot signal memory 52 and a received signal in the frequency domain; and the estimation result is transformed to a channel frequency response in a DFT (Discrete Fourier Transform) circuit. An SNR estimation circuit 61 estimates the SNR of the down-link as the transmission route information using the received signal of the frequency domain and the estimated channel frequency response; and outputs it. A feedback signal creation circuit 62 receives the estimated channel frequency response, the SNR, and a known pilot signal for an up-link which is output from an up-link pilot signal memory 54; multiplexes the SNR, the channel frequency response, and the pilot signal for the up-link; and carries out feedback of them. <P>COPYRIGHT: (C)2011,JPO&INPIT

Description

  The present invention relates to wireless communication such as a mobile phone system, and more particularly to a MIMO (Multiple Input Multiple Output) system that performs spatial multiplexing transmission using a plurality of transmission / reception antennas.

In wireless communication such as a cellular phone system, MIMO transmission, which performs spatial multiplexing transmission using a plurality of transmission / reception antennas, is attracting attention as a technique for increasing the transmission speed without expanding the frequency band. As a transmission technique for further improving the characteristics of this MIMO transmission, a precoding technique is known in which, when transmission path information such as a channel impulse response is known on the transmission side, linear processing is performed on a transmission signal according to the transmission path. .
In the case of FDD (Frequency Division Duplex) with different frequency channels on the upper and lower lines, transmission path information cannot be obtained on the transmission side, so it is necessary to feed back the transmission path information from the reception side to the transmission side to operate precoding. There is.

  In the following, a MIMO transceiver that performs feedback of transmission path information will be described. For simplicity, the transmitting side is a base station, the receiving side is a mobile terminal, and precoding is used in the downlink from the base station to the mobile terminal. The transmission path information is fed back on the uplink from the mobile terminal to the base station.

FIG. 1 shows a configuration in which precoding is operated using the fed back transmission path information in a conventional MIMO single carrier transmission radio transmitter. The number of transmission antennas _NT was 2. First, a transmission bit sequence is input from the input terminal 1 to the modulation circuit 2, and a complex symbol sequence that is a modulation signal is generated. A complex symbol is a digital signal and has two components, an in-phase component and a quadrature component, but is regarded as one signal. In the following, all baseband signals are expressed in a complex display with the in-phase component as the real part and the quadrature component as the imaginary part. The complex symbol sequence that is the output of the modulation circuit 2 is input to the serial-to-parallel converter 3 and divided into sequences of the number of transmission streams M _S. Here, M _S = 2, and each complex symbol sequence is input to the linear processing circuit 6 that performs precoding. The linear processing circuit 6 includes multipliers 4-1 to 4-4 that perform complex multiplication and adders 5-1 to 5-2 that perform complex addition, and synthesizes by multiplying the complex symbol by a weighting coefficient, A transmission signal having the number of transmission antennas N _T is generated. The weighting coefficient is controlled by the precoding matrix control circuit 21 and is output through the terminals 20-1 to 20-4. This control will be described later. The transmission signals that are the outputs of the linear processing circuit 6 are respectively input to the corresponding up-converters 12-1 and 12-2, frequency-converted to the RF frequency band, and then passed through the input terminals 24-1 and 24-2. The signals are amplified by the corresponding transmission amplifiers 9-1 and 9-2 and transmitted by the transmission antennas 11-1 and 11-2. The up-converter 12-1 includes a D / A converter 7 and a multiplier circuit 8. The D / A converter 7 converts the in-phase component and the quadrature component of the transmission signal into an analog signal. The multiplication circuit 8 multiplies the in-phase component of the analog signal by the carrier wave of the RF frequency output from the oscillator 10, multiplies the quadrature component of the analog signal by the carrier wave whose phase is rotated by 90 degrees, and adds the multiplication results to output as a transmission wave. To do.

  The received waves including the fed back transmission path information are received by the receiving antennas 13-1 and 13-2, amplified by the receiving amplifiers 14-1 and 14-2, and then output to the output terminals 25-1 and 25-2. Is output. Further, the signals are input to the down converters 18-1 and 18-2, frequency-converted from the RF frequency band to the baseband, and then output as a received signal. The down converter 18-1 includes a multiplier circuit 15, a low-pass filter 16, and an A / D converter 17. The multiplier circuit 15 converts the carrier wave output from the oscillator 22 to the amplified received wave and the phase of the carrier wave by 90 degrees. Multiply each of the rotated ones and output two multiplication results. After the high frequency component is removed from the multiplication result by the low-pass filter 16, the in-phase component and the quadrature component of the received signal that is the baseband signal are extracted. The A / D converter 17 converts the received signal into a digital signal and outputs it. The received signal passes through terminals 19-1 and 19-2 and is input to the precoding matrix control circuit 21.

  Although FIG. 1 shows a configuration example using separate antennas for transmission and reception, a configuration using a transmission / reception shared antenna is also possible. A configuration in the case of using a transmission / reception shared antenna is shown in FIG. The transmission wave input from the input terminal 24-3 is amplified by the transmission amplifier 9-3, passes through the circulator 75, and is transmitted by the transmission / reception shared antenna 23. This signal is not input to the receiving amplifier 14-3. On the other hand, the received wave is received by the transmission / reception shared antenna 23, passes through the circulator 75, and is output only to the reception amplifier 14-3. The amplified received wave is output from the output terminal 25-3.

The precoding matrix control circuit 21 in FIG. 1 is roughly divided into three types of configurations.
The first configuration is a case where transmission path information is quantized and the quantized bits are fed back, and the precoding matrix control circuit 21 has the configuration shown in FIG. First, a received signal including quantized transmission path information is input to the digitized transmission path information extraction circuit 76 through the terminals 19-1 and 19-2. The digitized transmission path information extraction circuit 76 extracts the quantized transmission path information from the received signal and inputs the value to the precoding matrix estimation circuit 77. In addition, since the number of transmission antennas _NT is 2, the transmission path information has a channel impulse response from the transmission antenna 11-1 and a reception from the transmission antenna 11-2 for each of the reception signals 19-1 and 19-2. There are two types of channel impulse responses. The precoding matrix estimation circuit 77 obtains a weighting coefficient based on the quantized transmission path information and outputs it to the terminals 20-1 to 20-4. Increasing the number of channel information quantization bits increases the amount of feedback information. Conversely, if the number of channel information quantization bits is decreased, the quantization error increases, and the precoding weighting coefficient is accurately determined. There is a problem that can not be.

  Therefore, as a second configuration, a code book method has been proposed so that the weighting coefficient can be obtained with a certain degree of accuracy even if the amount of feedback information is reduced (see Non-Patent Document 1). In this method, a plurality of candidates are determined in advance for a precoding matrix having weighting factors as elements. This set of candidates is called a code book. The codebook information is shared by the transmitter / receiver, the optimal precoding matrix candidate is selected on the receiving side, and the number (index) is fed back to the transmitting side as transmission path information.

  FIG. 4 shows the configuration of the precoding matrix control circuit 21 in the codebook method. First, a received signal including an index passes through terminals 19-1 and 19-2 and is input to the precoding / index extraction circuit 26. The precoding index extraction circuit 26 extracts an index from the received signal and inputs the value to the precoding matrix selection circuit 27. The precoding matrix selection circuit 27 selects a precoding matrix selected on the receiving side from the code book, and outputs the matrix elements as weighting coefficients to the terminals 20-1 to 20-4.

  In order to reduce the amount of feedback information, the code book method limits the size of the code book, that is, the number of precoding matrix candidates. For this reason, there is a problem that the effect of improving transmission characteristics by precoding is impaired. In order to overcome this problem, an analog feedback method as a third configuration has been proposed (see Non-Patent Document 2). In this method, transmission path information is fed back as an analog signal, and there is a possibility that transmission path information can be sent with high accuracy while reducing the amount of feedback information. Note that the strict definition of an analog signal is a continuous amount of signal in both time and signal value, but an analog feedback “analog signal” is a “digital signal” that does not introduce new quantization. In the following description, the expression “as an analog signal” is used in this sense.

  FIG. 5 shows the configuration of the precoding matrix control circuit 21 in the analog feedback system. First, a received signal including transmission path information is input to the analog transmission path information extraction circuit 28 through terminals 19-1 and 19-2. The analog transmission path information extraction circuit 28 extracts the transmission path information of the analog signal from the received signal, and inputs the value from the terminals 81-1 and 81-2 to the precoding matrix estimation circuit 77. The precoding matrix estimation circuit 77 obtains a weighting coefficient based on the transmission path information and outputs it to the terminals 20-1 to 20-4.

  FIG. 6 shows the configuration of the analog transmission path information extraction circuit 28. First, received signals are input to the channel estimation circuit 79 from the terminals 19-1 and 19-2. This received signal is a feedback signal received from the receiver side, and includes a known uplink pilot signal in addition to analog signal transmission path information. The channel estimation circuit 79 estimates an uplink channel impulse response using the received signal and a known uplink pilot signal output from the uplink pilot signal memory 48. Here, channel estimation circuit 79 and uplink pilot signal memory 48 correspond to channel estimation means. The estimated uplink channel impulse response is input to the impulse response extraction circuit 80 together with the received signal. The impulse response extraction circuit 80 extracts a downlink channel impulse response corresponding to the transmission path information from the estimated uplink channel impulse response and the received signal that is the received feedback signal. Output to 81-2. Here, the impulse response extraction circuit 80 corresponds to transmission line information extraction means.

Next, FIG. 7 shows a configuration in which precoding is operated using the fed back transmission path information in a conventional MIMO-OFDM (Orthogonal Frequency Division Multiple) transmission wireless transmitter. The number of transmission antennas _NT was 2. First, a transmission bit sequence is input from the input terminal 1 to the modulation circuit 2, and a complex symbol sequence that is a modulation signal is generated. The complex symbol sequence that is the output of the modulation circuit 2 is input to the serial-to-parallel converter 29 and divided into N subcarrier number N sequences. Further, the signals are input from the serial / parallel converter 3-1 to 3-2, and are divided into sequences of the number of transmission streams M _S. Here, M _S = 2 and each complex symbol sequence is changed from the linear processing circuit (# 1) 6-1 that performs precoding according to the subcarrier number n (1 ≦ n ≦ N) to the linear processing circuit ( #N) is input to 6-2. The weighting coefficients from the linear processing circuit (# 1) 6-1 to the linear processing circuit (#N) 6-2 are controlled by the OFDM precoding matrix control circuit 36 and output through terminals 35-1 to 35-2. This control will be described later. The transmission signals output from the linear processing circuit (# 1) 6-1 to the linear processing circuit (#N) 6-2 are respectively input to corresponding IFFT (Inverse Fast Fourier Transform) circuits 30-1 and 30-2. , Converted to time domain signal by IFFT. GI (Guard Interval) adding circuits 31-1 and 31-2 copy the tail signal of the time domain signal and add it to the head to generate an OFDM transmission signal. By this operation, periodicity is maintained even in a multipath environment, and intersubcarrier interference can be made zero. The OFDM transmission signal is frequency-converted to the RF frequency band by the up-converters 12-1 and 12-2 using the carrier wave output from the oscillator 10, and then amplified by the corresponding transmission amplifiers 9-1 and 9-2. Transmitted by the transmitting antennas 11-1 and 11-2.

  The received waves including the fed back transmission path information are received by the receiving antennas 13-1 and 13-2, amplified by the receiving amplifiers 14-1 and 14-2, and then output to the output terminals 25-1 and 25-2. Is output. Further, the signals are input to the down converters 18-1 and 18-2 using the carrier wave output from the oscillator 22, are frequency-converted from the RF frequency band to the baseband, and then output as a received signal. The received signal passes through terminals 19-1 and 19-2 and is input to GI removal circuits 32-1 and 32-2. The GI removal circuits 32-1 and 32-2 remove the copied and added GI signal components, and output them to FFT (Fast Fourier Transform) circuits 33-1 and 33-2. The FFT circuits 33-1 and 33-2 convert a time domain signal into a frequency domain signal by FFT, which is the inverse operation of IFFT. This frequency domain signal passes through terminals 34-1 and 34-2 and is input to the OFDM precoding matrix control circuit 36.

  The OFDM precoding matrix control circuit 36 shown in FIG. 7 is roughly classified into three types. The first configuration is a case where transmission path information is quantized and the quantized bits are fed back, and the OFDM precoding matrix control circuit 36 has the configuration shown in FIG. First, the received signal in the frequency domain including the quantized transmission path information is input to the digitized transmission path information extraction circuit 37 through the terminals 34-1 and 34-2. The digitized transmission path information extraction circuit 37 extracts a channel impulse response as quantized transmission path information from the frequency domain received signal. The extracted channel impulse response is converted into a channel frequency response by DFT in DFT (Discrete Fourier Transform) circuits 38-1 and 38-2. The channel frequency response of the frequency corresponding to the subcarrier number n is the precoding matrix estimation corresponding to n in the precoding matrix estimation circuit (# 1) 39-1 to the precoding matrix estimation circuit (#N) 39-2. Input to the circuit. The precoding matrix estimation circuit (# 1) 39-1 estimates the weighting coefficient of the linear processing circuit (# 1) 6-1 in FIG. 7 and outputs it to the terminal 35-1. On the other hand, the precoding matrix estimation circuit (#N) 39-2 estimates the weighting coefficient of the linear processing circuit (#N) 6-2 in FIG. 7 and outputs it to the terminal 35-2.

  Next, FIG. 9 shows a configuration of the OFDM precoding matrix control circuit 36 in the codebook system which is the second configuration. First, the received signal in the frequency domain including the index passes through terminals 34-1 and 34-2 and is input to the precoding index extraction circuit 40. The precoding index extraction circuit 40 extracts an index indicating the precoding matrix selected in each subcarrier from the received signal in the frequency domain, and precodes the value from the precoding matrix selection circuit (# 1) 41-1. Of the matrix selection circuit (#N) 41-2, the precoding matrix selection circuit corresponding to the subcarrier number n is input. The precoding matrix selection circuit (# 1) 41-1 to the precoding matrix selection circuit (#N) 41-2 select a precoding matrix selected on the receiving side from the codebook, and weight the elements of the matrix The coefficient is output to terminals 35-1 to 35-2.

  Further, FIG. 10 shows the configuration of the OFDM precoding matrix control circuit 36 in the analog feedback system which is the third configuration. First, the received signal in the frequency domain including the transmission path information is input to the analog transmission path information extraction circuit 42 through the terminals 34-1 and 34-2. The analog transmission path information extraction circuit 42 extracts a channel impulse response as analog signal transmission path information from the received signal in the frequency domain, and outputs the value to terminals 81-1 and 81-2. The extracted channel impulse response is converted into a channel frequency response by DFT in the DFT circuits 38-1 to 38-2. The channel frequency response of the frequency corresponding to the subcarrier number n is the precoding matrix estimation corresponding to n in the precoding matrix estimation circuit (# 1) 39-1 to the precoding matrix estimation circuit (#N) 39-2. Input to the circuit. The precoding matrix estimation circuit (# 1) 39-1 estimates the weighting coefficient of the linear processing circuit (# 1) 6-1 in FIG. 7 and outputs it to the terminal 35-1. On the other hand, the precoding matrix estimation circuit (#N) 39-2 estimates the weighting coefficient of the linear processing circuit (#N) 6-2 in FIG. 7 and outputs it to the terminal 35-2.

  FIG. 11 shows the configuration of the analog transmission path information extraction circuit 42. First, received signals in the frequency domain are input to the channel estimation circuit 82 from the terminals 34-1 and 34-2. This received signal in the frequency domain is obtained by receiving a feedback signal from the receiver side, and includes a known uplink pilot signal in addition to the transmission path information of the analog signal. The channel estimation circuit 82 estimates an uplink channel impulse response using the received signal and a known uplink pilot signal output from the uplink pilot signal memory 48. Here, channel estimation circuit 82 and uplink pilot signal memory 48 correspond to channel estimation means. The estimated uplink channel impulse response is input to the impulse response extraction circuit 83 together with the frequency domain received signal. The impulse response extraction circuit 83 extracts a downlink channel impulse response corresponding to the transmission path information from the estimated uplink channel impulse response and the received signal in the frequency domain that is the received feedback signal, and outputs a terminal 81 -1 and 81-2. Here, the impulse response extraction circuit 83 corresponds to transmission line information extraction means.

  Below, the receiver which produces | generates a feedback signal is demonstrated.

FIG. 12 shows a configuration in which transmission path information is fed back as an analog signal in a conventional MIMO single carrier transmission radio receiver. This is used in a pair with the MIMO single carrier transmission radio transmitter shown in FIG. The number of receiving antennas N _R is 2. First, the reception antennas 13-3 and 13-4 receive a transmission wave that has propagated through the downlink, that is, a reception wave. The received waves are amplified by the receiving amplifiers 14-3 and 14-4, respectively, and then input to the down converters 18-3 and 18-4 that use the carrier wave output from the oscillator 10 to convert the frequency from the RF frequency band to the baseband. Is output as a received signal. This received signal is input to the channel estimation circuit 43 and the signal detection circuit 45. The received signal includes a known downlink pilot signal, and the channel estimation circuit 43 uses the known downlink pilot signal output from the downlink pilot signal memory 44 and the received signal to obtain the downlink. The channel impulse response which is the transmission path information is estimated. Here, the channel estimation circuit 43 and the downlink pilot signal memory 44 correspond to transmission path information estimation means. The signal detection circuit 45 performs signal detection using the received signal and the estimated channel impulse response, determines the transmission bit sequence, and outputs the determination bit sequence to the output terminal 46. A feedback signal generation circuit 47 corresponding to feedback signal generation means receives, as inputs, a downlink channel impulse response, which is estimated transmission path information, and a known uplink pilot signal output from the uplink pilot signal memory 48. The downlink channel impulse response is not newly quantized and is multiplexed with the uplink pilot signal as an analog signal to generate and output a feedback signal. Here, the channel estimation circuit 43, the downlink pilot signal memory 44, the feedback signal generation circuit 47, and the uplink pilot signal memory 48 constitute a channel estimation / feedback signal generation circuit 49. The feedback signal output from the feedback signal generation circuit 47 is divided into N _R (= 2) signal series by the serial / parallel converter 50, and the up-converters 12-3 and 12-4 using the carrier wave output from the oscillator 22. And is frequency-converted to an RF frequency band, amplified by corresponding transmission amplifiers 9-3 and 9-4, and transmitted by transmission antennas 11-3 and 11-4.

  In addition, as shown in FIG. 2, there is a configuration using a transmission / reception shared antenna.

  FIG. 13 shows the configuration of the feedback signal output from the transmission antenna 11-3 of FIG. First, the receiving antennas 13-3 and 13-4 in FIG. 12 are referred to as first and second receiving antennas, respectively, and the transmitting antennas 11-1 and 11-2 in FIG. 1 are respectively referred to as first and second transmitting antennas. I will call it. The downlink channel is assumed to be flat fading. The feedback signal in FIG. 13 time-multiplexes a known uplink pilot signal and a channel impulse response which is downlink transmission path information. First, a known uplink pilot signal is arranged, followed by a channel impulse response h11 between the first receiving antenna and the first transmitting antenna, and finally a channel impulse response h12 between the first receiving antenna and the second transmitting antenna. Place. In the case of multipath fading, h11 and h12 are replaced with a sequence of complex envelopes for each path. In the feedback signal output from the transmission antenna 11-4 in FIG. 12, h11 is replaced with a channel impulse response h21 between the second reception antenna and the first transmission antenna, and h12 is between the second reception antenna and the second transmission antenna. The channel impulse response h22 may be replaced.

FIG. 14 shows a configuration in which transmission path information is fed back as an analog signal in a conventional MIMO-OFDM transmission wireless receiver (Non-Patent Document 3). This is used in combination with the MIMO-OFDM transmission wireless transmitter shown in FIG. The number of receiving antennas N _R is 2. First, the reception antennas 13-3 and 13-4 receive a transmission wave that has propagated through the downlink, that is, a reception wave. The received waves are amplified by the receiving amplifiers 14-3 and 14-4, respectively, and then input to the down converters 18-3 and 18-4 that use the carrier wave output from the oscillator 10 to convert the frequency from the RF frequency band to the baseband. Is output as a received signal. The received signal is input to the GI removal circuits 32-3 and 32-4. The GI removal circuits 32-3 and 32-4 remove the copied and added GI signal components, and the FFT circuits 33-3 and 33-4. Output to. The FFT circuits 33-3 and 33-4 convert a time domain signal into a frequency domain signal by FFT, which is the inverse operation of IFFT. The frequency domain received signal passes through terminals 59-1 and 59-2 and is input to the channel estimation circuit 51 and the signal detection circuit 57. The received signal in the frequency domain includes a known downlink pilot signal, and the channel estimation circuit 51 receives the known downlink pilot signal output from the downlink pilot signal memory 52, the received signal in the frequency domain, and Is used to estimate a channel impulse response which is downlink transmission path information. The estimated downlink channel impulse responses are converted into channel frequency responses by the DFT circuits 38-3 and 38-4, respectively. Here, the channel estimation circuit 51, the downlink pilot signal memory 52, and the DFT circuits 38-3 and 38-4 correspond to transmission path information estimation means. The signal detection circuit 57 performs signal detection using the frequency domain received signal and the estimated channel frequency response, determines the transmission bit sequence, and outputs the determination bit sequence to the output terminal 46. The linear synthesis circuit 53 receives the estimated channel frequency response as an input, linearly synthesizes the channel frequency response, and outputs it. The feedback signal generation circuit 55 receives the linear combination value, which is the estimated transmission path information, and the known uplink pilot signal output from the uplink pilot signal memory 54, and the linear combination value is newly quantized. The feedback signal is generated and output by multiplexing with the uplink pilot signal as it is without being converted into an analog signal. The linear synthesis circuit 53, the uplink pilot signal memory 54, and the feedback signal generation circuit 55 correspond to feedback signal generation means. The channel estimation circuit 51, the downlink pilot signal memory 52, the DFT circuits 38-3 and 38-4, the linear synthesis circuit 53, the uplink pilot signal memory 54, and the feedback signal generation circuit 55 are a channel estimation / feedback signal generation circuit. 56 is configured. The feedback signal output from the feedback signal generation circuit 55 to the terminal 60 is divided into N _R (= 2) signal series by the serial / parallel converter 58 and input to the corresponding IFFT circuits 30-3 and 30-4. , Converted to time domain signal by IFFT. The GI addition circuits 31-3 and 31-4 copy the tail signal of the time domain signal and add it to the head to generate an OFDM transmission signal. The OFDM transmission signal is frequency-converted to the RF frequency band by the up-converters 12-3 and 12-4 using the carrier wave output from the oscillator 22, and then amplified by the corresponding transmission amplifiers 9-3 and 9-4. Transmitted by antennas 11-3 and 11-4.
In addition, as shown in FIG. 2, there is a configuration using a transmission / reception shared antenna.

FIG. 15 shows the configuration of the feedback signal output from the transmission antenna 11-3 of FIG. First, the receiving antennas 13-3 and 13-4 in FIG. 14 are referred to as first and second receiving antennas, respectively, and the transmitting antennas 11-1 and 11-2 in FIG. 7 are respectively referred to as first and second transmitting antennas. I will call it. The horizontal axis in FIG. 15 is time in units of OFDM symbols, and the vertical axis is frequency in units of subcarrier frequency intervals. The hatched portion is an uplink pilot signal and is arranged when the subcarrier number n is an even number. When n is an odd number, a linear combination of downlink channel frequency responses is arranged. Here, H _lk (n) is a channel frequency response between the l-th (= 1, 2) reception antenna and the k-th (= 1, 2) transmission antenna at the frequency of the n-th subcarrier. a _lk (n) corresponds to a linear combination coefficient, and is a complex symbol known between transmission and reception. In the prior art, the feedback signal is transmitted only from one transmission antenna, and the feedback signal is not transmitted from the transmission antenna 11-4 in FIG.

  If a linear combination of channel frequency responses is sent as a feedback signal as described above, there is a problem that estimation accuracy deteriorates when individual channel impulse responses are separated and extracted from the received feedback signal.

  Also, precoding has been proposed that can minimize BER (Bit Error Rate) when performing MLD (Maximum Likelihood Detection), which is optimal reception, on the receiving side (Non-Patent Document 4). The transmission path information required to operate this BER minimum standard precoding requires not only the channel impulse response and equivalent frequency response, but also the downlink signal-to-noise ratio and SNR (Signal to Noise Ratio). Become. However, in the conventional analog feedback system, this SNR information is not fed back as transmission path information.

D. J. Love, R. W. Heath, W. Santipach, and M. L. Honig, "What is the value of limited feedback for MIMO channels?", IEEE Comm. Mag. , pp. 54-59, October 2004. E. Chiu and P. Ho, "Transmit beamforming with analog channel state information feedback," IEEE Trans. Wireless Commun. , vol. 7, no. 3, pp. 878-887, March 2008. T. A. Thomas, K. L. Baum, and P. Sartori, "Obtaining channel knowledge for closed-loop multi-stream broadband MIMO-OFDM communications using direct channel feedback," IEEE GLOBECOM '05, vol. 6, pp. 3907-3911, December 2005. B. Pitakdumrongkija, K. Fukawa, H. Suzuki, and T. Higiwara, "MIMO-OFDM precoding technique for minimizing BER upper bound of MLD," IEICE Trans. Commun. , vol. E91-B, no. 7, pp. 2287-2298, July 2008. Simon Haykin, Adaptive Filter Theory Third Edition Prentice-Hall, 1996.

  As described above, the conventional analog feedback method feeds back a linear combination of channel frequency responses, so that there is a problem that estimation accuracy deteriorates when individual channel impulse responses are separated and extracted from the feedback signal. In addition, there is a problem that the signal-to-noise ratio information required for precoding of the minimum BER criterion is not fed back.

  The present invention has been made in view of such a problem, and provides a transmission path information feedback method that can accurately estimate transmission path information such as a channel impulse response from a feedback signal and can also send signal-to-noise ratio information. The purpose is to do.

  According to the transmission path information feedback system of the present invention, the above object is achieved by the means described in the claims. That is, the transmission path information feedback system of the present invention comprises (i) transmission path information estimation means for estimating downlink transmission path information using a downlink reception signal and a known downlink pilot signal, (ii) Feedback signal generation means for generating a feedback signal by multiplexing the estimated transmission path information as an analog signal without linear synthesis, and (iii) a pilot received through an uplink channel Channel estimation means for estimating an uplink channel impulse response from the signal and a known uplink pilot signal, and (iv) transmission path information from the estimated uplink channel impulse response and the received feedback signal. It comprises transmission line information extracting means for extracting. The difference from the prior art is that the estimated transmission path information is not linearly synthesized, but is multiplexed with a known uplink pilot signal to generate a feedback signal.

  Also, the transmission path information fed back by the transmission path information feedback system of the present invention is a downlink signal-to-noise ratio and a channel impulse response.

  Also, the transmission path information fed back by the transmission path information feedback system of the present invention is a downlink signal-to-noise ratio and a channel frequency response.

  Further, (ii) feedback signal generation means of the transmission path information feedback system of the present invention generates a feedback signal by time-multiplexing downlink transmission path information and a known uplink pilot signal.

  Also, (ii) feedback signal generation means of the transmission path information feedback system of the present invention generates a feedback signal by frequency-time multiplexing downlink transmission path information and a known uplink pilot signal.

  Also, (i) transmission path information estimation means of the transmission path information feedback system of the present invention estimates downlink transmission path information by a least square method using a downlink reception signal and a known downlink pilot signal. To do.

  Further, (iv) transmission path information extraction means of the transmission path information feedback system of the present invention extracts the transmission path information from the estimated uplink channel impulse response and the received feedback signal by the least square method. .

  Further, (ii) feedback signal generation means of the transmission path information feedback system of the present invention nonlinearly compresses the amplitude of the downlink transmission path information, and sets the value as an analog signal.

  Furthermore, (i) transmission path information estimation means of the transmission path information feedback system of the present invention estimates transmission path information of interference waves as well as desired waves.

  The transmission path information feedback means (iv) transmission path information extraction means of the present invention extracts transmission path information of interference waves as well as desired waves.

  In addition, (iv) transmission path information extraction means of the transmission path information feedback system of the present invention linearly predicts the current transmission path information from the transmission path information extracted in the past.

  Furthermore, (i) transmission path information estimation means of the transmission path information feedback system of the present invention linearly predicts future transmission path information from the estimated transmission path information.

  The present invention has the following effects.

  According to the transmission path information feedback system of the first form, it is possible to accurately separate and extract individual transmission path information from the received feedback signal.

  According to the transmission path information feedback system of the second form, the signal-to-noise ratio and the channel impulse response can be fed back as the transmission path information.

  According to the transmission line information feedback system of the third form, the signal-to-noise ratio and the channel frequency response can be fed back as transmission line information.

  According to the transmission path information feedback system of the fourth mode, it is possible to generate a feedback signal by time-multiplexing transmission path information and a known uplink pilot signal.

  According to the transmission path information feedback system of the fifth embodiment, the transmission path information and the known uplink pilot signal can be frequency-time multiplexed to generate a feedback signal.

  According to the transmission path information feedback system of the sixth embodiment, transmission path information can be accurately estimated by the least square method.

  According to the transmission path information feedback system of the seventh embodiment, transmission path information fed back by the least square method can be extracted with high accuracy.

  According to the transmission path information feedback system of the eighth embodiment, the amplitude of the downlink transmission path information is nonlinearly compressed and the value is fed back, thereby improving resistance to noise and accurately extracting the transmission path information.

  According to the transmission path information feedback system of the ninth embodiment, it is possible to estimate transmission path information of interference waves as well as desired waves.

  According to the transmission path information feedback system of the tenth embodiment, it is possible to extract transmission path information of the desired wave and the interference wave that are fed back.

  According to the transmission path information feedback system of the eleventh aspect, the current transmission path information can be linearly predicted from the transmission path information extracted in the past, and this is effective when the downlink channel fluctuates over time.

  According to the transmission path information feedback system of the twelfth aspect, future transmission path information can be linearly predicted from past and current transmission path information, and is effective when the downlink channel fluctuates over time.

The block block diagram of the MIMO single carrier radio | wireless transmitter using the conventional precoding. The block diagram in the case of sharing a transmission / reception antenna. The block diagram of the pre-coding matrix control circuit 21 of FIG. The block diagram of the pre-coding matrix control circuit 21 of FIG. The block diagram of the pre-coding matrix control circuit 21 of FIG. FIG. 7 is a block diagram of the analog transmission line information extraction circuit 28 in FIG. 6. The block block diagram of the MIMO-OFDM radio transmitter using the conventional precoding. FIG. 8 is a configuration diagram of the OFDM precoding matrix control circuit 36 in FIG. 7. FIG. 8 is a configuration diagram of the OFDM precoding matrix control circuit 36 in FIG. 7. FIG. 8 is a configuration diagram of the OFDM precoding matrix control circuit 36 in FIG. 7. FIG. 11 is a block configuration diagram of an analog transmission line information extraction circuit in FIG. 10. The block block diagram of the MIMO single carrier radio | wireless receiver which performs the conventional feedback. The block diagram of the feedback signal of FIG. The block block diagram of the MIMO-OFDM radio receiver which performs the conventional feedback. The block diagram of the feedback signal of FIG. The block block diagram of the channel estimation and the feedback signal generation circuit 56 of FIG. 14 in the 3rd example by this invention. The block diagram of the feedback signal of this invention. The block block diagram of the OFDM precoding matrix control circuit 36 of FIG. 7 in the 3rd example by this invention. FIG. 20 is a block configuration diagram of an analog transmission line information extraction circuit 63 in FIG. 19. The block block diagram of the channel estimation and the feedback signal generation circuit 56 of FIG. 14 in the 8th example by this invention. The block block diagram of the OFDM precoding matrix control circuit 36 of FIG. 7 in the 8th example by this invention. The block block diagram of the channel estimation and the feedback signal generation circuit 56 of FIG. 14 in the 9th example by this invention. The block block diagram of the precoding matrix control circuit 36 for OFDM of FIG. 7 in the 10th example by this invention. FIG. 10 is a block diagram of the OFDM precoding matrix control circuit 36 of FIG. 7 in an eleventh example according to the present invention.

Hereinafter, the best mode for carrying out the present invention will be described taking MIMO-OFDM transmission as an example.
FIG. 16 shows a configuration related to claims 3 and 5 at the time of filing of the channel estimation / feedback signal generation circuit 56 of FIG. First, frequency domain received signals are input to the channel estimation circuit 51 and the SNR estimation circuit 61 from the terminals 59-1 and 59-2. The received signal in the frequency domain includes a known downlink pilot signal, and the channel estimation circuit 51 receives the known downlink pilot signal output from the downlink pilot signal memory 52, the received signal in the frequency domain, and Is used to estimate a channel impulse response which is downlink transmission path information. The estimated downlink channel impulse responses are converted into channel frequency responses by the DFT circuits 38-3 and 38-4, respectively. The SNR estimation circuit 61 estimates and outputs the downlink SNR as downlink transmission path information using the frequency domain received signal and the estimated channel frequency response. Here, the channel estimation circuit 51, the downlink pilot signal memory 52, the SNR estimation circuit 61, and the DFT circuits 38-3 and 38-4 correspond to transmission path information estimation means. The feedback signal generation circuit 62 receives the estimated channel frequency response and SNR and the known uplink pilot signal output from the uplink pilot signal memory 54, and inputs the SNR, the channel frequency response, and the uplink pilot signal. Are multiplexed to generate a feedback signal and output it to the terminal 60. Here, uplink pilot signal memory 54 and feedback signal generation circuit 62 correspond to feedback signal generation means.

Next, channel impulse response estimation in the channel estimation circuit 51 and SNR estimation in the SNR estimation circuit 61 will be described in detail using mathematical expressions.
First, the received signals in the frequency domain input from the terminals 59-1 and 59-2 are Y ^D ₁ (i, n) and Y ^D ₂ (i, n), respectively. Here, i and n are an OFDM symbol number and a subcarrier number, respectively. Y ^D _l (i, n), l (= 1, 2) is

と表わすことができる。ここで、a^D _k（i，n）は第k（＝1、2）送信アンテナから送信される、第iシンボル、第nサブキャリアの変調信号で、平均電力を１とする。N^D _l（i，n）は下り回線の雑音信号のFFTである。H^D _lk（n）は下り回線のチャネル周波数応答で、第k送信アンテナと第l受信アンテナ間のものである。このチャネルインパルス応答h^D _lk（τ）を

Can be expressed as Here, a ^D _k (i, n) is a modulated signal of the i-th symbol and the n-th subcarrier transmitted from the k-th (= 1, 2) transmission antenna, and the average power is 1. N ^D _l (i, n) is the FFT of the downlink noise signal. H ^D _lk (n) is a downlink channel frequency response between the k-th transmitting antenna and the l-th receiving antenna. This channel impulse response h ^D _lk (τ)

と仮定する。ただし、δ（τ）はデルタ関数であり、Δ_tはFFTのサンプリング間隔である。h^D _lk，mは遅延時間mΔ_tのパスの複素包絡線であり、最大遅延時間MΔ_tはGI長以下とする。第nサブキャリアの周波数を2πn／（NΔ_t）とすると、H^D _lk（n）は

Assume that Here, δ (τ) is a delta function, and _Δt is an FFT sampling interval. h ^D _{lk, m} is the complex envelope of the path of the delay time mΔ _t, maximum delay time Emuderuta _t is less GI length. If the frequency of the nth subcarrier is 2πn / (NΔ _t ), H ^D _lk (n) is

となる。数式１に数式３を代入して、変形すると

It becomes. Substituting Equation 3 into Equation 1 and transforming it

となる。ただし、^Hは複素共役転置を表わす。h^D _lとa^D（i，n）は以下で定める2（M＋1）次元ベクトルである。

It becomes. Where ^H is the complex conjugate transpose. h ^D _l and a ^D (i, n) are 2 (M + 1) -dimensional vectors defined below.

ただし、^Tは転置を表わす。

^T represents transposition.

h ^D _l is a downlink channel impulse response, which is estimated by the method of least squares (this relates to claim 6 at the time of filing). First, pilot subcarriers that are downlink pilots are in i _D1 to i _D2 symbols, and a set of subcarrier numbers n of pilot subcarriers is set to S in the i th (i _D1 ≦ i ≦ i _D21 ) symbol. _{Let D} (i). Therefore, the least squares evaluation function J _l is

と定めることができる。ただし、λ（0＜λ≦1）は忘却係数である。J_lを最小にするh^D _lを求め、これをh^D _lの推定値h^D _l，eとする。h^D _l，eは正規方程式の解であり、次式で与えられる（これについては、非特許文献５参照。）。

Can be determined. However, λ (0 <λ ≦ 1) is a forgetting factor. Seeking h ^D _l that minimizes the J _l, which estimates of h ^D _l h ^D _l, which as _e. h ^D _{l, e} is a solution of a normal equation and is given by the following equation (refer to Non-Patent Document 5 for this).

ただし、*は複素共役を表わす。

However, * represents a complex conjugate.

Estimate of _{^{H D lk (n) ^ H}} D lk (n) is obtained from Equation 3 as follows.

なお、^h^D _lk，mはh^D _lk，mの推定値であり、h^D＊ _l，eの要素である。^H^D _lk（n）は^h^D _lk，mのDFTと等価である。

_Here , ^ h ^D _{lk, m} is an estimated value of h ^D _{lk, m} and is an element of h ^{D *} _{l, e} . ^ H ^D _lk (n) is equivalent to the DFT of ^ h ^D _{lk, m} .

The estimated SNR ^ γ of the downlink is _{derived from} the estimated value of h ^D _{lk, m using} the least square method ^ h ^D _{lk, m}

と求める。ただし、^σ_D ²はN^D _l（i，n）の平均電力の推定値であり、数式１から

I ask. However, ^ σ _D ² is an estimated value of the average power of N ^D _l (i, n).

と求める。ただし、N_DはS_D（i）に含まれるnの数である。

I ask. However, N _D is the number of n included in S _D (i).

Next, the feedback signal generated by the feedback signal generation circuit 62 of FIG. 16 will be described in detail. First, the feedback signal X _l (i, n) at the l-th antenna, i-th symbol, and n-th subcarrier must send ^ H ^D _lk (n) and ^ γ information, plus the normal Also serves as a pilot signal for channel estimation. Therefore, X _l (i, n) is determined as follows.

ここで、フィードバック信号はi_U1からi_U2シンボル内に存在するものとし、S_kl（i）は^H^D _lk1（n）を送るサブキャリア番号の集合、S_Γ（i）はlog₁₀^γ／C_γを送るサブキャリア番号の集合、S_P（i）はパイロット信号を送るサブキャリア番号の集合である。なお、C_γは定数であり、a^U _l（i，n）は平均電力１の複素シンボルで、送受信間で既知とする。αは正の規格化定数であり、送信電力一定の拘束条件を満足するように求める。なお、周波数応答を定数倍しても、^γが分かれば定数倍の不確定は問題にならず、αをフィードバック先で推定する必要はない。

Here, it is assumed that the feedback signal exists in i _U1 to i _U2 symbols, S _kl (i) is a set of subcarrier numbers to send ^ H ^D _lk1 (n), and S _Γ (i) is log ₁₀ ^ γ A set of subcarrier numbers for sending / _Cγ , and S _P (i) is a set of subcarrier numbers for sending pilot signals. Note that C _γ is a constant, and a ^U _l (i, n) is a complex symbol with an average power of 1 and is known between transmission and reception. α is a positive normalization constant, and is obtained so as to satisfy the constraint condition of constant transmission power. Even if the frequency response is multiplied by a constant, if ^ γ is known, the uncertainty of the constant multiplication will not be a problem, and α need not be estimated at the feedback destination.

The above is a method of feeding back the channel frequency response of the downlink as the transmission path information. However, these may be converted to the time axis and fed back instead of ^ H ^D _lk (n) as the channel impulse response. (This relates to claim 2 as filed).

FIG. 17 shows the configuration of the feedback signal. FIG. 4A shows a feedback signal of the transmission antenna 11-3 (transmission antenna 1), and FIG. 4B shows a feedback signal of the transmission antenna 11-4 (transmission antenna 2). n∈S _kl (i) satisfies (i + n) mod 4 = k ₁ and n∈S _Γ (i) is (i + n) mod 4 = 3 and [(i + n) / 4 ”= 0,1 ( ["Represents a Gaussian symbol). All other than this, S _P (i). For estimation of ^ γ, two subcarriers per symbol are sufficient, and it is necessary to accurately perform uplink channel estimation, so as many subcarriers as possible are allocated to pilot signals.

  FIG. 18 shows a configuration related to claim 3 at the time of filing of the OFDM precoding matrix control circuit 36 shown in FIG. First, a frequency domain received signal including transmission path information passes through terminals 34-1 and 34-2, and is input to the analog transmission path information extraction circuit 63 and the SNR information extraction circuit 87. The analog transmission path information extraction circuit 63 extracts the downlink channel impulse response as transmission path information from the frequency domain received signal, and outputs it to the DFT circuits 38-1 and 38-2 from the terminals 84-1 and 84-2. . In addition, the channel impulse response of the uplink is estimated, and the estimated value is input to the SNR information extraction circuit 87 from the terminals 84-3 and 84-4. The extracted channel impulse response of the downlink is converted into a channel frequency response by DFT in the DFT circuits 38-1 and 38-2. The channel frequency response of subcarrier number n is input to precoding matrix estimation circuit corresponding to n among precoding matrix estimation circuit (# 1) 64-1 to precoding matrix estimation circuit (#N) 64-2. The On the other hand, the SNR information extraction circuit 87 extracts the downlink SNR as the transmission path information from the frequency domain received signal and the estimated value of the uplink channel impulse response, and the precoding matrix estimation circuit (# 1) 64-1. To the precoding matrix estimation circuit (#N) 64-2. The precoding matrix estimation circuit (# 1) 64-1 to the precoding matrix estimation circuit (#N) 64-2 perform control of the minimum BER criterion precoding, and the precoding matrix estimation circuit (# 1) 64-1 The weighting coefficient of the linear processing circuit (# 1) 6-1 in FIG. 7 is estimated and output to the terminal 35-1. On the other hand, the precoding matrix estimation circuit (#N) 64-2 estimates the weighting coefficient of the linear processing circuit (#N) 6-2 in FIG. 7 and outputs it to the terminal 35-2.

  The configuration of the analog transmission line information extraction circuit 63 in FIG. 18 is shown in FIG. First, received signals in the frequency domain are input to the channel estimation circuit 85 from the terminals 34-1 and 34-2. This received signal in the frequency domain is obtained by receiving a feedback signal from the receiver side, and includes a known uplink pilot signal in addition to the transmission path information of the analog signal. Channel estimation circuit 85 estimates and outputs an uplink channel impulse response using this received signal and a known uplink pilot signal output from uplink pilot signal memory 48. Here, channel estimation circuit 85 and uplink pilot signal memory 48 correspond to channel estimation means. The estimated uplink channel impulse response is input to the impulse response extraction circuit 86 together with the frequency domain received signal. The impulse response extraction circuit 86 extracts a downlink channel impulse response corresponding to the transmission path information from the estimated uplink channel impulse response and the received signal in the frequency domain, which is the received feedback signal, and outputs a terminal 84. -1 and 84-2. Here, the impulse response extraction circuit 86 and the SNR extraction circuit 62 of FIG. 18 correspond to transmission path information extraction means.

  Next, the estimation of the uplink channel impulse response in the channel estimation circuit 85 in FIG. 19, the extraction of the downlink SNR in the SNR extraction circuit 62 in FIG. 18, and the downlink channel in the impulse response extraction circuit 86 in FIG. The extraction of the impulse response will be described in detail using mathematical expressions.

First, the received signals of the feedback signals input from the terminals 34-1 and 34-2 are assumed to be Y ^U ₁ (i, n) and Y ^U ₂ (i, n), respectively. This Y ^U _k (i, n), k = 1, 2

と表わすことができる。ただし、N^U _k（i，n）は上り回線における雑音信号のFFTである。H^U _kl（n）は上り回線のチャネル周波数応答であり、第k受信アンテナと第l送信アンテナ間のものである。このチャネルインパルス応答h^U _kl（τ）を

Can be expressed as However, N ^U _k (i, n) is the FFT of the noise signal in the uplink. H ^U _kl (n) is the channel frequency response of the uplink, and is between the k th receive antenna and the l th transmit antenna. This channel impulse response h ^U _kl (τ)

と仮定すると、H^U _kl（n）は

Assuming that H ^U _kl (n) is

となる。

It becomes.

First, Y ^U _k (i, n) in S _P (i) (i _U1 ≤ i ≤ i _U2 ) is used to estimate the channel impulse response of the uplink. From Equation 12, n∈S _P (i) (i _U1 ≤ i ≤ i _U2 )

となる。数式１６に数式１５を代入して、変形すると

It becomes. Substituting Equation 15 into Equation 16 and transforming it

となる。ただし、h^U _kとa^U（i，n）は以下で定める2（M＋1）次元ベクトルである。

It becomes. However, h ^U _k and a ^U (i, n) are 2 (M + 1) -dimensional vectors defined below.

最小２乗法によりh^U _kを推定し、その推定値をh^U _k，eとする。数式８と同様、h^U _k，eは正規方程式の解となり、次式で与えられる。

H ^U _k is estimated by the method of least squares, and the estimated value is set to h ^U _{k, e} . Similar to Equation 8, h ^U _{k, e} is the solution of the normal equation and is given by the following equation.

なお、この値は、周波数領域の受信信号Y^U _k（i，n）とパイロット信号a_l（i，n）から計算でき、チャネル推定回路８５の出力に相当する。

This value can be calculated from the frequency domain received signal Y ^U _k (i, n) and the pilot signal a _l (i, n), and corresponds to the output of the channel estimation circuit 85.

Next, in order to extract the SNR of the downlink, using N∈S _gamma a _{(i) (i U1 ≦ i} ≦ i U2) in _{^{Y U k (i, n)}} . From Equation 12 and Equation 13, the Y ^U _k (i, n) is putting the _{Γ = log 10 ^ γ / C} γ

となる。さらに、H^U _kl（n）を以下の推定値^H^U _kl（n）で置換える。

It becomes. Furthermore, replace H ^U _kl (n) with the following estimated value ^ H ^U _kl (n).

ただし、^h^U _kl，mはh^U＊ _k，eの要素である。置換えを行うと

Here, ^ h ^U _{kl, m} is an element of h ^{U *} _{k, e} . When replacing

が得られる。

Is obtained.

n∈S using _gamma a _{(i) (i U1 ≦ i} ≦ i U2) in _{^{Y U k (i, n)}} , to estimate the gamma ^* based on Equation 23 with the least squares method (which, according to the time of application (Related to item 7). If the estimated value is Γ ^* _e

が得られる。Γ_eを実数に限定し、Γ_e＝Re（Γ^* _e）とする。すなわち

Is obtained. Γ _e is limited to a real number and Γ _e = Re (Γ ^* _e ). Ie

となる。結局、^γの推定値γ_eは以下のように与えられる。

It becomes. After all, the estimated value γ _e of ^ γ is given by

最後に、下り回線のチャネルインパルス応答を抽出するため、n∈S_kl（i）（i_U1≦i≦i_U2）におけるY^U _k（i，n）を用いる。数式１２と数式１３から、このY^U _k（i，n）は

Finally, to extract the channel impulse response of the _{^{downlink, Y U k (i, n}} ) in _{n∈S kl (i) (i U1} ≦ i ≦ i U2) is used. From Equation 12 and Equation 13, this Y ^U _k (i, n) is

と表わすことができる。ここで、H^U _kl（n）を数式２２の^H^U _kl（n）で置換えた。さらに

Can be expressed as Here, replaced with H ^U _kl (n) the formula _{^{22 ^ H U kl (n)}} . further

として、~h^D _lk1，mを推定する。ここで、~h^D _lk1，mは本来の下り回線のチャネルインパルス応答のα倍であるが、~h^D _lk1，mを推定するのは、SNRが分ればチャネルインパルス応答を定数倍してもよいからである。

_˜h ^D _{lk1, m} is estimated as follows. Here, ~ h ^D _{lk1, m} is α times the original downlink channel impulse response, but ~ h ^D _{lk1, m} is estimated by multiplying the channel impulse response by a constant if SNR is known. Because it is good.

  Substituting Equation 29 into Equation 28 and transforming it similarly to Equation 17

となる。ただし、~h^D _k1と~a^U _k（i，n）は以下で定める2（M＋1）次元ベクトルである。

It becomes. However, ~ h ^D _k1 and ~ a ^U _k (i, n) are 2 (M + 1) -dimensional vectors defined below.

~h^D _k1を最小２乗法で推定する（これは、出願時の請求項７に関連する）。その推定値を~h^D _k1，eとすると、正規方程式の解

~ h ^D _k1 is estimated by the method of least squares (this relates to claim 7 as filed). If the estimated value is ~ h ^D _{k1, e} ,

が得られる。なお、この値は下り回線のチャネルインパルス応答に相当し、周波数領域の受信信号Y^U _k（i，n）と上り回線のチャネルインパルス応答の推定値から求められる。

Is obtained. This value corresponds to the downlink channel impulse response, and is obtained from the frequency domain received signal Y ^U _k (i, n) and the estimated value of the uplink channel impulse response.

  FIG. 20 shows a configuration related to claim 8 at the time of filing of the channel estimation / feedback signal generation circuit 56 of FIG. 16 is different from the channel estimation / feedback signal generation circuit 56 shown in FIG. 16 in that the downlink channel impulse response output from the channel estimation circuit 51 is a log compression circuit 65-1 and 65-2 which is a kind of nonlinear compression. It only compresses and inputs the value to the DFT circuits 38-5 and 38-6. Here, the channel estimation circuit 51, the downlink pilot signal memory 52, the SNR estimation circuit 61, the log compression circuits 65-1 and 65-2, and the DFT circuits 38-3 to 38-6 correspond to transmission path information estimation means. To do.

The operations of the log compression circuits 65-1 and 65-2 will be described using mathematical expressions. First, the estimated value of the channel impulse response for the downlink ^ h ^D _{lk, m} = e ^jθ | ^ h ^D _{lk, m} |

と変換する。ただし、θ＝arg（^h^D _lk，m）である。なお、~αは正の規格化定数であり

And convert. However, θ = arg (^ h ^D _{lk, m} ). Note that ~ α is a positive normalization constant.

を満足するように定める。C_Hは定数であり、送信電力の拘束条件を満足するよう予め定めておく。

To satisfy. C _H is a constant and is determined in advance so as to satisfy the constraint condition of transmission power.

The DFT circuit 38-5 and 38-6 performs DFT on the converted data as shown in Equation 34, and feeds it back instead of ^ H ^D _lk (n).

  When log compression, which is nonlinear compression, is performed, it is necessary to change the OFDM precoding matrix control circuit 36 of FIG. 7 from the configuration of FIG. The changed configuration is shown in FIG. The difference from the configuration of the OFDM precoding matrix control circuit 36 shown in FIG. 18 is that the downlink transmission path information output from the terminals 84-1 and 84-2 of the analog transmission path information extraction circuit 63 is subjected to inverse log compression. In the circuits 66-1 and 66-2, the inverse operation of the log compression circuits 65-1 and 65-2 in FIG. This inverse operation will be described below using mathematical expressions.

_Let ξ _{lk1, m} be the [(l−1) (M + 1) + m + 1] element of ~ h ^D _{k1, e in} Equation 33. This is the downlink transmission line information output from the terminals 84-1 and 84-2 of the analog transmission line information extraction circuit 63 in FIG. This was subject to the following operations in reverse log compression circuit 66-1 and 66-2, obtains an estimated value of ^{_{h D lk1, m ~ h D}} lk1, m.

このようにlog圧縮を行うことで、雑音に対する耐性を高めることができ、精度良く下り回線のチャネルインパルス応答を抽出することができる。なお、チャネルインパルス応答の代わりにチャネル周波数応答をlog圧縮することも、容易にできる。

By performing log compression in this way, it is possible to increase resistance to noise and to extract a downlink channel impulse response with high accuracy. Note that the log compression of the channel frequency response can be easily performed instead of the channel impulse response.

  Precoding that minimizes the BER under the same channel interference condition requires not only the channel frequency response of the desired wave but also the channel frequency response of the interference wave as transmission path information. For this reason, in order to operate the above precoding, it is also necessary to feed back the channel frequency response of the interference wave. The configuration of the channel estimation / feedback signal generation circuit 56 shown in FIG. 14 that enables such feedback is shown in FIG. 22 (this relates to claim 9 at the time of filing). Note that the interference wave is one wave for simplicity.

  The differences between the configurations shown in FIGS. 22 and 16 are as follows. First, (i) the desired wave channel estimation circuit 67 and the interference wave channel estimation circuit 68 have, in addition to the downlink pilot signal, the downlink pilot signal of the interference wave output from the interference wave downlink pilot signal memory 69. It is used to simultaneously estimate the downlink channel impulse response of the desired wave and the interference wave. This can be easily done by extending the above-mentioned estimation by the method of least squares. Next, (ii) the channel impulse response of the desired wave is converted into a channel frequency response by the DFT circuits 38-3 and 38-4, and the channel impulse response of the interference wave is channeled by the DFT circuits 38-7 and 38-8. Converted to frequency response. The frequency responses of the desired wave and the interference wave are input to the feedback signal generation circuit 78 and fed back. (Iii) The SNR estimation circuit 70 estimates the SNR using not only the channel frequency response of the desired wave but also the channel frequency response of the interference wave.

Here, desired wave channel estimation circuit 67, interference wave channel estimation circuit 68, downlink pilot signal memory 52, interference wave downlink pilot signal memory 69, SNR estimation circuit 70, and DFT circuits 38-3, 38-4, 38 −7 and 38-8 correspond to transmission path information estimation means. The uplink pilot signal memory 54 and the feedback signal generation circuit 78 are
This corresponds to feedback signal generation means.

  If the downlink pilot signal of the interference wave is not known, the interference wave downlink pilot signal memory 69 may be replaced with a circuit that estimates the downlink pilot signal of the interference wave. Even when there are two or more interference waves, it can be easily realized by extending the above configuration. Further, the downlink channel impulse response of the estimated interference wave can be used for coefficient control in linear processing for interference suppression, and deterioration due to interference in downlink signal detection can be suppressed.

  The configuration of the OFDM precoding matrix control circuit 36 of FIG. 7 that can correspond to the configuration change of FIG. 22 is shown in FIG. 23 (this relates to claim 10 at the time of filing). First, a received signal in the frequency domain including transmission path information passes through terminals 34-1 and 34-2, a desired wave analog transmission path information extraction circuit 71, an interference wave analog transmission path information extraction circuit 72, and an SNR information extraction circuit 86. Is input. The desired wave analog transmission line information extraction circuit 71 extracts the channel impulse response of the desired wave downlink as transmission line information from the received signal in the frequency domain, and the DFT circuits 38-1 and 38- from the terminals 84-1 and 84-2. Output to 2. In addition, the channel impulse response of the desired wave uplink is estimated, and the estimated value is input to the SNR information extraction circuit 86 from the terminals 84-3 and 84-4. The interference wave analog transmission path information extraction circuit 72 extracts the channel impulse response of the interference wave downlink as transmission path information from the received signal in the frequency domain, and the DFT circuits 38-9 and 38- from the terminals 84-5 and 84-6. 10 is output. In addition, the channel impulse response of the interference wave uplink is estimated, and the estimated value is input to the SNR information extraction circuit 86 from the terminals 84-7 and 84-8. The channel impulse response extracted from the desired wave downlink is converted into a channel frequency response by DFT in the DFT circuits 38-1 and 38-2. The extracted channel impulse response of the interference wave downlink is converted into a channel frequency response by DFT in the DFT circuits 38-9 and 38-10. The channel frequency response of the subcarrier number n is input to the precoding matrix estimation circuit corresponding to n among the precoding matrix estimation circuit (# 1) 73-1 to the precoding matrix estimation circuit (#N) 73-2. The On the other hand, the SNR information extraction circuit 86 extracts the downlink SNR as transmission path information from the frequency domain received signal and the estimated value of the uplink channel impulse response, and precoding matrix estimation circuit (# 1) 73-1 To the precoding matrix estimation circuit (#N) 73-2. The precoding matrix estimation circuit (# 1) 73-1 to the precoding matrix estimation circuit (#N) 73-2 control the minimum BER reference precoding under the co-channel interference condition, and the precoding matrix estimation circuit ( # 1) 73-1 estimates the weighting coefficient of the linear processing circuit (# 1) 6-1 in FIG. 7 and outputs it to the terminal 35-1. On the other hand, the precoding matrix estimation circuit (#N) 73-2 estimates the weighting coefficient of the linear processing circuit (#N) 6-2 in FIG. 7 and outputs it to the terminal 35-2.

  Note that the estimated uplink channel impulse response of the interference wave can be used for coefficient control in linear processing for interference suppression, and deterioration due to interference in uplink signal detection can be suppressed.

  In the description so far, it has been implicitly assumed that the channel impulse response of the transmission path is time-invariant. In an actual transmission path, the channel impulse response varies with time due to Doppler variations. When the speed fluctuates so fast that it cannot be ignored, the fed back transmission path information becomes old, and precoding control cannot be performed with high accuracy, resulting in a problem that precoding characteristics deteriorate. In order to suppress this degradation, it is effective to linearly predict current transmission path information from past transmission path information. The configuration of the OFDM precoding matrix control circuit 36 of FIG. 7 that performs linear prediction is shown in FIG. 24 (this relates to claim 11 at the time of filing).

  24 differs from the configuration of FIG. 18 in the following points. First, (i) the downlink channel impulse response output from the analog transmission path information extraction circuit 63 is input to the linear prediction circuit 74. Further, (ii) the linear prediction circuit 74 estimates the current channel impulse response from the information of the past downlink channel impulse response and the extracted downlink SNR, and sends it to the DFT circuits 38-1 and 38-2. Output.

  The linear prediction circuit 74 linearly predicts the current channel impulse response by multiplying the past downlink channel impulse response by a prediction coefficient and adding them together. The prediction coefficient may be a fixed value obtained by calculation in advance or a value estimated by an adaptive algorithm. Note that downlink SNR information is used, and this is used to eliminate the uncertainty to some extent because the output of the analog transmission path information extraction circuit 63 has an uncertainty of a constant multiple.

  In addition, this time fluctuation can be followed by performing linear prediction of transmission path information in the channel estimation means. This can be realized by using the input of the linear prediction circuit 74 of FIG. 24 as the output of the channel estimation circuit 51 of FIG. 16 and DFT of the output of the linear prediction circuit 74 (this is related to claim 12 at the time of filing). To do).

  In the above, the best mode for carrying out the present invention has been described by taking MIMO-OFDM transmission as an example, but it can be easily realized even in the case of MIMO single carrier transmission. In this case, as transmission path information, a downlink signal-to-noise ratio and a channel impulse response may be time-multiplexed with a known uplink pilot signal to perform feedback (this is claimed in claim 2 and claim at the time of filing). (Related to item 4).

  In addition, the description has been made with the transmission side as a base station and the reception side as a mobile terminal, but it is naturally possible to use the transmission side as a mobile terminal and the reception side as a base station. It is only necessary to replace the “downlink”.

  The present invention is not limited to the best mode for carrying out the above-described invention, and various other configurations can be adopted without departing from the gist of the present invention.

1 input terminal, 2 modulation circuit, 3 serial / parallel converter, 4 multiplier, 5 adder, 6 linear processing circuit, 7 D / A converter, 8 multiplication circuit, 9 transmission amplifier, 10 oscillator, 11 transmission antenna, 12 Up converter, 13 receiving antenna, 14 receiving amplifier, 15 multiplication circuit, 16 low-pass filter, 17 A / D converter, 18 down converter, 19 terminal, 20 terminal, 21 precoding matrix control circuit, 22 oscillator, 23 transmitting / receiving antenna , 24 input terminals, 25 output terminals, 26 precoding index extraction circuit, 27 precoding matrix selection circuit, 28 analog transmission line information extraction circuit, 29 serial / parallel converter, 30 IFFT circuit, 31 guard interval addition circuit, 32 guards Interval removal circuit, 33FFT circuit, 34 terminals, 35 terminals, 36OFDM preco Coding matrix control circuit, 37 digitized transmission line information extraction circuit, 38 DFT circuit, 39 precoding matrix estimation circuit, 40 precoding index extraction circuit, 41 precoding matrix estimation circuit, 42 analog transmission line information extraction circuit, 43 channel estimation Circuit, 44 downlink pilot signal memory, 45 signal detection circuit, 46 output terminal, 47 feedback signal generation circuit, 48 uplink pilot signal memory, 49 channel estimation / feedback signal generation circuit, 50 serial / parallel converter, 51 channel estimation Circuit, 52 downlink pilot signal memory, 53 linear synthesis circuit, 54 uplink pilot signal memory, 55 feedback signal generation circuit, 56 channel estimation / feedback signal generation circuit, 58 serial / parallel converter, 59 terminal Child, 60 terminals, 61 SNR estimation circuit, 62 feedback signal generation circuit, 63 analog transmission line information extraction circuit, 64 precoding matrix estimation circuit, 65 log compression circuit, 66 inverse log compression circuit, 67 desired wave channel estimation circuit, 68 interference wave Channel estimation circuit, 69 interference wave downlink pilot signal memory, 70 SNR estimation circuit, 71 desired wave analog transmission line information extraction circuit, 72 interference wave analog transmission line information extraction circuit, 73 precoding matrix estimation circuit, 74 linear prediction circuit, 75 Circulator, 76 digitized transmission line information extraction circuit, 77 precoding matrix estimation circuit, 78 feedback signal generation circuit, 79 channel estimation circuit, 80 impulse response extraction circuit, 81 terminal, 82 channel estimation circuit, 83 impulse response extraction circuit, 84 Terminal, 85 channel estimation circuit 86 impulse response extraction circuit, 87SNR information extraction circuit

Claims (12)

Transmission path information estimation means for estimating downlink transmission path information using a downlink received signal and a known downlink pilot signal;
A plurality of the estimated transmission path information as analog signals independently, a feedback signal generating means for generating a feedback signal by multiplexing with a known uplink pilot signal;
Channel estimation means for estimating an uplink channel impulse response from the feedback signal received through the uplink channel and the known uplink pilot signal;
A transmission path information feedback system comprising transmission path information extraction means for extracting the transmission path information from the estimated channel impulse response of the uplink and the received feedback signal.   The transmission path information feedback system according to claim 1, wherein the transmission path information includes a downlink signal-to-noise ratio and a channel impulse response.   2. The transmission path information feedback system according to claim 1, wherein the transmission path information includes a downlink signal-to-noise ratio and a channel frequency response.   2. The transmission path information feedback system according to claim 1, wherein the feedback signal generation means generates a feedback signal by time-multiplexing the transmission path information and the known uplink pilot signal.   2. The transmission path information feedback system according to claim 1, wherein said feedback signal generation means frequency feedback multiplexes said transmission path information and said known uplink pilot signal to generate a feedback signal.   The transmission path according to claim 1, wherein the transmission path information estimation means estimates the transmission path information by a least square method using the downlink reception signal and the known downlink pilot signal. Information feedback system.   2. The transmission according to claim 1, wherein the transmission path information extracting unit extracts the transmission path information from the estimated channel impulse response of the uplink and the received feedback signal by a least square method. Road information feedback system.   2. The transmission path information feedback system according to claim 1, wherein the feedback signal generation means nonlinearly compresses the amplitude of the transmission path information and uses the value as an analog signal.   2. The transmission path information feedback system according to claim 1, wherein said transmission path information estimation means estimates transmission path information of not only a desired wave but also an interference wave.   2. The transmission path information feedback system according to claim 1, wherein the transmission path information extracting unit extracts transmission path information of not only a desired wave but also an interference wave.   2. The transmission path information feedback system according to claim 1, wherein the transmission path information extracting means linearly predicts current transmission path information from the transmission path information extracted in the past.   The transmission path information feedback system according to claim 1, wherein the transmission path information estimation means linearly predicts future transmission path information from the estimated transmission path information.

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