JP2011103610A - Reception method, reception device, noncontact ic card, and reader/writer - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To properly receive a Manchester encoded radio signal while adaptively equalizing turbulence of reception waveform caused by an increase in communication rate. <P>SOLUTION: When half-sampling a Manchester encoded signal, the shortest wavelength and the longest wavelength correspond to one pulse portion and a DC component, respectively, and thereby a bandwidth is 1/2 of a Manchester code, and is distributed around 0 MHz. Even when the number of taps of an FIR filter is the same, density of frequency positions controllable in the vicinity of the central distribution of a spectrum increases, an inverse characteristic of a frequency characteristic can be further faithfully expressed, and excellent adaptive equalization can be performed. <P>COPYRIGHT: (C)2011,JPO&INPIT

Description

  The present invention relates to a receiving method and receiving apparatus, a non-contact IC card, and a reader / writer for receiving and processing a radio signal encoded by Manchester in a non-contact communication system using electromagnetic coupling, and in particular, to increase the communication rate. The present invention relates to a receiving method and a receiving apparatus, a non-contact IC card, and a reader / writer that receive and process a Manchester-encoded radio signal while adaptively equalizing a received waveform disturbance accompanying the.

  Non-contact communication is a wireless technology that transmits data at a transmission distance of about 0 to several tens of centimeters, and is applied to an RFID system including a non-contact IC card and a reader / writer, for example. Depending on the communication direction, communication from the reader / writer to the card and communication from the card to the reader / writer can be divided into two types. In the present specification, the former is called “downlink” and the latter is called “uplink”. In either communication direction, the reader / writer always oscillates the carrier frequency, and the card performs transmission processing and reception processing based on the power obtained from this carrier frequency.

  Non-contact communication methods include an electrostatic coupling method, an electromagnetic induction method, a radio wave communication method, and the like. Among these, in the electromagnetic induction method, data communication is performed by magnetic coupling of the coil between the primary coil on the reader / writer side and the secondary coil on the card side (that is, the operation of the two coils as an LC resonance circuit). The reader / writer performs downlink data transmission by amplitude-modulating the magnetic field generated by the primary coil, that is, the carrier, and detects this on the transponder side. On the other hand, load modulation is used for the uplink, and the load resistance of the secondary coil is switched based on transmission information in the card. On the reader / writer side, when the load of the secondary coil that is electromagnetically coupled changes, the input impedance of the electromagnetic coupling changes, and as a result, the output level of the carrier frequency changes. Therefore, the transmission information from the card can be read by looking at this level fluctuation.

  For example, ISO / IEC IS 18092 (NFC IP-1) is a non-contact communication standard that defines the specifications of a reader / writer, which became an international standard in December 2003. The standard originally inherited Sony's “FeliCa” and Philips' “Mifare”, which are widely used as contactless IC cards. Among them, in the Felica format, a Manchester code is used, and the same packet structure is used in the downlink and the uplink. FIG. 17 shows a packet structure of Felica format. The illustrated packet is composed of three parts, “preamble”, “sink”, and “data”. The preamble is composed of a “0” sequence of 6 bytes, and the sync is a known sequence “0xB24D” of 2 bytes. The data includes a 1-byte LEN indicating the packet length, a (LEN-1) -byte data body (payload), and a 2-byte CRC (Cyclic Redundancy Check) code. All three parts are Manchester encoded.

  Here, for example, when a binary value “0” is transmitted, the Manchester code is changed from a low level to a high level at the center of the bit interval (input 0 is changed to “01”), while the binary value “0” is set. Conversely, when sending 1 ″, it is changed from the high level to the low level at the center of the bit interval (input 1 is changed to “10”). In other words, one bit interval is divided into a front cell and a rear cell at the center, and when the front cell is low level and the rear cell is high level, the logical value is “0”, and the front cell is high. When the level and the rear cell are at a low level, the code format is a logical value “1”. Manchester code can also be said to convert 1 input bit into 2 bits (or transmit 1 bit with 2 pulses (2T)), and the communication rate is halved by expanding to 2 times the band. However, the DC component of the transmission signal is eliminated.

  In the preamble part, 6 bytes of 0 are Manchester encoded. Therefore, “01” becomes a continuous waveform lasting 48 times. Further, the sync part is composed of a pattern obtained by Manchester encoding “0xB24D”. In the data part, transmission information, Length information (LEN), and CRC are combined and Manchester encoded.

  On the packet receiving side, clock (sampling timing) is extracted based on the preamble portion which is a continuous waveform. In this specification, this operation is called “timing synchronization”. Subsequently, a sync portion including a pattern obtained by Manchester encoding “0xB24D” is detected, and the start position of the subsequent data portion is estimated. In this specification, this operation is called “frame synchronization”. Then, the data portion is decoded based on this start position.

  Several proposals have been made for a receiving circuit that decodes a Manchester code into an NRZ (Non Return to Zero: NRZ) code (see, for example, Patent Documents 1 to 3).

  By the way, in the Felica format, 424 kbps, 848 kbps, 1.7 Mbps, 3.4 Mbps, etc., which are multiples of 212 kbps, are defined as communication rates. As the communication rate increases, the frequency band of the transmission signal increases in proportion thereto. As the frequency band of the signal becomes wider, the influence of the frequency characteristics of the transmission path, transmission RF analog circuit, and reception RF analog circuit increases. In general, the higher the frequency of these frequency characteristics, the greater the attenuation. Also, the higher the frequency, the greater the disturbance of the phase characteristics. For this reason, the higher the communication rate signal, the greater the disturbance of the received waveform.

  An adaptive equalization process can be cited as one method for compensating for a disturbance of a received signal in high-speed communication or the like (see, for example, Patent Documents 4 to 6). The adaptive equalization circuit includes, for example, an FIR (Finite Impulse Response) filter and a learning circuit. FIG. 18 schematically shows the configuration of the FIR filter. The FIR filter has a delay line in which a plurality of delay elements are connected in series, and the time-series input data corresponding to the number of arranged delay elements is multiplied by tap coefficients corresponding to the characteristics of the filter. After weighting, an equalized signal can be obtained by accumulating and averaging them. A learning signal known to the receiving side is transmitted from the transmitting side. A random pattern is usually used for the learning signal. When the learning circuit on the receiving side receives a disturbed learning signal through the transmission line, it adjusts the tap coefficient of the filter so that the equalized signal output from the FIR filter approaches the desired signal.

  In order to perform adaptive equalization, it is necessary to transmit a random pattern sequence having a length sufficient to learn tap coefficients of the FIR filter. On the other hand, in order to decode the data portion in the packet from the beginning, it is necessary to complete the learning of the FIR filter at an earlier stage.

  In order to complete the learning of the FIR filter before the data part arrives, a method of inserting a random pattern long enough for learning between the sink part and the data part, or a special packet for learning as a normal packet A method of transmitting prior to the transmission can be considered. However, in order to realize these methods, a packet format different from the Felica format defined in the NFC IP-1 standard is used, which may cause a problem in compatibility. In addition, since the time for transmitting information is reduced due to the time for transmitting a learning random pattern that is a known signal, the communication rate is also lowered.

Japanese Patent Laid-Open No. 11-146022 JP-A-11-251916 JP 2005-160042 A JP 2004-64681 A JP 2008-22422 A JP 2008-27270 A

  An object of the present invention is to provide an excellent receiving method and receiving apparatus, non-contact IC card, and reader / writer capable of suitably receiving and processing a radio signal encoded by Manchester in a non-contact communication system using electromagnetic coupling. Is to provide.

  A further object of the present invention is to suitably receive and process a Manchester encoded radio signal in a non-contact communication system using electromagnetic coupling while adaptively equalizing reception waveform disturbance accompanying an increase in communication rate. An object of the present invention is to provide an excellent receiving method and receiving apparatus, non-contact IC card, and reader / writer.

The present application has been made in consideration of the above problems, and the invention according to claim 1
An inclination calculation step for calculating an inclination of the received signal;
A sampling step of ½ sampling the received signal whose slope has been calculated;
An adaptive equalization step for adaptively equalizing the received signal after the ½ sampling,
It is a receiving method characterized by having.

The invention according to claim 2 of the present application is a receiving method for receiving a packet with learning bits encoded by Manchester,
A detection step of detecting a start position of the learning bit;
A delay time giving step for giving a delay to the reception signal so that the head of the learning bit is not output until the detection of the learning bit is confirmed in the detection step;
A slope calculation step for calculating a difference between the received signal given the delay time and a received signal further given a delay time equivalent to one clock in Manchester code;
A 1/2 sampling step of extracting from the head of the learning bit that has been subjected to the slope calculation at intervals of once every two Manchester code clocks;
An adaptive equalization step for performing adaptive equalization using the learning bits sampled in half.
A binary determination step of performing binary determination on the adaptively equalized reception signal and decoding the Manchester code into an NRZ code;
It is a receiving method characterized by having.

  According to claim 3 of the present application, in the delay time adding step in the reception method according to claim 2, the received learning bit is stored, and a predetermined number of repetitions is started from the timing at which the learning bit is detected. In the only read and adaptive equalization step, the received learning bit is learned by the number of repetitions.

The invention according to claim 4 of the present application is
An inclination calculation unit for calculating an inclination of the received signal;
A sampling unit that ½ samples the received signal subjected to the inclination calculation;
An adaptive equalization unit that adaptively equalizes the received signal after the ½ sampling,
It is a receiver characterized by comprising.

The invention according to claim 5 of the present application is a non-contact communication system using electromagnetic coupling, which is a Manchester encoded packet composed of a preamble portion comprising a continuous waveform, a sync portion comprising a specific pattern, and a data portion. A non-contact IC card that receives
A preamble detector that detects a preamble portion from the received signal and extracts a sampling timing based on the continuous waveform; and
A sync detector that detects a sync portion from a received signal based on the sampling timing and outputs a timing signal indicating a start position of the sync portion;
A delay time giving unit that gives a delay to the reception signal so that the head of the sync unit is not output until the detection of the sync unit is confirmed in the sync detection step;
A slope calculator that calculates a difference between the received signal given the delay time and a received signal further given a delay time equivalent to one clock in Manchester code;
A 1/2 sampling unit that extracts from the head of the sync unit for which the inclination is calculated at an interval of once every two Manchester code clocks;
An adaptive equalization unit that performs adaptive equalization using a ½ sampled sync unit based on the timing signal;
A binary determination unit for performing binary determination on the adaptively equalized reception signal and decoding the Manchester code into an NRZ code;
It is a non-contact IC card characterized by comprising.

The invention according to claim 6 of the present application is a non-contact communication system using electromagnetic coupling, which is a Manchester encoded packet composed of a preamble part composed of a continuous waveform, a sync part composed of a specific pattern, and a data part. Is a reader / writer that receives non-contact IC cards,
A preamble detector that detects a preamble portion from the received signal and extracts a sampling timing based on the continuous waveform; and
A sync detector that detects a sync portion from a received signal based on the sampling timing and outputs a timing signal indicating a start position of the sync portion;
A delay time giving unit that gives a delay to the reception signal so that the head of the sync unit is not output until the detection of the sync unit is confirmed in the sync detection step;
A slope calculator that calculates a difference between the received signal given the delay time and a received signal further given a delay time equivalent to one clock in Manchester code;
A 1/2 sampling unit that extracts from the head of the sync unit for which the inclination is calculated at an interval of once every two Manchester code clocks;
An adaptive equalization unit that performs adaptive equalization using a ½ sampled sync unit based on the timing signal;
A binary determination unit for performing binary determination on the adaptively equalized reception signal and decoding the Manchester code into an NRZ code;
A reader / writer characterized by comprising:

  According to the present invention, in a non-contact communication system using electromagnetic coupling, an excellent receiving method and receiving apparatus, non-contact IC card, and reader / writer capable of suitably receiving and processing a Manchester-encoded radio signal Can be provided.

  In addition, according to the present invention, in a non-contact communication system using electromagnetic coupling, it is possible to suitably receive and process a Manchester encoded radio signal while adaptively equalizing reception waveform disturbance accompanying an increase in communication rate. It is possible to provide an excellent receiving method and receiving apparatus, contactless IC card, and reader / writer.

  According to the invention described in claims 1 to 3, 5, and 6 of the present application, it is possible to perform adaptive equalization to improve more complicated frequency characteristics while using a FIR filter having a smaller number of taps, and use electromagnetic coupling. By applying to a non-contact communication system, data communication at a high communication rate can be performed stably. Further, since 1/2 sampling is performed before adaptive equalization, the circuit constituting the adaptive equalization unit only needs to operate at 1/2 speed, which leads to reduction in power consumption.

  According to the invention described in claim 4 of the present application, the number of bits of the sync part used for learning of the tap coefficient by the adaptive equalization is reduced by half as the 1/2 sampling is performed before the adaptive equalization. The tap coefficient convergence performance can be maintained by learning tap coefficients for a predetermined number of repetitions.

  Other objects, features, and advantages of the present invention will become apparent from more detailed description based on embodiments of the present invention described later and the accompanying drawings.

FIG. 1 is a diagram illustrating a configuration example of a contactless communication system. FIG. 2 is a flowchart showing a processing procedure executed by the reader / writer 10 when performing data transmission / reception processing in the communication system shown in FIG. FIG. 3 is a flowchart showing a processing procedure executed by the non-contact IC card 30 when data transmission / reception processing is performed in the communication system shown in FIG. FIG. 4 is a diagram schematically showing a configuration example of a receiving circuit that performs adaptive equalization using the Felica format as it is. FIG. 5 is a diagram illustrating an internal configuration example of the adaptive equalization unit 44. FIG. 6 is a diagram illustrating an internal configuration example of the Manchester decoding unit 45. FIG. 7 is a diagram for explaining a procedure for decoding a Manchester-encoded signal. FIG. 8A is a diagram illustrating a spectrum of a Manchester encoded signal having a communication rate of 3.4 Mbps (provided that the communication rate is 3.4 Mbps). FIG. 8B is a diagram illustrating a spectrum of an NRZ-encoded signal having a communication rate of 3.4 Mbps (provided that the communication rate is 3.4 Mbps). FIG. 9 is a diagram exemplifying frequency positions at which each tap of the FIR filter having 5 taps can be expressed within the baseband bandwidth 6.8 MHz. FIG. 10 is a diagram illustrating a configuration example of a receiving circuit that enables adaptive equalization that improves a more complicated frequency characteristic with an FIR filter having a smaller number of taps. FIG. 11 is a diagram for explaining a procedure for decoding the Manchester-encoded signal into the NRZ code by the subsequent processing from the delay buffer 103 in the receiving circuit shown in FIG. FIG. 12 is a diagram illustrating frequency positions at which each tap of an FIR filter with 5 taps can be expressed within a bandwidth of 3.4 MHz. FIG. 13 is a diagram showing an equalized signal point distribution when the Manchester encoded signal output from the delay buffer 43 is directly adaptively equalized in the receiving circuit shown in FIG. FIG. 14 is a diagram illustrating the equalized signal point distribution when the Manchester encoded signal is ½ sampled and then adaptively equalized in the receiving circuit illustrated in FIG. 10. FIG. 15 is a diagram illustrating a configuration example of a receiving circuit in which the problem of a decrease in the number of bits of the sync unit used for learning due to 1/2 sampling is further solved. FIG. 16 is a diagram illustrating a data string output from the RAM control unit 903 to the subsequent circuit when the number of repetitions of the sync unit is 2. FIG. 17 is a diagram illustrating a packet structure of Felica format. FIG. 18 is a diagram schematically showing the configuration of the FIR filter.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

  FIG. 1 shows a configuration example of a non-contact communication system. The illustrated system includes a reader / writer 10 and a non-contact IC card 30, and for example, packets of FeliCa format (described above) are exchanged between the two by a predetermined communication procedure.

  The reader / writer 10 includes a control unit 11, a transmission circuit 12, a reception circuit 13, and an antenna resonance circuit unit 14. On the other hand, the non-contact IC card 30 includes a control unit 31, an antenna resonance circuit unit 32, and a load switching modulation circuit unit 33. In the illustrated example, the control unit 34 includes a transmission circuit, a reception circuit, a logic circuit, and a nonvolatile memory such as an EEPROM (Electrically Erasable and Programmable Memory).

  The control unit 11 controls each unit in the reader / writer 10 to perform processing for transmitting and receiving data. At the time of downlink data transmission, the transmission circuit 12 includes a carrier oscillator (not shown). After transmission data is Manchester-encoded, the carrier is amplitude-modulated, for example, and data is transmitted. Further, the carrier continues to be transmitted from the transmission circuit 12 when receiving uplink data.

Antenna resonant circuit section 14 of the reader writer 10 side is constituted by a parallel resonance circuit consisting of coil L ₁₀ and the capacitor C _10, which acts as a primary coil. The resonance frequency is set in the vicinity of the carrier frequency generated by the transmission circuit 12.

On the other hand, the non-contact IC card 30 side antenna resonance circuit section 32 is composed of a coil L ₃₀ and the capacitor C _30, which acts as a secondary coil. The resonant frequency of the antenna resonant circuit section 32 is set to a predetermined value by the inductance of the capacitance and the coil L ₃₀ of the capacitor C _30. Normally, the antenna resonance circuit unit 32 is set around the carrier frequency, so that it is electromagnetically coupled to the antenna resonance circuit unit 14 on the reader / writer 10 side. Coil L ₁₀ and the coil L ₃₀ is magnetically coupled with a coupling coefficient K _13, its value becomes larger as both positions approaches.

A load switching modulation circuit unit 33 is connected to the antenna resonance circuit unit 32 in parallel. The load switching modulation circuit unit 33 includes a resistor R ₃₁ connected in series and a transistor switch Q ₃₁ made of a MOS (Metal Oxide Semiconductor). By turning on / off the transistor switch Q ₃₁ , antenna resonance occurs. Load modulation of the circuit unit 32 is performed.

  At the time of downlink data reception, the antenna resonance circuit unit 32 supplies a reception signal from the antenna resonance circuit unit 14 of the reader / writer 10 to the control unit 34. In the control unit 34, after demodulating the received signal, Manchester decoding is performed to reproduce the original transmission data. Note that adaptive equalization processing is necessary before Manchester decoding in order to compensate for disturbance of a received signal in high-speed communication or the like. Details of this point will be described later.

Further, at the time of uplink data transmission, the control unit 34 performs Manchester encoding on the transmission data. Carriers continue to be transmitted from the reader / writer 10 side, and a magnetic field is generated in the coil L ₃₀ of the antenna resonance circuit unit 12. Load switching modulation circuit 33, a magnetic field in response to the transmission data by switching the ON / OFF of the MOS switch Q ₃₁ in accordance with the Manchester encoded bit sequence consists of 1 and 0 are supplied from the control unit 34 The load is modulated and data is transmitted to the antenna resonance circuit unit 12 of the reader / writer 10.

The reader writer 10 side, the load of the secondary coil L ₃₀ which is electromagnetically coupled are changed, the input impedance of the electromagnetic coupling varies, as a result, the output level of the carrier frequency varies. Therefore, the receiving circuit 13 can read the transmission data from the non-contact IC card 30 by seeing the level fluctuation, and when the original decoding data is reproduced by the Manchester decoding process, it is passed to the control unit 11. Note that adaptive equalization processing is necessary before Manchester decoding in order to compensate for disturbance of a received signal in high-speed communication or the like. Details of this point will be described later.

  Next, data transmission / reception processing in the communication system shown in FIG. 1 will be described with reference to the flowcharts shown in FIGS. However, the flowchart of FIG. 2 shows a processing procedure executed by the reader / writer 10, and the flowchart of FIG. 3 shows a processing procedure of the non-contact IC card 30.

  The transmission circuit 12 of the reader / writer 10 generates a carrier frequency (step S1).

  The transmission circuit 12 encodes the transmission data acquired from the control unit 11 into a Manchester code (step S2).

  Next, the transmission circuit 12 amplitude-modulates the carrier generated in step S1 based on the encoded data (step S3).

  The modulation signal amplitude-modulated in step S3 is supplied to the antenna resonance circuit unit 14 (step S4). The antenna resonance circuit unit 14 generates a magnetic field according to the supplied modulation signal.

  As a result of the magnetic coupling by the magnetic field generated in step S4, an electromotive force is induced in the antenna resonance circuit unit 32 on the non-contact IC card 30 side (step S21).

  The non-contact IC card 30 includes an IC power generation circuit (not shown), forms a power circuit based on the electromotive force induced in step S21, and supplies necessary power to each unit (step S22). . The non-contact IC card 30 extracts a clock component from the electromotive force induced in step S21 (step S23).

  The receiving circuit in the control unit 34 demodulates the modulation signal subjected to amplitude modulation based on the voltage amplitude change of the electromotive force induced in step S21 (step S24).

  The signal demodulated in step S24 is encoded into Manchester code. The receiving circuit in the control unit 34 performs Manchester decoding on the demodulated signal (step S25). Note that adaptive equalization processing is necessary before Manchester decoding in order to compensate for disturbance of a received signal in high-speed communication or the like. Details of this point will be described later.

  A logic circuit (not shown) in the control unit 34 stores decoded data in an EEPROM (not shown) or the like in accordance with a predetermined program set in advance, or stores data stored in a nonvolatile manner. Read or delete. Also, transmission data to the reader / writer 10 is created by this logic circuit (step S26).

  The encoding / decoding circuit 39 encodes the transmission information created in step S26 into Manchester code and supplies it to the load switching modulation circuit unit 33 (step S27).

Carriers continue to be transmitted from the reader / writer 10 side, and a magnetic field is generated in the coil L ₃₀ of the antenna resonance circuit unit 12. Load switching modulation circuit 33, by switching on / off of the MOS switch Q ₃₁ according to the bit sequence of Manchester coded data, by changing the impedance of the antenna resonant circuit section 32, which load modulation of the magnetic field ( Step S28). Thereby, the Manchester encoded data is transmitted to the antenna resonance circuit unit 12 of the reader / writer 10 (step S29).

  An unmodulated carrier flows in the antenna resonance circuit section 14 of the reader / writer 10, and a voltage amplitude change corresponding to the impedance change generated in step S29 is induced in the carrier. The antenna resonance circuit unit 14 receives the signal from the non-contact IC card 30 by detecting this change (step S5).

  The receiving circuit 13 demodulates the load-modulated signal based on the voltage amplitude change induced in step S5 (step S6). Since the demodulated signal is encoded into Manchester code, the receiving circuit 13 further performs Manchester decoding on the demodulated signal (step S7), reproduces the transmission data, and supplies it to the control unit 11. Note that an adaptive equalization process is necessary before Manchester decoding in order to compensate for disturbances in received signals in high-speed communication or the like (same as above).

  As already mentioned, Manchester codes are used in Felica, and the same packet format (see FIG. 17) is used in the downlink and uplink.

  Further, when the communication rate between the reader / writer 10 and the non-contact IC card 30 is increased to 424 kbps, 848 kbps, 1.7 Mbps, and 3.4 Mbps, which are multiples of 212 kbps, the frequency band of the transmission signal is proportionally increased. As the frequency becomes wider, the influence of the frequency characteristics of the RF analog circuit in the transmission path, the transmission circuit, and the reception circuit increases, and the disturbance of the reception waveform increases. That is, if it is intended to realize high-speed communication in non-contact communication using electromagnetic coupling, adaptive equalization on the receiving side is required to compensate for the deterioration of frequency characteristics.

  Therefore, in the present embodiment, for example, at least one receiving circuit of the reader / writer 10 or the non-contact IC card 30 employs adaptive equalization to compensate for disturbance of the received signal.

  In addition, a sufficiently long signal sequence is required for learning in order to perform adaptive equalization. However, when such a signal sequence is transmitted, the Felica format (FIG. 17) defined in the NFC IP-1 standard is used. There is a possibility that the communication rate may decrease due to a compatibility problem or an increase in overhead.

  Therefore, in the present embodiment, the Felica format is used as it is, and adaptive equalization is performed using the packet sync part as a learning bit. This avoids compatibility problems and increased overhead. Actually, by providing a delay buffer in the receiving circuit, both frame synchronization using the sync unit and adaptive equalization using the same sync unit are realized.

  FIG. 4 schematically shows a configuration example of a receiving circuit that performs adaptive equalization using the Felica format as it is. The illustrated receiving circuit includes a preamble detection unit 41, a sync detection unit 42, a delay buffer 43, an adaptive equalization unit 44, and a Manchester decoding unit 45. Hereinafter, the operation of each unit when receiving a packet having the Felica format shown in FIG. 17 and performing adaptive equalization using the sync unit as a learning bit will be described.

  The received signal is first input to the preamble detector 41. When the preamble detection unit 41 detects a preamble portion that is a continuous waveform from the received signal, the preamble detection unit 41 extracts sampling timing based on the continuous waveform and obtains timing synchronization.

  Next, the received signal is input to the sync detector 42. Based on the received signal and the sampling timing extracted by the preamble detection unit 41, the sync detection unit 42 detects a sync unit including a bit sequence in which “0xB24D” is Manchester-encoded, thereby obtaining frame synchronization. Generally, methods such as pattern matching and cross-correlation are used for detection of the sync part. Each of the detection methods seeks the identity with a known specific pattern of the sink part. Usually, in order to increase noise resistance, the pattern identity is confirmed using almost the entire sink portion. When the sync detector 42 detects the sync portion from the received signal, the sync detector 42 outputs a timing signal indicating the start position of the sync portion.

  On the other hand, the received signal is also input to the delay buffer 43. The delay buffer 43 gives a delay equal to or longer than the length of the sync unit to the input reception signal. More precisely, a delay is provided so that the head of the sync unit is not output until the sync detection unit 42 determines the detection of the sync unit. For example, if the sync detection unit 42 can achieve frame synchronization using only the first half of the sync unit, the delay amount of the delay buffer 43 can be about half that of the sync unit. Alternatively, if the sync detection unit 42 uses the entire sync unit to perform frame synchronization, the delay amount of the delay buffer 43 needs to be equal to the entire sync unit.

  The received signal delayed by the delay buffer 43 is input to the adaptive equalization unit 44. The downstream Manchester decoding unit 45 then Manchester-decodes the equalized output signal from the adaptive equalization unit 44 to reproduce the original binary information bits.

  The adaptive equalization unit 44 is a learning type equalization circuit composed of an FIR filter and a learning circuit. Based on the timing signal indicating the start position of the sync unit output from the sync detection unit 42, the adaptive equalization unit 44 internally includes Compare with the sync pattern you have. Then, the tap coefficient of the FIR filter is adjusted so that the difference becomes small.

  FIG. 5 shows an internal configuration example of the adaptive equalization unit 44. The illustrated adaptive equalization unit 54 includes an FIR filter and a learning circuit, and NLMS (Normalized Least Mean Square) is used as a learning algorithm. Hereinafter, the adaptive equalization processing will be described with reference to FIG.

  The FIR filter shown in the figure has a delay line in which the number of taps is M and (M−1) delay elements (D) 51-1, 51-2,... Are connected in series (however, in FIG. In order to simplify the drawing, it is drawn as M = 5). Each of the delay elements 51-1, 51-2,... Has a delay time D corresponding to the sample period (one clock in Manchester code).

  Here, if the sampling time is n and the received signal at time n is u (n), time-series input data u (n), u (n−1),. n-M + 1) is obtained.

Further, multipliers 52-1 and 52-2 of the number of taps, ... will have the tap coefficients w ₁ in accordance with the characteristics of the filter (n), w ₂ (n), ..., w _M (n) of each The M pieces of input data u (n), u (n−1),..., U (n−M + 1) are respectively weighted and multiplied.

  Then, the accumulator 53 adds the time series input data weighted by the corresponding tap coefficients and performs an averaging process to obtain an equalized output signal r (n) at time n. The above equalization processing can be expressed as the following equation (1).

  Next, learning of tap coefficients will be described. A reference signal d (n) is input to the adder 54 together with the equalized output signal r (n), and an error signal e (n) is output by taking the difference between them. The reference signal d (n) corresponds to a pattern obtained by Manchester encoding “0xB24D” that the adaptive equalization unit 44 has in advance.

When the learning circuit 55 inputs time-series input data u (n), u (n−1),..., U (n−M + 1) at time n and the error signal e (n), the learning circuit 55 uses the NLMS algorithm. Then, tap coefficients w ₁ (n + 1), w ₂ (n + 1),..., W _M (n + 1) of the FIR filter at the next time are determined so that the equalized output signal r from the FIR filter approaches the reference signal d. Are supplied to the multipliers 52-1, 52-2,. The error signal e (n) and the tap coefficient update formula are expressed as the following formula (2).

By repeatedly executing the update equation shown in the above equation (2), each tap coefficient w ₁ (n), w ₂ (n),..., W _M (n) of the FIR filter is converted into an error signal e (n). Converge so that becomes smaller.

  Here, α in the above equation (2) is a step size, and 0 <α <2. When α is close to 1, it converges at high speed, but the variation in error increases. On the other hand, when α is close to 0, convergence is slow, but error variation is small.

  In the first half of the sync unit, the learning circuit 55 sets the step size α to a value close to 1 to perform high-speed learning while allowing error variation. In the subsequent second half of the sync unit, the variation in error is reduced by performing low-speed learning with the step size α set to a value close to zero. In this way, the learning circuit 55 realizes learning of adaptive equalization with high speed convergence and small convergence error as a whole.

  In the above description, the learning algorithm of the adaptive equalization unit 44 is a combination of an NLMS and an FIR filter, and is called NLMS-LE (Lenear Equalizer). However, the gist of the present invention is not limited to a specific learning algorithm. For example, other algorithms such as LMS (Least Mean Square) and RLS (Recursive Last Square) are used. Can be used.

  FIG. 6 illustrates an internal configuration example of the Manchester decoding unit 45.

  The delay element 61 has a delay time D corresponding to a sample period (one clock in Manchester code). The adder 62 calculates the difference between the received signal one clock before and the current received signal. The delay element 61 and the adder 62 operate as a differential filter that calculates the slope of the received signal and emphasizes the high frequency component.

  The 1/2 sampling unit 63 extracts from the head of the sync unit at intervals of once every two Manchester code clocks, and the subsequent binary determination unit 64 performs binary determination on the 1/2 sampling signal. The Manchester code is decoded into the NRZ code.

  A procedure for decoding a Manchester-encoded signal by the Manchester decoding unit 45 will be described with reference to FIG.

  For example, the data “B24D” has a bit sequence of “1011 0010 0100 1101” in the NRZ code (first stage from the top in FIG. 7).

  The Manchester code divides one bit section into a front cell and a rear cell at the center, and when the input 1 bit is a logical “0”, the front cell is low level and the rear cell is high level, That is, it is converted into 2 bits of “01”. Further, when the input 1 bit is a logical value “1”, the front cell is converted to a high level and the rear cell is converted to a low level, that is, “10”. Therefore, when the NRZ code including the bit sequence “1011 0010 0100 1101” of the data “B24D” is Manchester encoded, the bit sequence (MAN) “10011010010110010110010110100110” is converted. Note that the received waveform corresponding to the bit sequence (MAN) is a quantized value, but for the sake of simplicity of explanation, only the code is handled (second stage from the top in FIG. 7). The adaptive equalization unit 44 equalizes the disturbance of the signal received on the transmission path from the received waveform.

  The input bit sequence from the adaptive equalization unit 44 to the Manchester decoding unit 45 is “1 −1 −1 1 1 −1 1 −1 −1 1 −1 1 1 −1 −1 1 1 −1 1 1 −1 −1 1 1” Assuming that −1 1 1 −1 1 −1 −1 1 1 −1 ″, the delay element 61 gives a delay corresponding to one clock of the Manchester code to this bit sequence (MAN) (D) (FIG. 7 from the top of 7). Then, the adder 62 subtracts a bit sequence delayed by one clock from these input bit sequences. As a result, the output bit sequence (D-MAN) of the adder 52 is “−1 2 0 −2 0 2 −2 2 0 −2 2 −2 0 2 0 −2 0 −2 0 2 0 −2 2 −. 2 0 2 −2 2 0 −2 0 1 ″ (fourth stage from the top in FIG. 7).

  The ½ sampling unit 63 extracts the output of the adder 62 at an interval of once every two Manchester code clocks, so that the output bit sequence is “2-2-22-2-2-22-2-2−”. 2 2-2 -2 2 2 -2 2 "(fifth stage from the top in FIG. 7). Therefore, it can be seen that the binary bit sequence 1 0 1 1 0 0 1 0 0 1 0 0 1 1 0 1 ″ (the sixth stage from the top in FIG. 7), that is, the data “B24D” is decoded.

  As described above, according to the reception circuit configuration shown in FIG. 4, by providing the delay buffer 43, the Felica format is used as it is, frame synchronization using the sync unit, adaptation using the same sync unit, and the like. Both can be realized. Accordingly, compatibility problems and an increase in overhead can be avoided.

  However, there is a concern that the sync unit has a Manchester encoded bit number of 32 bits at most, which is insufficient as the number of learning times.

  For example, in the adaptive equalization unit 44 shown in FIG. 5, the learning circuit 55 sets the step size α to a value close to 1 in the first half of the sync unit, thereby allowing high-speed learning while allowing error variations. In the second half of the sync section, the fast step-by-step convergence and the adaptive equalization with a small convergence error are achieved by reducing the variation in error by performing low-speed learning with a value close to 0. A method for realizing this learning is also conceivable.

  In order to obtain a good learning result until the beginning of the data part arrives, the number of taps of the FIR filter is limited (if the number of taps increases, the time required for learning increases. The circuit scale is enlarged and the cost is increased.)

  On the other hand, when the number of taps of the FIR filter is small, it is difficult to express fine frequency characteristics. For this reason, depending on the influence of the frequency characteristic that the received signal has received on the transmission line, even if adaptive equalization is performed, the reception characteristic may not be sufficiently improved.

Manchester code is a coding method that doubles the band to remove the DC component from the signal, and the communication rate is halved (described above). For example, when the communication rate is 3.4 Mbps, the Manchester encoded channel rate is 6.8 Mbps, and the bandwidth of the baseband signal is 6.8 MHz. Further, the shortest wavelength of the signal encoded by Manchester is 1 pulse (1T _man ), the frequency is 3.4 MHz, and the longest wavelength is 2 pulses (2T _man ), and the frequency is 1.7 MHz. From this, it is assumed that the spectrum of a Manchester encoded signal having a communication rate of 3.4 Mbps is distributed around a band of 1.7 MHz to 3.4 MHz as shown in FIG. 8A. On the other hand, the shortest wavelength of an NRZ signal having a communication rate of 3.4 Mbps is equivalent to one pulse (1T _NRZ ) and has a frequency of 1.7 MHz, and the longest wavelength is a DC component and has a frequency of 0 MHz. In addition, the bandwidth is ½ of the Manchester code, and is assumed to be distributed around 0 MHz to 1.7.

  In the configuration of the receiving circuit illustrated in FIG. 4, the adaptive equalization unit 44 adaptively equalizes the Manchester code (output from the delay buffer 43). As described above, the spectrum of the signal encoded by Manchester is distributed around the band of 1.7 MHz to 3.4 MHz. On the other hand, when the number of taps of the FIR filter used in the adaptive equalization unit 44 is 5, as shown by a circle in FIG. 9, five frequency positions within the baseband bandwidth 6.8 MHz are expressed. Is possible.

Here, when the spectrum shown in FIG. 8A and the frequency position shown in FIG. 9 are superimposed, the band of 1.7 MHz to 3.4 MHz (that is, 1T _man and 2T _man ) corresponding to the center distribution of the spectrum is obtained. There is only one frequency position that can be controlled between them. For this reason, when the frequency characteristic of the transmission line is complicated between 1T _man and 2T _man , it is difficult for the FIR filter to express the inverse characteristic of the frequency characteristic, and good adaptive equalization is performed. Is difficult.

  If the number of taps of the FIR filter is increased, it becomes easier to express the inverse characteristics of the complex frequency characteristics in the band of 1.7 MHz to 3.4 MHz corresponding to the center distribution of the spectrum, and good adaptive equalization can be performed. become. However, as the number of taps increases, the time required for learning increases. Further, the circuit scale is enlarged according to the number of taps, resulting in an increase in cost.

  In the non-contact communication system using electromagnetic coupling, the adaptive equalization for improving more complicated frequency characteristics is achieved by using an FIR filter with a small number of taps in order to cope with an increase in communication rate. I think it is necessary.

  Next, FIG. 10 shows a configuration example of a receiving circuit that enables adaptive equalization that improves a more complicated frequency characteristic with an FIR filter having a smaller number of taps. The receiving circuit shown in the figure includes a preamble detection unit 101, a sync detection unit 102, a delay buffer 103, a delay element 104, an adder 105, a 1/2 sampling unit 106, an adaptive equalization unit 107, a binary value. A determination unit 108 is provided. Hereinafter, the operation of each unit when receiving a packet having the Felica format shown in FIG. 17 and performing adaptive equalization using the sync unit as a learning bit will be described.

  The received signal is first input to the preamble detector 101. When the preamble detection unit 101 detects a preamble portion that is a continuous waveform from the received signal, the preamble detection unit 101 extracts a sampling timing based on the continuous waveform and obtains timing synchronization.

  Next, the received signal is input to the sync detector 102. Based on the received signal and the sampling timing extracted by the preamble detection unit 101, the sync detection unit 42 detects a sync unit composed of a bit sequence in which “0xB24D” is Manchester-encoded to obtain frame synchronization. Generally, methods such as pattern matching and cross-correlation are used for detection of the sync part. Each of the detection methods seeks the identity with a known specific pattern of the sink part. Usually, in order to increase noise resistance, the pattern identity is confirmed using almost the entire sink portion. When the sync detection unit 102 detects the sync unit from the received signal, the sync detection unit 102 outputs a timing signal indicating the start position of the sync unit.

  On the other hand, the received signal is also input to the delay buffer 103. The delay buffer 103 gives a delay so that the head of the sync unit is not output until the sync detection unit 102 determines the detection of the sync unit.

  The delay element 104 gives a delay time D corresponding to the sample period (one clock in Manchester code) to the reception signal output from the delay buffer 103. The adder 105 calculates the difference between the received signal one clock before and the current received signal. The delay element 104 and the adder 105 function as a differential filter that calculates the slope of the received signal and emphasizes the high frequency component.

  The 1/2 sampling unit 106 extracts from the head of the sync unit at an interval of once every two Manchester code clocks.

  The adaptive equalization unit 107 is a learning-type equalization circuit including an FIR filter and a learning circuit. The adaptive equalization unit 107 starts from a timing signal indicating the start position of the sync unit output from the sync detection unit 102 and receives the received signal and the internal equalization circuit. Compare with sync pattern. Then, the tap coefficient of the FIR filter is adjusted so that the difference becomes small. Since the internal configuration of the adaptive equalization unit 107 may be the same as that shown in FIG. 5, for example, the description thereof is omitted here.

  The binary determination unit 108 decodes the Manchester code into the NRZ code by performing binary determination on the 1/2 sampling signal after adaptive equalization.

  A procedure for decoding the Manchester-encoded signal into the NRZ code in the subsequent processing from the delay buffer 103 will be described with reference to FIG.

  The output of the delay buffer 103 is, for example, a Manchester encoded NRZ code (first stage from the top in FIG. 11) consisting of data “B24D”, that is, the bit sequence “1011 0010 0100 1101”, and the bit sequence (MAN) “ 10011010010110010110010110100110 ″ (second stage from the top in FIG. 11). Although the received waveform is a quantized value, only the code is handled here for the sake of simplicity.

  The delay element 104 gives a delay corresponding to one clock of the Manchester code to the bit sequence (MAN) (D) (third stage from the top in FIG. 11). The adder 105 subtracts a bit sequence delayed by one clock from these input bit sequences. As a result, the output bit sequence (D-MAN) of the adder 52 is “−1 2 0 −2 0 2 −2 2 0 −2 2 −2 0 2 0 −2 0 −2 0 2 0 −2 2 −. 2 0 2 −2 2 0 −2 0 2 ″ (fourth stage from the top in FIG. 11).

  The 1/2 sampling unit 106 extracts the output of the adder 105 at an interval of once every two Manchester code clocks. Therefore, the output bit sequence is “2-2-2 2-2-2-2 2-2-2-22-2-2 2 2-2 2” (fifth stage from the top in FIG. 11).

  The adaptive equalization unit 107 equalizes the disturbance of the signal received on the transmission path from the output waveform of the ½ sampling unit 106. Here, the reception circuit shown in FIG. 4 is compared with adaptive equalization.

  In the configuration of the receiving circuit illustrated in FIG. 4, the adaptive equalization unit 44 adaptively equalizes the Manchester code (output from the delay buffer 43). In this case, if the number of taps of the FIR filter is 5, there is only one frequency position that can be controlled in the band of 1.7 MHz to 3.4 MHz corresponding to the center distribution of the spectrum. It is difficult to express and there is a concern that good adaptive equalization cannot be performed (described above).

On the other hand, in the configuration of the receiving circuit illustrated in FIG. 10, the adaptive equalization unit 107 equalizes the output waveform of the ½ sampling unit 106. As with the NRZ code, the output waveform of the ½ sampling unit 106 is one pulse (1T _NRZ ), the shortest wavelength is 3.4 MHz, and the longest wavelength is a DC component, as shown in FIG. 8B. In addition, the bandwidth is ½ of the Manchester code and is distributed around 0 MHz. In this case, even if the number of taps of the FIR filter used in the adaptive equalization unit 107 is also five, if five frequency positions within a bandwidth of 3.4 MHz are expressed as indicated by ◯ in FIG. Often, there are three controllable frequency positions within the 0 MHz to 1.7 MHz band corresponding to the center distribution of the spectrum. Therefore, even if the frequency characteristic of the transmission path is complicated by the center distribution of the spectrum, the FIR filter can more accurately express the inverse characteristic of the frequency characteristic, and good adaptive equalization can be performed. become able to.

  When the adaptive equalization unit 107 can equalize the output waveform of the 1/2 sampling unit 106 more accurately, the determination accuracy of the binary determination unit 108 in the subsequent stage is also improved, and the binary bit sequence 1 0 1 1 0 0 1 0 0 1 0 0 1 1 0 1 ″ (sixth stage from the top in FIG. 11), that is, data “B24D” can be obtained.

  FIG. 13 shows the equalized signal point distribution when the Manchester encoded signal output from the delay buffer 43 is directly adaptively equalized in the receiving circuit shown in FIG. FIG. 14 shows an equalized signal point distribution when the Manchester encoded signal is ½ sampled and then adaptively equalized in the receiving circuit shown in FIG. In the result shown in FIG. 13, since many signal points are distributed in the vicinity of the threshold value for binary determination, it is easy to make an erroneous determination when performing binary determination in the Manchester decoding unit 45, and the SNR becomes low. On the other hand, in the result shown in FIG. 14, the number of signal points arranged in the vicinity of the threshold for binary determination is reduced, so that binary determination can be performed without error after adaptive equalization. That is, it can be said that the received signal can be equalized well.

  As described above, in the receiving circuit configuration shown in FIG. 10, by reducing the bandwidth to 1/2 of the Manchester code by sampling 1/2 the Manchester encoded signal, the number of taps (or Even with the same number of taps as in the prior art, the inverse characteristic of the complex frequency characteristic in the vicinity of the center distribution of the spectrum can be suitably expressed, and good adaptive equalization can be realized. Further, since 1/2 sampling is performed before adaptive equalization, the circuit constituting the adaptive equalization unit only needs to operate at 1/2 speed, which leads to reduction in power consumption. On the other hand, with 1/2 sampling before adaptive equalization, the number of bits of the sync unit used for learning tap coefficients by adaptive equalization is also halved. This affects the convergence performance of the tap coefficient. If the tap coefficient does not converge to an appropriate value by the beginning of the data part, a good equalization result cannot be obtained and the reception performance deteriorates.

  FIG. 15 shows a configuration example of a receiving circuit in which the problem of a decrease in the number of bits of the sync unit used for learning due to 1/2 sampling is further solved. The illustrated receiving circuit includes a preamble detection unit 151, a sync detection unit 152, a RAM control unit 153, a RAM 159, a delay element 154, an adder 155, a 1/2 sampling unit 156, and an adaptive equalization unit 157. And a binary determination unit 158. The delay buffer 103 is different from the configuration example shown in FIG. 10 in that the RAM control unit 153 and the RAM 159 are replaced. Hereinafter, the operation of each unit when receiving a packet having the Felica format shown in FIG. 17 and performing adaptive equalization using the sync unit as a learning bit will be described.

  The received signal is first input to the preamble detector 151. When the preamble detection unit 151 detects a preamble portion that is a continuous waveform from the received signal, the preamble detection unit 151 extracts a sampling timing based on the continuous waveform and obtains timing synchronization.

  Next, the received signal is input to the sync detector 152. Based on the received signal and the sampling timing extracted by the preamble detection unit 151, the sync detection unit 152 detects a sync unit composed of a bit sequence in which “0xB24D” is Manchester-encoded to obtain frame synchronization.

  The RAM control unit 153 writes the reception signal input from the sync detection unit 152 into the RAM 159. In addition, the RAM control unit 153 reads the received signal from the RAM 159 and outputs it to the subsequent circuit.

  The RAM control unit 153 stores the address at which the head data of the sync unit is stored, starting from the detection timing of the sync unit determined by the sync detection unit 152. When the final data of the sync unit is output, the read address is set again as the storage address of the head data of the sync unit, and the sync unit is read. When reading of the predetermined number of repetitions of the sync part is completed, the data part is read. FIG. 16 shows a data string output from the RAM control unit 903 to the subsequent circuit when the number of repetitions of the sync unit is 2.

  The delay element 154 gives a delay time D corresponding to the sample period (one clock in Manchester code) to the reception signal output from the delay buffer 153. The adder 155 calculates the difference between the received signal one clock before and the current received signal. The delay element 154 and the adder 105 act as a differential filter that emphasizes the high-frequency component of the received signal. Then, the 1/2 sampling unit 156 extracts from the head of the sync unit at an interval of once every two Manchester code clocks.

  The adaptive equalization unit 157 is a learning type equalization circuit composed of an FIR filter and a learning circuit. The adaptive equalization unit 157 compares the received signal with an internal sync pattern, and reduces the difference between the FIR filter and the internal sync pattern. Adjust the tap factor. The internal configuration of the adaptive equalization unit 157 may be the same as that shown in FIG. 5, for example, and will not be described here. In the illustrated example, the adaptive equalization unit 157 uses the timing signal indicating the start position of the sync unit output from the sync detection unit 152 as a starting point, and tap coefficients corresponding to the number of repetitions of the sync unit determined in advance (described above). Since learning is performed, the convergence performance of the tap coefficient can be maintained.

  The binary determination unit 108 decodes the Manchester code into the NRZ code by performing binary determination on the 1/2 sampling signal after adaptive equalization.

  The procedure for decoding the Manchester-encoded signal into the NRZ code is as shown in FIG. 11, and a description thereof is omitted here.

  Along with performing 1/2 sampling before adaptive equalization, the number of bits of the sync part used for learning tap coefficients by adaptive equalization is also halved. On the other hand, the receiving circuit shown in FIG. 15 is configured to store the sync unit, read out the predetermined number of repetitions of the sync unit, and learn tap coefficients corresponding to the number of repetitions. The convergence performance of the tap coefficient can be maintained despite the halving of the number of bits of the sync part due to 1/2 sampling.

  The present invention has been described in detail above with reference to specific embodiments. However, it is obvious that those skilled in the art can make modifications and substitutions of the embodiment without departing from the gist of the present invention.

  In the present specification, the embodiment in which the present invention is applied to a contactless communication system according to the NFC IP-1 standard has been mainly described, but the gist of the present invention is not limited to this. The present invention can be similarly applied to a communication system compliant with various standards that performs communication using modulation by switching the change direction of the electrical load.

  In short, the present invention has been disclosed in the form of exemplification, and the description of the present specification should not be interpreted in a limited manner. In order to determine the gist of the present invention, the claims should be taken into consideration.

DESCRIPTION OF SYMBOLS 10 ... Reader writer 11 ... Control part 12 ... Transmission circuit 13 ... Reception circuit 14 ... Antenna resonance circuit part 30 ... Non-contact IC card 31 ... Control part 32 ... Antenna resonance circuit part 33 ... Load switching modulation circuit part 34 ... Control part 41 ... preamble detection unit 42 ... sync detection unit 43 ... delay buffer 44 ... adaptive equalization unit 45 ... Manchester decoding unit 41 ... preamble detection unit 42 ... sync detection unit 43 ... delay buffer 44 ... adaptive equalization unit 45 ... Manchester decoding unit 51 ... delay element 52 ... multiplier 53 ... accumulator 54 ... adder 55 ... learning circuit 61 ... delay element 62 ... adder 63 ... 1/2 sampling unit 64 ... binary determination unit 101 ... preamble detection unit 102 ... sink detection unit 103 ... Delay buffer 104 ... Adaptive equalization unit 105 ... Adder 106 ... 1/2 sump Ring unit 107 ... Adaptive equalization unit 108 ... Binary determination unit 151 ... Preamble detection unit 152 ... Sink detection unit 153 ... RAM control unit 154 ... Adaptive equalization unit 155 ... Adder 156 ... 1/2 sampling unit 157 ... Adaptation etc. 158: binary determination unit 159 ... RAM

Claims (6)

An inclination calculation step for calculating an inclination of the received signal;
A sampling step of ½ sampling the received signal whose slope has been calculated;
An adaptive equalization step for adaptively equalizing the received signal after the ½ sampling,
A receiving method comprising: A reception method for receiving a packet with a learning bit encoded by Manchester,
A detection step of detecting a start position of the learning bit;
A delay time giving step for giving a delay to the reception signal so that the head of the learning bit is not output until the detection of the learning bit is confirmed in the detection step;
A slope calculation step for calculating a difference between the received signal given the delay time and a received signal further given a delay time equivalent to one clock in Manchester code;
A 1/2 sampling step of extracting from the head of the learning bit that has been subjected to the slope calculation at intervals of once every two Manchester code clocks;
An adaptive equalization step for performing adaptive equalization using the learning bits sampled in half.
A binary determination step of performing binary determination on the adaptively equalized reception signal and decoding the Manchester code into an NRZ code;
A receiving method comprising: In the delay time giving step, the received learning bit is stored and read out a predetermined number of repetitions starting from the timing at which the learning bit is detected,
In the adaptive equalization step, the received learning bit is learned by the number of repetitions.
The receiving method according to claim 2, wherein: An inclination calculation unit for calculating an inclination of the received signal;
A sampling unit that ½ samples the received signal subjected to the inclination calculation;
An adaptive equalization unit that adaptively equalizes the received signal after the ½ sampling,
A receiving apparatus comprising: In a non-contact communication system using electromagnetic coupling, a non-contact IC card that receives a Manchester encoded packet composed of a preamble part consisting of a continuous waveform, a sync part consisting of a specific pattern, and a data part,
A preamble detector that detects a preamble portion from the received signal and extracts a sampling timing based on the continuous waveform; and
A sync detector that detects a sync portion from a received signal based on the sampling timing and outputs a timing signal indicating a start position of the sync portion;
A delay time giving unit that gives a delay to the reception signal so that the head of the sync unit is not output until the detection of the sync unit is confirmed in the sync detection step;
A slope calculator that calculates a difference between the received signal given the delay time and a received signal further given a delay time equivalent to one clock in Manchester code;
A 1/2 sampling unit that extracts from the head of the sync unit for which the inclination is calculated at an interval of once every two Manchester code clocks;
An adaptive equalization unit that performs adaptive equalization using a ½ sampled sync unit based on the timing signal;
A binary determination unit for performing binary determination on the adaptively equalized reception signal and decoding the Manchester code into an NRZ code;
A non-contact IC card comprising: In a non-contact communication system using electromagnetic coupling, a reader / writer that receives a Manchester encoded packet composed of a preamble part consisting of a continuous waveform, a sync part consisting of a specific pattern, and a data part from a non-contact IC card. ,
A preamble detector that detects a preamble portion from the received signal and extracts a sampling timing based on the continuous waveform; and
A sync detector that detects a sync portion from a received signal based on the sampling timing and outputs a timing signal indicating a start position of the sync portion;
A delay time giving unit that gives a delay to the reception signal so that the head of the sync unit is not output until the detection of the sync unit is confirmed in the sync detection step;
A slope calculator that calculates a difference between the received signal given the delay time and a received signal further given a delay time equivalent to one clock in Manchester code;
A 1/2 sampling unit that extracts from the head of the sync unit for which the inclination is calculated at an interval of once every two Manchester code clocks;
An adaptive equalization unit that performs adaptive equalization using a ½ sampled sync unit based on the timing signal;
A binary determination unit for performing binary determination on the adaptively equalized reception signal and decoding the Manchester code into an NRZ code;
A reader / writer characterized by comprising:

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