JP2011101516A - Multi-output switching power supply apparatus - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a multi-output switching power supply apparatus for stabilizing a plurality of inexpensive output voltages without using a step-up chopper circuit. <P>SOLUTION: The multi-output switching power supply apparatus includes a series circuit of a first switching element Q1 and a second switching element Q2, which are connected to output ends of a DC power supply 2, a control circuit 10a which alternately turns on/off the first switching element and the second switching element and converts DC voltage of the DC power supply into an AC voltage, a series circuit of a primary winding P1 of a reactor Lr and a transformer T1 and a current resonance capacitor Cri, which is connected to both ends of the first switching element or the second switching element, first rectifying/smoothing circuits D1 and C1 rectifying/smoothing a voltage generated in a first secondary winding S1 of the transformer and for output of a first output voltage Vo1, and second rectifying/smoothing circuits D2 and C2 rectifying/smoothing voltage generated in a second secondary winding S2 of the transformer and for output of second output voltage Vo2. The control circuit controls an on-period of the first switching element based on first output voltage and controls an on-period of the second switching element based on second output voltage. <P>COPYRIGHT: (C)2011,JPO&INPIT

Description

  The present invention relates to a multi-output switching power supply device having a plurality of outputs.

  FIG. 1 is a circuit diagram showing a conventional multi-output switching power supply device. In this multi-output switching power supply device, the control circuit 10 controls the duty of the switching elements Q1 and Q2 based on the feedback signal from the output voltage Vo1, and the booster chopper is controlled by the control circuit 11a based on the feedback signal from the output voltage Vo2. The output voltage Vo2 is controlled by controlling the output voltage of the circuit.

  The control circuit 10 alternately turns on / off the switching element Q1 and the switching element Q2 based on the detection signal of the output voltage Vo1 from the feedback circuit 5. That is, the control circuit 10 changes the charging voltage of the current resonance capacitor Cri by performing PWM control or frequency control of the switching element Q1 or the switching element Q2 based on the detection signal of the output voltage Vo1 from the feedback circuit 5, thereby switching the switching element Q2 The amount of energy sent to the output voltage Vo1 is controlled during the ON period. Here, the control circuit 10 controls the ON width of the switching element Q1 based on the output voltage Vo1 from the feedback circuit 5.

  The feedback circuit 6 is connected to a connection point between the smoothing capacitor C2 and the diode D2, detects the output voltage Vo2 of the smoothing capacitor C2, and outputs the detected voltage to the control circuit 11a. The control circuit 11a controls the output voltage Vo2 by controlling the voltage across the smoothing capacitor C3 in the boost chopper circuit based on the detected voltage from the feedback circuit 6.

  The control circuit 11a multiplies the detected voltage from the feedback circuit 6 by the divided voltage obtained by dividing the full-wave rectified voltage from the full-wave rectifier circuit 2 by the resistors R11 and R12, An error voltage signal is generated by amplifying an error voltage from a voltage proportional to the input current detected by the current detection resistor R13. The control circuit 11a generates a pulse signal by comparing the generated error voltage signal and the triangular wave signal, and applies the pulse signal to the switching element Q3.

  As described above, according to the conventional multi-output switching power supply apparatus, the switching elements Q1 and Q2 are switched by the feedback signals of the two outputs taken out from the two secondary windings S1 and S2 wound with opposite polarities. By controlling the duty and the output voltage of the boost chopper circuit in the previous stage, it is possible to stabilize a plurality of outputs even if one output voltage or both output voltages are variable.

JP 2007-28751 A

  However, in the conventional multi-output switching power supply apparatus, in order to control the output voltage Vo1 and the output voltage Vo2, the control circuit 10 of the DC-DC converter and the control circuit 11a of the step-up chopper circuit are required. For this reason, the multi-output switching power supply device is expensive.

  The step-up chopper circuit has a power factor improving function. In order to improve this power factor, the response to the output voltage of the step-up chopper circuit, that is, the phase compensation must be sufficiently slow with respect to the AC power supply frequency of the AC power supply 1.

  On the other hand, the output voltage Vo2 is stabilized by advancing the response to the output voltage fluctuation of the boost chopper circuit in conjunction with the voltage fluctuation of the output voltage Vo2. However, accelerating the response of the boost chopper circuit to fluctuations in output voltage has a problem of deteriorating the power factor.

  An object of the present invention is to provide a multi-output switching power supply apparatus that can stabilize a plurality of output voltages at low cost without using a step-up chopper circuit.

  In order to solve the above problems, the present invention provides a DC power supply, a first series circuit connected to both ends of the output of the DC power supply, and a first switching element and a second switching element connected in series; A control circuit for converting a DC voltage of the DC power source into an AC voltage by alternately turning on and off one switching element and the second switching element, a primary winding, a first secondary winding, and a second A transformer having a secondary winding and a second series connected to both ends of the first switching element or the second switching element, and a reactor, the primary winding of the transformer, and a current resonance capacitor connected in series A circuit, a first rectifying / smoothing circuit that rectifies and smoothes a voltage generated in the first secondary winding of the transformer and outputs a first output voltage, and is generated in the second secondary winding of the transformer And a second rectifying / smoothing circuit that rectifies and smoothes the output voltage and outputs a second output voltage, and the control circuit turns on the first switching element based on the first output voltage from the first rectifying and smoothing circuit. A period is controlled, and an ON period of the second switching element is controlled based on the second output voltage from the second rectifying and smoothing circuit.

  According to the present invention, the control circuit controls the ON period of the first switching element based on the first output voltage from the first rectifying and smoothing circuit, and performs the second switching based on the second output voltage from the second rectifying and smoothing circuit. Since the ON period of the element is controlled, a plurality of output voltages can be stabilized at low cost without using a boost chopper circuit.

It is a circuit block diagram which shows the conventional multiple output switching power supply device. It is a circuit block diagram which shows the multiple output switching power supply apparatus of Example 1 of this invention. It is a circuit block diagram which shows the control circuit in the multiple output switching power supply device of Example 1 of this invention. It is a timing chart which shows operation | movement of each part when raising the setting voltage for 1st output voltage in the multiple output switching power supply device of Example 1 of this invention. It is a timing chart which shows operation | movement of each part when raising the setting voltage for 2nd output voltage in the multiple output switching power supply device of Example 1 of this invention. 4 is a timing chart showing how the ON period of the switching element Q2 is controlled so that the bottom voltage of the amplitude voltage of the current resonance capacitor is lowered when the second output voltage is raised in the multi-output switching power supply device of Example 1 of the present invention. is there. 6 is a timing chart showing how the ON period of the switching element Q1 is controlled so that the amplitude voltage of the current resonance capacitor is increased when the first output voltage is raised in the multi-output switching power supply device according to the first embodiment of the present invention. It is a circuit block diagram which shows the multiple output switching power supply device of Example 2 of this invention. It is a circuit block diagram which shows the multiple output switching power supply device of Example 3 of this invention. It is a circuit block diagram which shows the multiple output switching power supply device of the modification of Example 3 of this invention.

  Hereinafter, embodiments of the multi-output switching power supply apparatus of the present invention will be described in detail with reference to the drawings.

  FIG. 2 is a circuit configuration diagram illustrating the multi-output switching power supply device according to the first embodiment. In the switching power supply device according to the first embodiment, the control circuit 10a changes the ON period of the switching element Q1 and the ON period of the switching element Q2, thereby changing the resonance voltage amplitude and the DC superimposed voltage of the current resonance capacitor Cri. The amount of energy sent from the primary side to the secondary side is controlled.

  Specifically, except for the control circuit 10a, the present invention is not significantly changed with respect to the configuration of the conventional switching power supply device, but the present invention has a capacity of the current resonance capacitor Cri that is sufficiently larger than the conventional capacity. The difference is that the resonance voltage amplitude and the DC superposition voltage of the current resonance capacitor Cri can be easily changed by making it smaller.

  In the multi-output switching power supply device of the first embodiment, the AC power supply 1 outputs an AC voltage to the rectifier 2. The rectifier 2 rectifies the AC voltage of the AC power source 1. The AC power source 1 and the rectifier 2 constitute a DC power source. Instead of using the AC power source 1 and the rectifier 2, a DC power source may be used.

  A smoothing capacitor C3 is connected to both ends of the output of the rectifier 2, and a series circuit of a switching element Q1 (first switching element) and a switching element Q2 (second switching element) is connected. The switching element Q1 and the switching element Q2 are made of, for example, MOSFETs. The drain of the switching element Q1 is connected to the positive electrode of the rectifier 2, and the source of the switching element Q2 is connected to the negative electrode of the rectifier 2.

  A voltage resonant capacitor Crv is connected between the drain and source of the switching element Q2, and a series circuit of the reactor Lr, the primary winding P1 of the transformer T1, and the current resonant capacitor Cri is connected.

  The transformer T1 has a primary winding P1 (the number of turns is N1), a first secondary winding S1 (the number of turns is N2), and a second secondary winding S2 (the number of turns is N3).

  A series circuit of a diode D1 and a smoothing capacitor C1 is connected to both ends of the first secondary winding S1 of the transformer T1. The diode D1 and the smoothing capacitor C1 constitute a first rectifying / smoothing circuit, rectifying and smoothing a voltage generated at both ends of the first secondary winding S1 of the transformer T1, and generating an output voltage Vo1 (first output voltage). Output.

  A series circuit of a diode D2 and a smoothing capacitor C2 is connected to both ends of the second secondary winding S2 of the transformer T1. The diode D2 and the smoothing capacitor C2 constitute a second rectifying / smoothing circuit, rectifying and smoothing the voltage generated at both ends of the second secondary winding S2 of the transformer T1, and generating an output voltage Vo2 (second output voltage). Output.

  In the first secondary winding S1 and the second secondary winding S2 of the transformer T1, the energy on the primary side of the transformer T1 is transmitted to the second secondary winding S2 and switched when the switching element Q1 is turned on. It is wound so that the primary energy is transmitted to the first secondary winding S1 when the element Q2 is turned on.

  That is, in the example shown in FIG. 2, the first secondary winding S1 is wound in the opposite phase to the primary winding P1 of the transformer T1, and the second secondary winding is wound around the primary winding P1 of the transformer T1. Winding S2 is wound in the same phase.

  The feedback circuit 5 outputs an error voltage between the output voltage Vo1 that is the voltage across the smoothing capacitor C1 and the first set voltage to the control circuit 10a as the first feedback signal FBL. The feedback circuit 6 outputs an error voltage between the output voltage Vo2 that is the voltage across the smoothing capacitor C2 and the second set voltage to the control circuit 10a as the second feedback signal FBH.

  The control circuit 10a rectifies the rectifier 2 by alternately turning on / off the switching element Q1 and the switching element Q2 based on the first feedback signal FBL from the feedback circuit 5 and the second feedback signal FBH from the feedback circuit 6. Convert voltage to pulse signal. Further, the control circuit 10a controls the on period of the switching element Q1 based on the first feedback signal FBL from the feedback circuit 5, and controls the on period of the switching element Q2 based on the second feedback signal FBH from the feedback circuit 6. .

  Furthermore, when the output voltage Vo1 is increased, the control circuit 10a extends the ON period of the switching element Q1 so that the upper limit voltage of the current resonance capacitor Cri increases. When increasing the output voltage Vo2, the control circuit 10a lengthens the ON period of the switching element Q2 so that the lower limit voltage of the current resonance capacitor Cri decreases.

  FIG. 3 is a circuit configuration diagram showing a control circuit in the multi-output switching power supply apparatus according to Embodiment 1 of the present invention. In FIG. 3, one end of the resistor R1 inputs the second feedback signal FBH from the feedback circuit 6, and one end of the resistor R2 inputs the first feedback signal FBL from the feedback circuit 5. The other end of the resistor R1 is connected to one end of the resistor R3, and the other end of the resistor R2 is connected to one end of the resistor R4.

  The current mirror circuit 101 includes a transistor Q4 and a transistor Q5. The base and collector of the transistor Q4 and the base of the transistor Q5 are connected to the other end of the resistor R1 and one end of the resistor R3. The emitter of the transistor Q4 and the emitter of the transistor Q5 are connected to the positive electrode of the power supply Vref. The collector of the transistor Q5 is connected to the non-inverting terminal of the comparator CP2 (second comparing means), one end of the capacitor C4 (second capacitor), and the drain of the MOSFET Q8.

  The current mirror circuit 102 includes a transistor Q6 and a transistor Q7. The base and collector of the transistor Q6 and the base of the transistor Q7 are connected to the other end of the resistor R2 and one end of the resistor R4. The emitter of the transistor Q6 and the emitter of the transistor Q7 are connected to the positive electrode of the power supply Vref. The collector of the transistor Q7 is connected to the non-inverting terminal of the comparator CP1 (first comparing means), one end of the capacitor C5 (first capacitor), and the drain of the MOSFET Q9.

  The negative electrode of the power supply Vref, the other end of the resistor R3, the other end of the resistor R4, the other end of the capacitor C4, the other end of the capacitor C5, the source of the MOSFET Q8, and the source of the MOSFET Q9 are grounded. The inverting terminal of the comparator CP1 and the inverting terminal of the comparator CP2 are connected to the positive electrode of the reference power supply Vth, and the negative electrode of the reference power supply Vth is grounded.

  The comparator CP2 compares the charging voltage charged in the capacitor C4 with the reference voltage of the reference power source Vth, and outputs an L level to the set terminal S of the flip-flop circuit FF1 when the charging voltage is less than the reference voltage. When is equal to or higher than the reference voltage, the first pulse signal is generated by outputting the H level to the set terminal S of the flip-flop circuit FF1.

  The comparator CP1 compares the charging voltage charged in the capacitor C5 with the reference voltage of the reference power supply Vth, and outputs an L level to the reset terminal R of the flip-flop circuit FF1 when the charging voltage is less than the reference voltage. When is equal to or higher than the reference voltage, the second pulse signal is generated by outputting the H level to the reset terminal R of the flip-flop circuit FF1.

  The flip-flop circuit FF1 outputs the first pulse signal input from the comparator CP2 to the set terminal S from the output terminal Q to the delay circuit 103 and the gate of the MOSFET Q8. The flip-flop circuit FF1 outputs the second pulse signal input from the comparator CP1 to the reset terminal R to the delay circuit 104 and the gate of the MOSFET Q9 from the inverted output terminal Qb.

The MOSFET Q8 (second discharge means) discharges the electric charge of the capacitor C4 when an H level first pulse signal is input from the output terminal Q of the flip-flop circuit FF1. The MOSFET Q9 (first discharging means) discharges the electric charge of the capacitor C5 when the H level second pulse signal is input from the inverting output terminal Qb of the flip-flop circuit FF1.
The delay circuit 103 delays the first pulse signal from the output terminal Q of the flip-flop circuit FF1 for a first predetermined time, and outputs the first delay pulse signal Hout to the gate of the switching element Q1 via the driver DR1.

  In the delay circuit 103, the input terminal of the buffer circuit BF1 is connected to the output terminal Q of the flip-flop circuit FF1, and the cathode of the diode D4 and one end of the resistor R5 are connected to the output terminal of the buffer circuit BF1. The anode of the diode D4 and the other end of the resistor R5 are connected to one end of the capacitor C6 and one end of the buffer circuit BF2, and the other end of the buffer circuit BF2 is connected to the input terminal of the driver DR1.

  The delay circuit 104 delays the second pulse signal from the inverting output terminal Qb of the flip-flop circuit FF1 for a second predetermined time, and outputs the second delay pulse signal Lout to the gate of the switching element Q2 via the driver DR2.

  In the delay circuit 104, the input terminal of the buffer circuit BF3 is connected to the inverting output terminal Qb of the flip-flop circuit FF1, and the cathode of the diode D5 and one end of the resistor R6 are connected to the output terminal of the buffer circuit BF3. The anode of the diode D5 and the other end of the resistor R6 are connected to one end of the capacitor C7 and one end of the buffer circuit BF4, and the other end of the buffer circuit BF4 is connected to the input terminal of the driver DR2.

The time constants R5 and C6 of the delay circuit 103 and the time constants R6 and C7 of the delay circuit 104 are determined so that the switching elements Q1 and Q2 are not turned on simultaneously.
Next, the operation of the multi-output switching power supply device of the first embodiment configured as described above will be described with reference to FIG.

  First, during the ON period of the switching element Q1, energy is stored in the current resonance capacitor Cri via the exciting inductance of the primary winding P1 of the transformer T1 and the reactor Lr (leakage inductance between the primary and secondary of the transformer T1). In addition, a resonance current by the reactor Lr and the current resonance capacitor Cri is sent to the output voltage Vo2.

  Further, during the ON period of the switching element Q1, the voltage obtained by subtracting the voltage of the current resonant capacitor Cri from the output voltage of the rectifier 2 is divided into the primary winding P1 by the inductance ratio of the exciting inductance of the primary winding P1 and the reactor Lr. A voltage corresponding to the excitation inductance ratio of the pressed primary winding P1 is applied.

  Then, the voltage of the primary winding P1 is clamped when it becomes (Vo2 + Vf) × N1 / N3, and the resonance current by the current resonance capacitor Cri and the reactor Lr is sent to the secondary side. Here, Vf is a forward voltage of the diode D2. When the voltage of the primary winding P1 is less than (Vo2 + Vf) × N1 / N3, energy to the secondary side of the transformer T1 is not transmitted, and the resonance operation is performed only on the primary side of the transformer T1.

  Next, during the ON period of the switching element Q2, the resonance current generated by the reactor Lr and the current resonance capacitor Cri is sent to the output voltage Vo1 via the secondary winding S1 of the transformer T1 by the energy stored in the current resonance capacitor Cri. Further, the excitation energy of the excitation inductance of the primary winding P1 is reset.

  Further, during the ON period of the switching element Q2, the primary winding P1 has an exciting inductance of the primary winding P1 obtained by dividing the voltage of the current resonance capacitor Cri by an inductance ratio of the exciting inductance of the primary winding P1 and the reactor Lr. A specific voltage is applied.

  And it clamps when the voltage of the primary winding P1 becomes (Vo1 + Vf) × N1 / N2, and the resonance current by the current resonance capacitor Cri and the reactor Lr is sent to the secondary side via the secondary winding S1. . Here, Vf is the forward voltage of the diode D1.

  When the voltage of the primary winding P1 is less than (Vo1 + Vf) × N1 / N2, no energy is transmitted to the secondary side of the transformer T1, and only the primary side of the transformer T1 performs a resonance operation.

  Next, the operation of the control circuit 10a, which is a feature of the multi-output switching power supply apparatus according to the first embodiment, will be described with reference to FIGS.

  First, the operation of each part when raising the set voltage for the output voltage Vo1 will be described with reference to the timing chart of FIG.

  In FIG. 4, Vr1 is a set voltage for the output voltage Vo1, Vr2 is a set voltage for the output voltage Vo2, IFBL is a current flowing in the current mirror circuit 102, IFBH is a current flowing in the current mirror circuit 101, and VC4 is The voltage across the capacitor C4, VC5 the voltage across the capacitor C5, Q the first pulse signal from the output terminal Q of the flip-flop circuit FF1, Qb the first pulse signal from the inverting output terminal Qb of the flip-flop circuit FF1, Lout Represents a second delayed pulse signal from the driver DR2, and Hout represents a first delayed pulse signal from the driver DR1.

  The set voltage Vr1 is a reference voltage in the feedback circuit 5, and the set voltage Vr2 is a reference voltage in the feedback circuit 6. These reference voltages are variable.

  First, at time t0, when an H level first pulse signal is output from the output terminal Q of the flip-flop circuit FF1 to the delay circuit 103, the delay circuit 103 delays the first pulse signal to delay the first delay pulse signal Hout. Is output to the switching element Q1. For this reason, the switching element Q1 is turned on. The MOSFET Q8 is turned on by the first pulse signal at the H level, the charge of the capacitor C4 is discharged, and the voltage VC4 becomes zero.

  When the set voltage Vr1 is increased by a constant value to increase the output voltage Vo1 at time t1, the current IFBL flowing through the current mirror circuit 102 decreases by a constant value.

  When the voltage VC5 reaches the reference voltage Vth at time t2, the second pulse signal from the inverting output terminal Qb of the flip-flop circuit FF1 becomes H level, and the MOSFET Q9 is turned on by the H level second pulse signal, so that the capacitor C5 Are discharged, and the voltage VC5 becomes zero.

  Further, the second pulse signal from the inverting output terminal Qb of the flip-flop circuit FF1 becomes H level. Since the delay circuit 104 delays the second pulse signal and outputs the second delayed pulse signal Lout to the switching element Q2, the switching element Q2 is turned on. Here, the MOSFET Q8 is turned off by the L level first pulse signal, and the voltage VC4 of the capacitor C4 rises linearly.

  When the voltage VC4 reaches the reference voltage Vth at time t3, the first pulse signal from the output terminal Q of the flip-flop circuit FF1 becomes H level, and the MOSFET Q8 is turned on by the H level first pulse signal. The electric charge is discharged, and the voltage VC4 becomes zero.

  Further, the second pulse signal from the inverting output terminal Qb of the flip-flop circuit FF1 becomes L level. For this reason, the MOSFET Q9 is turned off, and the voltage VC5 of the capacitor C5 rises linearly.

  At this time, since the current IFBL flowing through the current mirror circuit 102 decreases, the current flowing through the capacitor C5 also decreases. For this reason, the time for the voltage VC5 of the capacitor C5 to reach the reference voltage Vth from zero is (time t4−time t3), which is longer than (time t2−time t0). For this reason, since the 1st delay pulse signal Hout also becomes long, the ON period of the switching element Q1 becomes long.

  FIG. 5 is a timing chart showing the operation of each unit when the set voltage for the output voltage Vo2 is raised. The operation of each part when raising the set voltage for the output voltage Vo2 shown in FIG. 5 is the same as the operation of each part when raising the set voltage for the output voltage Vo1 shown in FIG. Only the operation will be described.

  When the set voltage Vr2 is increased by a certain value to increase the output voltage Vo2 at time t11, the current IFBH flowing through the current mirror circuit 101 is decreased by a certain value.

  When the voltage VC5 reaches the reference voltage Vth at time t12, the second pulse signal from the inverting output terminal Qb of the flip-flop circuit FF1 becomes H level, the MOSFET Q9 is turned on, and the voltage VC5 of the capacitor C5 becomes zero. Further, the first pulse signal from the output terminal Q of the flip-flop circuit FF1 becomes L level. For this reason, the voltage VC4 of the capacitor C4 increases from zero.

  At this time, since the current IFBH flowing through the current mirror circuit 101 has decreased, the current flowing through the capacitor C4 also decreases. For this reason, the time for the voltage VC4 of the capacitor C4 to reach the reference voltage Vth from zero is (time t13-time t12), which is longer than (time t10-time t9). For this reason, since the second delay pulse signal Lout is also lengthened, the ON period of the switching element Q2 is lengthened.

  Next, referring to the timing charts of FIGS. 6A to 6B, the ON period of the switching element Q2 is set so that the bottom voltage of the amplitude voltage of the current resonance capacitor Cri is lowered when the output voltage Vo2 is increased. A method for controlling the above will be described. 6A shows the operation of each part when the output voltage Vo2 is not increased, and FIG. 6B shows the operation of each part when the output voltage Vo2 is increased.

  In FIG. 6, Vcr2 is the voltage across the current resonance capacitor Cri, Q2ds is the drain-source voltage of the switching element Q2, Q2id is the drain current of the switching element Q2, and Q1id is the drain current of the switching element Q1.

  When the output voltage Vo2 is increased by increasing the output current on the output voltage Vo2 side or changing the output voltage, as shown in FIG. 6B, the second secondary of the transformer T1 is turned on during the ON period of the switching element Q1. Energy is transmitted to winding S2.

  That is, the voltage applied to the primary winding P1 of the transformer T1 is increased during the ON period of the switching element Q1 (OFF period of the switching element Q2). Further, since the voltage on the output voltage Vo1 side should not change, the charging voltage of the current resonance capacitor Cri must be constant.

  For this reason, the control circuit 10a keeps the upper limit voltage of the voltage Vcr2 across the current resonance capacitor Cri as it is, and the lower limit voltage (bottom voltage) VL2 becomes lower than the lower limit voltage VL1 shown in FIG. The ON period of the switching element Q2 is lengthened. In FIG. 6, it can be seen that the period of the current Q2id flowing through the switching element Q2 is long. As a result, the output voltage Vo2 can be increased.

  In order to keep the output voltage Vo constant, the ON period of the switching element Q1 is shortened to suppress the upper limit voltage of the both-ends voltage Vcr2 of the current resonance capacitor Cri.

  Next, referring to the timing charts of FIGS. 7A to 7B, the ON period of the switching element Q1 is set so that the upper limit voltage of the amplitude voltage of the current resonance capacitor Cri increases when the output voltage Vo1 is increased. A method for controlling the above will be described. FIG. 7A shows the operation of each part when the output voltage Vo1 is not raised, and FIG. 7B shows the operation of each part when the output voltage Vo1 is raised.

  As shown in FIG. 7B, the voltage applied to the primary winding P1 of the transformer T1 is increased when the output voltage Vo1 is increased by increasing the output current on the output voltage Vo1 side or changing the output voltage. There is a need. In this case, energy is transmitted to the first secondary winding S1 of the transformer T1 during the ON period of the switching element Q2.

  Therefore, the control circuit 10a extends the ON period of the switching element Q1 so that the upper limit voltage VH2 of the both-ends voltage Vcr1 of the current resonance capacitor Cri becomes larger than the upper limit voltage VH1 shown in FIG. In FIG. 7, it can be seen that the period of the current Q1id flowing through the switching element Q1 is long. As a result, the output voltage Vo1 can be increased.

  Here, the lower limit voltage VL of the voltage Vcr2 across the current resonance capacitor Cri is not changed because the output voltage setting on the output voltage Vo2 side is not changed.

  FIG. 8 is a circuit configuration diagram showing a multi-output switching power supply apparatus according to Embodiment 2 of the present invention. In the multi-output switching power supply apparatus according to the first embodiment, the series circuit of the reactor Lr, the primary winding P1 of the transformer T1, and the current resonance capacitor Cri is connected between the drain and source of the switching element Q2.

  On the other hand, the multi-output switching power supply apparatus according to the second embodiment is characterized in that a series circuit of a reactor Lr, a primary winding P1 of a transformer T1, and a current resonance capacitor Cri is connected to a switching element Q1.

  In this case, the first secondary winding S1 and the second secondary winding S2 of the transformer T1 are such that when the switching element Q1 is turned on, the energy on the primary side of the transformer T1 is the first secondary winding S1. And when the switching element Q2 is turned on, the primary energy is wound so as to be transmitted to the second secondary winding S2.

  That is, the first secondary winding S1 is wound in the reverse phase with respect to the primary winding P1 of the transformer T1, and the second secondary winding S2 is wound in the same phase with respect to the primary winding P1 of the transformer T1. It has been turned.

  Further, since the second feedback signal FBH from the feedback circuit 6 becomes a feedback signal of the output voltage Vo1, and the first feedback signal FBL from the feedback circuit 5 becomes a feedback signal of the output voltage Vo2, each of the feedback circuits 5 and 6 The connection is switched.

  As described above, the multi-output switching power supply apparatus according to the second embodiment can achieve the same effects as the multi-output switching power supply apparatus according to the first embodiment.

  FIG. 9 is a circuit diagram showing a multi-output switching power supply device according to Embodiment 3 of the present invention. The multi-output switching power supply of Example 3 shown in FIG. 9 is further different from the multi-output switching power supply of Example 1 shown in FIG. 2 in that a logic circuit or the like is provided between the output voltage Vo1 terminal and the ground GND terminal. A load RL is connected, and a load composed of a plurality of light emitting diodes LED connected in series is connected between the output voltage Vo2 terminal and the ground GND terminal. A current detection resistor Rs is connected between a connection point between the second secondary winding S2 and the capacitor C2 and the ground GND terminal.

  The feedback circuit 5 receives a voltage across the load RL such as a logic circuit from the output voltage Vo terminal, and outputs a feedback signal FBL corresponding to the voltage across the load RL to the control circuit 10a. The control circuit 10a performs constant voltage control based on the feedback signal FBL corresponding to the voltage across the load RL, and applies a constant voltage to the load RL. For this reason, the load RL can be operated stably.

  The feedback circuit 6 inputs a current flowing through the current detection resistor Rs, and outputs a feedback signal FBH corresponding to the input current to the control circuit 10a. The control circuit 10a performs constant current control based on the feedback signal FBH corresponding to the input current, and causes a constant current to flow through the plurality of light emitting diodes LED. For this reason, luminance unevenness of the light emitting diode LED is eliminated.

  As a modification of the third embodiment, for example, in the case of a load such as a light emitting diode LED, a circuit as shown in FIG. 10 may be used. In this case, a photocoupler PC, an operational amplifier OP1, a feedback resistor Rf, and a feedback capacitor Cf are further provided in the configuration shown in FIG.

  The inverting terminal of the operational amplifier OP1 and one end of the feedback resistor Rf are connected to the connection point between the current detection resistor Rs and the light emitting diode LED, and the reference voltage Vref-cc is connected to the non-inverting terminal of the operational amplifier OP1. The other end of the feedback resistor Rf is connected to the output terminal of the operational amplifier OP1 via the feedback capacitor Cf. The output terminal of the operational amplifier OP1 is connected to the output voltage Vo2 terminal through the resistor Ro and the diode of the photocoupler PC. The phototransistor of the photocoupler PC is connected to the feedback circuit 6.

  According to this configuration, when the current flowing through the light emitting diode LED increases, the voltage of the current detection resistor Rs increases, and the voltage of the inverting terminal of the operational amplifier OP1 increases. For this reason, since the output of the operational amplifier OP1 decreases, the current flowing through the diode of the photocoupler PC and the resistor Ro increases. For this reason, the current flowing through the phototransistor of the photocoupler PC also increases. That is, since a current corresponding to the current flowing through the current detection resistor Rs is input to the feedback circuit 6, constant current control of the light emitting diode LED can be performed by the control circuit 10a via the feedback circuit 6.

  The present invention is applicable to switching power supply devices such as a DC-DC converter and an AC-DC converter.

DESCRIPTION OF SYMBOLS 1 AC power supply 2 Full wave rectifier circuit 5, 6 Feedback circuit 10, 10a, 11a Control circuit 101, 102 Current mirror circuit 103, 104 Delay circuit D1-D5 Diode C1-C3 Smoothing capacitor R1-R6, R11-R13 Resistor Q1, Q2, Q3 switching element Q4-Q7 transistor Q8, Q9 MOSFET
CP1, CP2 Comparator FF1 Flip-flop circuit BF1-BF4 Buffer circuit DR1, DR2 Driver circuit Cri Current resonance capacitor
Crv voltage resonance capacitor Lr reactor T1 transformer P1 primary winding S1 first secondary winding S2 second secondary winding
LED Light emitting diode RL Load
Rs Current detection resistor PC Photocoupler OP1 Operational amplifier Rf Feedback resistor
Cf feedback capacitor

Claims (6)

DC power supply,
A first series circuit that is connected to both ends of the output of the DC power source and in which a first switching element and a second switching element are connected in series;
A control circuit that converts a DC voltage of the DC power source into an AC voltage by alternately turning on and off the first switching element and the second switching element;
A transformer having a primary winding, a first secondary winding and a second secondary winding;
A second series circuit connected to both ends of the first switching element or the second switching element, wherein a reactor, the primary winding of the transformer, and a current resonance capacitor are connected in series;
A first rectifying / smoothing circuit that rectifies and smoothes a voltage generated in the first secondary winding of the transformer and outputs a first output voltage;
A second rectifying / smoothing circuit that rectifies and smoothes a voltage generated in the second secondary winding of the transformer and outputs a second output voltage;
The control circuit controls an ON period of the first switching element based on the first output voltage from the first rectifying / smoothing circuit, and controls the second output voltage from the second rectifying / smoothing circuit based on the second output voltage. A multi-output switching power supply device for controlling an ON period of a switching element.   2. The multi-output switching power supply apparatus according to claim 1, wherein when the first output voltage is increased, the control circuit extends the ON period of the first switching element so that an upper limit voltage of the current resonance capacitor is increased.   3. The multi-output switching power supply apparatus according to claim 2, wherein when the second output voltage is increased, the control circuit extends the ON period of the second switching element so that a lower limit voltage of the current resonance capacitor decreases. The control circuit includes a first capacitor that is charged by the first output voltage;
A first comparing means for generating a second pulse signal by comparing a charging voltage charged in the first capacitor with a reference voltage, and outputting the second pulse signal to the second switching element;
First discharging means for discharging the charge of the first capacitor by the second pulse signal from the first comparing means;
A second capacitor charged by the second output voltage;
A second comparing means for generating a first pulse signal by comparing a charging voltage charged in the second capacitor and the reference voltage, and outputting the first pulse signal to the first switching element;
Second discharge means for discharging the charge of the second capacitor by a first pulse signal from the second comparison means;
The on-time of the first pulse signal from the second comparing means is determined based on a charging time obtained by comparing the charging voltage charged in the first capacitor with a reference voltage, and the second capacitor is charged. An on-time determining means for determining an on-time of a second pulse signal from the first comparing means based on a charging time obtained by comparing a charging voltage with the reference voltage;
The multi-output switching power supply device according to claim 3.   In the first secondary winding and the second secondary winding of the transformer, energy on the primary side of the transformer is transmitted to the second secondary winding when the first switching element is turned on. 5. The multi-output switching power supply device according to claim 4, wherein the primary output is wound so that the energy on the primary side is transmitted to the first secondary winding when the second switching element is turned on. For each of the load connected to the output terminal of the first rectifying / smoothing circuit and the load connected to the output terminal of the second rectifying / smoothing circuit, the both-end voltage of the load is detected according to the type of the load, or the Having detection means for detecting the current flowing through the load;
The control circuit performs constant voltage control of the load when the voltage across the load is detected by the detection means, and controls the load when the current flowing through the load is detected by the detection means. The multi-output switching power supply device according to any one of claims 1 to 5, wherein current control is performed.

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