JP2010245905A - Transmission apparatus and communication system - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To eliminate a 90-degree phase shifter and/or an orthogonal oscillator or the like for orthogonalizing I-axis and Q-axis to simplify a circuit while providing high precision and high-speed transmission. <P>SOLUTION: In a transmission apparatus, a baseband signal Si is modulated into a modulated signal Sa by a BPSK modulation part 212. The modulated signal Sa is multiplied with a carrier signal Sc by a mixer 214, up-converted, and supplied to an input point 218. A baseband signal Sq is modulated into a modulated signal Sb by a BPSK modulation part 213. The modulated signal Sb is multiplied with the carrier signal Sc by a mixer 215, up-converted, and supplied to an input point 219. The input point 218 and input point 219 are set deviating each other by a distance of L1=(1/4+N)λ wavelength in carrier direction of a waveguide 220. This will cause a phase difference between the modulated signals Sa, Sb received by an output point 258 to deviate by λ/4 wavelength to provide an IQ orthogonal axis. <P>COPYRIGHT: (C)2011,JPO&INPIT

Description

  The present invention relates to a transmission device and a communication system. Specifically, a first input point for inputting the first transmission signal to the waveguide and a second input point for inputting the second transmission signal to the waveguide are referred to as the first transmission signal and the second transmission signal. Is shifted by a distance that gives a predetermined phase difference between the two and the high-speed transmission by the IQ orthogonal axis.

  In recent years, high-speed transmission technology using a high-frequency band such as millimeter waves has been actively studied aiming at low-power consumption and low-cost (small circuit scale) signal transmission by using CMOS technology. This is because a digital circuit module and a millimeter-wave RF front-end module can be realized on a single chip by using a CMOS process suitable for mass production. A conventional high-speed baseband signal transmission device includes a substrate with a small dielectric loss, first and second components mounted on the substrate, and a waveguide that couples the first and second components. Has been. By reducing mutual interference, high-speed signal transmission between the first and second components on the substrate is realized.

By the way, in the above-described high-speed baseband signal transmission apparatus and the like, phase modulation using orthogonal IQ axes has been widely used for the purpose of further increasing the transmission speed. Phase modulation is a modulation method in which the phase of a carrier wave is discretely changed according to a digital code in digital transmission, for example, and performs mapping of a digital code with respect to a phase component θ given in (1) below. is there.
S (t) = A (t) cos (2πfc + θ (t)) (1)
Here, A (t) is the amplitude and θ (t) is the phase.

  For example, when QPSK modulation which is a kind of phase modulation is considered, it can be realized by expressing a 2-bit digital code with one phase. An example of the mapping between the digital code and the phase θ is shown in FIG. 16, and how each digital code is mapped on the complex baseband signal is shown in FIG. As described above, a technique that can increase the speed by placing multi-bit information on one phase information has been performed for a long time. As other techniques for transmitting multiple bits, QAM (Quadrature Amplitude Modulation) transmission using both amplitude and phase has been actively performed in recent years.

  However, in order to perform these transmissions, a 90 ° phase shifter or the like that makes the I axis and Q axis orthogonal to each other in the complex plane is required. It is required to be realized with. Therefore, various accuracy compensation methods have been considered so far.

  For example, the invention described in Patent Document 1 concerns the accuracy of the quadrature modulation circuit, and by increasing or decreasing the DC bias so that the detection voltage generated by the detector becomes a predetermined value or less, the carrier leak can be reduced to a desired value. A transmitter including a compensation circuit that performs suppression control below a value is disclosed.

  Further, in the invention described in Patent Document 2, when accuracy is poor and IQ orthogonality is unnecessary, switching to the BPSK system is performed, and switching to the QPSK system is performed when accuracy is good, so that transmission can be performed rather than maintaining the orthogonal accuracy of IQ. A wireless communication device that ensures channel maintenance is disclosed.

Japanese Patent Laid-Open No. 10-70582 (page 5, FIG. 1) JP 2007-150646 A (page 8, Fig. 4)

  However, in the inventions disclosed in Patent Documents 1 and 2, there are the following problems when trying to realize high-speed transmission using IQ orthogonal axes using a 90 ° phase shifter or the like. That is, in the invention described in Patent Document 1, it is necessary to separately provide a compensation circuit including a calculation unit, a detection unit, and the like in addition to the transmission path including the original signal modulation unit and mixer. For this reason, there is a problem in that the circuit of the transmitter is reduced in size, which leads to an increase in cost. In the invention described in Patent Document 2, it is clear that the transmission rate is lowered when BPSK modulation is applied, and stable high-speed transmission cannot be expected.

  Therefore, the present invention has been made in view of the above-described problems, and has achieved high accuracy and high speed while simplifying the circuit by reducing a 90 ° phase shifter, a quadrature oscillator, or the like that makes the I axis and Q axis orthogonal. An object of the present invention is to provide a transmission device and a communication system using a waveguide that realize transmission.

  A transmission apparatus according to the present invention receives a first transmission unit that modulates a carrier signal having a predetermined frequency based on a first signal input thereto and outputs the first transmission signal, and a carrier signal having a predetermined frequency. And a second transmission unit that modulates based on the second signal and outputs a second transmission signal, and inputs the first transmission signal output from the first transmission unit to the waveguide. And a second input point for inputting the second transmission signal output from the second transmission unit to the waveguide is a predetermined position between the first transmission signal and the second transmission signal. It is shifted by a distance that gives a phase difference.

  The communication system according to the present invention also includes a first transmitter that modulates a carrier signal having a predetermined frequency based on a first signal input thereto and outputs a first transmission signal, and a carrier signal having a predetermined frequency. A second transmission unit that modulates the second transmission signal and outputs a second transmission signal, a first transmission signal output from the first transmission unit, and a second transmission unit A waveguide to which the second transmission signal output from the transmission unit is input, and the first and second transmission signals transmitted through the waveguide are received, and the first and second transmission signals are determined in advance. A receiving device having a receiving unit that demodulates based on a frequency carrier signal to obtain a received signal, and a first input point that inputs the first transmission signal output by the first transmitting unit to the waveguide; The second transmission signal output from the second transmitter is input to the waveguide. That a second input point, in which are shifted by a distance that gives a predetermined phase difference between the first transmission signal and the second transmit signal.

  In the present invention, a carrier wave signal having a predetermined frequency is modulated by the first transmission unit based on the input first signal and is output as the first transmission signal. A carrier wave signal having a predetermined frequency is modulated by the second transmitter based on the input second signal and is output as a second transmission signal. Examples of the modulation method include phase modulation and amplitude modulation, and more specifically, modulation methods such as BPSK, QPSK, 8-phase PSK, and QAM.

  The signal-processed first transmission signal is input to the waveguide via the first input point, and the second transmission signal is input to the waveguide via the second input point. In the present invention, the first input point and the second input point are shifted by a distance that gives a predetermined phase difference between the first transmission signal and the second transmission signal. Therefore, for example, when the distance between the first and second input points is shifted by (1/4 + N) λ wavelengths, the phase difference between the first transmission signal and the second transmission signal is set to 90 °. Since it can be controlled, high-speed transmission can be realized by using the IQ orthogonal axis. Here, N is an integer, and λ is the wavelength of the carrier signal. In addition, since a highly accurate IQ orthogonal axis can be realized without using a complicated system, a high-order modulation method such as multi-level modulation can be realized.

  According to the present invention, the first input point and the second input point are provided in the waveguide while being shifted by a distance that gives a phase difference between the first transmission signal and the second transmission signal. IQ quadrature transmission can be realized without using a quadrature oscillator or a 90 ° phase shifter. As a result, it is possible to reduce the circuit scale of the transmission device and thus reduce the cost.

It is a figure which shows the structural example of the high frequency transmission system which concerns on the 1st Embodiment of this invention. It is a figure which shows the block structural example of a high frequency transmission system. It is a figure which shows the circuit structural example of a high frequency transmission system. It is a figure which shows the phase relationship example of the modulation signal on a waveguide. It is a figure which shows the structural example of the high frequency transmission system which concerns on the modification of 1st Embodiment. It is a figure which shows the structural example of the antenna member of the high frequency transmission system which has a calibration function which concerns on the 2nd Embodiment of this invention. It is a figure which shows the modification of an antenna member. It is a figure which shows the structural example of the high frequency transmission system which has a calibration function. It is a figure which shows the circuit structural example of the high frequency transmission system which has a calibration function which concerns on the modification of the 2nd Embodiment of this invention. It is a figure which shows the structural example of the high frequency transmission system which has a calibration function which concerns on the 3rd Embodiment of this invention. It is a figure which shows the structural example of the phase control part of the high frequency transmission system which has a calibration function which concerns on the 4th Embodiment of this invention. It is a figure which shows the structural example of the phase control part of the high frequency transmission system which has a calibration function which concerns on the 5th Embodiment of this invention. It is a figure which shows the structural example of the phase control part of the high frequency transmission system which has a calibration function which concerns on the 6th Embodiment of this invention. It is a flowchart which shows the operation example of the high frequency transmission system which has a calibration function which concerns on the 7th Embodiment of this invention. It is a flowchart which shows the operation example of the high frequency transmission system which has a calibration function which concerns on the 8th Embodiment of this invention. It is a figure for demonstrating QPSK modulation (the 1). It is a figure for demonstrating QPSK modulation (the 2).

Hereinafter, the best mode for carrying out the invention (hereinafter referred to as an embodiment) will be described. The description will be given in the following order.
1. 1st Embodiment (example which implement | achieves IQ orthogonal axis in a high frequency transmission system)
2. 2. Modification of first embodiment Second Embodiment (Example in which calibration is performed using an antenna member)
4). 4. Modification of second embodiment Third Embodiment (Example of performing amplitude calibration)
6). Fourth Embodiment (Example of performing calibration using a liquid crystal layer)
7). Fifth Embodiment (Example of performing calibration using a delay element)
8). Sixth Embodiment (Example of performing calibration using a phase shift element)
9. Seventh Embodiment (Example in which calibration is performed in conjunction with a transmission device and a reception device)
10. Eighth Embodiment (Example in which calibration is performed in conjunction with a transmission device and a reception device)

<1. First Embodiment>
[Configuration example of high-frequency system]
As shown in FIG. 1, the high-frequency transmission system 100A according to the present invention is an example of a communication system, and can transmit a millimeter-wave signal having a frequency of a transmitted signal of 30 GHz to 300 GHz at high speed, for example. The high-frequency transmission system 100A is composed of a part A, a part B, a part C, a part D, and a waveguide 20 that couples the parts A, B, C, and D mounted on the substrate 40. .

  Each of the parts A to D has a built-in millimeter-wave transmission / reception module. In this example, the parts A and C are configured as a transmitting apparatus, and the parts B and D are configured as a receiving apparatus. The component A and the component B are connected via the waveguide 20 and perform high-speed signal transmission such as an image signal and an audio signal between the components A and B. The component C and the component D are connected via the waveguide 20 and perform high-speed signal transmission such as an image signal and an audio signal between the components C and D. In the following, signal transmission between the component A (transmitting device 200) and the component B (receiving device 300) will be described as an example.

[Example of block configuration of high-frequency system]
As shown in FIG. 2, the high-frequency transmission system 100A includes a transmission device 200, a reception device 300, and a waveguide 20 that couples them. The transmission apparatus 200 includes a signal input unit 10, a modulation circuit 12, a frequency conversion circuit 14, an amplifier 16, and a coupling circuit 18 for a substrate. The receiving apparatus 300 includes a circuit board coupling circuit 22, an amplifier 24, a frequency conversion circuit 26, a demodulation circuit 28, and a signal output unit 30.

  A predetermined signal generated in the transmission apparatus 200 is input to the signal input unit 10. The signal input to the signal input unit 10 is supplied to the modulation circuit 12. The modulation circuit 12 modulates the supplied signal and supplies it to the frequency conversion circuit 14. The frequency conversion circuit 14 up-converts the modulated signal to a desired frequency band (millimeter wave) and supplies it to the amplifier 16. The amplifier 16 amplifies the up-converted signal and supplies it to the coupling circuit 18. The coupling circuit 18 transmits the amplified signal to the receiving device 300 through the waveguide 20.

  The coupling circuit 22 of the receiving device 300 receives a signal transmitted from the transmitting device 200 side via the waveguide 20 and supplies the signal to the amplifier 24. The amplifier 24 amplifies the received signal to compensate for attenuation, and supplies the amplified signal to the frequency conversion circuit 26. The frequency conversion circuit 26 down-converts the amplified signal and supplies it to the demodulation circuit 28. The demodulation circuit 28 demodulates the downconverted signal to obtain a baseband signal. Finally, the signal output unit 30 outputs a data string based on the demodulated baseband signal, whereby signal transmission from the transmission device 200 to the reception device 300 is completed. If the transmitter 200 and the receiver 300 that perform communication are close to each other, the amplifiers 16 and 24 on the transmission side and the reception side may be omitted.

[Circuit configuration example of high-frequency transmission system]
Next, a circuit configuration example of the above-described high-frequency transmission system 100A will be described. In FIG. 3, BPSK modulation sections 212 and 213 correspond to the modulation circuit 12 shown in FIG. 2, and the carrier wave signal generation section 216 and the mixers 214 and 215 correspond to the frequency conversion circuit 14 shown in FIG. Reference numerals 218 and 219 correspond to the coupling circuit 18 shown in FIG. In FIG. 3, the output point 258 corresponds to the coupling circuit 22 shown in FIG. 2, and the QPSK demodulator 250 corresponds to the demodulation circuit 28 shown in FIG.

  As shown in FIG. 3, each signal to be transmitted to receiving apparatus 300 is assigned to an I phase and a Q phase. The baseband signal Si assigned to the I phase is supplied to the BPSK modulation unit 212. The BPSK modulation unit 212 generates a modulation signal Sa by performing BPSK modulation (mapping) according to the allocated bits and supplies the modulated signal Sa to the mixer 214. The carrier wave signal generation unit 216 generates a carrier wave signal Sc having a predetermined frequency and supplies it to the mixer 214. The mixer 214 multiplies the modulation signal Sa generated by the BPSK modulation unit 212 and the carrier signal Sc generated by the carrier signal generation unit 216 to frequency-convert (up-convert) the modulation signal Sa, and the frequency-converted modulation A signal Sa is supplied to the input point 218. The baseband signal Si is an example of the first signal, and the modulation signal Sa is an example of the first transmission signal. The input point 218 is an example of a first input point, and the BPSK modulation unit 212 and the mixer 214 constitute an example of a first transmission unit.

  The baseband signal Sq assigned to the Q phase is supplied to the BPSK modulation unit 213. The BPSK modulation unit 213 generates a modulation signal Sb by performing BPSK modulation (mapping) according to the assigned bits, and supplies the modulated signal Sb to the mixer 215. The carrier wave signal generation unit 216 generates a carrier wave signal Sc having a predetermined frequency and supplies it to the mixer 215. The mixer 215 multiplies the modulation signal Sb generated by the BPSK modulation unit 213 and the carrier signal Sc to frequency-convert (up-convert) the modulation signal Sb, and supplies the frequency-converted modulation signal Sb to the input point 219. . The baseband signal Sq is an example of the second signal, and the modulation signal Sb is an example of the second transmission signal. The input point 219 is an example of a second input point, and the BPSK modulation unit 213 and the mixer 215 constitute an example of a second transmission unit.

The input points 218 and 219 are provided on one end side of the waveguide 220, and each of the input points 218 and 219 is provided with a dipole antenna, a loop antenna, a small aperture coupling element (slit antenna) or the like as described later. The input point 218 and the input point 219 are shifted by a distance L1 that satisfies the relationship of the following expression (2) that gives a predetermined phase difference between the modulation signal Sa and the modulation signal Sb in the conveyance direction of the waveguide 220. Yes.
L1 = (1/4 + N) λ (2)
Here, λ = c / √ε _r f, C is the high speed in vacuum, f is the frequency of the carrier signal, εr is the relative dielectric constant of the waveguide, and N is an integer.

  As a result, as shown in FIG. 4B, the phase of the modulated signal Sa transmitted from the input point 218 and the modulated signal Sb transmitted from the input point 219 are shifted by at least 90 ° on the complex plane. . That is, by shifting the distance L1 between the input point 218 and the input point 219 by (1/4 + N) λ, the phases of the modulation signals Sa and Sb can be made orthogonal without using a 90 ° phase shifter as in the prior art. become able to. When this state is observed at one point on the waveguide 220 (for example, the output point 258), it is equal to the relationship between the sin wave and the cos wave, and it is as if a QPSK modulated signal has arrived. FIG. 4A shows the phase relationship between the modulation signals Sa and Sb before the phase is shifted.

  Returning to FIG. 3, the waveguide 220 has an elongated casing partitioned by a conductive member, and is mounted between the transmitter 200 and the receiver 300 mounted on the substrate 40. A dielectric having a predetermined relative dielectric constant, such as air or epoxy resin (the same material as the substrate 40), is enclosed in the housing of the waveguide 220, for example. In addition, as the substrate 40, for example, a copper foil is attached to both surfaces with a glass epoxy resin as an insulating base, for example, the dielectric loss tangent (tan δ) is 0.01 or more, and conventionally, transmission loss is large in the millimeter wave band, A substrate having a large loss that is not suitable for millimeter wave transmission is used.

  The output point 258 is provided on the other end side of the waveguide 220. The output point 258 is provided with a dipole antenna, a loop antenna, a small aperture coupling element (slit antenna), or the like as will be described later. The output point 258 receives the modulation signals Sa and Sb transmitted through the waveguide 220 and supplies them to the QPSK demodulator 250.

  The QPSK demodulator 250 performs QPSK modulation on the orthogonally-modulated modulation signals Sa and Sb received by the output point 258, thereby generating a baseband signal Sirx based on the modulation signal Sa and a baseband signal Sqrx based on the modulation signal Sb. Get.

  As described above, in the present embodiment, the distance L1 between the input point 218 and the input point 219 is set to (1/4 + N) λ and coupled to the waveguide 220. Thereby, the phase difference between the first modulation signal Sa and the second modulation signal Sb can be set to 90 °, and IQ orthogonal transmission can be realized without using a quadrature oscillator or a 90 ° phase shifter. Become. As a result, it is possible to reduce the circuit scale of the transmission apparatus 200, and hence the cost. Furthermore, in the high-frequency transmission system 100A using the waveguide 220, high-speed transmission can be realized by using the IQ orthogonal axis described above.

<2. Modification of First Embodiment>
In the modification of the first embodiment, the receiving device 300 of the high-frequency transmission system 100B adopts a configuration different from the receiving device 300 of the high-frequency transmission system 100A described in the first embodiment. In addition, the same code | symbol is attached | subjected to the component which is common in the high frequency transmission system 100A of 1st Embodiment mentioned above, and detailed description is abbreviate | omitted.

  As illustrated in FIG. 5, the reception device 300 includes output points 260 and 261, mixers 254 and 255, and BPSK demodulation units 252 and 253. The output point 261 is provided on the other end side of the waveguide 220, and a dipole antenna or the like is provided at the output point 261, for example. The output point 261 receives the modulation signal Sa transmitted via the waveguide 220 and supplies it to the mixer 254. The carrier wave signal generation unit 256 generates a carrier wave signal Scrx and supplies it to the mixer 254. The mixer 254 multiplies the modulation signal Sa and the carrier signal Scrx to frequency-convert (down-convert) the modulation signal Sa, and supplies the frequency-converted modulation signal Sa to the BPSK demodulator 252. The BPSK demodulator 252 demodulates (maps) the modulated signal Sa to obtain a baseband signal Sirx.

  The output point 260 is provided on the other end side of the waveguide 220. The output point 260 is provided with, for example, a dipole antenna. The output point 260 is provided at a position shifted from the output point 261 by a distance L1 that satisfies the relationship of the above expression (2). Therefore, the modulation signal Sb transmitted through the waveguide 220 can be received in a state of being shifted from the modulation signal Sa by (¼ + N) λ wavelength. The received modulation signal Sa is supplied to the mixer 255. The mixer 255 multiplies the modulation signal Sb received via the waveguide 220 and the carrier signal Scrx to frequency-convert (down-convert) the modulation signal Sb, and sends the frequency-converted modulation signal Sa to the BPSK demodulator 253. Supply. The BPSK demodulator 253 demodulates (maps) the modulated signal Sb to obtain a baseband signal Sqrx.

  Even if the receiving apparatus 300 is configured in the same manner as the transmitting apparatus 200 as in this modification, the same operational effects as in the first embodiment can be obtained. For example, the phase difference between the first modulation signal Sa and the second modulation signal Sb can be set to 90 °, and IQ orthogonal transmission can be realized without using a quadrature oscillator or a 90 ° phase shifter. Therefore, it is possible to reduce the circuit scale of the transmission apparatus 200, and thus reduce the cost.

<3. Second Embodiment>
[Example of high-frequency transmission system with phase calibration function]
In the second embodiment, a high-frequency transmission system 100C having a calibration function for adjusting the phase difference between the I-phase modulation signal Sa and the Q-phase modulation signal Sb to 90 ° will be described. Note that the high-frequency transmission system 100C is obtained by adding a calibration function to the above-described high-frequency transmission system 100A, and thus description of common components and operations is omitted.

(Configuration example of antenna member)
First, a configuration example of the antenna member of the high-frequency transmission system 100C having a calibration function will be described. As shown in FIGS. 6 and 8, the I-phase input point 218 is provided with a dipole-type antenna member 218a having a predetermined length based on the wavelength λ of a millimeter wave signal, for example. The antenna member 218 a is coupled to one end side of the waveguide 220, and radiates the test modulation signal Sat output from the mixer 254 into the waveguide 220.

  The Q-phase input point 219 is provided with, for example, four antenna members 219a to 219d of a dipole type or the like having a predetermined length based on the wavelength λ of the millimeter wave signal, and each is guided at a predetermined interval. It is coupled to one end side of the waveguide 220. Any one of the antenna members 219a to 219d is electrically connected to the switch unit 274 in response to switching, and receives the test modulation signal Sat radiated from the antenna member 218a in the calibration mode. In the above-described example, the four antenna members 219a to 219d are used. However, the number may be two, or five or more. By increasing the number of antenna members, finer adjustment of the phase difference becomes possible.

  Here, of the antenna members 219a to 219d, for example, the antenna member 219b is coupled to a position shifted by (1/4 + N) λ from the I-phase antenna member 218a, and the other antenna members 219a, 219c, and 219d are antenna members 413. It is combined with the position near. The position (phase shift) information of the antenna members 219a to 219d is stored in a position rotation amount calculation unit 270, which will be described later, and the optimum antenna member 219a to 219a to 219a is based on the calculated shift amount and the position information of the antenna members 219a to 219d. 219d can be selected.

(Another configuration example of the antenna member)
In the above-described example, the antenna members 218a to 218d are configured by dipole antennas, but may be configured by slit-shaped antennas instead. As shown in FIG. 7, the slit antenna 318 a is a bottom surface portion on one end side of the waveguide 220 and is formed along the short direction. The slit antennas 319a to 319d are bottom portions on one end side of the waveguide 220, and are formed at predetermined intervals along the short direction.

  Any of the slit antennas 319a to 319d (for example, the slit antenna 319b) is formed at a position shifted from the slit antenna 318a by (1/4 + N) λ. The other slit antennas 319a, 319c, and 319d are formed in the vicinity of the slit antenna 319b. Even with such a configuration, the phase difference between the modulation signals Sa and Sb can be finely adjusted.

(Configuration example of high-frequency transmission system)
Next, the high-frequency transmission system 100C including the antenna members 218a and 219a to 219d described above will be described. The high-frequency transmission system 100C has a calibration mode for performing phase control and a communication mode for performing normal communication. Switching between these modes can be arbitrarily set by the user or automatically by a control unit (not shown). When the calibration mode is set, the high-frequency transmission system 100C forms a loop Lp that receives the test modulation signal Sat transmitted from the input point 218 at the input point 219, as shown in FIG. Perform calibration. That is, phase calibration is performed only on the transmitting apparatus 200 side.

  The BPSK modulation unit 212 on the I-phase side generates a test modulation signal Sat by performing BPSK modulation according to the assigned test baseband signal Sit (bits) and supplies the test modulation signal Sat to the mixer 214. The carrier wave signal generation unit 216 generates a carrier wave signal Sct and supplies it to the mixer 214. The mixer 214 multiplies the modulation signal Sat generated by the BPSK modulation unit 212 by the carrier signal Sct to frequency-convert (upconvert) the modulation signal Sat, and supplies the frequency-converted modulation signal Sat to the input point 218. The modulation signal Sat supplied to the input point 218 is transmitted to the inside of the waveguide 220 through the antenna member.

  Any of the antenna members 219a to 219d provided at the input point 219 also functions as an antenna member that receives the test modulation signal Sat in the calibration mode, and receives the modulation signal Sat transmitted from the input point 218. To the mixer 215. Here, it is assumed that the distance L1 between the input points 218 and 219 is set to (1/4 + N) λ as described above. The mixer 215 multiplies the modulation signal Sat received by the antenna member at the input point 219 by the carrier signal Sct generated by the carrier signal generation unit 216, thereby frequency-converting (down-converting) the modulation signal Sat and receiving the received signal. Get Sarx. The frequency-converted received signal Sarx is supplied to the position rotation amount calculation unit 270 via the switch unit 272. In the calibration mode, the switch unit 272 is switched to the terminal b side by the position rotation amount calculation unit 270 or a control unit (not shown).

  The position rotation amount calculation unit 270 is an example of a control unit, and determines whether or not the phase difference between the test reception signal Sarx output from the mixer 215 and the reference signal St is 90 ° (1/4 + N) λ. to decide. This is because, since the phase difference between the input points 218 and 219 is set to (1/4 + N) λ, the phase difference between the reference signal St and the reception signal Sarx is theoretically 90 °. As the reference signal St, the carrier signal Sct supplied from the carrier signal generation unit 216 or the modulation signal Sat modulated by the BPSK modulation unit 212 and transmitted from the input point 218 in advance is used. When the position rotation amount calculation unit 270 determines that the phase difference is not 90 °, how much the phase of the reception signal Sarx is shifted when the rotation amount of the phase, specifically 90 ° is used as a reference. Calculate the amount of deviation. Then, the position rotation amount calculation unit 270 selects any one of the antenna members 219a to 219d based on the calculated deviation amount, generates a switching signal corresponding to the selection, and supplies the switching signal to the switch unit 274.

  The switch unit 274 switches to the antenna members 219a to 219d corresponding to the switching signal based on the switching signal supplied from the position rotation amount calculation unit 270 (see FIG. 6). As a result, the input points 218 and 219 are finely adjusted to (1/4 + N) λ in the calibration mode, so that the phase difference between the I-phase modulation signal Sa and the Q-phase modulation signal Sb is set to 90 in the communication mode. It becomes possible to set to °. As a result, high-speed transmission can be realized by a highly accurate IQ orthogonal axis.

<4. Modification of Second Embodiment>
[Example of high-frequency transmission system with phase calibration function]
In the modification of the second embodiment, a case where the antenna members 219a to 219d are calibrated using the test carrier signal Sci will be described. In FIG. 9, the configuration of the BPSK modulators 212, 213, etc. is omitted for convenience.

  As shown in FIG. 9, the carrier signal generation unit 216 of the high-frequency transmission system 100D generates a test carrier signal Sci having a predetermined frequency. The test carrier signal Sci generated by the carrier signal generator 216 is radiated from the antenna member 218 a at the input point 218 to the inside of the waveguide 220.

  The antenna member 219A at the input point 219 receives the test carrier signal Sci radiated from the antenna member 218a at the input point 218 and supplies it to the mixer 215. In the following description, when any one of the antenna members 219a to 219d is indicated, it is referred to as an antenna member 219A (see FIG. 6). The carrier signal generation unit 216 generates a test carrier signal Scq having a predetermined frequency and supplies it to the mixer 215. It is assumed that the test carrier signal Sci and the carrier signal Scq are set to the same frequency. The mixer 215 multiplies the test carrier signal Sci and the carrier signal Scq, and supplies the output signal Smx obtained by the multiplication to the frequency analysis unit 280.

  In the frequency analysis unit 280, the frequency of the output signal Smx output from the mixer 215 and observed by the frequency analysis unit 280 is twice the frequency of the test carrier signals Sci and Scq generated by the carrier signal generation unit 216. It is determined whether or not there is (see FIG. 9B). If the distance L1 between the input points 218 and 219 is (1/4 + N) λ, that is, if the phase difference between the test carrier signals Sci and Scq is 90 °, the mixer 215 multiplies the carrier signals Sci and Scq. This is because a double frequency component can be obtained. The frequency analysis unit 280 constitutes an example of a control unit.

  When the frequency analysis unit 280 determines that the frequency of the output signal Smx is twice, the frequency analysis unit 280 determines that the distance L1 between the input points 218 and 219 is (1/4 + N) λ, and is currently set. The antenna members 219a to 219d are maintained. On the other hand, if it is determined that the frequency of the output signal Smx is not double, it is determined that the distance L1 between the input points 218 and 219 is not (1/4 + N) λ, and the antenna members 219a to 219d shown in FIG. The switch unit 587 is switched to any of the above. While repeating such operations, one of the antenna members 219a to 219d having a frequency component twice that of the carrier wave signals Sci and Scq is selected, and the distance L1 between the input points 218 and 219 is set to (1/4 + N). Set to λ. Thereby, the phase of the modulation signal Sa transmitted from the antenna member 218a at the input point 218 and the phase of the modulation signal Sb transmitted from the antenna member 219A at the input point 219 can be orthogonalized.

<5. Third Embodiment>
[Example of high-frequency transmission system with amplitude calibration function]
In the third embodiment, a method of adjusting the amplitude value in addition to the phase difference between signals transmitted from the antenna members 218a and 219A at the input points 218 and 219 in the calibration mode will be described. The operation until the modulation signal is transmitted from the input point 218 on the I-phase side is the same as that in the second embodiment, and the description thereof is omitted.

  As shown in FIG. 10, the antenna member 219A of the input point 219 of the high frequency transmission system 100E also functions as an antenna member that receives the modulation signal Sat in the calibration mode, and the modulation transmitted from the antenna member 218a of the input point 218 is performed. The signal Sat is received. The modulated signal Sat received by the antenna member 218a is supplied to the mixer 215. Here, it is assumed that the distance L1 between the input points 218 and 219 is set to (1/4 + N) λ as described above.

  The mixer 215 multiplies the modulation signal Sat received by the antenna member 219A at the input point 219 and the carrier signal Sc generated by the carrier signal generation unit 216 to frequency-convert the modulation signal Sat to obtain a reception signal Sarx. . The frequency-converted received signal Sarx is supplied to the amplitude value measuring unit 370 via the switch unit 276 set on the terminal d side.

  The amplitude value measurement unit 370 measures the amplitude value of the received signal Sarx output from the mixer 215, and is stored in advance in the memory and the carrier signal Sct supplied from the carrier signal generation unit 216 and the measured amplitude value of the received signal Sarx. The difference from the amplitude value of the modulated signal Sat before attenuation is calculated. Then, the amplitude value measuring unit 370 generates a control signal based on this difference and supplies it to the amplitude value control unit 378.

  Based on the control signal from the amplitude value measurement unit 370, the amplitude value control unit 378 converts the amplitude value of the modulation signal Sat received by the antenna member 219A at the input point 219 into the amplitude of the modulation signal Sat and the carrier signal Sct before attenuation. Amplitude control is performed to match the value. For example, when the received modulation signal Sat is attenuated, amplitude control is performed to increase the amplitude value. The amplitude-controlled test reception signal Sarx is supplied to the position rotation amount calculation unit 270 via the switch unit 276 set on the terminal b side.

  The position rotation amount calculation unit 270 performs the phase calibration as described above based on the test reception signal Sarx in which the attenuation by the waveguide 220 is corrected, and selects the optimal antenna members 219a to 219d. The I-phase side amplitude value control unit 380 and the Q-phase side amplitude value control unit 378 are also used as amplifiers for amplifying the modulation signals Sa and Sb in the normal communication mode.

  As described above, in the present embodiment, the amplitude of the modulation signal Sat due to the influence of the waveguide 220 between the input points 218 and 219 is measured by measuring the amplitude value of the test reception signal Sarx in the calibration mode. The value decay rate can be calculated. Thereby, an error in the amplitude direction can be corrected, and more accurate phase calibration can be performed.

<6. Fourth Embodiment>
[Example of high-frequency transmission system with phase calibration function]
In the fourth embodiment, the phase calibration is not performed by switching the plurality of antenna members 219a to 219d described above, but by using a dielectric whose dielectric constant is variable when electric or optical energy is applied, the I-phase is obtained. And adjust the phase difference between the Q-phase signals. In the following example, a case where liquid crystal is used as a dielectric having a variable dielectric constant will be described.

  As shown in FIGS. 11A and 11B, the transmission side end of the waveguide 220 is for adjusting the distance L between the input points 218 and 219 to control the phase difference between the two signals. The phase controller 400 is provided. The phase control unit 400 includes a case 402, a liquid crystal layer 406 sealed in the case 402, electrodes 410 and 412 provided on the upper surface and the lower surface of the case 402, and electrodes 410 and 412. And a voltage control unit 414 for applying the voltage.

  As the liquid crystal layer 406, for example, a nematic liquid crystal is preferably used. It has been clarified that the relative dielectric constant ε of the nematic liquid crystal changes from 3.0 to 3.5 depending on the applied voltage even in a millimeter wave band such as 60 GHz. Therefore, by enclosing such a liquid crystal layer 406 in the waveguide 220, it is possible to finely adjust the wavelength while applying a voltage.

  The voltage control unit 414 is connected to the position rotation amount calculation unit 270 shown in FIG. 8, calculates the applied voltage based on the rotation amount (deviation amount) calculated by the position rotation amount calculation unit 270, and calculates the calculated application voltage. Is applied to the electrodes 410, 412. Thereby, the transmittance of the signal passing through the liquid crystal layer 406 is adjusted in accordance with the applied voltage value.

  The antenna member 218X at the input point 218 is coupled to the end of the housing 402 opposite to the waveguide 220, and radiates the modulation signal Sa to the waveguide 220 via the liquid crystal layer 406. The antenna member 219 </ b> Y at the input point 219 is coupled to the end of the waveguide 220 on the phase control unit 400 side outside the housing 402, and radiates the modulation signal Sb to the waveguide 220. In this example, it is assumed that the distance L between the input points 218X and 219Y is slightly deviated from (1/4 + N) λ.

  In the high-frequency transmission system 100F configured as described above, the signal assigned to the I phase is radiated from the antenna member 218X at the input point 218 into the liquid crystal layer 406 given by the relative dielectric constant ε. In this example, the voltage control unit 414 electrically controls the applied voltage, thereby changing the dielectric constant of the liquid crystal layer 406 and changing the transmittance of the signal passing through the liquid crystal layer 406. Thus, the modulation signal Sa radiated from the I-phase side antenna member 218X and the modulation signal Sb radiated from the Q-phase side antenna member 219Y are independent of the distance L between the input points 218 and 219. The phase difference can be set to 90 °.

In addition to the liquid crystal layer 406, substances that change the dielectric constant include those that use magnetic energy and light energy, and those that use thermal energy and mechanical energy, which are applied to the system shown in the present invention. I can easily imagine that. As an example of changing the dielectric constant using magnetic energy or light energy, as described in Japanese Patent Laid-Open No. 2003-209266, for example, it is composed of a quantum paraelectric material (SrTiO ₃ , CaTiO ₃ , KTaO _3, etc.). Listed substances. Examples of thermal energy include fluorine-based ferroelectric polymers. In this case, the dielectric constant can be changed by changing the temperature of the fluorinated ferroelectric polymer with a heat sink. Furthermore, examples of mechanical energy include lithium niobate. In this case, the dielectric constant can be changed by applying pressure to the lithium niobate with a fastening member such as a screw.

<7. Fifth embodiment>
[Example of high-frequency transmission system with phase calibration function]
In the fifth embodiment, phase calibration is not performed by switching the plurality of antenna members 219a to 219d described above, but the phase difference between the I-phase and Q-phase signals is adjusted by using a delay element. .

  As shown in FIG. 12, the phase control unit 500 is configured by a delay element group such as a buffer, for example, and is provided between the switch unit 274 and the input point 219. The phase control unit 500 includes, for example, a plurality of stages (n stages), and n delay elements connected in series according to the number of stages are arranged in each stage. For example, one delay element is provided in the first stage, two delay elements are provided in series in the second stage, and n delay elements are connected in series in the n stage. Provided. The delay element disposed on one end side of each stage is electrically connected to the switch unit 274 in accordance with the switching of the switch unit 274.

  In the high-frequency transmission system 100G configured as described above, the position rotation amount calculation unit 270 illustrated in FIG. 8 has a phase difference of 90 ° based on the test modulation signal Sat in the calibration mode, for example. Such a delay element is selected. Then, the position rotation amount calculation unit 270 generates a switching signal corresponding to the selected delay element and supplies it to the switch unit 274. Based on the switching signal supplied from the position rotation amount calculation unit 270, the switch unit 274 switches to the optimal delay element that makes the phase difference between the signals 90 °.

  As described above, according to the present embodiment, by providing phase control unit 500, the phase of Q-phase modulation signal Sb can be delayed in a plurality of stages. Thereby, the phase difference between the modulation signal Sa radiated from the antenna member 218X of the input point 218 on the I-phase side and the modulation signal Sb radiated from the antenna member 219Y of the input point 219 on the Q-phase side is adjusted with high accuracy. And high-speed transmission using IQ orthogonal axes can be realized. Note that the phase control unit 500 can be provided not on the Q phase side but on the I phase side, or on both the I phase and the Q phase.

<8. Sixth Embodiment>
[Example of high-frequency transmission system with phase calibration function]
In the sixth embodiment, the phase calibration is not performed by switching the plurality of antenna members 219a to 219d described above, but the phase difference between the I-phase and Q-phase signals is obtained by using a phase shift element. adjust.

  As illustrated in FIG. 13, the phase control unit 600 includes, for example, a resistor (R), an inductor (L), a capacitor (C), and the like, and is provided between the switch unit 274 and the input point 219. The phase control unit 600 includes, for example, a plurality of stages (n stages). In each stage, n phase shift elements connected in series or in parallel are arranged according to the number of stages. For example, an inductor and a resistor are provided in series at the first stage, and a capacitor and a resistor are provided in parallel at the second stage.

  In the high-frequency transmission system 100H configured as described above, the position rotation amount calculation unit 270 illustrated in FIG. 8 has a phase difference of 90 ° based on the test modulation signal Sat in the calibration mode, for example. Such a phase shift element is selected. Then, the position rotation amount calculation unit 270 generates a switching signal corresponding to the selected phase shift element and supplies it to the switch unit 274. Based on the switching signal supplied from the position rotation amount calculation unit 270, the switch unit 274 switches to the optimal delay element that makes the phase difference between the signals 90 °.

  As described above, according to the present embodiment, by providing phase control unit 600, the phase of Q-phase modulation signal Sb can be delayed in a plurality of stages. Thereby, the phase difference between the modulation signal Sa radiated from the antenna member 218X of the input point 218 on the I-phase side and the modulation signal Sb radiated from the antenna member 219Y of the input point 219 on the Q-phase side is adjusted with high accuracy. And high-speed transmission using IQ orthogonal axes can be realized. The phase control unit 600 can be provided on the I phase side instead of the Q phase side, or can be provided on both the I phase and the Q phase.

<9. Seventh Embodiment>
[Example of high-frequency transmission system with phase calibration function]
In the seventh embodiment, calibration is performed in conjunction between the transmission device 200 and the reception device 300. The high-frequency transmission system 100I according to the present embodiment has the communication function of the high-frequency transmission system 100A shown in FIG. 3, and the reception device 300 also has a transmission function in addition to the reception function. Hereinafter, each of the plurality of antenna members 219a to 219d shown in FIG. 1-No. 4 is referred to.

  As shown in FIG. 14, when the high-frequency transmission system 100I is set to the calibration mode in step S100, the receiving apparatus 300 transmits the test IQ signal from the antenna member at the output point 258 via the waveguide 220. 200.

  Next, in step S102, when the transmission apparatus 200 is set to the calibration mode, the transmission unit 200 sets the switch unit 274 to the antenna member 219a (No. 1) among the antenna members 219a to 219d.

  In step S <b> 104, the antenna member 219 a receives the test IQ signal transmitted from the receiving apparatus 300 and supplies it to the position rotation amount calculation unit 270. Further, the antenna member 218 a receives the test IQ signal transmitted from the receiving device 300 and supplies it to the position rotation amount calculation unit 270. The position rotation amount calculation unit 270 acquires signal point information from each test IQ signal and stores it in a memory unit (not shown).

  After acquiring the signal point information in the antenna member 219a, the receiving apparatus 300 transmits the test IQ signal from the antenna member at the output point 258 to the transmitting apparatus 200 again via the waveguide 220 in step S106. Next, in step S108, the transmitting apparatus 200 switches the switch unit 274 from the antenna member 219a to the antenna member 219b.

  In step S <b> 110, the antenna member 419 b receives the test IQ signal transmitted from the receiving device 300 and supplies it to the position rotation amount calculation unit 270. Further, the antenna member 218 a receives the test IQ signal transmitted from the receiving device 300 and supplies it to the position rotation amount calculation unit 270. The position rotation amount calculation unit 270 acquires signal point information from each test IQ signal and stores it in a memory unit (not shown).

  Such a calibration operation is performed for each of the antenna members 219c and 219d, and signal point information in the test IQ signal received by each of the antenna members 219c and 219d is acquired and stored in the memory unit.

  In step S112, the position rotation amount calculation unit 270 determines that the phase difference between the modulation signals Sa and Sb is 90 ° based on the signal point information in the test IQ signals of the antenna members 219a to 219d stored in the memory unit. The antenna members 219a to 219d are determined. Then, the position rotation amount calculation unit 270 generates a switching signal corresponding to the determined antenna members 219a to 219d and supplies the switching signal to the switch unit 274.

  In step S114, the switch unit 274 switches to the optimum antenna members 219a to 219d in which the phase difference between the modulation signals Sa and Sb is 90 ° based on the switching signal supplied from the position rotation amount calculation unit 270. By such a series of calibration operations, the phase difference between the modulation signals Sa and Sb can be adjusted with high accuracy, and high-speed transmission using the IQ orthogonal axis can be realized. In addition to the signal point information described above, it can be easily realized by error bit information (error correction signal) added to the IQ signal, for example.

<10. Eighth Embodiment>
[Example of high-frequency transmission system with phase calibration function]
In the eighth embodiment, as in the seventh embodiment, calibration is performed in conjunction between the transmission device 200 and the reception device 300. As illustrated in FIG. 15, in step S200, the transmission device 200 of the high-frequency transmission system 100J sets the switch unit 274 to the antenna member 219a (No. 1) among the antenna members 219a to 219d. In step S202, the transmission apparatus 200 transmits a test IQ signal from each of the antenna members 218a and 219a to the reception apparatus 300 via the waveguide 220.

  In step S204, the receiving apparatus 300 receives each of the test IQ signals via the antenna member of the output point 258 shown in FIG. The receiving apparatus 300 is provided with the position rotation amount calculation unit 270 shown in FIG. 8, and the position rotation amount calculation unit 270 obtains signal point information (phase rotation amount of IQ signal) from each test IQ signal. Acquired and stored in a memory unit (not shown).

  Next, in step S206, the transmission apparatus 200 switches the switch unit 274 from the antenna member 219a (No. 1) to the antenna member 219b (No. 2). In step S208, a test IQ signal is transmitted from each of the antenna members 218a and 219a to the receiving apparatus 300 via the waveguide 220.

  In step S <b> 210, receiving apparatus 300 receives each of the test IQ signals via the antenna member at output point 258. The position rotation amount calculation unit 270 acquires signal point information (the rotation amount of the phase of the IQ signal) from each test IQ signal and stores it in a memory unit (not shown).

  Such a calibration operation is also performed on each of the antenna members 219c and 219d, and signal point information in the test IQ signal received by each of the antenna members 219c and 219d is acquired and stored in the memory unit.

  In step S212, the receiving apparatus 300 sets the phase difference between the modulation signals Sa and Sb to 90 ° based on the signal point information in the test IQ signals of the antenna members 219a to 219d stored in the memory unit. The antenna members 219a to 219d are determined.

  When the optimal antenna members 219a to 219d are determined in step S214, the reception device 300 feeds back a switching signal based on the antenna members 219a to 219d determined by the position rotation amount calculation unit 270 to the transmission device 200. The transmission apparatus 200 receives the switching signal via any one of the antenna member 218a at the input point 218 and the antenna members 219a to 219d at the input point 219.

  In step S216, based on the switching signal supplied from the reception device 300, the transmission device 200 switches to the optimal antenna members 219a to 219d in which the phase difference between the modulation signals Sa and Sb is 90 °. By such a series of calibration operations, the phase difference between the modulation signals Sa and Sb can be adjusted with high accuracy, and high-speed transmission using the IQ orthogonal axis can be realized.

  It should be noted that the technical scope of the present invention is not limited to the above-described embodiments, and includes those in which various modifications are made to the above-described embodiments without departing from the spirit of the present invention.

  For example, in the above-described embodiments, BPSK modulation has been described. However, the present invention is not limited to this, and the present invention can also be applied to a QPSK (4-phase PSK) modulation system, an 8-phase PSK modulation system, and the like. . Further, it is obvious that an amplitude control device is added before and after the above-described BPSK modulators 212 and 213 so that it can be applied to a QAM system in which a digital code is superimposed also in the amplitude direction. According to the QAM transmission, the orthogonality of the IQ axis has a great influence on the transmission characteristics, so that the high-precision IQ orthogonal axis transmission can be performed by utilizing the characteristics of the high-frequency transmission system.

100A, 100B, 100C, 100D, 100E, 100F, 100G, 100H, 00I, 100J ... high frequency transmission system, 200 ... transmission device, 212, 213 ... BPSK modulator, 214, 215 ... mixer , 216, 256... Carrier wave signal generation unit, 218, 219... Input point, 218 a, 218 X, 219 a to 219 d, 219 Y... Antenna member, 220. , 252 ... 253 ... BPSK modulator, 254,255 ... mixer, 258,260 ... output point, 270 ... position rotation amount calculator, 280 ... frequency analyzer, 370 ... Amplitude value measuring unit, 376... Switch unit, 378... Amplitude value control unit, 300... Receiving device, 400, 500, 600. Control unit

Claims (14)

A first transmitter that modulates a carrier signal of a predetermined frequency based on an input first signal and outputs a first transmission signal;
A second transmission unit that modulates a carrier signal of a predetermined frequency based on an input second signal and outputs a second transmission signal;
A first input point for inputting the first transmission signal output from the first transmission unit to the waveguide, and a second input signal output from the second transmission unit to the waveguide. And a second input point that is shifted by a distance that gives a predetermined phase difference between the first transmission signal and the second transmission signal.   2. The transmission device according to claim 1, wherein when N is an integer and λ is the wavelength of the carrier signal, the first input point and the second input point are shifted by (1/4 + N) λ wavelengths.   The transmission apparatus according to claim 2, wherein the first and second transmission signals output by the first and second transmission units are millimeter band signals.   The transmission device according to claim 3, wherein a dielectric having a predetermined dielectric constant is used for the waveguide.   The transmission apparatus according to claim 1, wherein a dipole antenna or a slit antenna for transmitting the first and second transmission signals is provided at each of the first and second input points. The second input point is provided with a plurality of antennas for receiving the first transmission signal,
The phase of the carrier wave signal is compared with the phase of the first transmission signal transmitted from the first input point and received by the second input point, and the first calculated by the comparison result is compared. The transmission apparatus according to claim 1, further comprising a control unit that selects any one of the plurality of antennas based on a phase shift amount of a transmission signal. The first transmitter is provided with a first amplifier for amplifying the first signal,
The second transmission unit includes a second amplification unit that amplifies the second signal,
The amplitude value of the first transmission signal transmitted from the first input point and received by the second input point is measured, and the amplitude value output from the second amplification unit based on the measurement result The transmission apparatus according to claim 6, further comprising an amplitude value measurement unit that adjusts an amplitude value of the second signal. Comparing the phase of the carrier wave signal with the phase of the first transmission signal transmitted from the first input point and received by the second input point, and based on the comparison result, A control unit for calculating a phase shift amount of the transmission signal;
The transmission apparatus according to claim 1, further comprising a phase adjustment unit that adjusts a phase of at least one of the first and second transmission signals based on the shift amount calculated by the control unit. The phase adjusting unit is
The transmission device according to claim 8, wherein the transmission device is formed of a substance that is provided at one end of the waveguide and whose dielectric constant changes due to electric, optical, magnetic, or thermal energy. The phase adjusting unit is
The transmission device according to claim 8, wherein the transmission device is provided outside the waveguide and includes at least one of a delay element, a resistor, an inductor, and a capacitor. The signal processing is phase modulation for modulating the phase of the first and second signals;
The transmission apparatus according to claim 1, wherein amplitude modulation of the first and second signals is performed in addition to the phase modulation. A first transmitter that modulates a carrier signal of a predetermined frequency based on an input first signal and outputs a first transmission signal;
A transmission device having a second transmission unit that modulates a carrier wave signal of a predetermined frequency based on an input second signal and outputs a second transmission signal;
A waveguide to which the first transmission signal output from the first transmission unit and the second transmission signal output from the second transmission unit are input;
A receiving unit that receives the first and second transmission signals transmitted through the waveguide, demodulates the first and second transmission signals based on a carrier signal of a predetermined frequency, and obtains a reception signal; And a receiving device having
A first input point for inputting the first transmission signal output from the first transmission unit to the waveguide and a second transmission signal output from the second transmission unit to the waveguide. A communication system in which an input second input point is shifted by a distance that gives a predetermined phase difference between the first transmission signal and the second transmission signal. The transmitter is
Transmitting the first and second transmission signals output from the first and second transmission units to the reception device via the waveguide;
The receiving device is:
The first input point and the second input point are received based on the received first and second transmission signals, the first and second transmission signals transmitted from the transmission device are received. The communication system according to claim 12, wherein it is determined whether or not a predetermined phase difference is shifted, and the determination result is fed back to the transmission device. The receiving device is:
Transmitting the first and second transmission signals to the transmission device via the waveguide;
The transmitter is
The first input point and the second input point are received based on the received first and second transmission signals, the first and second transmission signals transmitted from the receiving device are received. The communication system according to claim 12, wherein it is determined whether or not a predetermined phase difference is shifted.

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