JP2010200395A - Driver and apparatus for motor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To realize small size and cost reduction by enabling the suppression of vibration and noise based on a voltage or current waveform distortion due to dead time and a torque ripple due to the current distortion. <P>SOLUTION: AC power is rectified via a small capacity reactor 4, and is smoothed via a small capacity capacitor 6 so as to get varying DC power. An inverter control means 8 includes a voltage command processor 13 which computes a voltage command from a frequency command, the output of a zero cross detector 9, the output of a bus voltage detector 10, and the output of a current detector 11, a dead time compensator 14 which generates an amount of compensation for compensating an error voltage between a voltage command and an output voltage caused by the dead time for prevention of arm short-circuit of a switching means from the output of a bus voltage detector 10 and the output of the current detector 11, a voltage command corrector 15 which corrects the voltage command of the voltage command processor 13 based on the output of the dead time compensator 14, and a PWM signal generator 16 which generates a PWM signal obtained from the voltage command corrector 15. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention relates to an electric motor drive device used for driving an electric motor, and devices such as a refrigeration air conditioner, a heat pump water heater, an electric vacuum cleaner, a hand dryer, and a ventilation air blower using the electric motor drive device. .

Conventionally, in order to prevent an arm short circuit of a switching element of a three-phase voltage type PWM inverter used for driving a motor of a refrigeration air conditioner, a short-circuit prevention time (dead time) including a turn-off time of the switching element is inserted. In order to compensate for the error voltage between the inverter PWM voltage command and the output voltage due to the short-circuit prevention time, the polarity of the square-wave compensation voltage added to the PWM voltage command is set to the output current of the inverter. In general, a method of changing in synchronization with the time point of passing through the zero cross point is performed. However, according to the conventional dead time compensation method, since the polarity of the compensation voltage is changed stepwise at the zero crossing point of the output current of the inverter, the ripple of the output current near the zero crossing point increases. There has been a problem that uneven rotation due to ripples becomes large.
In order to solve this problem, for example, a comparator is used to compare the inverter output current with the reference value for each phase, and the time width corresponding to the magnitude of the comparison result (error) near the zero cross point of the output current The compensation amount calculation unit calculates the dead time compensation amount (compensation amount amplitude) based on the PWM frequency, dead time, and DC power supply voltage, and multiplies the comparator output and the compensation amount calculation unit output. The multiplier output is integrated by an integrator to generate a trapezoidal waveform compensation amount signal. The compensation amount signal is added to the command value from the PWM voltage command calculator, and the PWM pulse calculator A dead time compensation method is disclosed in which a dead time after PWM calculation is added to the addition result to output a drive signal to the inverter main circuit. With this configuration, the polarity change of the compensation voltage that is performed as the inverter output current passes through the zero cross point can be performed with a predetermined gradient from the change start time before the zero cross point of the output current. Thus, the problem of an increase in the output current ripple at the zero crossing point can be solved (see, for example, Patent Document 1).
Further, a technique is disclosed in which the dead time compensation voltage is an AC component of an odd-order frequency in the electric frequency of the inverter determined from the rotation speed of the motor. By using this technology, in an inverter composed of a small-capacity reactor and a small-capacitance capacitor, not only can voltage and current distortion due to inverter output voltage error due to dead time be eliminated, but it is also defined by IEC, JIS, etc. It can be adapted to a harmonic standard or the like (see, for example, Patent Document 2).
A single-phase diode full-wave rectifier circuit using a single-phase AC power supply as input, a small-capacity smoothing capacitor of about 1/100 of a smoothing capacitor for a conventional diode full-wave rectifier circuit connected thereto, a control PWM inverter, Control of motor drive inverter that realizes improvement of input power factor and waveform of diode full-wave rectifier circuit by controlling motor torque at twice the frequency of power supply in advance by control circuit composed of motor An apparatus technology is disclosed (for example, see Patent Document 3).

Japanese Patent No. 3738883 (2nd to 4th pages, FIG. 1 and FIG. 2) JP 2007-104813 A (pages 8 to 9, FIG. 6) JP 2002-51589 A (pages 4 to 5)

However, since the conventional dead time compensation device disclosed in Patent Document 1 is configured as described above, harmonics remain in the dead time compensation voltage of the trapezoidal waveform, and the specific order harmonics are problematic as noise. There was.
In addition, when low noise driving is required, the noise level near the PWM frequency is high, and there are significant restrictions in use particularly when the PWM frequency must be set in the audible range. In addition, since the gradient of the compensation voltage near the zero cross changes depending on the magnitude of the output current, the ratio of the specific order harmonics changes, and the driving range is limited for motor driving where noise is a problem.
In addition, since a comparator is used, the tolerance to external noise is weak, and since an integrator is included in the calculation of the compensation voltage, there are problems such as overflow, which makes it difficult to implement with a simple system. .
Further, in the technology disclosed in Patent Document 2, since the bus voltage pulsates in an inverter constructed with a small-capacity reactor or a small-capacitance capacitor, when dead time compensation is performed when the bus voltage is near zero, The bus voltage may be boosted by the regenerative voltage, leading to breakdown of the inverter's breakdown voltage, and the influence of voltage error due to dead time is determined by the polarity of the current. Distortion of the voltage / current waveform due to the occurrence of noise has caused problems such as generation of noise and unstable control.
Further, in the technique disclosed in Patent Document 3, when the inverter is operated at a low speed or a high carrier frequency, the voltage command value and the actual inverter are caused by the influence of dead time for preventing a short circuit of the switching element of the inverter. As a result, an error occurs in the output voltage, resulting in torque ripple due to current distortion, and noise and vibration from the motor.

  The present invention has been made to solve the above-described problems, and suppresses voltage and current waveform distortion due to dead time even in an inverter constituted by a small-capacity reactor or a small-capacitance capacitor. An object of the present invention is to obtain a motor drive device that can suppress vibration and noise caused by torque ripple due to current distortion as well as a small size and low cost, and a device using the motor drive device.

An electric motor drive device according to the present invention includes:
Rectifying means for full-wave rectification of an AC power supply via a reactor;
A capacitor for smoothing the output of the rectifying means and outputting a bus voltage pulsating at a frequency approximately twice that of the AC power supply;
An inverter having switching means for applying a voltage to the electric motor using the bus voltage output from the capacitor as a power source;
Inverter control means for controlling the inverter;
Zero-cross detection means for detecting zero-cross of the AC power supply;
A bus voltage detecting means for detecting a voltage of the bus power supply;
Current detecting means for detecting a current of two or more phases flowing through the electric motor;
Rotation phase detection means for detecting the rotation phase of the electric motor,
The inverter control means calculates a voltage command based on a frequency command, an output of the zero cross detection means, an output of the bus voltage detection means, an output of the current detection means, and an output of the rotation phase detection means. When,
Based on the output of the bus voltage detection means and the output of the current detection means, a voltage command from the voltage command calculation means and an output voltage from the inverter due to a dead time for preventing an arm short circuit of the switching means Dead time compensation means for generating a compensation amount for compensating the error voltage between,
Voltage command correcting means for correcting the output of the voltage command calculating means based on the output of the dead time compensating means;
PWM signal generating means for generating a PWM signal obtained by the voltage command correcting means.

  According to the motor drive device of the present invention, voltage and current waveform distortion due to dead time can be suppressed even in a small and low-cost inverter constituted by a small-capacity reactor or a small-capacitance capacitor. In addition to reducing unpleasant noise due to torque ripple, it is possible to suppress the influence of regenerative voltage by suppressing dead time compensation when the bus voltage is near zero, and to obtain a highly reliable motor drive device Can do.

It is a figure which shows the structure of the drive device of the electric motor which concerns on Embodiment 1. FIG. It is a figure showing the conventional dead time compensation. It is a figure which shows operation | movement of a PWM signal generation means. FIG. 6 is a diagram illustrating an operation of dead time compensation in the first embodiment. FIG. 5 is a diagram illustrating a configuration of a drive device for an electric motor according to a second embodiment. It is a figure which shows the coordinate relationship with respect to rotation of an electric motor. FIG. 4 is a diagram showing a configuration of dead time compensation means 14. It is a figure which shows the phase difference of the phase of a dq axis coordinate, and the phase of an electric current vector. It is a figure which shows the relationship between the electric current of UVW phase with respect to rotation phase (theta) of an electric motor, and a dead time compensation amount. FIG. 6 is a diagram showing another configuration of the dead time compensation means 14. It is a figure which shows the general voltage command correction value of dead time compensation. FIG. 10 is a diagram illustrating an operation of dead time compensation in the second embodiment.

Embodiment 1 FIG.
FIG. 1 is a diagram illustrating a drive device for an electric motor according to a first embodiment of the present invention. As shown in FIG. 1, the electric motor system includes an AC power source 1, an electric motor driving device 2 according to the present invention, and an electric motor 3. The motor driving device 2 drives the motor 3 by supplying power from the AC power source 1.
In addition, the electric motor drive device 2 has a sufficiently smaller capacity reactor (hereinafter simply referred to as “reactor”) 4, a rectifying means 5, and a generally used condenser than a generally used reactor. A small-capacitance capacitor (hereinafter simply referred to as a capacitor) 6, an inverter 7, inverter control means 8, zero cross detection means 9, bus voltage detection means 10, current detection means 11, and magnetic pole position detection means 12 are provided.

  The AC power supply 1 is full-wave rectified by the rectifying means 5 via the reactor 4 and smoothed by the capacitor 6 to generate a bus voltage that is a power supply for the inverter 7. Here, since the power source of the inverter 7 is considerably smaller than that of the capacitor 6 in general, a waveform obtained by rectifying the AC power source 1 substantially full wave (a voltage that fluctuates at a frequency approximately twice that of the AC power source). Waveform or current waveform).

  The inverter 7 has switching elements such as IGBTs and MOSFETs as switching means, and each switching element includes a free-wheeling diode connected in parallel, and is an inverter control means constituted by an electronic circuit such as a microcomputer, DSP, or logic circuit. 8 controls the inverter 7 to supply a voltage to the motor 3.

  The zero cross detection means 9 detects the zero cross of the AC power supply 1 and outputs a zero cross signal ZC to the inverter control means 8. As a method of detecting the zero crossing, for example, when the AC power source 1 is switched from negative to positive, it is HIGH (5 V), and when it is switched from positive to negative, it is LOW (0 V). It can be easily detected by importing into. Needless to say, there is no problem even if HIGH and LOW are reversed, and the voltage is not limited to 0 V or 5 V, and a zero cross can be detected.

  The bus voltage detection means 10 detects the bus voltage Vdc, which is a voltage smoothed by the capacitor 6, and outputs it to the inverter control means 8. As a method for detecting the bus voltage Vdc, the voltage is reduced by using a voltage dividing resistor or the like and taken into a microcomputer or the like constituting the inverter control means 8 and converted in consideration of the reduced voltage inside the microcomputer or the like. Thus, the bus voltage can be easily detected.

The current detection unit 11 detects at least two or more of the U-phase current Iu, the V-phase current Iv, and the W-phase current Iw flowing into the electric motor 3 and outputs them to the inverter control unit 8. Current detection methods include ACCT and DCCT, which are general current detection elements, a method of detecting a current flowing on the power source side of the inverter 7 to restore a three-phase current, and a lower switching element of the inverter 7 There are methods for detecting a flowing current, and any of them can be detected by taking it into a microcomputer or the like constituting the inverter control means 8. Here, when only two phases of Iu and Iw are detected, the current Iv not detected is from Kirchhoff's law Iu + Iv + Iw = 0.

で求めることができる。

Can be obtained.

  The magnetic pole position detection means 12 detects the rotational phase θ that is the magnetic pole position of the rotor of the electric motor 3 and outputs it to the inverter control means 8. The magnetic pole position detection method can be detected by a rotary encoder, a resolver, a hall sensor, or the like. Although not shown, it is also possible to detect the magnetic pole position based on the output of the current detection means 11 and the voltage-current equation obtained from the characteristics of the motor, and it is widely used as position sensorless control. It is also possible to use a method. The magnetic pole position detection means 12 constitutes a rotation phase detection means.

  Next, the inverter control means 8 will be described. The inverter control unit 8 includes a voltage command calculation unit 13, a dead time compensation unit 14, a voltage command correction unit 15, and a PWM signal generation unit 16.

  The voltage command calculation means 13 is a frequency command inputted from the outside, a magnetic pole position (rotation phase) θ as an output of the magnetic pole position detection means, a zero cross signal ZC as an output of the zero cross detection means, and a bus voltage as an output of the bus voltage detection means. With Vdc and UVW phase currents Iu, Iv and Iw as outputs of current detection means as inputs, VW *, Vv * and Vw * as UVW phase voltage commands are output. As a method for obtaining the UVW phase voltage commands Vu *, Vv *, and Vw *, for example, there is a method shown in Patent Document 3.

  The dead time compensation means 14 receives the bus voltage Vdc, which is the output of the bus voltage detection means, and the UVW phase currents Iu, Iv, Iw, which are the outputs of the current detection means, as inputs, and UVW voltage command correction values ΔVu, ΔVv, ΔVw is obtained.

  Here, a method of obtaining the UVW phase voltage command correction values ΔVu, ΔVv, ΔVw will be described. The error ΔV between the voltage command value based on the dead time and the actually output voltage varies depending on the polarity of the flowing current, but as an absolute value, using the dead time Td, the carrier frequency fc, and the bus voltage Vdc, 1).

電流の極性を考慮すると、例えばＵ相電圧指令補正値ΔＶｕはＵ相電流Ｉｕの極性を用いて次式（２）で得られる。

Considering the polarity of the current, for example, the U-phase voltage command correction value ΔVu is obtained by the following equation (2) using the polarity of the U-phase current Iu.

ただし、sign関数は次式（３）の通りである。

However, the sign function is as shown in the following equation (3).

ここで、ｘはＩｕ、Ｉｖ、Ｉｗのいずれかを指す。
同様にＶ相、Ｗ相の電圧指令補正値ΔＶｖ、ΔＶｗについては次式（４）及び（５）となる。

Here, x indicates any of Iu, Iv, and Iw.
Similarly, the V-phase and W-phase voltage command correction values ΔVv and ΔVw are expressed by the following equations (4) and (5).

As described above, the voltage command correction values ΔVu, ΔVv, and ΔVw can be obtained.
For simplicity of explanation, voltage command correction values ΔVu, ΔVv, and ΔVw when the bus voltage Vdc is constant are shown in FIG. (Note that, as will be described later, in the present invention, the bus voltage Vdc pulsates instead of being constant, so that the voltage command correction values ΔVu, ΔVv, ΔVw handled in the present invention are different from those in FIG. 2.)

  Next, the voltage command correction means 15 receives the corrected voltage command with the outputs of Vu *, Vv *, Vw * as outputs of the voltage command calculation means 13 and ΔVu, ΔVv, ΔVw as outputs of the dead time compensation means 14 as inputs. Values Vu **, Vv **, and Vw ** are created and output to the PWM signal generating means 16.

  The corrected voltage command values Vu **, Vv **, and Vw ** can be obtained by using the following equations (6) to (8).

  The PWM signal generating means 16 outputs the PWM signals UP, UN, VP, VN, WP, WN for driving the switching elements of the inverter 7 with the corrected voltage command values Vu **, Vv **, Vw ** as inputs. To do.

  FIG. 3 is a diagram showing the operation of the PWM signal generating means 16 in the U phase. The corrected U-phase voltage command value Vu ** is compared with the carrier of the frequency fc. When the corrected U-phase voltage command value Vu ** is larger than the carrier, it is HIGH, and when it is small, it is LOW. The arm PWM signal UP is output. For the U-phase lower arm PWM signal, a signal having a logic opposite to that of UP is output. In addition, although it is necessary to reverse HIGH and LOW depending on the kind of inverter 7, it cannot be overemphasized that basic operation does not change.

  Furthermore, to prevent the UP and UN from being turned on at the same time, the output of the UP or UN PWM signal is reduced by the dead time to prevent the inverter 7 from being short-circuited. An error occurs in the voltage output to the motor 3, and the current applied by the voltage applied to the motor 3 is distorted, so that the torque ripple is generated, and there is a problem that noise and vibration are generated due to the torque ripple. There are problems such as unstable control. Therefore, as described above, the above-mentioned problem can be solved by compensating the voltage command by the dead time compensation means 14 or the like.

  However, as described above, the bus voltage Vdc has a substantially full-wave rectified waveform (pulsation waveform). Therefore, as can be seen from Equation (1), the voltage command correction value ΔV is the same as the pulsation frequency of the bus voltage Vdc or is AC. Since correction as shown in FIG. 4 can be performed so as to cause pulsation at a frequency that is approximately twice the frequency of the power supply 1, it can be made substantially the same as the pulsating bus voltage Vdc, which is more accurate than in the past. Time compensation is possible, and not only controllability is improved, but also vibration and noise due to voltage error due to dead time can be reduced, and a highly reliable motor drive device can be obtained.

  However, when the rotational speed is small and the phase current is small, when performing dead time compensation using Equation (2), Equation (4), and Equation (5), the vicinity of the zero cross of the current of each phase is positive or negative. Since the determination is difficult and the sign of each voltage command correction value is not determined, accurate dead time compensation cannot be performed, and control may become unstable near the zero cross.

  Therefore, the dead time compensation means 14 has a ratio K (x) corresponding to the current value in the voltage command correction value (where x is any one of Iu, Iv, and Iw) as shown in equations (9) to (11). By multiplying by (), stable operation is possible even near the zero cross where the current is small.

Ｋ（ｘ）については、例えば電流値が０Ａから最大定格の２０％程度までは０〜１００％と段階的に変化させ、それ以上の電流値になったら１００％の電圧指令補正値にするなどの方式や、０Ａから最大定格の２０％程度までは０％で、それ以上は１００％にする方法などが挙げられる。また、電流値以外にも回転数で電圧指令補正値の割合を変化させても何ら問題ないことは言うまでもない。この回転数で電圧指令補正値の割合を変化させる場合も電流で制御する方法と同じ方法を適用することができる。

For K (x), for example, when the current value is from 0 A to about 20% of the maximum rating, it is changed in steps from 0 to 100%, and when the current value exceeds that, the voltage command correction value is set to 100%. And a method of 0% from 0A to about 20% of the maximum rating and 100% beyond that. Needless to say, there is no problem even if the ratio of the voltage command correction value is changed by the rotation speed other than the current value. Even when the ratio of the voltage command correction value is changed at this rotational speed, the same method as the method of controlling by the current can be applied.

  The above dead time compensation means can be realized by a microcomputer, a DSP (Digital Signal Processor), an electronic circuit, or the like.

Embodiment 2. FIG.
In the first embodiment, the method for compensating the dead time according to the polarity of the current has been described.
In the second embodiment of the present invention, a method for accurately detecting the zero crossing of the current of each phase and performing accurate dead time compensation will be described.
FIG. 5 is a diagram showing the configuration of the drive device for the electric motor according to Embodiment 2 of the present invention.
Since the second embodiment is the same as the first embodiment except that the rotational phase θ is input to the dead time compensation means 14, the same parts as those of the first embodiment are denoted by the same reference numerals and the description thereof is omitted.

  Here, FIG. 6 is an analysis model diagram of the electric motor 3. FIG. 6 shows U-phase, V-phase, and W-phase armature winding fixed axes. Reference numeral 17 denotes a permanent magnet constituting the rotor of the electric motor 3. In a rotating coordinate system that rotates at the same speed as the magnetic flux produced by the permanent magnet 17, the direction of the magnetic flux produced by the permanent magnet 17 is taken as the d-axis, and the estimated control axis corresponding to the d-axis is taken as the γ-axis. Although not shown, the q axis is taken as a phase advanced by 90 degrees in electrical angle from the d axis, and the estimated δ axis is taken as phase advanced by 90 degrees in electrical angle from the γ axis. The coordinate axis of the rotating coordinate system in which the d axis and the q axis are selected as the coordinate axes is referred to as the dq axis. The rotating coordinate system for control by the inverter is a coordinate system in which the γ axis and the δ axis are selected as coordinate axes, and the coordinate axes are called γδ axes.

  The dq axis is rotating, and the rotation speed of the electric motor 3 that is the rotation speed is referred to as an electric motor rotation speed ω. The γδ axis is also rotating, and the rotation speed is called an inverter rotation speed ω1. In addition, in the dq axis rotating at a certain moment, the phase of the d axis is represented by the motor rotation phase θ with reference to the U-phase armature winding fixed axis. Similarly, in the γδ axis that is rotating at a certain moment, the phase of the γ axis is expressed by the inverter rotation phase θ1 with reference to the U-phase armature winding fixed axis. Then, the axial error Δθ between the d-axis and the γ-axis is

で表される。

It is represented by

  When a sensor for detecting the magnetic pole position of the rotor of the electric motor 3 is provided, it is possible to discuss on the dq axis because an accurate magnetic pole position can be detected. However, when there is no sensor for detecting the magnetic pole position, a method of estimating the magnetic pole position as the γδ axis by the current flowing through the electric motor 3 is generally used, but an axial error Δθ occurs in the dq axis and the γδ axis. However, in actual operation, the operation may be performed where the axis error Δθ is small, and the dq axis and the γδ axis can be regarded as the same. For this reason, the following description will be made assuming that the dq axis is used unless otherwise specified. Needless to say, the γδ axis can be realized with the same configuration. In this case, it goes without saying that the motor rotational speed ω may be replaced with the inverter rotational speed ω1 and the motor rotational phase θ may be replaced with the inverter rotational phase θ1.

  FIG. 7 is a block diagram showing the operation of the dead time compensation means 14. The dead time compensation unit 14 includes a coordinate conversion unit 18, a phase difference detection unit 19, and a compensation amount calculation unit 20.

  The coordinate conversion unit 18 uses the rotational phase θ obtained from the magnetic pole position detection unit 12 to calculate the current (Iu, Iv, Iw) output from the current detection unit 11 using the equation (12). Both of the function of converting current to two-phase current and the function of converting the two-phase current obtained by the conversion into rotational coordinates are converted into d-axis current Id and q-axis current Iq and output.

  Here, a method of obtaining the phase difference Δθi between the rotation phase θ, which is the phase of the magnetic pole position, and the phase of the phase current by the phase difference detection means 19 will be described. As shown in FIG. 8, the phase difference θi between the current vector I and the d axis that is the magnetic pole axis can be obtained by the following equation (13) using Id and Iq.

  Therefore, since the rotational phase θ and the current phase difference Δθi are obtained, the compensation amount calculation means 20 uses the equations (14) to (16), so that the voltage command correction with high stability can be achieved even near the zero cross of each phase current. Values ΔVu, ΔVv, ΔVw can be obtained. However, the current phase θi is

とする。

And

  FIG. 9 specifically shows the above relationship. For example, when θi is 0 °, the dead time compensation amount of each phase of UVW is ΔVu: positive, ΔVv: Negative, ΔVw: negative is obtained.

  The current phase difference Δθi uses not only the currents Id and Iq on the dq axis but also the currents Iα and Iβ of the fixed coordinate system (αβ coordinate system) obtained from the equation (17) which is a three-phase to two-phase conversion means. It is also possible to obtain it using equation (18).

  The sign of the current phase difference Δθi and rotation phase θ is reversed depending on the direction of rotation of the motor 3 and how the rotating coordinate system is used, but dead time compensation is achieved in the same way by using different signs depending on the system. Needless to say, you can. In addition, since the current phase difference Δθi may vary depending on the system, in this case, the varying component may be suppressed using an LPF (Low Pass Filter).

  By determining whether the voltage command correction value is positive or negative using the current phase difference Δθi as described above, it is possible to accurately perform dead time compensation near the zero cross where it is difficult to determine whether the phase current is positive or negative. Not only can a high motor drive system be obtained, but also noise and vibration generated by torque ripple due to voltage errors due to dead time can be reduced, and a low noise, low vibration motor drive device can be obtained. Become.

  Here, in the case of the dead time compensating means 14 shown in FIG. 7, the voltage command correction values ΔVu, ΔVv, ΔVw are waveforms including a plurality of higher-order harmonic components, as can be seen from FIGS. These higher-order harmonic components may be generated as sound from the motor 3, and harmonic components are generated in the current flowing in the AC power source 1, which can meet the harmonic standards defined in JIS and IEC. There is a risk of disappearing.

  Therefore, a method of performing dead time compensation that can solve the above problem by reducing the influence of higher-order harmonic components by using only the fundamental frequency of the voltage command correction value will be described. As a configuration, a compensation amount characteristic order component calculating unit 21 is provided instead of the compensation amount calculating unit 20 as shown in FIG. In addition to the function of the compensation amount calculation unit 20, the compensation amount characteristic order component calculation unit 21 has a function of performing dead time compensation using only the fundamental frequency of the voltage command correction value.

  As shown in FIG. 2, in general dead time compensation, if the bus voltage Vdc is constant, the voltage command correction value becomes a rectangular wave. Therefore, a method of extracting a fundamental wave component based on the voltage command correction value will be described. Do.

  FIG. 11 shows the U-phase voltage command correction value ΔVu when rotating at θi. When this waveform is expanded into an angular frequency component by Fourier series expansion, the following equation (19) is obtained.

  Similarly, ΔVv and ΔVw having a phase difference of ± 120 ° compared to ΔVu are expressed by the following equations (20) and (21).

  Therefore, in the equations (19) to (21), when only the fundamental wave component is extracted by substituting n = 1, the voltage command correction values become equations (22) to (24), respectively, and are obtained in the obtained current phase θi. By performing the dead time compensation based on the dead time compensation, it is possible to perform the dead time compensation with less harmonic components and low noise.

  Here, when the small-capacitance capacitor 6 is used, the bus voltage Vdc pulsates at about twice the frequency of the AC power supply 1, but the compensation amount characteristic order component calculation means 21 uses the detected bus voltage Vdc in the equation (22). ) To (24). As a result, as shown in FIG. 12, a voltage command correction value substantially similar to the current is obtained, and a motor drive device that has less harmonic components and generates less noise than the rectangular wave dead time compensation is obtained. .

Further, in addition to the fundamental wave component, the frequency is 5 times (frequency obtained by substituting n = 3 in equations (18) to (20)) or 7 times (in equations (18) to (20), n = 4) (frequency obtained by substituting 4) (excluding frequencies obtained by substituting n = 2 or n = 5 in the equations (18) to (20)). By performing the dead time compensation by adding an odd multiple of the voltage command correction value component to the fundamental wave component, it is possible to obtain a motor drive device with a smaller voltage error and high reliability.
The reason for excluding the frequency that is a multiple of 3 is as follows. The UVW phase voltages are different from each other in phase by 2π / 3, and the harmonics of multiples of 3 of the UVW phase voltage are also different in multiples of 2π because the phase shift is also a multiple of 3. As a result, the phase of the UWW phase voltage becomes the same. Since the phase is the same and the amplitude is the same, all the harmonics of multiples of 3 of the UVW phase are the same signal, and the line voltage is zero. Therefore, the frequency components of multiples of 3 are excluded because they are meaningless and do not contribute to dead time compensation.

  However, if a small-capacitance capacitor 6 is used, if dead time compensation is performed in a section where the bus voltage is substantially zero, unnecessary PWM may be output, and the rotational energy of the motor may return to the inverter. In addition to unstable control, the inverter may be destroyed by regeneration due to rotational energy, so the above problem can be solved by taking measures such as not performing dead time compensation near the bus voltage is zero. Can be avoided. In this case, the zero cross of the bus voltage may be determined based on the output of the zero cross detection means 9.

In addition, when the small-capacitance capacitor 6 is used, the bus voltage pulsates at a frequency approximately twice that of the AC power supply 1, so that the peak value of the bus voltage is detected when obtaining the voltage command correction value. A value that maximizes the command correction value may be stored, and the voltage command correction value may be controlled to pulsate at a frequency that is approximately twice that of the AC power source 1 based on the output of the zero cross detection means 9.
Thus, the corrected voltage command values Vu **, Vv **, and Vw ** corrected by the voltage command correction values ΔVu, ΔVv, and ΔVw can be made substantially the same as the pulsating bus voltage Vdc.

Embodiment 3 FIG.
Examples of the use of the motor drive device described in the first and second embodiments include a refrigeration air conditioner such as a refrigerator and an air conditioner, a heat pump water heater, a vacuum cleaner, and a hand dryer.

  Refrigeration air conditioners and heat pump water heaters typified by refrigerators and air conditioners are premised on long-term use exceeding 10 years, and the electrolytic capacitor 6 that is a life component is an obstacle to long life. Therefore, it is possible to extend the service life by using the motor drive device using the small-capacitance capacitor 6 of the present invention.

  Further, by using the dead time compensation according to the present invention for driving an outdoor fan motor of an air conditioner or a heat pump water heater, noise can be reduced.

  Furthermore, by applying the present invention to a vacuum cleaner, it is possible to reduce the size of reactors and electrolytic capacitors, which are large parts that have been indispensable when driving an inverter, and are light in weight and user-friendly It is possible to provide a vacuum cleaner that can reduce the above.

When the motor is driven by an inverter, the maximum rotation speed of the motor is 60000 rpm / (number of poles / 2) or more. For example, the size of the blades and motors for obtaining the same air volume and static pressure by driving the motor at a rotation speed exceeding 60000 rpm for a 2-pole motor with an inverter and exceeding 30000 rpm for a 4-pole motor. Therefore, a small and light vacuum cleaner can be obtained. However, in order to stably drive the electric motor, it is desirable to set the rotation speed at which the control is performed at least 10 times in one electrical angle cycle as the upper limit.

  Furthermore, by applying the present invention to a hand dryer, a reactor or an electrolytic capacitor can be replaced with a small-capacity reactor or capacitor, so that the size and weight can be reduced. Moreover, since a large air volume and a high static pressure can be obtained by rotating at high speed with an inverter, a structural design with a higher degree of freedom is possible.

Furthermore, by applying the present invention to a ventilation fan such as a ventilation fan or a fan, the reactor or electrolytic capacitor can be replaced with a small-capacity reactor or capacitor, so that the size and weight can be reduced. In addition, by applying the dead time compensation of the present invention, torque ripple due to output voltage error reduction is reduced, so that noise can be reduced.
A permanent magnet type brushless DC motor can be applied as the electric motor.

  DESCRIPTION OF SYMBOLS 1 AC power supply, 2 Motor drive device, 3 Electric motor, 4 Reactor, 5 Rectification means, 6 Condenser, 7 Inverter, 8 Inverter control means, 9 Zero cross detection means, 10 Bus voltage detection means, 11 Current detection means, 12 Magnetic pole position Detection means, 13 Voltage command calculation means, 14 Dead time compensation means, 15 Voltage command correction means, 16 PWM signal generation means, 17 Permanent magnet, 18 Coordinate conversion means, 19 Phase difference detection means, 20 Compensation amount calculation means, 21 Compensation Quantity specific order component calculation means.

Claims (19)

Rectifying means for full-wave rectification of an AC power supply via a reactor;
A capacitor for smoothing the output of the rectifying means and outputting a bus voltage pulsating at a frequency approximately twice that of the AC power supply;
An inverter having switching means for applying a voltage to the electric motor using the bus voltage output from the capacitor as a power source;
Inverter control means for controlling the inverter;
Zero-cross detection means for detecting zero-cross of the AC power supply;
A bus voltage detecting means for detecting a voltage of the bus power supply;
Current detecting means for detecting a current of two or more phases flowing through the electric motor;
Rotation phase detection means for detecting the rotation phase of the electric motor,
The inverter control means calculates a voltage command based on a frequency command, an output of the zero cross detection means, an output of the bus voltage detection means, an output of the current detection means, and an output of the rotation phase detection means. When,
Based on the output of the bus voltage detection means and the output of the current detection means, a voltage command from the voltage command calculation means and an output voltage from the inverter due to a dead time for preventing an arm short circuit of the switching means Dead time compensation means for generating a compensation amount for compensating the error voltage between,
Voltage command correcting means for correcting the output of the voltage command calculating means based on the output of the dead time compensating means;
And a PWM signal generating means for generating a PWM signal obtained by the voltage command correcting means. Rectifying means for full-wave rectification of an AC power supply via a reactor;
A capacitor for smoothing the output of the rectifying means and outputting a bus voltage pulsating at a frequency approximately twice that of the AC power supply;
An inverter having switching means for applying a voltage to the electric motor using the bus voltage output from the capacitor as a power source;
Inverter control means for controlling the inverter;
Zero-cross detection means for detecting zero-cross of the AC power supply;
A bus voltage detecting means for detecting a voltage of the bus power supply;
Current detecting means for detecting current of two or more phases flowing through the electric motor,
The inverter control means includes a voltage command calculation means for calculating a voltage command based on a frequency command, an output of the zero cross detection means, an output of the bus voltage detection means, and an output of the current detection means,
Based on the output of the bus voltage detection means and the output of the current detection means, a voltage command from the voltage command calculation means and an output voltage from the inverter due to a dead time for preventing an arm short circuit of the switching means Dead time compensation means for generating a compensation amount for compensating the error voltage between,
Voltage command correcting means for correcting the output of the voltage command calculating means based on the output of the dead time compensating means;
And a PWM signal generating means for generating a PWM signal obtained by the voltage command correcting means.   The dead time compensation means includes a compensation amount calculation means for generating the compensation amount based on a carrier frequency of the switching means, an output of the bus voltage detection means, and a predetermined dead time. The drive device for an electric motor according to claim 1 or 2. The dead time compensation means includes a phase difference detection means for calculating a phase difference between a rotation phase of the electric motor and a phase of a phase current based on an output of the current detection means;
And a compensation amount calculating means for generating the compensation amount based on a carrier frequency of the switching means, an output of the bus voltage detecting means, a predetermined dead time, and an output of the phase difference detecting means. The drive device for an electric motor according to claim 1 or 2. The dead time compensation means includes a phase difference detection means for calculating a phase difference between a rotation phase of the electric motor and a phase of a phase current based on an output of the current detection means;
Create a dead time compensation voltage based on the carrier frequency of the switching means and the output of the bus voltage detection means and a predetermined dead time, calculate a specific order component of the dead time compensation voltage,
3. The electric motor according to claim 1, further comprising: a compensation amount specific order component calculation unit that generates the compensation amount based on an output of the phase difference detection unit and the calculated specific order component. Drive device.   6. The motor driving apparatus according to claim 5, wherein the dead time compensation means performs dead time compensation using at least one of the specific order components calculated by the compensation amount specific order component calculation means.   The dead time compensation means obtains at least one frequency component other than a multiple component of 3 among the harmonic components of the specific order of the dead time compensation voltage calculated by the compensation amount specific order component calculation means, and performs dead time compensation. The motor drive device according to claim 6, wherein: Coordinate conversion means for converting the phase current detected by the current detection means into a biaxial component on a stator coordinate system (αβ coordinate system) and then converting it into rotational coordinates,
8. The phase difference detection unit calculates a phase difference between a rotation phase of the electric motor and a phase of a phase current based on an output of the coordinate conversion unit. 8. Motor drive device. The biaxial component of the phase current on the coordinate system (dq or γδ coordinate system) that rotates the phase current detected by the current detection means at the same frequency as the electrical angular frequency of the rotor based on the output of the rotation phase detection means A coordinate conversion means for converting to rotational coordinates after conversion to
8. The phase difference detection unit according to claim 4, wherein the phase difference detection unit calculates a phase difference between the rotation of the electric motor and a phase of a phase current based on an output of the coordinate conversion unit. Electric motor drive device.   2. The dead time compensation means pulsates the compensation amount at a frequency approximately twice that of the AC power supply based on the output of the bus voltage detection means and the output of the zero cross detection means. The drive device for the electric motor according to claim 9.   The dead time compensation means performs dead time compensation by reducing the compensation voltage at a constant rate when the electric motor rotates at a lower speed than a predetermined value or the output of the current detection means is smaller than a predetermined value. The motor drive device according to any one of claims 1 to 10.   The motor drive device according to claim 11, wherein the rotational speed is 20% or less of a maximum rating.   The motor driving device according to claim 11, wherein the current is 20% or less of a maximum rating.   14. The dead time compensation means sets the compensation amount to zero near the zero cross of the AC power supply based on the detection of the zero cross detection means or the output of the bus voltage detection means. The electric motor drive device according to any one of the above.   The motor drive device according to any one of claims 1 to 14, wherein the rotation phase detection unit detects a rotation phase of the motor based on an output of the current detection unit.   The motor drive device according to any one of claims 1 to 15, wherein the electric motor is a permanent magnet type brushless DC motor.   The motor drive device according to any one of claims 1 to 16, wherein the maximum rotation speed of the electric motor is (60000 ÷ (number of poles / 2)) rpm or more.   The motor drive device according to any one of claims 1 to 17, wherein the inverter control means includes a microcomputer, a DSP, or an electronic circuit.   An apparatus comprising the electric motor drive device according to any one of claims 1 to 18.

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