JP2010154489A - Array antenna - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To reduce a space correlation by utilizing a parameter other than an antenna element interval. <P>SOLUTION: A phase directivity pattern of antenna elements RA1, RA2, that an array antenna includes, is set to meet a relation where a phase of the phase directivity pattern of the antenna element at which a signal arrives first, is delayed from a phase of the phase directivity pattern of the antenna element at which a signal arrives later, so that a phase difference caused by the directivity pattern is added to a phase difference caused by a layout of the antenna elements. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention relates to an array antenna composed of a plurality of antenna elements.

  MIMO (Multiple Input Multiple Output) is a technique for simultaneously transmitting a plurality of data streams in the same band by preparing a plurality of independent spatial transmission paths (hereinafter referred to as “channels”). MIMO is adopted as a standard in IEEE802.11n, IEEE802.16e, and the like. Further, in next-generation broadband wireless communication systems such as LTE (Long Term Evolution), MIMO with a multiplicity exceeding 2 is planned.

In MIMO transmission, it is necessary to dispose a plurality of antenna elements on both the transmitting and receiving sides and to widen the antenna element spacing on both the transmitting and receiving sides to such an extent that the plurality of channels can be regarded as independent from each other. This is because the transmission rate of the MIMO transmission can be improved up to the sum of the maximum transmission rates of each channel because the channels are independent of each other.
There is a spatial correlation as an index indicating whether or not the channels are independent from each other (see, for example, Non-Patent Document 1), and the spatial correlation becomes smaller as the channels are independent from each other. For example, the spatial correlation ρ _{12 in} the case of two multiplexing and two antenna elements on the transmitting side and two on the receiving side is expressed as follows when the channel response vectors are expressed by the following equations (1) and (2), respectively. (3)

但し、ｈ_１１，ｈ_１２，ｈ_２１，ｈ_２２の夫々はチャネル応答であり（図１参照。）、振幅及び位相を含んだ複素数である。また、正規化後の空間相関は０から１の間の値となる。なお、上添え字^Ｔは転置を示し、上添え字^Ｈは複素共役転置を示し、上添え字^＊は複素共役を示す。

However, h ₁₁ , h ₁₂ , h ₂₁ , and h ₂₂ are channel responses (see FIG. 1), which are complex numbers including amplitude and phase. The normalized spatial correlation is a value between 0 and 1. The superscript ^T indicates transposition, the superscript ^H indicates complex conjugate transpose, and the superscript ^* indicates complex conjugate.

In order to reduce the spatial correlation, it is necessary to increase the phase difference between h ₁₁ and h ₂₁ and the phase difference between h ₁₂ and h _22. For this purpose, it is necessary to widen the antenna element interval. For example, the antenna elements are arranged so that the interval is 0.5 wavelength. By the way, when the 2.5 GHz band is used for wireless communication, the 0.5 wavelength at 2.5 GHz is 6 cm. Therefore, in order to avoid the deterioration of spatial correlation, the distance between the two antenna elements is 6 cm. It is arranged to become.

Tanaka, Ogane, Ogawa, “Channel Allocation Criteria for SDMA with Adaptive Array”, IEICE Transactions (B), vol.J82-B, no.11, pp.2133-2141, Nov.1999.

  However, if the multiplicity increases, the number of necessary antenna elements increases accordingly. For example, in the case of four-multiplex MIMO, four antenna elements are required. For this reason, for example, when four-multiplex MIMO is applied to a communication device such as a mobile phone, it is difficult to arrange the four antenna elements with an interval of 0.5 wavelengths due to restrictions imposed by the size of the communication device. It is assumed that

Also, even in the case of two multiplexing, for example, depending on the size of the communication device and depending on the band used for wireless communication, two antenna elements may be arranged with an interval of 0.5 wavelength. It can be difficult.
The same can be said for the antenna element on the receiving side in wireless communication other than MIMO such as SIMO (Simple Input Multiple Output).

  Therefore, an object of the present invention is to provide an array antenna that can reduce spatial correlation using parameters other than the antenna element spacing.

  In order to achieve the above object, an array antenna of the present invention is an array antenna including M (M is an integer of 2 or more) antenna elements, and each of the antenna elements is a directional antenna. The phase directivity pattern of each antenna element is set so that the phase directivity patterns are different from each other.

  According to the array antenna described above, since the phase directivity patterns between the antenna elements are different from each other, there is a possibility that the spatial correlation can be reduced even with the same antenna element spacing.

The system block diagram of the radio | wireless communications system of 1st Embodiment. The block diagram of the mobile telephone of FIG. The schematic diagram of the electronically controlled waveguide antenna used for the antenna element of FIG. The figure which shows the relationship of the reactance value of the varactor diode of the electronically controlled waveguide antenna used for the antenna element of FIG. (A) is a figure which shows the amplitude directivity pattern of the antenna element of FIG. 1, (b) is a figure which shows the phase directivity pattern. The figure for demonstrating the phase difference of the received signal of antenna element A1 and antenna element A2 of FIG. The figure which shows the other directivity of an antenna element. (A) is a figure which shows the amplitude directivity pattern of four antenna elements, (b) is a figure which shows the phase directivity pattern. The schematic diagram of the electronically controlled waveguide antenna used for the antenna element with which the receiver of 2nd Embodiment is provided. The figure which shows the relationship of the reactance value of the varactor diode of the electronically controlled waveguide antenna used for the antenna element with which the receiver of 2nd Embodiment is provided. (A) is a figure which shows the amplitude directivity pattern of the antenna element with which the receiver of 2nd Embodiment is provided, (b) is a figure which shows the phase directivity pattern.

<< First Embodiment >>
Hereinafter, a first embodiment of the present invention will be described with reference to the drawings.
<Wireless communication system>
A radio communication system according to the present embodiment will be described with reference to FIG. FIG. 1 is a system configuration diagram of a radio communication system according to the present embodiment.

The base station T includes antenna elements TA1 and TA2, and the mobile phone R includes antenna elements RA1 and RA2. However, the antenna elements TA1 and TA2 constitute an array antenna, and the antenna elements RA1 and RA2 constitute an array antenna.
In the wireless communication from the base station T to the mobile phone R, four channels shown in FIG. 1 are formed, and the respective channel responses are represented by h ₁₁ , h ₂₁ , h ₁₂ , and h ₂₂ as shown in FIG. . The channel responses h ₁₁ , h ₂₁ , h ₁₂ , and h ₂₂ are complex numbers that indicate fluctuation amounts of amplitude and phase between transmission and reception points, respectively, and include amplitude characteristics and phase characteristics of antenna elements on the transmission side and reception side. Is also included. In this embodiment, in order to increase the phase difference between the channel response h ₁₁ and the channel response h ₂₁ and the phase difference between the channel response h ₁₂ and the channel response h ₂₂ to reduce the spatial correlation, the antenna elements RA1, RA2 The phase characteristics of the light have different directivities.

In the following description, attention will be focused on downlink communication from the base station T to the mobile phone R, and further focusing on the amplitude directivity and phase directivity of the antenna elements RA1 and RA2 of the mobile phone R on the receiving side.
<Mobile phone>
Hereinafter, the configuration of the mobile phone R in FIG. 1 will be described with reference to FIG. FIG. 2 is a configuration diagram of the mobile phone R in FIG.

  The mobile phone R includes antenna elements RA1 and RA2, front end units 11 and 12, a channel estimation unit 13, a weight generation unit 14, a separation / synthesis unit 15, and a demodulation unit 16. The cellular phone R can use a general configuration except for the antenna elements RA1 and RA2, and the details of the antenna elements RA1 and RA2 will be described later. Here, the outline of each component will be described. stop.

The antenna elements RA1 and RA2 are directional antennas, and electronically controlled waveguide antennas that are one of variable directional antennas are used as will be described later. However, the antenna element RA1 and the antenna element RA2 are arranged, for example, with an interval of 0.25 wavelengths.
The front end units 11 and 12 perform gain adjustment, frequency conversion, and the like on RF (Radio Frequency) signals received by the antenna elements RA1 and RA2, and output, for example, baseband reception signals.

The channel estimation unit 13 estimates channel responses h ₁₁ , h ₂₁ , h ₁₂ , and h ₂₂ using the preambles in the received signals output from the front end units 11 and ₁₂ . Then, the weight generation unit 14 generates weights using the channel responses h ₁₁ , h ₂₁ , h ₁₂ , h ₂₂ estimated by the channel estimation unit 13. The weight is generated by using, for example, maximum ratio combining (MRC) or minimum mean square error (MMSE).

The separation / combination unit 15 performs signal separation and synthesis by weighting the reception signals output from the front end units 11 and 12 using the weights generated by the weight generation unit 14. Two transmitted signals are estimated. The demodulator 16 demodulates the two transmission signals output from the separation / combination unit 15.
<Antenna element>
(Construction)
The structure of the electronically controlled waveguide antenna used for the antenna elements RA1 and RA2 in FIG. 1 will be described with reference to FIG. FIG. 3 is a schematic diagram showing the structure of an electronically controlled waveguide antenna used for the antenna elements RA1 and RA2 of FIG.

  The electronically controlled waveguide antenna used for the antenna elements RA1 and RA2 is a variable directivity antenna using inter-element electromagnetic coupling. Electronically controlled waveguide antennas are described in, for example, “Hashiguchi, Hodori, Iigusa, TAILLEFER, Hirata, Ohira,“ Design and Trial Production of ESPAR Antenna for Wireless Ad Hoc Networks, ”IEICE Transactions (B), vol. J85-B, no.12, pp.2245-2256, Dec. 2002. ”.

  The electronically controlled waveguide antenna has linear conductors (peripheral elements) 33, 35, and 37 with varactor diodes 34, 36, and 38 around a monopole feeding element (central element) 31 connected to an AC power source 32. It has an arranged structure. The electronically controlled waveguide antenna changes the directivity pattern by changing the inter-element electromagnetic coupling by changing the reactance values of the varactor diodes 34, 36, and 38. However, in this embodiment, it is assumed that the linear conductors 33, 35, and 37 are arranged at intervals of 120 ° (2π / 3) in the direction along the circumference.

  The feed element 31 and the linear conductors 33, 35, and 37 are preferably arranged at a distance of about 0.25 wavelength in free space, but a dielectric is provided between the feed element 31 and the linear conductors 33, 35, and 37. By filling with the body 39, the distance can be reduced to about 0.025 wavelength in free space. In addition, when using a 2.5 GHz band, the space | interval of the electric power feeding element 31 and the linear conductors 33, 35, and 37 is 3 mm.

(Direction of electronically controlled waveguide antenna and received signal at feeding element)
The directivity of the electronically controlled waveguide antenna whose structure is shown in FIG. 3 and the received signal at the feed element will be described.
The directivity pattern D (θ, φ) in the direction of the elevation angle θ and the azimuth angle φ of the electronically controlled waveguide antenna is expressed by the following equation (4). The reception signal y (t) at the element is expressed by the following equation (5). In the equations (4) and (5), the superscript ^T indicates transposition.

但し、ａ（θ，φ）は電子制御導波器アンテナのモードベクトルであり、下記の式（６）で表される。ｗ（ｘ_１，ｘ_２，ｘ_３）は電子制御導波器アンテナの素子間相互結合を含んだ等価ウェイトベクトルであり、下記の式（７）で表され、ｘ_１，ｘ_２，ｘ_３はバラクタダイオード３４，３６，３８のリアクタンス値である。ｓ（ｔ）は送信信号であり、ｈ_ｓは送信点から電子制御導波器アンテナの給電素子までのチャネル応答（アンテナ素子の指向性を除いたチャネル応答）である。

However, a (θ, φ) is a mode vector of the electronically controlled waveguide antenna and is expressed by the following equation (6). w (x ₁ , x ₂ , x ₃ ) is an equivalent weight vector including the mutual coupling between elements of the electronically controlled waveguide antenna, and is expressed by the following equation (7), and x ₁ , x ₂ , x ₃ Is the reactance value of the varactor diodes 34, 36, 38. s (t) is a transmission signal, and h _s is a channel response (a channel response excluding the directivity of the antenna element) from the transmission point to the feeding element of the electronically controlled waveguide antenna.

但し、βは伝搬定数（＝２π／λ）、ｄは給電素子と線状導体間の間隔である。

Where β is a propagation constant (= 2π / λ), and d is the distance between the feed element and the linear conductor.

但し、ｚ_ｓは送受信機の入出力インピーダンスである。Ｚは給電素子及び線状導体間のインピーダンス行列であり、下記の式（８）で表される。Ｘは送受信機の入出力インピーダンスとバラクタダイオードのリアクタンス値を成分とする対角行列であり、下記の式（９）で表される。ｕ_０は単位ベクトルであり、下記の式（１０）で表される。なお、式（１０）において、上添え字^Ｔは転置を示す。

Where z _s is the input / output impedance of the transceiver. Z is an impedance matrix between the feed element and the linear conductor, and is represented by the following formula (8). X is a diagonal matrix whose components are the input / output impedance of the transceiver and the reactance value of the varactor diode, and is represented by the following equation (9). u ₀ is a unit vector and is expressed by the following equation (10). In equation (10), the superscript ^T indicates transposition.

但し、Ｚの各要素は複素パラメータであり、ｚ_００は給電素子の自己入力インピーダンス、ｚ_０１は給電素子と線状導体間結合インピーダンス、ｚ_１１は線状導体の自己入力インピーダンス、ｚ_１２は隣接する線状導体間結合入力インピーダンスである。

However, each element of the Z is a complex parameter, z ₀₀ is self-input impedance of the feed element, z ₀₁ is between the feed element and the linear conductor coupling impedance, z ₁₁ is self-input impedance of the linear conductors, z ₁₂ adjacent It is a coupling input impedance between linear conductors.

上記のｗ（ｘ_１，ｘ_２，ｘ_３）はリアクタンス値ｘ_１，ｘ_２，ｘ_３の関数であるため、ｗ（ｘ_１，ｘ_２，ｘ_３）はリアクタンス値ｘ_１，ｘ_２，ｘ_３によって変化し、指向性パターンＤ（θ，φ）はリアクタンス値ｘ_１，ｘ_２，ｘ_３によって変化する（式（４）参照。）。従って、リアクタンス値ｘ_１，ｘ_２，ｘ_３を調整することによって、所望の指向性パターンＤ（θ，φ）を得ることが可能である。つまり、リアクタンス値ｘ_１，ｘ_２，ｘ_３を調整することによって、所望の振幅指向性パターンを得ることができ、また、所望の位相指向性パターンを得ることができる。

Since w (x ₁ , x ₂ , x ₃ ) is a function of the reactance values x ₁ , x ₂ , x ₃ , w (x ₁ , x ₂ , x ₃ ) is a reactance value x ₁ , x ₂ , x ₃ , varies with x _3, the directivity pattern D (theta, phi) is changed by the reactance values _{x 1, x 2, x 3} ( formula (4) reference.). Therefore, it is possible to obtain a desired directivity pattern D (θ, φ) by adjusting the reactance values x ₁ , x ₂ , x ₃ . That is, by adjusting the reactance values x ₁ , x ₂ , and x ₃ , a desired amplitude directivity pattern can be obtained, and a desired phase directivity pattern can be obtained.

(Specific example of directivity of electronically controlled waveguide antenna)
The directivity patterns of the antenna elements RA1 and RA2 used in the present embodiment will be described with reference to FIGS. FIG. 4 is a diagram showing a relationship between reactance values of the varactor diodes 34, 36, and 38 of the antenna element RA1 and the antenna element RA2. FIG. 5A is a diagram showing amplitude directivity patterns of the antenna elements RA1 and RA2, and FIG. 5B is a diagram showing phase directivity patterns of the antenna elements RA1 and RA2. However, the horizontal plane azimuth described below corresponds to the azimuth φ in the above (directivity of the electronically controlled waveguide antenna).

The linear conductors 33, 35, and 37 connected to the varactor diodes 34, 36, and 38 of the antenna elements RA1 and RA2 have horizontal plane azimuth angles φ of 90 ° (π / 2), 210 ° (7π / 6), and 330 °. It is arranged at a position of (11π / 6).
As shown in FIG. 4, the reactance values of the varactor diodes 34, 36, and 38 of the antenna element RA1 are set to x _a , x _b , and x _c (x ₁ = x _a , x ₂ = x _b , x ₃ = x _c ). However, as described above, the desired amplitude directivity pattern and phase directivity pattern can be obtained by adjusting the reactance values of the varactor diodes 34, 36, and 38. Therefore, the reactance values of the varactor diodes 34, 36, and 38 are adjusted so that the amplitude directivity pattern of the antenna element RA1 is as shown in FIG. 5A and the phase directivity pattern is as shown in FIG. 5B. As a result, it is assumed that x _a , x _b and x _c are obtained as reactance values of the varactor diodes 34, 36 and 38.

Also, the reactance value of the varactor diode 34, 36, 38 _x a of the antenna elements _RA2, x c, is set to _{x b (x 1 = x a} , x 2 = x c, x 3 = x b). As a result, the amplitude directivity pattern and the phase directivity pattern of the antenna element RA2 shown in FIGS. 5A and 5B are obtained. The reactance values of the varactor diodes 34, 36, and 38 of the antenna element RA2 are set so that the amplitude directivity pattern and the phase directivity pattern of the antenna element RA2 become the patterns shown in FIGS. 5 (a) and 5 (b). It only needs to be adjusted.

Hereinafter, the amplitude directivity pattern and phase directivity pattern of the antenna elements RA1 and RA2 will be described.
The amplitude directivity patterns of the antenna elements RA1 and RA2 are substantially omnidirectional as shown in FIG.
As shown in FIG. 5B, in the phase directivity pattern of the antenna element RA1, the phase monotonously increases when the horizontal plane azimuth angle φ is in the range of 0 ° to 180 °, and the horizontal plane azimuth φ is 180 ° to 360 °. It is a monotonic decrease in the range of. On the other hand, in the phase directivity pattern of the antenna element RA2, the phase monotonously decreases when the horizontal plane azimuth angle φ is 0 ° or more and 180 ° or less, and monotonically increases when the horizontal plane azimuth angle φ is 180 ° or more and 360 ° or less. It is. Thus, the phase directivity pattern of the antenna element RA1 and the phase directivity pattern of the antenna element RA2 are opposite to each other. In addition, by making the phase directivity pattern monotonously increase and monotonously decrease, a phase difference caused by the difference in the phase directivity pattern of the antenna elements RA1 and RA2 is generated for radio waves arriving from almost all directions. Is possible.

  When the horizontal plane azimuth angle φ is 0 ° or more and less than 90 ° and greater than 270 ° and 360 ° or less, the phase of the phase directivity pattern of the antenna element RA2 is more advanced than that of the antenna element RA1, and the horizontal plane azimuth φ is 90 °. When the angle is larger than 270 °, the phase of the phase directivity pattern of the antenna element RA1 is more advanced than that of the antenna element RA2. That is, in the phase directivity pattern of the antenna elements RA1 and RA2, the phase of the phase directivity pattern of the antenna element where the signal arrives with a delayed phase is the phase of the phase directivity pattern of the antenna element where the signal arrives with a advanced phase. The phase is later.

(Phase difference of received signals between antenna element RA1 and antenna element RA2)
Hereinafter, the phase difference between the reception signals of the antenna element RA1 and the antenna element RA2 will be described with reference to FIG. FIG. 6 is a diagram for explaining a phase difference of received signals between the antenna element RA1 and the antenna element RA2. However, the “reception signal” here is a signal after the phase according to the phase directivity pattern of the antenna elements RA1 and RA2 is added. Note that the base station T performs two-stream transmission, but the same can be said in any case. Therefore, here, description will be given focusing on only one stream.

Assume that the distance between the antenna element RA1 and the antenna element RA2 is L, and the signal A arrives at the antenna elements RA1 and RA2 from the direction of the horizontal plane azimuth angle φ. The antenna element RA1 is the antenna element RA2, the phase difference h _.phi.1 delayed signal A corresponding to the distance Lcosφ due to the arrangement of the antenna element RA1 and the antenna element RA2 arrives (refer to FIG. 6 (a)).
At the horizontal plane azimuth angle φ, the phase of the phase directivity pattern of the antenna element RA1 is delayed by a phase difference h _φ2 from that of the antenna RA2 (see FIG. 6B).

A reception signal received by the antenna element RA1, from the received signal received by the antenna elements RA2, the phase difference h _.phi.2 caused by the phase directivity pattern on the phase difference h _.phi.1 due to the arrangement of the antenna elements are summed position The phase difference (h _φ1 + h _φ2 ) is delayed (see FIG. _6A ).
If the phase directivity pattern of the antenna elements RA1 and RA2 is omnidirectional, there will be no phase difference due to the phase directivity pattern between the antenna element RA1 and the antenna element RA2. For this reason, the reception signal received by the antenna element RA1 is delayed from the reception signal received by the antenna element RA2 by the phase difference h _φ1 due to the arrangement of the antenna elements.

Therefore, when the phase directivity patterns of the antenna elements RA1 and RA2 are set as shown in FIG. 5B, the phase difference due to the phase directivity pattern is added to the phase difference due to the arrangement of the antenna elements. Therefore, as compared with the case where the phase directional pattern of the antenna element RA1, RA2 is omnidirectional, the phase difference between the phase difference and the channel response _{h 12} and the channel response _{h 22} of the channel response _{h 11} and the channel response _{h 21} Can be increased. Thus, the array antenna of the present embodiment can reduce the spatial correlation even when the distance between the antenna elements RA1 and RA2 is the same as that of the conventional array antenna.

(Spatial correlation)
The spatial correlation when the antenna elements RA1 and RA2 have different phase directivity patterns will be described.
The channel response h (h ₁₁ , h ₁₂ , h ₂₁ , h _{22 in} FIG. 1) is from the directivity pattern D (θ, φ) of the antenna elements RA 1 and RA 2 to the feeding element of the electronically controlled waveguide antenna. Is expressed by the following equation (11) multiplied by the channel response h _{s of} In equation (11), the superscript ^T indicates transposition.

アンテナ素子ＲＡ１、ＲＡ２に図５（ｂ）に示す逆の位相指向性パターンが与えられた場合の、ＭＩＭＯにおけるアンテナ素子ＴＡ１及びアンテナ素子ＴＡ２に対するチャネル応答ベクトルｈ_{１，ｄｉｒｅｃｔｉｏｎａｌ}，ｈ_{２，ｄｉｒｅｃｔｉｏｎａｌ}は、夫々、下記の式（１２）、（１３）で与えられる。なお、式（１２），（１３）において、上添え字^Ｔは転置を示す。

Antenna element RA1, when RA2 reverse phase directivity pattern shown in FIG. 5 (b) given to the channel response vector _h 1 for antenna element TA1 and the antenna element TA2 in _{MIMO, directional, h 2, directional} is They are given by the following equations (12) and (13), respectively. In the equations (12) and (13), the superscript ^T indicates transposition.

但し、ｈ、ｈ_ｓ、θ、φの夫々の下添え字の２つの数字のうちの１つ目の“１”は受信側のアンテナ素子ＲＡ１を、“２”はアンテナ素子ＲＡ２を示し、当該下添え字の２つ目の“１”は送信側のアンテナ素子ＴＡ１を、“２”はアンテナ素子ＴＡ２を示す。ψ_１１及びψ_１２は、夫々、アンテナ素子ＴＡ１からの信号到来方向に対するアンテナ素子ＲＡ１及びアンテナ素子ＲＡ２の指向性パターンの位相を示す。また、ψ_１２及びψ_２２は、夫々、アンテナ素子ＴＡ２からの信号到来方向に対するアンテナ素子ＲＡ１及びアンテナ素子ＲＡ２の指向性パターンの位相を示す。なお、指向性パターンの振幅は大きな減衰がない無指向性に近いものであるとしているため、空間相関の計算においては無視できるものとして、チャネル応答ベクトルの要素から省略している。

However, the first “1” of the two subscript numbers of h, h _s , θ, and φ represents the antenna element RA1 on the receiving side, and “2” represents the antenna element RA2. The second subscript “1” indicates the antenna element TA1 on the transmission side, and “2” indicates the antenna element TA2. ψ ₁₁ and ψ ₁₂ indicate the phase of the directivity pattern of the antenna element RA1 and the antenna element RA2 with respect to the signal arrival direction from the antenna element TA1, respectively. Further, ψ ₁₂ and ψ ₂₂ indicate the phase of the directivity pattern of the antenna element RA1 and the antenna element RA2 with respect to the signal arrival direction from the antenna element TA2, respectively. The amplitude of the directivity pattern is assumed to be close to omnidirectional with no significant attenuation, and is omitted from the channel response vector elements as being negligible in the calculation of spatial correlation.

The spatial correlation ρ ₁₂ ^directional when a directional antenna is used as the receiving antenna is expressed by the following equation (14). In equation (14), the superscript ^H indicates complex conjugate transposition.

式（３）と式（１４）とを比較すると、受信側に指向性アンテナを使用し適切な位相指向性パターンを形成することによって、空間相関を低下させることが可能となることが分かる。なお、このことは、受信側のみならず、送信側のみに可変指向性アンテナを使用した場合であっても、また、送受信側双方に可変指向性アンテナを使用した場合であっても同様に成立する。

Comparing equation (3) and equation (14), it can be seen that the spatial correlation can be lowered by forming an appropriate phase directivity pattern using a directional antenna on the receiving side. This is true even when the variable directional antenna is used not only on the receiving side but also on the transmitting side, and also when the variable directional antenna is used on both the transmitting and receiving sides. To do.

<< Modification of First Embodiment >>
<First Modification>
Instead of the phase directivity pattern (see FIG. 5B) of the antenna elements RA1 and RA2 provided in the mobile phone R of the first embodiment, the antenna elements RA1 and RA2 provided in the mobile phone are shown in FIG. Such a phase directivity pattern is provided.

<Second Modification>
Unlike the mobile phone R according to the first embodiment, the mobile phone according to the second modification has four antenna elements A1 to A4, and the antenna elements A1 to A4 have amplitude directivity shown in FIG. And the phase directivity pattern shown in FIG. However, it is assumed that the antenna elements A1, A2, A3, and A4 are aligned in that order.

As shown in FIG. 8A, the amplitude directivity patterns of the antenna elements A1, A2, A3, and A4 are set to be substantially omnidirectional.
As shown in FIG. 8B, in each of the phase directivity patterns of the antenna elements A1 and A2, the phase monotonically increases when the horizontal plane azimuth φ is in the range of 0 ° to 180 °, and the horizontal azimuth φ is 180 ° or more. In the range of 360 ° or less, it decreases monotonously. On the other hand, in the phase directivity pattern of the antenna elements A3 and A4, the phase monotonously decreases when the horizontal plane azimuth φ is in the range of 0 ° or more and 180 ° or less, and in the range where the horizontal plane azimuth φ is 180 ° or more and 360 ° or less. It is a monotonous increase.

  When the horizontal plane azimuth angle φ is 0 ° or more and less than 90 ° and greater than 270 ° and 360 ° or less, the phase of the phase directivity pattern advances in the order of the antenna elements A4, A3, A2, and A1, and the horizontal plane azimuth φ is If it is greater than 90 ° and less than 270 °, the phase of the phase directivity pattern advances in the order of the antenna elements A1, A2, A3, and A4. That is, in the phase directivity pattern of antenna elements A1, A2, A3, and A4, the phase directivity pattern of the antenna element from which the signal arrives with a delayed phase is the phase directivity of the antenna element from which the signal arrives with a advanced phase. The phase is delayed from the phase of the sex pattern.

<< Second Embodiment >>
Hereinafter, a second embodiment of the present invention will be described with reference to the drawings. The mobile phone according to the present embodiment includes four antenna elements B1 to B4 as in the second modification of the first embodiment, but the phase directivity of the antenna elements B1 to B4. The relationship between the patterns is different from the relationship between the phase directivity patterns of the antenna elements A1 to A4 according to the second modification of the first embodiment (see FIG. 8B).

<Antenna element>
The structure of the electronically controlled waveguide antenna used for the antenna elements B1 to B4 of the present embodiment will be described with reference to FIG. FIG. 9 is a schematic diagram showing the structure of an electronically controlled waveguide antenna used for the antenna elements B1 to B4 of the present embodiment.
The electronically controlled waveguide antenna used for the antenna elements B1 to B4 is a linear conductor with varactor diodes 104, 106, 108, 110 around a monopole feeding element (central element) 101 connected to an AC power source 102. (Peripheral elements) 103, 105, 107, 109 are arranged. However, in the present embodiment, it is assumed that the linear conductors 103, 105, 107, 109 are arranged at intervals of 90 ° (π / 2) in the direction along the circumference.

(Direction of electronically controlled waveguide antenna and received signal at feeding element)
The directivity of the electronically controlled waveguide antenna whose structure is shown in FIG. 9 and the received signal at the feed element will be described.
The directivity pattern D (θ, φ) in the direction of the elevation angle θ and the azimuth angle φ of the electronically controlled waveguide antenna is expressed by the following equation (15). The reception signal y (t) at the element is expressed by the following equation (16). In the equations (15) and (16), the superscript ^T indicates transposition.

但し、ａ（θ，φ）は電子制御導波器アンテナのモードベクトルであり、下記の式（１７）で表される。ｗ（ｘ_１，ｘ_２，ｘ_３，ｘ_４）は電子制御導波器アンテナの素子間相互結合を含んだ等価ウェイトベクトルであり、下記の式（１８）で表され、ｘ_１，ｘ_２，ｘ_３，ｘ_４はバラクタダイオード１０４，１０６，１０８，１１０のリアクタンス値である。ｓ（ｔ）は送信信号であり、ｈ_ｓは送信点から電子制御導波器アンテナの給電素子までのチャネル応答（アンテナ素子の指向性を除いたチャネル応答）である。

However, a (θ, φ) is a mode vector of the electronically controlled waveguide antenna and is expressed by the following equation (17). w (x ₁ , x ₂ , x ₃ , x ₄ ) is an equivalent weight vector including the mutual coupling between elements of the electronically controlled waveguide antenna, and is expressed by the following equation (18), and x ₁ , x ₂ , X ₃ , x ₄ are reactance values of the varactor diodes 104, 106, 108, 110. s (t) is a transmission signal, and h _s is a channel response (a channel response excluding the directivity of the antenna element) from the transmission point to the feeding element of the electronically controlled waveguide antenna.

但し、βは伝搬定数（＝２π／λ）、ｄは給電素子と線状導体間の間隔である。

Where β is a propagation constant (= 2π / λ), and d is the distance between the feed element and the linear conductor.

但し、ｚ_ｓは送受信機の入出力インピーダンスである。Ｚは給電素子及び線状導体間のインピーダンス行列であり、下記の式（１９）で表される。Ｘは送受信機の入出力インピーダンスとバラクタダイオードのリアクタンス値を成分とする対角行列であり、下記の式（２０）で表される。ｕ_０は単位ベクトルであり、下記の式（２１）で表される。なお、式（２１）において、上添え字^Ｔは転置を示す。

Where z _s is the input / output impedance of the transceiver. Z is an impedance matrix between the feed element and the linear conductor, and is represented by the following equation (19). X is a diagonal matrix whose components are the input / output impedance of the transceiver and the reactance value of the varactor diode, and is represented by the following equation (20). u ₀ is a unit vector and is represented by the following equation (21). In the formula (21), the superscript ^T indicates transposition.

但し、Ｚの各要素は複素パラメータであり、ｚ_００は給電素子の自己入力インピーダンス、ｚ_０１は給電素子と線状導体間結合インピーダンス、ｚ_１１は線状導体の自己入力インピーダンス、ｚ_１２は隣接する線状導体間結合入力インピーダンスであり、ｚ_１３は対角にある線状導体間結合入力インピーダンスである。

However, each element of the Z is a complex parameter, z ₀₀ is self-input impedance of the feed element, z ₀₁ is between the feed element and the linear conductor coupling impedance, z ₁₁ is self-input impedance of the linear conductors, z ₁₂ adjacent The linear conductor-to-line coupling input impedance z ₁₃ is the diagonal linear conductor-to-linear coupling input impedance.

上記のｗ（ｘ_１，ｘ_２，ｘ_３，ｘ_４）はリアクタンス値ｘ_１，ｘ_２，ｘ_３，ｘ_４の関数であるため、ｗ（ｘ_１，ｘ_２，ｘ_３，ｘ_４）はリアクタンス値ｘ_１，ｘ_２，ｘ_３，ｘ_４によって変化し、指向性パターンＤ（θ，φ）はリアクタンス値ｘ_１，ｘ_２，ｘ_３，ｘ_４によって変化する（式（１５）参照。）。従って、リアクタンス値ｘ_１，ｘ_２，ｘ_３，ｘ_４を調整することによって、所望の指向性パターンＤ（θ，φ）を得ることが可能である。つまり、リアクタンス値ｘ_１，ｘ_２，ｘ_３，ｘ_４を調整することによって、所望の振幅指向性パターンを得ることができ、また、所望の位相指向性パターンを得ることができる。

Since w (x ₁ , x ₂ , x ₃ , x ₄ ) is a function of the reactance values x ₁ , x ₂ , x ₃ , x ₄ , w (x ₁ , x ₂ , x ₃ , x ₄ ) Changes according to reactance values x ₁ , x ₂ , x ₃ , x ₄ , and the directivity pattern D (θ, φ) changes according to reactance values x ₁ , x ₂ , x ₃ , x ₄ (see equation (15)). .) Therefore, it is possible to obtain a desired directivity pattern D (θ, φ) by adjusting the reactance values x ₁ , x ₂ , x ₃ , x ₄ . That is, by adjusting the reactance values x ₁ , x ₂ , x ₃ , and x ₄ , a desired amplitude directivity pattern can be obtained and a desired phase directivity pattern can be obtained.

(Specific example of directivity of electronically controlled waveguide antenna)
The directivity patterns of the antenna elements B1 to B4 used in the present embodiment will be described with reference to FIGS. FIG. 10 is a diagram showing the relationship of reactance values of the varactor diodes 104, 106, 108, and 110 of the antenna elements B1 to B4. FIG. 11A is a diagram showing amplitude directivity patterns of the antenna elements B1 to B4, and FIG. 11B is a diagram showing phase directivity patterns of the antenna elements B1 to B4. However, the horizontal plane azimuth described below corresponds to the azimuth φ in the above (directivity of the electronically controlled waveguide antenna).

  The antenna elements B1 to B4 are respectively arranged on the vertices of a square, that is, the horizontal plane azimuth angle φ is 225 ° (5π / 4), 315 ° (7π / 4), 45 ° (π / 4), 135 °. It is arranged at a position of (3π / 4). Further, the linear conductors 103, 105, 107, 109 to which the varactor diodes 104, 106, 108, 110 of the antenna elements B1 to B4 are connected have a horizontal plane azimuth φ of 225 ° (5π / 4), 315 ° (7π / 4), 45 ° (π / 4), and 135 ° (3π / 4).

As shown in FIG. 10, the reactance value of the varactor diode 104, 106, 108, 110 of the antenna element _{B1 x a, x b, x} c, is set to _{x d (x 1 = x a} , x 2 = x b , X ₃ = x _c , x ₄ = x _d ). However, as described above, the desired amplitude directivity pattern and phase directivity pattern can be obtained by adjusting the reactance values of the varactor diodes 104, 106, 108, and 110. Therefore, the reactance values of the varactor diodes 104, 106, 108, and 110 are set so that the amplitude directivity pattern of the antenna element B1 is as shown in FIG. 11A and the phase directivity pattern is as shown in FIG. As a result, it is assumed that x _a , x _b , x _c and x _d are obtained as reactance values of the varactor diodes 104, 106, 108 and 110.

Also, the reactance value _x d of the varactor diodes 104, 106, 108, 110 of the antenna element _B2, x _a, x b, are set to _{x c (x 1 = x d} , x 2 = x a, x 3 = x _b , x ₄ = x _c ). As a result, the amplitude directivity pattern and the phase directivity pattern of the antenna element B2 shown in FIG. 11A and FIG. 11B are obtained. The reactance values of the varactor diodes 104, 106, 108, 110 of the antenna element B2 are the patterns shown in FIGS. 11A and 11B in the amplitude directivity pattern and the phase directivity pattern of the antenna element B2. As long as it is adjusted.

Further, the reactance values of the varactor diodes 104, 106, 108, and 110 of the antenna element B3 are set to x _c , x _d , x _a , and x _b (x ₁ = x _c , x ₂ = x _d , x ₃ = x _a , x ₄ = x _b ). Thereby, the amplitude directivity pattern and the phase directivity pattern of the antenna element B3 shown in FIGS. 11A and 11B are obtained. The reactance values of the varactor diodes 104, 106, 108, and 110 of the antenna element B3 are the patterns shown in FIGS. 11A and 11B in the amplitude directivity pattern and the phase directivity pattern of the antenna element B3. As long as it is adjusted.

Furthermore, the reactance value of the varactor diode 104, 106, 108, 110 of the antenna element _{B4 x b, x c, x} d, is set to _{x a (x 1 = x b} , x 2 = x c, x 3 = x _d , x ₄ = x _a ). As a result, the amplitude directivity pattern and phase directivity pattern of the antenna element B4 shown in FIGS. 11A and 11B are obtained. The reactance values of the varactor diodes 104, 106, 108, 110 of the antenna element B4 are such that the amplitude directivity pattern and the phase directivity pattern of the antenna element B4 are the patterns shown in FIGS. As long as it is adjusted.

Below, it describes below about the amplitude directivity pattern and phase directivity pattern of antenna element B1-B4.
The amplitude directivity patterns of the antenna elements B1 to B4 are substantially omnidirectional as shown in FIG.
In the phase directivity pattern of the antenna element B1, as shown in FIG. 11B, when the horizontal plane azimuth is in the range of 0 ° to 45 ° and in the range of 0 ° to 225 ° to 360 °, it decreases monotonously, and the horizontal plane azimuth is 45. It is monotonically increasing from ° to 225 °. The phase directivity pattern of the antenna element B1 is obtained by sliding a later-described phase directivity pattern of the antenna element B4 by 90 ° in the positive direction of the horizontal plane azimuth angle. However, this 90 ° is obtained by dividing 360 ° by the number of antenna elements “4”.

  The phase directivity pattern of the antenna element B2 is obtained by sliding the phase directivity pattern of the antenna element B1 by 90 ° in the positive direction of the horizontal plane azimuth. The phase directivity pattern of the antenna element B3 is obtained by sliding the phase directivity pattern of the antenna element B2 by 90 ° in the positive direction of the horizontal plane azimuth. Furthermore, the phase directivity pattern of the antenna element B4 is obtained by sliding the phase directivity pattern of the antenna element B3 by 90 ° in the positive direction of the horizontal plane azimuth.

That is, the phase directivity pattern of the antenna elements B1, B2, B3, and B4 is 90 degrees in the positive direction of the horizontal plane azimuth angle with respect to the phase directivity pattern of the previous antenna element B4, B1, B2, and B3 counterclockwise. It has been slid once.
Further, in the phase directivity patterns of the antenna elements B1 to B4, the phase of the phase directivity pattern advances in the order of the antenna elements B3, B2, B4, and B1 in which the horizontal plane azimuth angle φ is greater than 0 ° and less than 45 °. When the horizontal plane azimuth φ is greater than 45 ° and less than 90 °, the phase of the phase directivity pattern advances in the order of the antenna elements B3, B4, B2, and B1. When the horizontal plane azimuth angle φ is greater than 90 ° and less than 135 °, the phase of the phase directivity pattern advances in the order of the antenna elements B4, B3, B1, and B2. When the horizontal plane azimuth φ is greater than 135 ° and less than 180 °, the phase of the phase directivity pattern advances in the order of the antenna elements B4, B1, B3, and B2.

  When the horizontal plane azimuth φ is greater than 180 ° and less than 225 °, the phase of the phase directivity pattern advances in the order of the antenna elements B1, B4, B2, and B3. When the horizontal plane azimuth φ is greater than 225 ° and less than 270 °, the phase of the phase directivity pattern advances in the order of the antenna elements B1, B2, B4, and B3. When the horizontal plane azimuth φ is greater than 270 ° and less than 315 °, the phase of the phase directivity pattern advances in the order of the antenna elements B2, B1, B3, and B4. When the horizontal plane azimuth φ is greater than 315 ° and less than 360 °, the phase of the phase directivity pattern advances in the order of the antenna elements B2, B3, B1, and B4.

That is, in the phase directivity pattern of the antenna elements B1 to B4, the phase directivity pattern of the antenna element from which the signal arrives with a delayed phase is different from the phase directivity pattern of the antenna element from which the signal arrives with the advanced phase. The phase is later.
<Supplement>
The present invention is not limited to the contents described in the above embodiment, and can be implemented in any form for achieving the object of the present invention and the object related thereto or incidental thereto. .

(1) In the second embodiment described above, the four antenna elements are arranged on the vertices of the square. However, the present invention is not limited to this, and for example, a square such as a rectangle, a rhombus, or a parallelogram is used. It does not have to be.
(2-1) The relationship of the phase directivity pattern between the antenna elements included in the array antenna is not limited to the relationship described in the first embodiment and the modifications thereof. For example, the array antenna includes M (M is an integer of 2 or more) antenna elements, the i-th phase directivity pattern of the i-th (i is an integer of 1 to M) antenna elements, and the j-th ( j may be 1 or more and M or less, and the relationship with the j-th phase directivity pattern of the antenna element (which is an integer other than i) may satisfy the following relationship.

  The i-th phase directivity pattern and the j-th phase directivity pattern have a phase difference generated based on the arrangement of the i-th antenna element and the j-th antenna element with respect to signals arriving from the same transmission source. What is necessary is just to set so that the phase difference which generate | occur | produces based on the phase directivity of an i-th antenna element and a j-th antenna element may be added. In other words, the i-th phase directivity pattern and the j-th phase directivity pattern of the antenna element from which the signal arrives at a delayed phase is the phase of the antenna element from which the signal arrives at a delayed phase. It is only necessary that the phase is set so as to be delayed from the phase of the directivity pattern.

In addition, the antenna element passes through the i-th antenna element (i is an integer from 1 to M) and the j-th antenna element (j is an integer other than i from 1 to M) and passes through the i-th antenna element to the j-th antenna. When the angle with respect to the axis whose positive direction is the direction toward the element is φ, φ is 0 ° or more and 360 ° or less, and 0 ° ≦ φ <φ _A1 ° (φ _A1 is a predetermined value smaller than 90) and φ _B1 ° (φ _B1 is a predetermined value larger than 270) <φ ≦ 360 °, the phase of the j-th phase directivity pattern of the j-th antenna element is the phase of the i-th phase directivity pattern of the i-th antenna element. more advances, the φ _A2 ° (φ _A2 is 90 larger than the predetermined _{value) <φ <φ B2 ° (} φ B2 is greater than phi _A2 270 is smaller than a predetermined value), the phase directivity pattern of the i phase So that the phase advances from the phase of the j-th phase directivity pattern Phase directivity pattern of the i and the j has only to be set. For example, φ _A1 and φ _A2 are approximately 90, and φ _B1 and φ _B2 are approximately 270.

In this case, as the phase directivity pattern of each antenna element, a pattern that monotonically increases when 0 ° ≦ φ ≦ 180 °, monotonously decreases when 180 ° ≦ φ ≦ 360 °, and monotonically decreases when 0 ° ≦ φ ≦ 180 °. Thus, the phase directivity pattern of each antenna element may be set so as to satisfy the above relationship as one of the patterns that monotonously increase when 180 ° ≦ φ ≦ 360 °.
Further, when the M antenna elements are divided into two antenna element pairs, the phase directivity of each antenna element is such that the phase directivity patterns of the two antenna elements of the same pair are opposite to each other. A pattern may be set. However, when M is an odd number, one antenna element is excluded. In the examples of FIGS. 5B and 7, the phase directivity patterns of the antenna elements RA1 and RA2 are opposite to each other. In the example of FIG. 8B, the phase directivity patterns of the antenna elements A1 and A4 are opposite to each other, and the phase directivity patterns of the antenna elements A2 and A3 are opposite to each other.

  (2-2) The relationship of the phase directivity pattern between the antenna elements included in the array antenna is not limited to the relationship described in the second embodiment. For example, the number of antenna elements included in the array antenna is M (M is an integer of 3 or more), and each array antenna element is counterclockwise on the vertex of an M-gon (a polygon having the number of vertices M). When the first, second,..., Mth antenna elements are arranged, the phase directivity pattern of each antenna element may satisfy the following relationship.

  The i-th phase directivity pattern of the i-th antenna element (i is an integer of 1 to M) and the j-th antenna element of the j-th antenna element (j is an integer of 1 to M and other than i). The phase directivity pattern refers to the phase difference generated based on the arrangement of the i-th antenna element and the j-th antenna element with respect to signals coming from the same transmission source, and the i-th antenna element and the j-th antenna element. It is sufficient that the phase difference generated based on the phase directivity is set to be added. In other words, the i-th phase directivity pattern and the j-th phase directivity pattern of the antenna element from which the signal arrives at a delayed phase is the phase of the antenna element from which the signal arrives at a delayed phase. It is only necessary that the phase is set so as to be delayed from the phase of the directivity pattern.

  Under the above conditions, the i-th phase directivity pattern of the i-th antenna element (i is an integer not less than 1 and not more than M) is the M-th phase directivity pattern of the M-th antenna element when i is 1. When i is other than 1, the (i-1) th phase directivity pattern of the (i-1) th antenna element may be slid in the horizontal plane azimuth direction. In this case, the sliding angle is approximately φ, and the value of φ may be a value obtained by dividing 360 ° by M (360 ° / M), for example.

  (3) In the first embodiment, the relationship between the arrangement of the linear conductors 33, 35, 37 of the antenna elements RA1, RA2 and the reactance values of the varactor diodes 34, 36, 38 is shown in FIG. However, it is not limited to this. For example, the arrangement of the linear conductor of each antenna element and the reactance value of the varactor diode are set so that the phase directivity pattern of each antenna element satisfies the relationship described in the first embodiment. Just do it.

  In the second embodiment, the relationship between the arrangement of the linear conductors 103, 105, 107, and 109 of the antenna elements B1 to B4 and the reactance values of the varactor diodes 104, 105, 108, and 110 is shown in FIG. It was supposed to be, but it is not limited to this. For example, the arrangement of the linear conductor of each antenna element and the reactance value of the varactor diode are set so that the phase directivity pattern of each antenna element satisfies the relationship described in the second embodiment. Just do it.

  (4) In the first embodiment, the interval between the antenna element RA1 and the antenna element RA2 is described as 0.25 wavelength. However, the interval between the antenna element RA1 and the antenna element RA2 is 0.25 wavelength. It is not limited to this, and may be determined in consideration of the size of the mobile phone on which the array antenna having the antenna elements RA1 and RA2 is mounted and the phase directivity pattern of the antenna elements RA1 and RA2. Even in the first and second modifications of the first embodiment, the second embodiment, and the supplementary explanation, the spacing between antenna elements is a cellular phone equipped with an array antenna. And the phase directivity pattern of each antenna element may be determined.

(5) In the first embodiment described above, the electronically controlled waveguide antenna is intended for the case where the number of linear conductors with varactor diodes around the feed element is three. The number of linear conductors is not limited to this, and may be two or four or more, for example.
For example, when an angle with respect to an axis passing through one antenna element and another antenna element and having a positive direction from one antenna element to another antenna element is φ, the one antenna element and the other antenna element Then, it is assumed that one of the linear conductors with varactor diodes is arranged at an azimuth angle φ of 90 °, and the other linear conductors with varactor diodes are arranged at regular intervals along the circumferential direction. . In this case, in one antenna element, the reactance values of the varactor diodes attached to the linear conductor are set in the order of X1, X2, X2, X2, in order from the linear conductor arranged at the position where the azimuth angle φ is 90 °. Let X3,... On the other hand, in other antenna elements, the reactance values of the varactor diodes attached to the linear conductors are set in the clockwise direction from the linear conductor arranged at the position where the azimuth angle φ is 90 °. Let X3,... The reactance values X1, X2, X3,... Are adjusted so that the phase directivity pattern of one antenna element becomes, for example, the phase directivity pattern of the antenna element RA1 shown in FIG. The

  (6) In the second embodiment, since the cellular phone has four antenna elements, the number of linear conductors with varactor diodes around the feeding element is four. For example, the following may be used. Assume that the cellular phone has M antenna elements (M is an integer of 3, 5 or more) antenna elements, and M is a linear conductor with a varactor diode around the power feeding element. In this case, for example, the reactance value of each varactor diode of each antenna element is set so that the reactance value of each varactor diode of the previous antenna element counterclockwise is rotated by one counterclockwise. Note that the number of antenna elements included in the mobile phone does not necessarily match the number of linear conductors with varactor diodes included in the antenna element.

  (7) In the above-described embodiment, the electronically controlled waveguide antenna is used for the antenna elements RA1 and RA2. However, the present invention is not limited to this, and an antenna element that can adjust the phase directivity pattern, such as a switching diversity antenna or metamaterial. An antenna may be used for the antenna elements RA1 and RA2. The same applies to the first and second modifications of the first embodiment, the second embodiment, and the antenna element described in the supplement.

  (8) In the above embodiment, the phase directivity patterns of the antenna elements RA1 and RA2 of the mobile phone R are different from each other. However, the present invention is not limited to this, and the antenna elements TA1 and TA2 of the base station T The phase directivity patterns may be different from each other. Further, the phase directivity patterns of the antenna elements RA1 and RA2 of the mobile phone R may be different from each other, and the phase directivity patterns of the antenna elements TA1 and TA2 of the base station T may be different from each other. Note that the method of setting the phase directivity pattern of the antenna elements TA1 and TA2 of the base station T is substantially the same as the method of setting the phase directivity pattern of the antenna elements RA1 and RA2 of the mobile phone R. Note that the first and second modified examples of the first embodiment, the second embodiment, and the phase directivity pattern setting method described in the supplement are applied to the base station T. Also good.

(9) The array antenna described in the above embodiment has been described by taking a base station and a mobile phone as examples. However, the present invention is not limited to this and can be applied to various communication devices.
(10) The array antenna described in the above embodiment can be applied to MIMO, for example, SIMO (Simple Input Multiple Output).

  The present invention can be used for an array antenna including a plurality of antenna elements.

T base station TA1, TA2 antenna element R mobile phone RA1, RA2 antenna element 11, 12 front end unit 13 channel estimation unit 14 weight generation unit 15 separation / combination unit 16 demodulation unit 31 feeding element 32 AC power source 33, 35, 37 linear Conductor 34, 36, 38 Varactor diode

Claims (10)

An array antenna having M antenna elements (M is an integer of 2 or more),
Each antenna element is a directional antenna;
The array antenna is characterized in that the phase directivity pattern of each antenna element is set so that the phase directivity patterns of the antenna elements are different from each other. The i-th phase directivity pattern of the i-th (i is an integer from 1 to M) antenna element and the j-th phase directivity pattern of the j-th (j is an integer other than i from 1 to M). Is
The phase difference generated based on the phase directivity of the i-th and j-th antenna elements is added to the phase difference generated based on the arrangement of the i-th and j-th antenna elements for signals coming from the same transmission source. The array antenna according to claim 1, wherein the array antenna is set so as to satisfy the following relationship. The i-th antenna element passes from the i-th antenna element to the j-th antenna element through the i-th (i is an integer from 1 to M) antenna element and the j-th (j is an integer from 1 to M and other than i) antenna elements. When the angle with respect to the axis whose forward direction is the positive direction is φ, φ is 0 ° to 360 °,
When 0 ° ≦ φ <φ _A1 ° (φ _A1 is a predetermined value smaller than 90) and φ _B1 ° (φ _B1 is a predetermined value larger than 270) <φ ≦ 360 °, the jth of the jth antenna element The phase of the phase directivity pattern is ahead of the phase of the i th phase directivity pattern of the i th antenna element;
φ _A2 ° (φ _A2 is 90 larger than the predetermined _{value) <φ <φ B2 ° (} φ B2 is 270 less than the predetermined value greater than phi _A2) in the phase of the phase directional pattern of a i is the j 2. The array antenna according to claim 1, wherein the i-th and j-th phase directivity patterns are set so as to be in a relation of advancing from the phase of the phase directivity pattern. phi _A1 and phi _A2 is approximately 90, the array antenna of claim 3, wherein the the phi _B1 and phi _B2 is approximately 270. The phase directivity pattern of each antenna element is
A pattern that monotonously increases when 0 ° ≦ φ ≦ 180 °, monotonously decreases when 180 ° ≦ φ ≦ 360 °, and monotonically decreases when 0 ° ≦ φ ≦ 180 °, and monotonically increases when 180 ° ≦ φ ≦ 360 °. The array antenna according to any one of claims 3 and 4, wherein the array antenna is any one of the following patterns. When the two antenna elements are paired, the phase directivity patterns of the antenna elements are set so that the phase directivity patterns of the two antenna elements are opposite to each other. The array antenna according to any one of claims 1 to 5. M is 2;
Each of the antenna elements is an electronically controlled waveguide antenna that includes a plurality of linear conductors with varactor diodes that are provided with a feeding element and are arranged around the circumference at regular intervals.
One of the linear conductors of each of the antenna elements passes through one antenna element and the other antenna element, and is 90 ° with respect to an axis whose forward direction is from one antenna element to the other antenna element. Placed in position,
The array antenna according to claim 1, wherein the counterclockwise arrangement of the reactance values of the varactor diodes of one antenna element is equal to the arrangement of the reactance values of the varactor diodes of the other antenna elements in the clockwise direction. The M antenna elements are arranged on an apex of an M square,
The array antenna according to claim 2, wherein the phase directivity pattern of each antenna element is obtained by sliding the phase directivity pattern of the previous antenna element counterclockwise in the azimuth direction. . The sliding angle is approximately φ,
The array antenna according to claim 8, wherein φ is a value obtained by dividing 360 by M. Each of the antenna elements is an electronically controlled waveguide antenna that includes a plurality of linear conductors with varactor diodes that are provided with a feeding element and are arranged around the circumference at regular intervals.
The reactance value of each varactor diode of each antenna element is obtained by rotating the reactance value of each varactor diode of the previous antenna element counterclockwise by one counterclockwise. The array antenna according to claim 1.

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