JP2009267450A - Receiving device and receiving method - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a receiving device and a receiving method that prevent the signal reception characteristics of a receiving device from becoming worse, due to inter-symbol interference, inter-carrier interference, and inter-code interference. <P>SOLUTION: The receiving device includes a replica generating unit which generates a replica signal as a replica of a transmitted signal, based on a received signal; an interference canceling unit which removes inter-code interference from the received signal, by using the replica signal; an arrival wave removing unit which removes an arrival wave from the received signal at each predetermined time band using the replica signal; a compositing unit which puts together signals, obtained after the arrival-wave removing unit removes the arrival waves in each predetermined time band; and a demodulating unit which performs demodulation processing with respect to the signal composited by the compositing unit. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention relates to a receiver and a receiving method.

MC-CDMA (Multi Carrier-Code Division: Code Division Multiplexing) is a combination of multi-carrier transmission schemes such as OFDM (Orthogonal Frequency Division Multiplexing) and CDM (Code Division Multiplexing). Carrier code division multiple access), MC-CDM (Multi Carrier Code Division Multiplexing), Spread-OFDM, and the like.
In these systems, good characteristics can be obtained in a multipath fading environment by obtaining frequency diversity effect by arranging data multiplied by coding and spreading codes over subcarriers.

FIGS. 19A and 19B are diagrams illustrating an example of a relationship between subcarriers and orthogonal codes corresponding to the subcarriers in the MC-CDM system. In FIG. 19A and FIG. 19B, the horizontal axis represents frequency.
FIG. 19A shows eight subcarriers in the MC-CDM system as an example. FIG. 19B shows three types of C _8,1 , C _8,2 , and C _8,7 as orthogonal codes corresponding to each subcarrier.
_{Here, C 8,1 = (1,1,1,1,1,1,1,1),} C 8,2 = (1,1,1,1, -1, -1, -1, - _{1), C 8,7 = (1} , -1, -1,1,1, -1, a -1).

One of the features of the MC-CDM system is that data can be multiplexed and communicated by multiplying data with these three types of orthogonal codes using the same time and the same frequency. Yes.
Note that the three types of orthogonal codes C _8,1 , C _8,2 and C _8,7 are all orthogonal codes with a period of 8, and data is separated between orthogonal codes by performing addition during one period. It can be performed. Note that SF _freq in FIG. 19A indicates the period of the orthogonal code.

20 (a) and 20 (b), FIG. 21 (a) and FIG. 21 (b) show the code C ′ _8, when the MC-CDM system signal propagates in the air and is received by the receiver _{. 1, C '8,2, C'} 8,7, C '' 8,1, C '' 8,2, which is a diagram illustrating a C _{'' 8,7.} In these figures, the horizontal axis represents frequency.
20A and 20B show a case where there is no frequency fluctuation in the period of the orthogonal code. In FIG. 20B, C ′ _8,1 = (0.5, 0.5, 0.5, 0.5, 0.5, 0.5, 0.5, 0.5), C ′ _{8 , 2} = (0.5, 0.5, 0.5, 0.5, -0.5, -0.5, -0.5, -0.5), C _8,7 = (0.5 , -0.5, -0.5, 0.5, 0.5, -0.5, -0.5, 0.5).
At this time, despreading is performed using the orthogonal code _C8,1 . That is, when the inner product with the orthogonal code C _8,1 is taken, that is, when all the values in the period SF _freq are added, the orthogonal code C ′ _8,1 becomes 4, and the orthogonal code C ′ _8,2 , C ′ _{8, 7} becomes 0. Such a situation is said to maintain the orthogonality between codes.

FIGS. 21A and 21B show a case where there is a frequency variation in the period of the orthogonal code. In FIG. _{21 (b), C ''} 8,1 = (1,1,1,1,0.25,0.25,0.25,0.25), C '' 8,2 = (1, 1,1,1, -0.25, -0.25, -0.25, -0.25), C '' 8,7 = (1,1,1,1,0.25, -0. 25, -0.25, 0.25).
At this time, when despreading with the orthogonal code C _8,1 , the orthogonal code C ″ _8,1 becomes 5, the orthogonal code C ″ _8,2 becomes 3, and the orthogonal code C ″ _8,7 becomes 0. Become. That is, orthogonal codes C '' _{8, 1} and orthogonal code C 'interference components between the' _8,2 is present, the situation where the orthogonality is not maintained between codes.
As described above, when the frequency fluctuation of the propagation path is fast (fluctuates fast in the frequency direction), in the MC-CDM system, inter-code interference (MCI: Multi-Code Interference) causes deterioration of characteristics.

  One method for improving characteristic degradation due to the loss of orthogonality between the codes is described in Patent Document 1, Non-Patent Document 1, and Non-Patent Document 2. In these prior arts, there is a difference between downlink and uplink, but in order to remove inter-code interference due to code multiplexing at the time of MC-CDM communication, it is desirable to use data after error correction or after despreading. The characteristics are improved by removing signals other than the code.

  On the other hand, in multi-carrier transmission, if there is a delayed wave exceeding a guard interval (GI), an inter-symbol interference (ISI) caused by a previous symbol entering an FFT (Fast Fourier Transform) section. : Inter Symbol Interference) and inter-carrier interference (ICI: Inter Carrier Interference) caused by a symbol break in the fast Fourier transform section, that is, a signal discontinuous section.

FIG. 22 is a diagram illustrating a signal reaching the receiver from the transmitter via the multipath environment. In FIG. 22, the horizontal axis represents time. A guard interval (GI) obtained by copying the second half of the symbol is added in front of the symbol.
When synchronizing with the direct wave s1 (the first wave that arrives) and performing FFT processing in the interval t3, the delay wave s2 indicates that the delay time falls within the delay t2 within the guard interval, and the delay wave s3 is the guard A delayed wave having a delay t3 exceeding the interval is shown. Direct waves and delayed waves are also referred to as incoming waves.

A hatched portion in front of the delayed wave s3 indicates a portion where the symbol preceding the desired symbol has entered the FFT section of the desired symbol, and the shaded portion is an intersymbol interference (ISI) component. Further, in the delayed wave s3, a symbol break occurs in the interval t3, which causes inter-carrier interference (ICI).
JP 2005-198223 A Y. Zhou, J. et al. Wang, M.M. Sawahashi, “Downlink Transmission of Broadband OFCDM Systems-Part I: Hybrid Detection,” IEEE Transactions on Communications, pp. 718-729, Vol. 53, no. 4, April 2005 Y. Zhou, J. et al. Wang, M.M. Sawahashi, “Downlink Transmission of Broadband OFCDM Systems-Part III: Turbo-Coded,” IEEE Journal on Selected Areas in Communications (JSAC), p. 132-140, Vol. 24, no. 1. January 2006

When a delayed wave exceeding the guard interval arrives at the receiver, in Non-Patent Document 1 and Non-Patent Document 2, signal removal other than the desired code is performed on the frequency domain signal (the signal after FFT processing). In addition, there is a problem that the accuracy of the replica used for removing signals other than the desired code is deteriorated due to intersymbol interference and intercarrier interference, and the characteristics are deteriorated.
In Patent Document 1, signal removal other than the desired code is performed on the signal in the time domain (the signal before the FFT processing), and intersymbol interference and intercarrier interference of the signal other than the desired code can be removed. Since the inter-symbol interference and inter-carrier interference of this signal remain, there is a problem that the characteristics deteriorate.

  As described above, in Patent Document 1 and Non-Patent Document 1, when inter-symbol interference, inter-carrier interference, and inter-code interference occur at the same time, inter-symbol interference and inter-carrier interference cannot be removed. There is a problem that inter-symbol interference or inter-carrier interference component is included in the replica, the replica accuracy is deteriorated, the performance of the canceller is deteriorated, and the reception characteristic of the signal in the receiver is deteriorated.

  The present invention has been made in view of the above circumstances, and a purpose thereof is to provide a receiver that can prevent signal reception characteristics from being deteriorated by inter-symbol interference, inter-carrier interference, and inter-code interference. It is to provide a receiving method.

(1) The present invention has been made to solve the above problems, and a receiver according to an aspect of the present invention includes a replica generation unit that creates a replica signal that is a replica of a transmission signal based on the received signal; An interference canceller that removes inter-code interference from the received signal using the replica signal, an incoming wave remover that removes an incoming wave from the received signal every predetermined time zone using the replica signal, and the incoming wave removal A synthesizing unit that synthesizes a signal from which an incoming wave is removed every predetermined time period, and a demodulating unit that demodulates the signal synthesized by the synthesizing unit.

(2) In addition, the interference canceller unit of the receiver according to one aspect of the present invention removes inter-code interference from the received signal for each predetermined time period using the replica signal.

(3) In addition, the incoming wave removal unit of the receiver according to one aspect of the present invention removes the incoming wave for each predetermined time period from the output signal of the interference canceller unit using the replica signal.

(4) In addition, the interference canceller unit of the receiver according to an aspect of the present invention removes inter-code interference in the frequency domain for each predetermined time zone using the replica created by the replica generation unit.

(5) In addition, the interference canceller unit of the receiver according to one aspect of the present invention removes inter-code interference in the time domain using the replica created by the replica generation unit.

(6) In addition, the incoming wave removal unit of the receiver according to one aspect of the present invention includes a delayed wave replica generation unit that creates a replica of an incoming wave for each predetermined time zone, and the delayed wave replica generation unit from a received signal Includes a subtracting unit that subtracts a replica of the incoming wave for each predetermined time period.

(7) In addition, the delayed wave replica generation unit of the receiver according to one aspect of the present invention sets the predetermined time zone based on the number of identified incoming waves.

(8) In addition, the delayed wave replica generation unit of the receiver according to one aspect of the present invention sets the predetermined time zone based on the time of the identified incoming wave.

(9) In addition, the delayed wave replica generation unit of the receiver according to one aspect of the present invention sets the predetermined time zone based on the received power of the identified incoming wave.

(10) In addition, the delayed wave replica generation unit of the receiver according to one aspect of the present invention sets the predetermined time zone so as to be a section that does not exceed the guard interval length.

(11) Further, the receiver according to one aspect of the present invention includes a signal determination unit that performs error correction decoding based on a result of the demodulation process performed by the demodulation unit and determines a signal for each bit, and the replica The generation unit creates a replica signal that is a replica of the transmission signal based on the determination value calculated by the signal determination unit.

(12) In addition, the signal determination unit of the receiver according to one aspect of the present invention performs error correction decoding based on a result of the demodulation process performed by the demodulation unit, and calculates a log likelihood ratio for each bit to be calculated. The judgment value.

(13) Further, a receiver according to an aspect of the present invention includes a propagation path / noise power estimation unit that estimates a noise power estimation value, and the synthesis unit is based on a channel impulse response estimation value and the noise power estimation value. Determine the filter coefficients.

(14) In addition, the combining unit of the receiver according to one aspect of the present invention includes the filter coefficient W _m represented by the formula (A) or the formula (B) or the filter coefficient W ′ _i represented by the formula (C). the _receiver of claim 13, wherein the use of _m (although, m is a natural number, H _{^ m} the transfer function of the m-th channel, ^{H ^} _{H m} is Hamiltonian H _{^ m, C mux} Is the number of multiplexed codes, σ ^ _N ² is an estimated value of noise power, i is a natural number equal to or less than the number of incoming wave removal units, and H ^ _{i, m} are transfer functions of the m-th propagation path in the i-th arrival wave removal unit , H ^ ^H _{i, m} is the Hamiltonian of H ^ _{i, m} ).

(15) In addition, the propagation path / noise power estimation unit of the receiver according to one aspect of the present invention generates a replica signal of the reception signal based on the replica signal generated by the replica signal generation unit and the channel impulse response estimation value. A reception signal replica generation unit to be generated; and a noise power estimation unit that estimates noise power by obtaining a difference between the replica signal generated by the reception signal replica generation unit and the reception signal.

(16) In addition, a reception method according to an aspect of the present invention includes a replica generation process of creating a replica signal that is a replica of a transmission signal based on the reception signal, and a predetermined time period from the reception signal using the replica signal. , An incoming wave removal process for removing the incoming wave, a synthesis process for synthesizing the signal from which the incoming wave is removed every predetermined time period in the incoming wave removal process, and a demodulation process for the signal synthesized in the synthesis process A demodulating process to be performed, and in the incoming wave removing process, inter-code interference is removed from the received signal for each predetermined time period using the replica signal.

(17) In addition, a reception method according to an aspect of the present invention includes a replica generation process of creating a replica signal that is a replica of a transmission signal based on the reception signal, and removing inter-code interference from the reception signal using the replica signal. An interference canceller process, an incoming wave removal process for removing an incoming wave from the output signal in the interference canceller process using a replica signal at a predetermined time period, and a predetermined time period at the incoming wave cancellation process. A synthesis process for synthesizing the signal from which the incoming wave has been removed; and a demodulation process for demodulating the signal synthesized in the synthesis process.

  In the receiver and the reception method of the present invention, it is possible to effectively prevent signal reception characteristics from being deteriorated by inter-symbol interference and inter-carrier interference in addition to inter-code interference.

(First embodiment)
First, a first embodiment of the present invention will be described. In the present embodiment, even when there is intersymbol interference (ISI) and interchannel interference (ICI) due to delayed waves exceeding the guard interval, and further there is intercode interference (MCI) due to frequency selectivity of the propagation path. A receiver capable of obtaining good characteristics will be described.

FIG. 1 is a schematic block diagram showing a configuration of a transmitter 100a according to the first embodiment of the present invention. The transmitter 100a includes an S / P (Serial / Parallel) conversion unit 1, code-by-code signal processing units 2-1 to 2-C _mux (C _mux is a code multiplexing number), a code multiplexing unit 8, and a pilot. Multiplexer 9, scrambling unit 10, IFFT (Inverse Fast Fourier Transform) unit 11, GI insertion unit 12, MAC (Media Access Control) unit 70, filtering processing unit 71, D / A A (Digital / Analog: digital / analog) converter 72, a frequency converter 73, and a transmission antenna 74 are provided.

Each code signal processing unit 2-1 to 2-C _mux includes an error correction coding unit 3, a bit interleaver unit 4, a modulation unit 5, a symbol interleaver unit 6, and a frequency-time spreading unit 7, respectively.

The information signal output from the MAC unit 70 is input to the S / P conversion unit 1, and the serial-parallel conversion output of the S / P conversion unit 1 is input to the code-by-code signal processing units 2-1 to 2 -C _mux . Is output.
The configuration of the code signal processing units 2-2 to 2-C _mux is the same as that of the code signal processing unit 2-1, so that the code signal processing unit 2-1 will be described as a representative. .

  The signal input to the code-by-code signal processing unit 2-1 is subjected to any error such as turbo coding, LDPC (Low Density Parity Check) coding, convolutional coding, or the like in the error correction coding unit 3. Correction coding processing is performed, and the output of the error correction coding unit 3 is bit-interleaved by the bit interleaver unit 4 in order to improve the occurrence of a burst error based on a drop in received power due to frequency selective fading. The order is changed in an appropriate order and output to the modulation unit 5.

  The output of the bit interleaver unit 4 is output from the BPSK (Binary Phase Shift Keying), QPSK (Quadrature Phase Shift Keying), 16QAM (16 Quadrature Amplitude Modulation Amplitude) in the modulation unit 5. Symbol modulation processing such as 16-value quadrature amplitude modulation) and 64 QAM (64 quadrature amplitude modulation) is performed and output to the symbol interleaver unit 6.

The output of the modulation unit 5 is changed in an appropriate order for each symbol by the symbol interleaver unit 6 in order to improve the burst error, and is output to the frequency-time spreading unit 7. The output of the symbol interleaver unit 6 is spread by a frequency-time spreading unit 7 with a predetermined spreading code (channelization code).
Here, orthogonal codes such as an OVSF (Orthogonal Variable Spread Factor) code and a Walsh-Hadamard code are used as the spreading code, but other spreading codes may be used.

Signals spread with different spreading codes are output as the outputs of the signal processing units 2-1 to 2-C _mux for each code, code multiplexed (addition processing) by the code multiplexing unit 8, and output to the pilot multiplexing unit 9 The Subsequently, the pilot multiplexing unit 9 inserts (multiplexes) a pilot signal (PICH: Pilot Channel) used for propagation path estimation or the like at a predetermined position. Insertion methods include time multiplexing, frequency multiplexing, code multiplexing, and the like.

Then, after being scrambled with a scrambling code unique to the transmitter 100 a in the scrambling unit 10, frequency-time conversion is performed in the IFFT unit 11 and output to the GI insertion unit 12.
After the guard interval is inserted in the GI insertion unit 12, a filtering process by the filtering unit 71, a digital / analog conversion process by the D / A conversion unit 72, a frequency conversion process by the frequency conversion unit 73, and the like are performed. To the receiver as a transmission signal.

In FIG. 1, both the bit interleaver unit 4 and the symbol interleaver unit 6 are arranged in the code-by-code signal processing units 2-1 to 2-C _mux , but only one of them may be arranged.
Moreover, it is not necessary to arrange both the bit interleaver unit 4 and the symbol interleaver unit 6 in the per-code signal processing units 2-1 to 2-C _mux .

FIG. 2 is a diagram illustrating an example of a frame format used in the first embodiment of the present invention. This figure shows the frame format of the multicarrier signal transmitted from the transmitter 100a to the receiver 200a (FIG. 3).
In FIG. 2, time is taken on the horizontal axis and received power is taken on the vertical axis. As shown in FIG. 2, the pilot signal (PICH) is arranged before and after the frame and in the middle.
Data traffic signals (DTCH: Data Traffic Channel) used for data transmission are arranged in the first half and the second half of a frame, and signals spread by C _mux different spreading codes are code-multiplexed. Here, the case of C _mux = 4 is shown when four pieces of data are stacked. Further, the ratio of the reception power of the pilot signal (PICH) and the reception power per code of the data traffic signal (DTCH) is represented by P _{PICH / DTCH} .

FIG. 3 is a schematic block diagram showing the configuration of the receiver 200a according to the first embodiment of the present invention. The receiver 200a includes a symbol synchronization unit 21, a propagation path / noise power estimation unit 22, a signal detection unit 23, code-by-code decoding units 24-1 to 24-C _mux (also referred to as a signal determination unit), a replica generation unit 28, A P / S (Parallel / Serial) converter 39, a receiving antenna 75, a frequency converter 76, and an A / D (Analog / Digital) converter 77 are provided.

The replica generation unit 28 includes code-specific symbol generation units 29-1 to 29 -C _mux , a code multiplexing unit 34, a pilot multiplexing unit 35, a scrambling unit 36, an IFFT unit 37, and a GI insertion unit 38.
The replica generation unit 28 generates a replica signal that is a replica of the transmission signal based on the reception signal r (t). More specifically, the replica generation unit 28 creates a replica signal that is a replica of the transmission signal based on the log likelihood ratio calculated by the decoding unit 26.

Each code symbol generation unit 29-1 to 29 -C _mux includes a bit interleaver unit 30, a symbol generation unit 31, a symbol interleaver unit 32, and a frequency-time spreading unit 33.
Each code decoding unit 24-1 to 24 -C _mux includes a bit deinterleaver unit 25, a decoding unit 26, and an adding unit 27.

The received signal received by the receiving antenna 75 is subjected to frequency conversion processing by the frequency conversion unit 76 and analog-digital conversion processing by the A / D conversion unit 77, and then symbol synchronization is performed in the symbol synchronization unit 21 as a digital reception signal r (t). Done.
The symbol synchronization unit 21 performs symbol synchronization using a correlation characteristic between a guard interval (GI) and an effective signal interval, and performs subsequent signal processing based on the result.
Subsequently, the propagation path estimation / noise power estimation unit 22 uses the pilot signal (PICH) included in the received signal to estimate the channel impulse response and the noise power estimation value.
As a propagation path estimation method, a replica signal of a pilot signal is created, and an RLS (Recursive Least Square) algorithm is performed so that the square error of the absolute value is minimized, or the received signal and the pilot signal are There are various methods for obtaining the cross-correlation with the replica by taking the time axis or the frequency axis, but the present invention is not limited to this.

  In addition, regarding the noise power estimation method, a method of creating a replica of a pilot signal using the estimated channel impulse response from the received pilot signal and obtaining the difference between these can be used, but the method is not limited thereto. It is not a thing.

The channel impulse response and the noise power estimation value output from the propagation path / noise power estimation unit 22 are input to the signal detection unit 23 and used to calculate the log likelihood ratio for each bit. Examples of the signal detection unit include a MAP detector (a maximum posterior probability detector, using a maximum posterior probability (MAP) decoding method (described later)).
At the first time, the signal detection unit 23 outputs a log likelihood ratio for each bit using the received signal, the channel impulse response, and the noise power estimation value. The log likelihood ratio is a value indicating whether the received bit is most likely 0 or 1 and is calculated based on the bit error rate of the communication channel.

In FIG. 3, C _mux outputs are output to the code-by-code decoding units 24-1 to 24-C _mux, which output log likelihood ratios of bits allocated to different spreading codes. .
Further, at the time of repetition to be described later, the log likelihood ratio for each bit is output using the received signal and the replica signal obtained from the decoding result, the channel impulse response, and the noise power estimation value.

Subsequently, in the code-by-code decoding units 24-1 to 24-C _mux , the bit deinterleaver unit 25 performs deinterleaving processing on the input signal for each bit. The deinterleaving process is the reverse of the interleaving process, and the order change by the interleaving process is restored.
The decoding unit 26 performs MAP decoding processing on the output of the bit deinterleaver unit 25. Specifically, the decoding unit 26 performs error correction decoding based on the result of the soft decision performed by the demodulation unit 50 (FIG. 4, which will be described later) of the signal detection unit 23, and calculates a log likelihood ratio for each bit. .
Note that the MAP decoding process is a soft decision such as log likelihood ratio including information bits and parity bits without performing hard decision at the time of normal error correction decoding such as turbo decoding, LDPC decoding, and Viterbi decoding. It is a method of outputting the result. That is, the hard decision is determined based on only 0 or 1 of the received signal, while the soft decision is determined based on information on how likely the soft decision is (soft decision information).

Subsequently, a difference λ 2 between the input of the decoding unit 26 and the output of the decoding unit 26 is calculated by the adding unit 27 and output to the replica generation unit 28.
The input to the replica generation unit 28 is input to the bit interleaver unit 30, and the bit interleaver unit 30 outputs λ 2 for each bit and outputs to the symbol generation unit 31.
The output of the bit interleaver unit 30 is subjected to symbol modulation processing in the symbol generation unit 31 with the same modulation scheme (BPSK, QPSK, 16QAM, 64QAM, etc.) as the transmitter 100a in consideration of the magnitude of λ2.
The output of the symbol generator 31 is changed in order for each symbol by the symbol interleaver 32, and the output of the symbol interleaver 32 is spread by the frequency-time spreader 33 with a predetermined spreading code (channelization code).

Note that the receiver 200a includes the code-decoding unit and the code-symbol generation unit for the code multiplexing number C _mux (C _mux is a natural number of 1 or more).
Signals (code channels) spread with different spreading codes are output from the symbol-by-code generation units 29-1 to 29 -C _mux, and are code multiplexed (added) by the code multiplexing unit 34. Further, the symbol-by-code generation units 29-1 to 29 -C _mux also input a signal (code channel) spread by the different spreading code to the signal detection unit 23. Subsequently, in the pilot multiplexing unit 35, a pilot signal used for propagation path estimation or the like is inserted (time multiplexed) at a predetermined position. Thereafter, the signal is scrambled by a scrambling code unique to the transmitter 100 a in the scrambling unit 36 and then input to the IFFT unit 37. The IFFT unit 37 performs frequency-time conversion, and the GI insertion unit 38 inserts a guard interval, and then outputs the signal to the signal detection unit 23. The output from the scrambling unit 36 is used for signal processing at the time of repetition. The
Note that after the above iterative decoding operation has been performed a predetermined number of times, the output of the decoding unit 26 is input to the P / S conversion unit 39, subjected to parallel-serial conversion, and then output to the MAC unit (not shown) as a decoding result. The

FIG. 4 is a schematic block diagram showing an example of the configuration of the signal detection unit 23 (FIG. 3) according to the first embodiment of the present invention. Here, a MAP detector is applied to the signal detector 23. The signal detection unit 23 includes soft canceller units 45-1 to 45-3 (also referred to as incoming wave removal units), an MMSE (Minimum- Mean Square-Error) filter unit 46 (also referred to as a synthesis unit), and a code unit. Demodulating units 47-1 to 47-C _mux are provided.

The soft canceller units 45-1 to 45-3 each include a delayed wave replica generation unit 41, an addition unit 42 (also referred to as a subtraction unit), a GI removal unit 43, an FFT unit 44, an MCI replica generation unit 51, and an addition unit 52. ing.
The delayed wave replica generation unit 41 and the addition unit 42 are also referred to as a delayed wave canceller unit 53. In addition, the MCI replica generation unit 51 and the addition unit 52 are also referred to as an interference canceller unit 54.

The soft cancellers 45-1 to 45-3 remove the delayed wave from the received signal r (t) every predetermined time zone using the time domain replica signal created by the replica generator 28.
The delayed wave replica generation unit 41 uses a channel impulse response estimation value that is a propagation path estimation value estimated from the received signal r (t) and a time domain replica signal s ^ (t generated by the replica generation unit 28 (FIG. 3). ) And a replica of the delayed wave for each predetermined time period is created.
The adder 42 subtracts the delayed wave replica for each predetermined time zone created by the delayed wave replica generator 41 from the received signal r (t). Further, the MCI replica generation unit 51 performs the frequency domain replica signal Sn ^ (k) output from each of the code-specific symbol generation units 29-1 to 29-C _mux of the replica generation unit 28 and the propagation path estimation. An MCI replica is generated based on the value transfer function.
The adder 52 removes MCI by subtracting the MCI replica from the output signal of the FFT unit 44.
Each code demodulator 47-1 to 47-C _mux includes a despreader 48, a symbol deinterleaver 49, and a demodulator 50.

The received signal r (t) input to the signal detection unit 23 is a delayed wave obtained based on the replica signal s ^ (t) input to the signal detection unit 23 and the channel impulse response estimated values h ^to (t). A difference from the output of the replica generation unit 41 is calculated by the addition unit 42, and this is output to the GI removal unit 43.
The guard interval (GI) is removed by the GI removal unit 43 and output to the FFT unit 44. The FFT unit 44 performs time frequency conversion on the input signal to obtain signals R ^to i. Further, the signals R 1 ^to i and the frequency domain replica signal S _n (k) (n = 1, 2,..., C _mux , k is a subcarrier) input to the signal detection unit 23, and The difference from the output from the MCI replica generation unit 51 obtained based on the transfer function H ^to (k) with respect ^{to the} channel impulse response estimated value h ^to (t) is calculated by the adding unit 52 and output to the MMSE filter unit 46 .
The signal detection unit 23 is provided with a soft canceller unit B (B is a natural number of 1 or more) blocks. Note that i is a natural number and 1 ≦ i ≦ B.
FIG. 4 is an example in the case of B = 3, and the number of soft cancellers is determined by the number of predetermined time zones in which delayed waves are removed (number of divided delayed waves into blocks).

Subsequently, the MMSE filter unit 46 synthesizes signals from which the delayed waves are removed by the soft canceller units 45-1 to 45-3 for each predetermined time period. Specifically, the MMSE filtering process is performed in the MMSE filter unit 46 using the outputs R ^to i ′ of the soft canceller unit, the channel impulse response estimated value, and the noise power estimated value, and the signal Y ′ is obtained.

Using this signal Y ′, the C _mux code-by-code decoding units 47-1 to 47-C _mux output a log likelihood ratio for each bit in each code. The signals input to the code-by-code decoding units 47-1 to 47-C _mux are signals obtained by descrambling the signal Y ′ with respect to the scramble unit 10 of the transmitter 100.
The despreading unit 48 performs a despreading process using each spreading code. The symbol deinterleaver unit 49 replaces the output of the despreading unit 48 for each symbol.
The demodulator 50 performs a soft decision on the signal synthesized by the MMSE filter unit 46. The demodulator 50 outputs the log likelihood ratio λ1 for each bit as the soft decision result with respect to the output of the symbol deinterleaver 49.
The demodulator 50 calculates the log likelihood ratio λ1 by using the following formulas (1) to (3). That is, if the output of the nth symbol of the symbol deinterleaver unit 49 is Zn, the soft decision result λ1 at the time of QPSK modulation can be expressed by the following equations (1) and (2).

  Here, R [] indicates a real part in parentheses, Im [] indicates an imaginary part in parentheses, and μ (n) indicates a reference symbol (amplitude of a pilot signal) of n symbols. The modulated signal can be expressed by the following formula (3).

  Although an example of QPSK modulation is shown here, the soft decision result (log likelihood ratio) λ1 for each bit can be similarly obtained in other modulation schemes.

3 and 4, the bit interleaver unit 30, the bit deinterleaver unit 25, the symbol interleaver unit 32, and the symbol deinterleaver unit 49 are both arranged. Only the bit deinterleaver unit 25 or only the symbol interleaver unit 32 and the symbol deinterleaver unit 49 may be used.
Further, all of the bit interleaver unit 30, the bit deinterleaver unit 25, the symbol interleaver unit 32, and the symbol deinterleaver unit 49 may not be arranged.

FIG. 5 is a flowchart showing a process of the receiver 200a according to the first embodiment of the present invention. First, the signal detection unit 23 determines whether or not the operation is an initial operation (step S1). If it is determined in step S1 that the operation is the first operation, the GI removal unit 43 removes the guard interval (GI) from the received signal r (t) (step S2).
Then, the FFT unit 44 performs FFT processing (time frequency conversion processing) (step S3). Next, the MMSE filter unit 46 performs normal MMSE filter processing (step S4).

Then, the despreading unit 48 performs a despreading process (step S5). Next, the symbol deinterleaver unit 49 performs symbol deinterleaver processing (step S6). Then, the demodulator 50 performs soft decision bit output processing (step S7).
Next, the bit deinterleaver unit 25 performs a bit deinterleaver process (step S8). And the decoding part 26 performs a decoding process (step S9). Next, it is determined whether or not the processes in steps S5 to S9 described above have been repeated a predetermined number of times (step S10).
As described with reference to FIG. 3, the processing may be performed in C _mux circuits arranged in parallel. The first MMSE filter process will be described later.

If it is determined in step S10 that the processes in steps S5 to S9 have not been repeated a predetermined number of times, the bit interleaver unit 30 bit-interleaves the log likelihood ratio using the decoding result λ2 for the C _mux code ( Step S11).
Then, the symbol generator 31 creates a modulated signal replica (step S12). Next, the symbol interleaver unit 32 performs symbol interleaver processing (step S13). Then, the frequency-time spreading unit 33 performs spreading processing using a predetermined spreading code (step S14).

After the processes in steps S11 to S14 described above are repeated C _mux times, the code multiplexing unit 34 performs code multiplexing (step S15). And the pilot multiplexing part 35 multiplexes a pilot signal (step S16).
Next, the scrambling unit 36 performs a scrambling process (step S17). Then, the IFFT unit 37 performs IFFT processing (step S18). Next, the GI insertion unit 38 inserts a guard interval (GI) (step S19). The time-domain replica signal in which the guard interval (GI) is inserted in step S19 and the frequency-domain replica signal after code multiplexing in step S15 are used during repeated demodulation.

If it is determined in step S1 that it is a repetition time, that is, it is not the first operation, the soft canceller units 45-1 to 45-3 remove all but the predetermined delay wave for each block (step S20).
Then, the GI removal unit 43 performs a guard interval (GI) removal process (step S21). Next, the FFT unit 44 performs an FFT process (step S22). Then, the inter-code interference (MCI) is removed by removing the MCI replica from the signal converted into the frequency domain by the FFT processing (step S23). The removal of the MCI replica in step S23 is performed for all code channels (for C _mux codes) that have been code-multiplexed.

After performing the above-described processing of steps S20 to S23 for B (B is a natural number) blocks, the MMSE filter unit 46 combines the output signals from the B block with the MMSE filter according to the least square error criterion. That is, MMSE filter processing is performed (step 24). In step S24 and subsequent steps, the process proceeds to step S5, and the same process as the initial process is performed.
Steps S1 to S9 and steps S11 to S24 are repeated until it is determined in step S10 that the above-described processing has been repeated a predetermined number of times.

Next, the processing of the soft canceller units 45-1 to 45-3 will be specifically described. Here, operations of the delayed wave replica generation unit 41 and the addition unit 42 of the i-th soft canceller unit 45-i will be described.
First, in the soft canceller unit 45-i, the delayed wave replica generation unit 41 generates h _i and performs a convolution operation with the replica signal s ^ (t) from the received signal r (t). To do. This is the output of the adder 42 (where i is a natural number where i ≦ B (the natural number B will be described later)).

FIG. 6 is a diagram illustrating an example of a channel impulse response estimation value according to the first embodiment of the present invention. In FIG. 6, the horizontal axis indicates time, and the vertical axis indicates received power.
Here, the case where the channel impulse response estimation value obtained from the propagation path / noise power estimation unit 22 is obtained will be described. A case where 6-path channel impulse response estimation values P1 to P6 are obtained will be described.
Here, in the soft canceller units 45-1 to 45-3, the six-path delayed wave is decomposed into three delayed waves of two paths. That is, P1 to P6 are decomposed into P1 and P2, P3 and P4, P5 and P6.

FIG. 7 is a diagram illustrating an example of a channel impulse response estimation value in the soft canceller unit 45-1 according to the first embodiment of the present invention. In FIG. 7, the horizontal axis indicates time, and the vertical axis indicates received power.
As shown in FIG. 7, first, in the soft canceller unit 45-1, the third path (P3), the fourth path (P4), the fifth path (P5), and the sixth path (P6) surrounded by a dotted line are performed. It is defined as h ₁ (t) and is generated by the delayed wave replica generation unit 41.

The output of the delayed wave replica generation unit 41 is obtained by convolution of h ₁ (t) and s ^ (t), and the output of the addition unit 42 is obtained from the received signal r (t) and h ₁ (t) The result of subtracting the convolution operation of s ^ (t) is obtained.
That is, when the replica is correctly generated, the output of the adder 42 can be considered as a signal received via the propagation path represented by (h (t) −h ₁ (t)). . As a result, the signals P1 and P2 received via the propagation path indicated by the solid line in FIG. t1 represents the maximum delay time of the signal output from the subtraction unit of the soft canceller unit 45-1.

FIG. 8 is a diagram illustrating a channel impulse response estimation value in the soft canceller unit 45-2 according to the first embodiment of the present invention. In FIG. 8, the horizontal axis indicates time, and the vertical axis indicates received power.
As shown in FIG. 8, in the soft canceller unit 45-2, the first path (P1), the second path (P2), the fifth path (P5), and the sixth path (P6) surrounded by dotted lines are h. ₂ (t), which is created by the delayed wave replica generation unit 41.
The output of the delayed wave replica generation unit 41 is obtained by convolution of h ₂ (t) and s ^ (t), and the output of the addition unit 42 is obtained from the received signal r (t) and h ₂ (t) The result of subtracting the convolution operation of s ^ (t) is obtained.

That is, when the replica is correctly generated, the output of the adding unit 42 can be considered as a signal received through the propagation path represented by (h (t) −h ₂ (t)). .
As a result, the signals P3 and P4 received through the propagation path indicated by the solid line in FIG. t2 represents the maximum delay time of the signal output from the subtraction unit of the soft canceller unit 45-2.

FIG. 9 is a diagram illustrating an example of a channel impulse response estimation value in the soft canceller unit 45-3 according to the first embodiment of the present invention. In FIG. 9, the horizontal axis indicates time, and the vertical axis indicates received power.
As shown in FIG. 9, in the soft canceller unit 45-3, the first path (P1), the second path (P2), the third path (P3), and the fourth path (P4) surrounded by dotted lines are h. ₃ (t), which is created by the delayed wave replica generation unit 41.

The output of the delayed wave replica generating unit 41 is obtained by convolution of h ₃ (t) and s ^ (t), and the output of the adding unit 42 is obtained from the received signal r (t) and h ₃ (t) The result of subtracting the convolution operation of s ^ (t) is obtained. That is, when the replica is correctly generated, the output of the adder 42 can be considered as a signal received through the propagation path represented by (h (t) −h ₃ (t)). .
Thus, the signals P5 and P6 received through the propagation path indicated by the solid line in FIG. t3 indicates the maximum delay time of the signal output from the subtraction unit of the soft canceller unit 45-3.

  In the description of FIGS. 7 to 9, the case where the soft canceller units 45-1 to 45-3 set a predetermined time zone based on the number of identified delayed waves has been described. That is, a case has been described in which the replica signal to be generated and subtracted is changed for each of the soft canceller units 45-1 to 45-3 based on the number of identified delayed waves based on the channel impulse response estimated value. 7 to 9 are examples in which the delayed wave is divided into two waves. In addition to this method, the following method can be used.

  For example, the soft cancellers 45-1 to 45-3 set a predetermined time zone based on the identified delayed wave time. That is, the arrival time of the delayed wave may be divided into B pieces, and it may be determined which soft canceller unit performs processing according to which time zone the delayed wave has reached. That is, the replica signal to be created and subtracted for each soft canceller unit may be changed based on the identified delayed wave time.

FIG. 10 is a diagram for describing a case where a predetermined time zone is set based on the time of the identified delayed wave. In FIG. 10, the horizontal axis represents time and the vertical axis represents received power. FIG. 10 shows a case where a predetermined time zone is set based on the guard interval (GI) length as an example. In FIG. 10, a time difference between a wave estimated to have arrived first and a wave estimated to have arrived last is assumed to be T _all .
Assuming that the predetermined time zone is set based on the guard interval (GI) length T _GI , the delay wave is divided for each guard interval length T _{GI in} order from the arrival time of the wave that has arrived first, and the divided time Each soft canceller performs processing for each band.

In FIG. 10, the delayed wave is divided into three time zones A to C so that each time zone has a guard interval length T _GI (example of B = 3), and time zone A is soft canceller 45-1. The time zone B is processed by the soft canceller unit 45-2, and the time zone C is processed by the soft canceller unit 45-3.

  Further, the soft canceller units 45-1 to 45-3 may set a predetermined time zone based on the received power of the identified delayed wave. That is, all received signals may be divided into B signals in order of arrival time so that the received signals included in the delay wave are substantially constant, and based on this, which soft canceller unit is to be processed may be determined. That is, the replica signal to be generated and subtracted for each soft canceller unit may be changed based on the received power of the identified delayed wave.

FIG. 11 is a diagram illustrating a case where all received signals are divided into three so that the received signals included in the delay wave are substantially constant in the order of arrival times. In FIG. 11, the horizontal axis represents time, and the vertical axis represents received power. FIG. 11 shows a channel impulse response estimated value when a propagation path is estimated that there is an incoming wave from time t ₁₀ when the received signal first arrived to time t ₁₁ delayed by time T _all .
When the total received power at the time T _all where the incoming wave exists is P _all , the impulse response is set to the time zone T1 so that the signal power values of the regions B1, B2, and B3 that can be divided are P _all / B. , T2, and T3.

Then, the time zone T1 is processed by the soft canceller unit 45-1, the time zone T2 is processed by the soft canceller unit 45-2, and the time zone T3 is processed by the soft canceller unit 45-3. .
The total received power can be calculated, for example, by integrating in the time axis direction at time T _all when the incoming wave exists.

  By setting the predetermined time zone as described above (FIG. 11), the maximum delay times T1 to T3 of the signal output from the subtracting section of each soft canceller section are set to be shorter than the guard interval (GI) length. Even when a long delay wave exceeding the guard interval (GI) arrives, intersymbol interference (ISI) and interchannel interference (ICI) can be reduced.

Next, the operation of the interference canceller unit that removes inter-code interference (MCI) of the signal detection unit 23 in FIG. 4 will be described.
The MCI replica generation unit 51 includes frequency-domain replica signals S _n ^ (k) output from each of the code generation units 29-1 to 29-C _mux of the replica generation unit 28 (where n = 1, 2, _.. , C _mux , k are subcarriers) and a transfer function converted to the frequency domain of the channel impulse response estimation value which is a propagation path estimation value, to generate an MCI replica.
When the soft canceller units 45-1 to 45-3 operate based on the example described with reference to FIGS. 7 to 9, the nth code channel (n = 1, 1) output from the MCI replica generation unit belonging to the soft canceller unit 45-1. 2,..., C _mux ) is represented by R _{1, n} ^ (k), and is expressed by the following equation (4), and all code channels (ie, code multiplexed) constituting the received signal r (t) ( MCI replicas are generated for each of (C _mux ).

However, fft [a] indicates that the signal a is converted from the time domain to the frequency domain.
Similarly, the soft canceller unit 45-2 or the MCI replica generation unit of the soft canceller unit 45-3 is expressed by the following equations (5) and (6).

Hereinafter, the MCI replicas for all code channels output from the MCI replica generation unit belonging to the soft canceller unit 45-b are collectively referred to as R ^ _b (k).

Next, the adding unit 52 subtracts the output signal of the MCI replica generation unit from the output signal from the FFT unit. Soft canceller units 45-1 to signals of the n code channel addition section 52 outputs R ^~ belonging to _{^1, n} '(k) and the following equation (7), and the code-multiplexed all code channels (C Subtraction is performed on each of ( _mux ). R ^~ ₁ (k) denotes the output signal from the FFT unit.

  Similarly, the output signals of the soft canceller unit 45-2 or the adder unit 52 of the soft canceller unit 45-3 are expressed by the following equations (8) and (9).

In the following description, signals for all code channels output from the adder 52 belonging to the soft canceller 45-b are collectively denoted as R ^to _b ′ (k).

  As described above, the delayed wave is removed so that the signal output from the delayed wave canceller 53 is in a time zone shorter than the guard interval length, and the FFT processing is performed on the signal in the time zone shorter than the guard interval length. By removing inter-code interference (MCI) in the frequency domain, even when a delay wave longer than the guard interval length arrives, the code is not affected by inter-symbol interference (ISI) and inter-channel interference (ICI). Interference (MCI) can be reduced.

FIG. 12A to FIG. 12C are diagrams showing channel impulse response estimation values and MMSE filter units in the initial processing according to the first embodiment of the present invention. Here, the MMSE filter unit 46 shown in FIG. 4 and the operations of steps S4 and S23 shown in FIG. 5 will be described.
First, the operation of the first MMSE filter unit 46 will be described. When the received signal is expressed in the frequency domain, the received signal R can be expressed as the following Expression (10).

  Here, H ^ represents the transfer function of the estimated propagation path, and can be represented by a diagonal matrix of Nc * Nc, assuming that only a delayed wave within the guard interval exists. Nc indicates the number of sub-carriers in spread-OFCDM. H ^ can be expressed as the following equation (11).

  S represents a transmission symbol, and can be represented by a vector of Nc * 1, as shown in the following equation (12).

  Similarly, the received signal R and the noise component N can be represented by a vector of Nc * 1, as shown in the following equations (13) and (14).

In Expressions (12) to (14), T used as a subscript represents a transposed matrix.
When such a received signal is received, the output Y of the MMSE filter unit 46 can be represented by a vector of Nc * 1, as shown in the following equation (15).

  The MMSE filter unit 46 determines the MMSE filter coefficient W based on the channel impulse response estimated value and the noise power estimated value. Here, the MMSE filter coefficient W can be represented by a diagonal matrix of Nc * Nc as shown in the following equation (16).

Further, each element of the MMSE filter coefficient W _m can be expressed by the following equation (17) at the time of frequency direction spreading.

  In addition,

Is the interference component from other codes when code is multiplexed,

Indicates the estimated noise power. The subscript H indicates Hamiltonian (conjugate transposition).
Further, each element of the MMSE filter coefficient W _m can be expressed by the following equation (18) on the assumption that the orthogonality between codes is maintained at the time direction spreading.

12 (a) to 12 (c) show how the signal that has passed through the propagation path shown in FIG. 6 is input to the MMSE filter unit 46 based on the coefficients in the initial processing.
FIG. 12A shows the channel impulse responses P1 to P6 shown in FIG. FIG. 12B shows a transfer function in which the channel impulse responses P1 to P6 are expressed on the frequency axis.

Next, the operation of the MMSE filter unit at the time of repetition will be described. First, the replica signal r ^ _i used in the i-th soft canceller unit 45-i at the time of repetitive demodulation can be expressed as the following Expression (19).

Here, h ^ _i is a delay profile obtained by extracting only the delayed wave to be processed in the i-th soft canceller unit 45-i. s ^ (t) is a replica signal calculated based on the log likelihood ratio λ2 obtained by the previous MAP decoding.

Indicates a convolution operation. Therefore, the output of the soft canceller unit 45-i, that is, the outputs R to _i of the i-th soft canceller unit 45 in FIG. 4 can be expressed as the following Expression (20).

  Here, Δ includes an error signal due to replica uncertainty and a thermal noise component. At this time, the output Y ′ of the MMSE filter unit 46 can be expressed by the following equation (21).

  Here, assuming that the replica signal is generated with high accuracy, and that Δ does not include a component due to the error of the replica but only includes a thermal noise component, a submatrix of the MMSE filter coefficient is expressed by the following equation (22). ) As a diagonal matrix.

  Further, the input signal to the MMSE filter unit 46 has a low frequency selectivity as described later, is in a state close to flat fading, and code interference is eliminated by the interference canceller unit. Assuming that there is no inter-code interference at the time, each element can be expressed by the following equation (23).

Incidentally, H ^ _{i ', m} is i' is a m-th transfer function of the propagation path in th soft canceller unit, H ^ _{i ',} ^{m H} the H ^ _i' is a _{Hamiltonian, m.}

FIGS. 13A to 13G are diagrams showing channel impulse response estimation values and MMSE filter units in the iterative processing according to the first embodiment of the present invention. FIGS. 13 (a) to 13 (g) show how the signals that have passed through the propagation paths shown in FIGS. 7 to 9 are input to the MMSE filter unit 46 based on the MMSE filter coefficients in the iterative processing. ing. Here, the number B of soft canceller block units is set to three.
The MMSE filter unit 46 uses the MMSE filter coefficient W _m represented by Expression (17) or Expression (18) at the time of the first demodulation, and uses the MMSE filter coefficient W ′ _{i, m} represented by Expression (23) at the time of repeated demodulation.

  As in FIG. 12A, FIGS. 13A, 13C, and 13E show the channel impulse responses P1 to P6 shown in FIGS. FIGS. 13B, 13D, and 13F show transfer functions in which the channel impulse responses P1 to P6 are expressed on the frequency axis. The horizontal axis represents frequency, and the vertical axis represents received power.

  FIG. 14 is a schematic block diagram showing the configuration of the propagation path / noise power estimation unit 22 (FIG. 3) according to the first embodiment of the present invention. The propagation path / noise power estimation unit 22 includes a propagation path estimation unit 61, a preamble replica generation unit 62, and a noise power estimation unit 63.

The propagation path estimation unit 61 estimates a channel impulse response using a pilot signal included in the received signal.
The preamble replica generation unit 62 creates a pilot signal replica signal using the channel impulse response estimation value obtained by the propagation path estimation unit 61 and the signal waveform of the pilot signal (PICH), which is known information.
The noise power estimation unit 63 estimates the noise power by taking the difference between the pilot signal part included in the received signal and the replica signal of the pilot signal output from the preamble replica generation unit 62.
Note that as a propagation path estimation method in the propagation path estimation unit 61, various methods such as a method of deriving based on a least square error criterion using an RLS algorithm or a method using a frequency correlation can be used.

According to the receiver 200a according to the first embodiment of the present invention, the delayed wave replica generating unit 41 uses the replica signal generated by the replica generating unit 28 to cause the delayed wave replica generating unit 41 to generate the delayed wave for each predetermined time period from the received signal r (t). Since the MMSE filter unit 46 synthesizes the signal from which the delayed wave is removed for each predetermined time zone, and the demodulator 50 makes a soft decision on the synthesized signal, the delayed wave is removed. It is possible to perform FFT processing on the received signal, and to reduce intersymbol interference (ISI) and interchannel interference (ICI) due to a delayed wave exceeding the guard interval length.
Furthermore, even in the case where a delayed wave exceeding the guard interval arrives by removing inter-code interference (MCI) in the frequency domain in the signal from which the delayed wave is removed for each predetermined time period, inter-symbol interference (ISI) Further, since the inter-code interference (MCI) can be removed using a replica that is less affected by the inter-channel interference (ICI), the removal accuracy is improved.
Further, since the inter-code interference (MCI) is removed in the frequency domain after the delay wave is removed for each predetermined time period, the magnitude of the inter-code interference MCI that changes depending on the size of the predetermined time period. Stable characteristics can be obtained without being affected. As a result, for example, the predetermined time zone can be flexibly set only by the GI length or the signal power for each time zone without considering the magnitude of the inter-code interference MCI.

(Modification of the first embodiment)
The radio receiver according to the modification of the present embodiment uses a different noise power estimation method compared to the first embodiment.

  FIG. 15 is a schematic block diagram illustrating a configuration of a wireless receiver according to a modification of the first embodiment of the present invention. The configuration of the wireless receiver is substantially the same as that of the wireless receiver according to the first embodiment (FIG. 3), but the propagation path / noise power estimation unit 22 in FIG. 3 is different. With respect to the input signal, only the received signal r (t) is input to the propagation path / noise power estimation unit 22 shown in FIG. 3, whereas the propagation path / noise power estimation unit 322 of FIG. r (t) and a replica signal s ^ (t) which is an output of the replica signal creation unit 28 are input.

FIG. 16 is a schematic block diagram showing a configuration of a propagation path / noise power estimation unit 322 (FIG. 15) according to a modification of the first embodiment of the present invention. The propagation path / noise power estimation unit 322 includes a propagation path estimation unit 61, a received signal replica generation unit 362, and a noise power estimation unit 363.
The propagation path estimation unit 61 estimates the channel impulse response using the PICH included in the received signal.
The reception signal replica generation unit 362 generates a replica of the reception signal r (t) based on the replica signal generated by the replica signal generation unit 28 and the channel impulse response estimated value. Specifically, the received signal replica generation unit 362 obtains the channel impulse response estimated value h ^ (t) obtained by the propagation path estimation unit 61, the PICH signal waveform that is known information, and the output of the MAP decoding unit 26. The replica signal s ^ (t) obtained from the obtained log likelihood ratio λ2 for each bit is used to create a PICH replica signal and a DTCH replica signal.
The noise power estimation unit 363 generates a replica signal generated by the reception signal replica generation unit 362.

And the received signal r (t), the noise power is estimated. Accordingly, the noise power estimation value calculated by the noise power estimation unit 363 can include both the error of the MAP decoding result and the Gaussian noise component, and the MMSE filter coefficient in the MMSE filter unit 46 can be obtained more appropriately.
Note that the noise power estimation method by the propagation path / noise power estimation unit 322 of the wireless receiver according to the present embodiment can also be applied to the wireless receiver according to the second embodiment.

(Second Embodiment)
Next, a second embodiment of the present invention will be described. In the second embodiment, when there is intersymbol interference (ISI) and interchannel interference (ICI) due to delayed waves exceeding the guard interval, and further there is intercode interference (MCI) due to frequency selectivity of the propagation path. A receiver capable of obtaining good characteristics will be described.
In addition, since the transmitter in 2nd Embodiment can use the same thing as the transmitter 100a (FIG. 1) by 1st Embodiment, the description is abbreviate | omitted.

  FIG. 17 is a schematic block diagram showing a configuration of a receiver 200b according to the second embodiment of the present invention. Compared to the first embodiment, the configurations of the replica generation unit and the signal detection unit are different. Hereinafter, a different part from 1st Embodiment is demonstrated.

The replica generation unit 128 in the receiver 200b of the second embodiment includes code-by-code symbol generation units 29-1 to 29-C _mux and code-by-code replica generation units 129-1 to 129-C _mux .
Each code symbol generation unit 29-1 to 29 -C _mux includes a bit interleaver unit 30, a symbol generation unit 31, a symbol interleaver unit 32, and a frequency-time spreading unit 33, which are the same as those in the first embodiment. By the operation, the replica signal S _n (k) (n = 1, 2,..., C _mux ) of each code channel is generated using the output signal from the decoding unit 26.

Each code replica generation section 129-1 to 129-C _mux includes a code multiplexing section 134, a pilot multiplexing section 135, a scrambling section 136, an IFFT section 137, and a GI insertion section 138.
Signals (code channels) S _n (k) spread by different spreading codes output from the symbol-by-code generation units 29-1 to 29 -C _mux (where n = 1, 2,..., C _mux , K are subcarriers) are code multiplexed (added) by the code multiplexing unit 134.
For example, the code multiplexing unit belonging to the code-by-code replica generation unit 129-1 code-multiplexes the code channel signals excluding the first code channel signal S ₁ (k). When the output signal of the code multiplexing unit belonging to the code-by-code replica generation unit 129-n is Sc ^ _n '(k), the following equation (24) is obtained.

Subsequently, in the pilot multiplexing unit 135, a pilot signal used for propagation path estimation or the like is inserted into a predetermined position in the output signal Sc ^ _n '(k) of the code multiplexing unit. Further, the pilot multiplexing unit 135 adds the S ^ to the output signal S ^ _n (k) from the symbol-by-code generation unit 29-n that becomes a desired signal in the code-specific replica generation unit 129-n to which the pilot multiplexing unit 135 belongs. Similar to _n ′ (k), a pilot signal used for propagation path estimation is inserted at a predetermined position. For example, pilot multiplexing section 135 belonging to code-by-code replica generation section 129-3 inserts a pilot signal into signal S ₃ (k) output from code-by-code symbol generation section 29-3.
Thereafter, in the scrambling unit 136, output signals corresponding to the two types of signals Sc _n '(k) and S _n (k) from the pilot multiplexing unit 135 are transmitted by scrambling codes specific to the transmitter. After being scrambled, it is output to IFFT section 137.
The IFFT unit 137 performs frequency-time conversion on the two types of signals, and the GI insertion unit 138 inserts a guard interval.
An output signal with respect to the two kinds of signals of the GI insertion unit 138 belonging to every code replica generation unit _{129-n sc ^ n (t} ), when the _{s ^} n (t), and becomes the following equation (25).

However, ift [A] indicates that the signal A is converted from the frequency domain to the time domain.
Through these processes, replicas for all code channels constituting the received signal are generated. In the following description, sc ^ (t) = (sc ^ ₁ (t), sc ^ ₂ each of two types of signals output from the GI insertion section 138 of the code-by-code replica generation sections 129-1 to 129-C _mux. (T), ..., sc ^ _n (t), ..., sc ^ _Cmux (t)), s ^ (t) = (s ^ ₁ (t), s ^ ₂ (t), ... ., S ^ _n (t), ..., s ^ _Cmux (t)).

FIG. 18 is a schematic block diagram illustrating a configuration of the signal detection unit 123 in the receiver 200b according to the second embodiment of this invention.
The signal detection unit 123 includes an MCI replica generation unit 151, an addition unit 152, soft canceller units 145-1 to 145-3 (also referred to as incoming wave removal units), an MMSE filter unit 46 (also referred to as a synthesis unit), and a code-by-code demodulation unit 47-1 to 47-C _mux . The MCI replica generation unit 151 and the addition unit 152 are also collectively referred to as an interference canceller unit 154.

The MCI replica generation unit 151 generates a replica signal sc ^ (t) output from each of the replica generation units 129-1 to 129-C _mux for each code of the replica generation unit 128 and a channel impulse response that is a channel estimation value. Based on this, an MCI replica is generated.
The adder 152 removes inter-code interference (MCI) by subtracting the MCI replica from the received signal r (t). If the MCI replica signal for the nth code channel (n = 1, 2,..., C _mux ) output from the MCI replica generation unit is r _n ^ (t), the following equation (26) is obtained, and the received signal r ( An MCI replica is generated for each code multiplexed code channel constituting t).

  However, h (t) is a channel impulse response which is a propagation path estimated value. Also,

Indicates a convolution operation.
If the signal for the n-th code channel output from the adder 152 is r _n ′ (t), the following equation (27) is obtained, and all the code-multiplexed code channels constituting the received signal r (t) are respectively Perform subtraction.

In the following, output signals for all code channels output from the adder 152 are collectively r ′ (t) = (r ₁ ′ (t), r ₂ ′ (t),..., R _n ′. (T),..., R _Cmux '(t)).
Interference between signals other than the desired code by removing the MCI replica in the time domain generated for each code-multiplexed signal constituting the received signal r (t) from the received signal by the interference canceller unit 154 described above. (MCI), inter-symbol interference (ISI), and inter-code interference (ICI) can be removed.

  Each of the soft canceller units 145-1 to 145-3 includes a delayed wave replica generation unit 141, an addition unit 142 (also referred to as a subtraction unit), a GI removal unit 43, and an FFT unit 44. The delayed wave replica generation unit 141 and the addition unit 142 are also referred to as a delayed wave canceller unit 153.

The soft cancellers 145-1 to 145-3 remove the delayed wave from the output signal r ′ (t) of the adder 152 for each predetermined time period using the replica signal created by the replica generator 128.
The delayed wave replica generation unit 141 generates a channel impulse response estimation value, which is a propagation path estimation value estimated from the received signal r (t), and a replica signal s ^ (t) generated by the replica generation unit 128 (FIG. 18). Based on this, a replica of the delayed wave for each predetermined time zone is created. The predetermined time zone is set so that each time zone does not exceed the guard interval length.

For example, when the soft canceller units 145-1 to 145-3 are operated as in the example described with reference to FIGS. 7 to 9, the delayed wave replica generation unit belonging to the soft canceller unit 145-1 is (h (t) − The result of the convolution operation between h ₁ (t)) and the replica signal s ^ (t) is output. The above convolution operation is performed on all code channels to generate delayed wave replicas for each predetermined time zone.
The adder 142 subtracts the result of the convolution operation of (h (t) −h ₁ (t)) and the replica signal s ^ (t) from the output signal r ′ (t) from the adder 152.
Similarly, in the soft canceller unit 45-2, the delayed wave replica generation unit to which the convolution operation of (h (t) -h ₂ (t)) and the replica signal s ^ (t) is performed is an output signal. The result of subtraction from r ′ (t) is the output of the adder 142.

In the soft canceller unit 45-3, a delayed wave replica generation unit to which the result of the convolution operation of (h (t) −h ₃ (t)) and the replica signal s ^ (t) is performed is an output signal r. The result of subtracting from '(t) is the output of the adder 142.
The GI removal unit 43, the FFT unit 44, the MMSE filter unit 46, and the per-code demodulation units 47-1 to 47-C _mux perform the same operations as in the first embodiment.
In the delay wave canceller described above, a delay wave replica for each predetermined time zone is subtracted from the signal from which the signal other than the desired code (MCI) has been removed by the interference canceller, and in a time zone that does not exceed the guard interval. By performing FFT processing, it is possible to reduce intersymbol interference (ISI) and intercode interference (ICI) of a signal of a desired code.

As described above, after the inter-code interference (MCI) is removed from the received signal r (t) using the time domain MCI replica, the signal output from the delay wave canceller unit 153 is shorter than the guard interval length. The interfering interference (MCI), the intersymbol interference (ISI), and the interchannel interference (ICI) are performed by removing the delayed wave so that Even in the case of simultaneous occurrence, it is possible to suppress characteristic deterioration due to inter-code interference (MCI), inter-symbol interference (ISI), and inter-channel interference (ICI).
Furthermore, since the receiver of this embodiment removes the delayed wave for each predetermined time period after removing the inter-code interference (MCI), the receiver is not affected by the magnitude of the inter-code interference MCI and is stable. It becomes possible to obtain characteristics. As a result, for example, the predetermined time zone can be flexibly set only by the GI length or the signal power for each time zone without considering the magnitude of the inter-code interference MCI.

The code multiplexing number C _mux can be obtained by notification to the receiving side using a control channel or the like or estimation based on the received signal.

  In the embodiment described above, a program for realizing the function of each unit of the transmitter (FIG. 1) and each unit of the receiver (FIGS. 3 and 17) is recorded on a computer-readable recording medium, The program recorded on the recording medium may be read into a computer system and executed to control the transmitter and the receiver. Here, the “computer system” includes an OS and hardware such as peripheral devices.

  The “computer-readable recording medium” refers to a storage device such as a flexible medium, a magneto-optical disk, a portable medium such as a ROM and a CD-ROM, and a hard disk incorporated in a computer system. Further, the “computer-readable recording medium” dynamically holds a program for a short time, like a communication line when transmitting a program via a network such as the Internet or a communication line such as a telephone line. In this case, it is also assumed that a server that holds a program for a certain time, such as a volatile memory inside a computer system that serves as a server or client. The program may be a program for realizing a part of the functions described above, and may be a program capable of realizing the functions described above in combination with a program already recorded in a computer system.

  The embodiment of the present invention has been described in detail with reference to the drawings. However, the specific configuration is not limited to this embodiment, and the design and the like within the scope of the present invention are also within the scope of the claims. include.

It is a schematic block diagram which shows the structure of the transmitting apparatus 100a by the 1st Embodiment of this invention. It is a figure which shows an example of the frame format used in the 1st Embodiment of this invention. It is a schematic block diagram which shows the structure of the receiver 200a by the 1st Embodiment of this invention. It is a schematic block diagram which shows an example of a structure of the signal detection part 23 (FIG. 3) by the 1st Embodiment of this invention. It is a flowchart which shows the process of the receiver 200a by the 1st Embodiment of this invention. It is a figure which shows an example of the channel impulse response estimated value by the 1st Embodiment of this invention. It is a figure which shows an example of the channel impulse response estimated value in the soft canceller part 45-1 by the 1st Embodiment of this invention. It is a figure which shows an example of the channel impulse response estimated value in the soft canceller part 45-3 by the 1st Embodiment of this invention. It is a figure which shows an example of the channel impulse response estimated value in the soft canceller part 45-3 by the 1st Embodiment of this invention. It is a figure explaining the case where a predetermined time slot | zone is set based on the time of the identified delay wave. It is a figure which shows the case where all the received signals are divided | segmented into 3 so that the received signal contained in a delay wave may become substantially constant in order of arrival time. It is a figure which shows the channel impulse response estimated value and MMSE filter part in the initial process by the 1st Embodiment of this invention. It is a figure which shows the channel impulse response estimated value and MMSE filter part in the iterative process by the 1st Embodiment of this invention. It is a schematic block diagram which shows the structure of the propagation path and noise electric power estimation part 22 (FIG. 3) by the 1st Embodiment of this invention. It is a schematic block diagram which shows the structure of the radio | wireless receiver by the modification of the 1st Embodiment of this invention. It is a schematic block diagram which shows the structure of the propagation path and noise electric power estimation part 322 (FIG. 15) by the modification of the 1st Embodiment of this invention. It is a schematic block diagram which shows the structure of the receiver 200b by the 2nd Embodiment of this invention. It is a schematic block diagram which shows the structure of the signal detection part 123 in the receiver 200b of the 2nd Embodiment of this invention. It is a figure which shows an example of the relationship of the orthogonal code corresponding to the subcarrier in MC-CDM system and each subcarrier. Signal MC-CDM system is propagated in the air, numeral C _'8,1, C' when received at the receiver _{8,2, C '8,7, C'} '8,1, C''8 _{, 2} , C ″ ₈ , ₇ . Signal MC-CDM system is propagated in the air, numeral C _'8,1, C' when received at the receiver _{8,2, C '8,7, C'} '8,1, C''8 _{, 2} , C ″ ₈ , ₇ . It is a figure which shows the signal which reaches | attains a receiver from a transmitter via a multipath environment.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 ... S / P conversion part, 2-1 to 2-C _mux ... Signal processing part for every code, 8 ... Code multiplexing part, 9 ... Pilot multiplexing part, 10 ... Scrambling part 11... IFFT section, 12... GI insertion section, 21... Symbol synchronization section, 22... Propagation path / noise power estimation section, 23. C _mux ... Decoding unit for each code, 28 ... Replica generation unit, 39 ... P / S conversion unit, 70 ... MAC unit, 71 ... Filtering processing unit, 72 ... D / A Conversion unit, 73 ... frequency conversion unit, 74 ... transmission antenna, 75 ... reception antenna, 76 ... frequency conversion unit, 77 ... A / D conversion unit, 100a ... transmitter, 200a, 200b ... Receiver

Claims (17)

A replica generation unit that creates a replica signal that is a replica of the transmission signal based on the received signal;
An interference canceller that removes inter-code interference from the received signal using the replica signal; an incoming wave remover that removes an incoming wave from the received signal every predetermined time period using the replica signal;
A synthesizer that synthesizes the signal from which the arriving wave removal unit has removed the arriving wave every predetermined time period; and
A demodulation unit that performs demodulation processing on the signal synthesized by the synthesis unit;
A receiver comprising:   The receiver according to claim 1, wherein the interference canceller removes inter-code interference from the received signal for each predetermined time period using the replica signal.   The receiver according to claim 1, wherein the incoming wave removal unit removes an incoming wave for each predetermined time period from the output signal of the interference canceller unit using the replica signal.   4. The interference canceller unit removes inter-code interference in a frequency domain for each predetermined time zone using the replica created by the replica generation unit. 5. Receiver.   4. The receiver according to claim 1, wherein the interference canceller unit removes inter-code interference in a time domain by using the replica created by the replica generation unit. The incoming wave removal unit
A delayed wave replica generation unit that creates a replica of an incoming wave for each predetermined time period; and
A subtracting unit for subtracting a replica of an incoming wave for each predetermined time zone created by the delayed wave replica generating unit from a received signal;
The receiver according to any one of claims 1 to 5, further comprising:   The receiver according to claim 6, wherein the delayed wave replica generation unit sets the predetermined time zone based on the number of identified incoming waves.   The receiver according to claim 6, wherein the delayed wave replica generation unit sets the predetermined time zone based on the time of the identified incoming wave.   The receiver according to claim 6, wherein the delayed wave replica generation unit sets the predetermined time zone based on the received power of the identified incoming wave.   The receiver according to any one of claims 7 to 9, wherein the delayed wave replica generation unit sets the predetermined time period to be a section that does not exceed a guard interval length. Based on the result of the demodulation processing performed by the demodulation unit, error correction decoding is performed, and a signal determination unit that determines a signal for each bit is provided.
The reception according to any one of claims 1 to 10, wherein the replica generation unit creates a replica signal that is a replica of a transmission signal based on the determination value calculated by the signal determination unit. Machine.   12. The signal determination unit according to claim 11, wherein the signal determination unit performs error correction decoding based on a result of the demodulation process performed by the demodulation unit, and uses a calculated log likelihood ratio for each bit as a determination value. Receiver. Providing a propagation path / noise power estimator for estimating the noise power estimate,
The receiver according to any one of claims 1 to 12, wherein the synthesis unit determines a filter coefficient based on a channel impulse response estimation value and the noise power estimation value. The combining unit, the filter coefficients W _m expressed by the formula (A) or formula _(B), or the filter coefficient W _'i of the formula _(C), according to claim 13, characterized by using the _m the receiver (where, m is a natural number, H _{^ m} the transfer function of the m-th channel, ^{H ^} _{H m} is Hamiltonian H _{^ m, C mux} is the number of multiplexed codes, σ _^ ^{N 2} the estimation of the noise power Value, i is a natural number equal to or less than the number of arrival wave removal units, H _{i, m} is a transfer function of the m th propagation path in the i th arrival wave removal unit, and H ^ ^H _{i, m} is H _{i i, m} Hamiltonian).

The propagation path / noise power estimation unit
A reception signal replica generation unit that generates a replica signal of the reception signal based on the replica signal and the channel impulse response estimation value generated by the replica signal generation unit;
A noise power estimation unit that estimates noise power by obtaining a difference between the replica signal and the reception signal created by the reception signal replica generation unit;
The receiver according to claim 13 or 14, further comprising: A replica generation process for creating a replica signal that is a replica of the transmission signal based on the reception signal;
An incoming wave removal process for removing an incoming wave from the received signal every predetermined time period using the replica signal,
A synthesizing process for synthesizing a signal from which the incoming wave is removed every predetermined time period in the incoming wave removing process;
A demodulation process for performing demodulation processing on the signal synthesized in the synthesis process,
In the incoming wave removal process, the inter-code interference is removed from the received signal at predetermined time intervals using the replica signal. A replica generation process for creating a replica signal that is a replica of the transmission signal based on the reception signal;
An interference canceller process for removing inter-code interference from the received signal using the replica signal;
An incoming wave removal process for removing an incoming wave from the output signal in the interference canceller process at predetermined time intervals using the replica signal;
A synthesizing process for synthesizing a signal from which the incoming wave is removed every predetermined time period in the incoming wave removing process;
A demodulation process for performing demodulation processing on the signal synthesized in the synthesis process;
A receiving method comprising:

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