JP2009081648A - Temperature compensation type crystal oscillator - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To solve a problem that output noise generated by a network for temperature compensation affects phase-noise characteristics of a temperature compensation type crystal oscillator. <P>SOLUTION: A temperature signal S12 which varies in potential along a linear function as ambient temperature varies and has a negative temperature gradient coefficient and a temperature signal S10 which varies in potential along a linear function as well and a positive temperature gradient coefficient are put together, and a temperature correction signal generating circuit 270 is controlled based upon a temperature signal having a temperature gradient coefficient amplified with two initial temperature signals to lower the signal amplification factor from a temperature sensor circuit of an input stage associated with temperature correction to the temperature signal generating circuit. <P>COPYRIGHT: (C)2009,JPO&INPIT

Description

  The present invention relates to a temperature-compensated crystal oscillator that compensates for the temperature characteristics of a crystal oscillator using a crystal resonator.

  A crystal oscillator used as a reference oscillator for mobile communications in recent years is equipped with an AT-cut crystal resonator that is superior to other oscillators in frequency stability due to temperature changes. A so-called temperature compensated crystal oscillator that compensates is widely used.

  The AT-cut crystal unit (crystal piece) indicates the initial cutting direction from the processing of the crystal block to the crystal plate and pieces, and the frequency with respect to temperature among the crystal units with various cutting directions. The variation is considered to be the smallest.

This temperature-compensated crystal oscillator is used as a reference oscillating source for a frequency synthesizer, such as a cellular phone device or a GPS receiver, or a reference source for a PLL that forms a local oscillator.
Therefore, the required accuracy is also high, and the frequency stability with respect to temperature change, the phase noise characteristic that is an index of noise power with respect to carrier (desired frequency) power, and the like are determined in detail.

Quartz resonators have a high Q value, and among others, AT-cut quartz resonators are most utilized at oscillation frequencies in the MHz band and have high frequency stability against temperature changes. However, it cannot be used in communication equipment without any correction, and requires some temperature correction.
The frequency-temperature characteristic of an AT-cut crystal resonator is approximately expressed by a function having a first-order component and a third-order component. To compensate for this change in frequency, the oscillator generates some function signal. In addition, a frequency modulation circuit for canceling the frequency temperature characteristic of the crystal resonator must be provided.

There are a direct compensation method and an indirect compensation method in the temperature compensation method of the crystal oscillator. In the direct compensation method, since the temperature compensation circuit unit is directly connected to the crystal unit, the effective Q value of the crystal resonance circuit is lowered as a resistance component. However, since the circuit is simple, there is an advantage that phase noise is low. is there.
On the other hand, in the indirect compensation method, since the capacitance component is voltage-controlled by the temperature compensation circuit, the effective Q value does not change greatly directly. Although temperature compensation with high accuracy is possible, the circuit is complicated and large-scale, and there is a drawback that downsizing and deterioration of phase noise characteristics become problems.

  In general, an indirect compensation type oscillator is widely used. The frequency voltage control type oscillator that adjusts the (series) resonance frequency of the crystal resonator by changing the capacitance component in the oscillation loop by voltage control is the most common. In recent years, there are many oscillators that perform temperature compensation by mounting an oscillation circuit unit, a frequency modulation circuit, and a memory circuit that stores an adjusted value in a semiconductor integrated circuit by various semiconductor manufacturing processes.

  As a reference oscillation source for communication equipment, of course, temperature compensation for frequency stabilization is essential, but phase noise is also extremely important. For example, it is an error in demodulated data in a mobile phone, or positioning in a GPS receiver. It depends on whether it is possible.

The phase noise characteristics of a temperature-compensated crystal oscillator vary depending on the Q value of the crystal resonator, but the frequency-dependent elements and circuit networks are caused by the current when the oscillation circuit section and the temperature compensation circuit section operate. It is affected by noise and noise independent of frequency. The phase noise is expressed by a ratio of noise power in the 1 Hz band at a discrete frequency from the fundamental frequency with reference to the signal power of the fundamental frequency (carrier), and has a characteristic that spreads in a broad manner around the spectrum of the fundamental frequency.

  In the phase noise characteristic, in the vicinity of several Hz to several kHz from the fundamental frequency to the discrete frequency, the flicker noise of the crystal resonator and the element and the noise due to the transfer function of the network are converted to modulate the frequency. It changes from -30 dB / dec to -20 dB / dec from the lower discrete frequency. Further, at a discrete frequency of several kHz to several hundred kHz, noise that does not depend on the frequency of the elements and circuits becomes dominant.

Therefore, a method of suppressing circuit noise by simplifying a circuit unit that generates a function signal related to temperature compensation in an indirect compensation type oscillator has been proposed (see, for example, Patent Document 1).
The prior art disclosed in Patent Document 1 uses a MIS (metal-insulating film-semiconductor) type variable capacitance element as a capacitance element capable of variable capacitance.

  FIG. 9 is a block diagram showing a schematic configuration of the temperature-compensated crystal oscillator disclosed in Patent Document 1. In FIG. 9 includes a first temperature detection circuit 208, a temperature gradient signal calculation circuit 200, a temperature signal selection circuit 300, a temperature third-order component correction signal generation circuit 260, and a temperature first-order component correction. The signal generation circuit 280, the external frequency control voltage generation circuit 160, the external input terminal 50, the frequency adjustment circuit 130, and the oscillation circuit 100 are configured.

As shown in FIG. 9, the frequency adjustment circuit 130 includes MIS variable capacitance elements 142a and 142b, resistance elements 135a, 135b, and 138, and DC-cut capacitance elements 132a, 132b, and 150.
The oscillation circuit 100 includes an AT-cut crystal resonator 120 that is a piezoelectric resonator, an inverting amplifier circuit 110, a feedback resistance element 106, and an attenuation resistance element 112.

The first temperature detection circuit 208 generates a first temperature signal S12 that changes in a linear function with respect to a change in ambient temperature, and the first approximated primary signal amplified by the temperature gradient signal calculation circuit 200. S14 and a second approximate primary signal S16 obtained by signal inversion amplification are generated.
The subsequent temperature signal selection circuit 300 is based on the first approximate primary signal S14 and the second approximate primary signal S16, and the temperature tertiary component correction signal generation circuit 260 and the temperature primary component correction signal. The first gate electrode signal S20b, the second gate electrode signal S20c, and the approximate primary correction signal S20a, which are temperature gradient adjusted to drive the generation circuit 280, are generated separately.

The first gate electrode signal S20b and the second gate electrode signal S20c are applied to the gate electrode (not shown) of the MOSFET in the temperature third-order component correction signal generation circuit 260, and the temperature third-order of the crystal resonator An approximate cubic function current for correcting the component is generated and converted into a voltage by the temperature primary component correction signal generation circuit 280.
One approximate primary correction signal S20a is a voltage signal for correcting the temperature primary component of the crystal resonator, and is applied to the temperature primary component correction signal generation circuit 280 to generate an approximate cubic function voltage signal. Is added.

  This approximate cubic function voltage signal is applied to the MIS variable capacitance element in the frequency adjustment circuit 130 while the ambient temperature changes from a low temperature to a high temperature, and cancels the frequency-temperature characteristic of the AT-cut crystal resonator. To work.

JP 2006-94262 A (pages 9 to 11, FIG. 1)

In the case of the conventional technique shown in Patent Document 1, if the temperature coefficient of the first temperature signal S12 is less than the expected temperature gradient, in order to sufficiently correct the temperature third-order component of the crystal resonator The current change in the approximate cubic function generated by the temperature third-order component correction signal generation circuit 260 is increased.
For example, the first temperature signal S12 is amplified by increasing the voltage conversion rate of the generated approximate cubic function current and amplifying the approximate cubic function current generated in the temperature third component correction signal generation circuit 260. For example, the temperature gradient may be increased by amplification.

  In general, it is assumed that the noise figure of the network is the first stage noise, but the phase noise of the temperature-compensated crystal oscillator is a circuit network related to temperature compensation, in addition to the circuit scale and the circuit network used, It is also necessary to consider element noise and mutual conductance. Therefore, it is ideal to reduce the amplification factor from the input side to the output side of the network.

  The present invention provides a technique that enables noise suppression even when the temperature coefficient of the temperature signal as described above does not have an expected temperature gradient.

  In order to achieve the above object, the temperature compensated crystal oscillator of the present invention adopts the following configuration.

An oscillation circuit unit including a crystal resonator having a temperature characteristic in which an oscillation frequency changes in a cubic function with respect to a temperature change, a frequency adjustment circuit, a temperature signal generation circuit, a temperature third-order component correction signal generation circuit, A temperature compensated crystal oscillator having a temperature primary component correction signal generation circuit,
A temperature signal is generated by synthesizing a plurality of temperature detection circuits that generate a temperature signal having a predetermined variation characteristic with respect to a temperature change, and a difference between the plurality of temperature signals generated by each temperature detection circuit. And a synthesis circuit, and drives the temperature third-order component correction signal generation circuit and the temperature first-order component correction signal generation circuit using the synthesized temperature signal to adjust the frequency of the oscillation circuit section.

  The temperature detection circuit generates a first temperature signal that changes in a linear function with respect to a temperature change, a first temperature detection circuit that changes in a linear function with respect to the temperature change, A second temperature detection circuit that generates a second temperature signal different from the temperature signal, and a synthesis circuit synthesizes the first temperature signal and the second temperature signal, And a temperature third-order component correction signal generation circuit and a temperature first-order component correction signal generation circuit are driven using the third temperature signal to adjust the frequency of the oscillation circuit unit. To do.

  The first temperature signal has a positive or negative primary coefficient with respect to a temperature change, and the second temperature signal has a polarity different from that of the first temperature signal with respect to the temperature change. It is characterized by having.

  The synthesizing circuit generates a potential of a difference between the first temperature signal and the second temperature signal.

According to the temperature-compensated crystal oscillator of the present invention, of the temperature signals from at least two temperature detection circuits, a signal having a temperature first order coefficient having a positive polarity and a temperature first order having a negative polarity. The potential of the difference from the signal having the coefficient is generated in the synthesis circuit.
At this time, even if the amplification factor of the synthesis circuit is not increased, the temperature signal which is the potential of the difference becomes a temperature signal having a temperature first-order coefficient larger than that of the underlying temperature signal.
This temperature signal is electrically equivalent to the first stage of the entire circuit system, and can suppress the amplification factor going to the latter stage of the circuit. As a result, the noise of the entire circuit system is suppressed, and the temperature compensated crystal oscillator It is possible to improve the phase noise characteristics.

  Exemplary embodiments of a temperature compensated crystal oscillator according to the present invention will be explained below in detail with reference to the drawings.

[Description of overall configuration: Fig. 1]
FIG. 1 is a block diagram showing the configuration of the main part of the temperature compensated crystal oscillator of the present invention. 230 is a temperature gradient signal generation circuit, 270 is a temperature correction signal generation circuit, 130 is a frequency adjustment circuit, 100 is an oscillation circuit, 160 is an external frequency control voltage generation circuit, and 50 is an external input terminal.

The configuration of the temperature gradient signal generation circuit 230 is as follows.
Reference numeral 208 denotes a first temperature detection circuit, 209 denotes a second temperature detection circuit, 200 denotes a temperature gradient signal calculation circuit, and 300 denotes a temperature signal selection circuit.
A first temperature signal S12 is generated from the first temperature detection circuit 208, and a second temperature signal S10 is generated from the second temperature detection circuit 209.
The temperature gradient signal calculation circuit 200 is a synthesis circuit that generates a potential corresponding to the difference between the first temperature signal S12 and the second temperature signal S10.
The temperature signal selection circuit 300 generates a plurality (n) of temperature gradient signal groups S20 for controlling the temperature correction signal generation circuit 270 at the subsequent stage.

The configuration of the temperature correction signal generation circuit 270 is as follows.
Reference numeral 260 denotes a temperature tertiary component correction signal generation circuit, and 280 denotes a temperature primary component correction signal generation circuit.
The temperature tertiary component correction signal generation circuit 260 and the temperature primary component correction signal generation circuit 280 generate the correction signal S30.

The configuration of the oscillation circuit 100 is as follows.
For example, an inverting amplifier 110 made of a CMOS inverter circuit and a feedback resistor 106 are connected in parallel to the AT cut crystal resonator 120, and an attenuation resistance element 112 is connected between the inverting amplifier 110 and the AT cut crystal resonator 120. Has been.
Although not shown, the output of the inverting amplifier 110 is output as an output signal of the oscillation circuit 100 through an amplifier such as a CMOS inverter circuit, for example.

The configuration of the frequency adjustment circuit 130 connected to the oscillation circuit 100 is as follows.
A series connection body formed by connecting a DC cut capacitive element 132a, a variable capacitive element such as a first MIS variable capacitive element 142a, and a DC cut capacitive element 150 in series is an input side of the inverting amplifier 110. And ground (potential).
Both electrodes of the first MIS variable capacitance element 142a are separated from the DC component of the oscillation circuit 100 and the ground by the DC cut capacitance elements 132a and 150.

In addition, a series connection body in which a DC cut capacitive element 132b, a variable capacitive element, for example, a second MIS variable capacitive element 142b, and a DC cut capacitive element 150 are connected in series with the same configuration is inverted. It is connected between the output side of the amplifier 110 and the ground (potential).
Both electrodes of the second MIS variable capacitance element 142b are separated from the DC component of the oscillation circuit 100 and the ground by the DC cut capacitance elements 132b and 150.

A gate potential Vg and a well potential Vw, which will be described later, are applied to the electrodes of the capacitive elements of the first MIS variable capacitive element 142a and the second MIS variable capacitive element 142b, and from the potential difference (Vg−Vw). Control the capacitance value.
In the present embodiment, one electrode of the first MIS variable capacitance element 142a is connected to the external frequency control voltage generation circuit 160 via the resistance element 135a. The other electrode is connected to the temperature correction signal generation circuit 270 via the resistance element 138, and controls the first MIS variable capacitance element 142a by the correction signal S30.

  Similarly, one electrode of the second MIS variable capacitance element 142b is connected to the external frequency control voltage generation circuit 160 via the resistance element 135b. The other electrode is connected to the temperature correction signal generation circuit 270 via the resistance element 138, and controls the second MIS variable capacitance element 142b by the correction signal S30.

[Detailed description of each element: FIGS. 1, 2, 5, 6, and 7]
FIG. 2 is a circuit diagram for explaining each element constituting the temperature compensated crystal oscillator of the embodiment of the present invention.
The temperature gradient signal generation circuit 230 includes a first temperature detection circuit 208, a second temperature detection circuit 209, a temperature gradient signal calculation circuit 200, and a temperature signal selection circuit 300. The temperature gradient signal arithmetic circuit 200 includes resistance elements 202, 204, 212, and 214 and operational amplifier circuits 206 and 216.

The first temperature signal S12 generated by the first temperature detection circuit 208 is connected to the differential (−) side terminal of the operational amplifier circuit 206 through the resistance element 202. The first approximated primary signal S14 output from the operational amplifier circuit 206 is fed back to the differential (−) side terminal of the operational amplifier circuit 206 via the resistance element 204, thereby constituting an inverting amplifier. .
The second temperature signal S10 generated by the second temperature detection circuit 209 is connected to the differential (+) side terminal of the operational amplifier circuit 206.
The potential fed back to the differential (−) side terminal of the operational amplifier circuit 206 is the second temperature due to an imaginary short between the differential (+) side terminal and the differential (−) side terminal of the operational amplifier circuit 206. The potential is controlled to be substantially the same as the signal S10.

The first approximate primary signal S <b> 14 is connected to the differential (−) side terminal of the operational amplifier circuit 216 via the resistance element 212. The second approximate primary signal S16 output from the operational amplifier circuit 216 is fed back to the differential (−) side terminal of the operational amplifier circuit 216 via the resistance element 214, thereby constituting an inverting amplifier. .
The first constant voltage signal S19 serving as a reference voltage is connected to the differential (+) side terminal of the operational amplifier circuit 216.
The potential fed back to the differential (−) side terminal of the operational amplifier circuit 216 is controlled to be substantially equal to the first constant voltage signal S19 input to the differential (+) side terminal of the operational amplifier circuit 216. . The first constant voltage signal S19 will be described later.

  The temperature signal selection circuit 300 receives the first approximate primary signal S14 and the second approximate primary signal S16, and based on these inputs, the approximate primary correction signal S20a, the first gate voltage signal S20b, 2 gate voltage signal S20c is generated and output.

[Description of First Temperature Detection Circuit: FIG. 5]
Next, the circuit configuration of the first temperature detection circuit 208 will be described with reference to FIG.
The first temperature detection circuit 208 includes an operational amplifier circuit 250, a PMOS transistor 252, and a resistance element 254.
The differential (+) side terminal of the operational amplifier circuit 250 inputs the first constant voltage signal S19.
A PMOS transistor 252 and a resistance element 254 are connected in series between the first temperature signal S12 and the ground node at the differential (−) side terminal of the operational amplifier circuit 250, and the PMOS transistor 252 and the resistance element 254 are connected to each other. The voltage signal S13 divided by is fed back.

  Here, the PMOS transistor 252 has a connection configuration in which the drain node voltage is fed back to the gate node, and the resistance element 254 is a resistance element having a low resistance such as polysilicon whose resistance value is extremely small.

  Electrically, the differential (−) side terminal of the operational amplifier circuit 250 is constant voltage controlled by the imaginary short circuit so that the voltage signal S13 and the first constant voltage signal S19 are substantially the same. A constant current with extremely small temperature dependence flows. Since the drain current of the PMOS transistor 252 serves as a current source for the resistance element 254, the PMOS transistor 252 is similarly controlled at a constant current. In the PMOS transistor 252, the threshold voltage and carrier mobility of the transistor change as the ambient temperature changes. As a result, the first temperature signal S12 changes approximately linearly as the ambient temperature changes.

The first constant voltage signal S19 is a constant voltage serving as a reference arbitrarily determined in circuit operation. The first constant voltage signal S19 generated here serves as a reference voltage for a subsequent circuit, for example, the temperature correction signal generation circuit 270.
In the example shown in FIG. 5, resistance elements 223, 224, and 225 are provided in series between the constant voltage source 220 and the ground node, and are generated by dividing the voltage. The first constant voltage signal S19 can be, for example, approximately half the voltage of the constant voltage source 220 and the ground node.

  Note that the first constant voltage signal S19 is not limited to the example shown in FIG. 5, and its configuration can be freely selected. For example, a known constant voltage source such as a regulator circuit can be used.

[Description of Second Temperature Detection Circuit: FIG. 6]
Next, the circuit configuration of the second temperature detection circuit 209 will be described with reference to FIG.
The second temperature detection circuit 209 has a configuration in which the PMOS transistor 252 and the resistance element 254 connected in series are replaced with the circuit configuration of the first temperature detection circuit 208.
Similar to the circuit configuration shown in FIG. 5, the second temperature detection circuit 209 includes an operational amplifier circuit 240, a resistance element 244, and a PMOS transistor 242.
The differential (+) side terminal of the operational amplifier circuit 240 includes, for example, a second constant voltage signal S18 that is divided by providing resistance elements 227, 228, and 229 in series between the constant voltage source 220 and the ground node. input. A resistor (244) and a PMOS transistor (242) are connected in series between the second temperature signal (S10) and the ground node at the differential (−) terminal of the operational amplifier circuit (240). The voltage signal S11 divided by is fed back.

  Here, the PMOS transistor 242 has a connection configuration in which the drain node voltage (here, the ground node voltage) is fed back to the gate node, and the resistance element 244 has a low resistance such as polysilicon whose resistance value is extremely small. The resistance element is used.

Electrically, the differential (−) side terminal of the operational amplifier circuit 240 is constant voltage controlled by the imaginary short circuit so that the voltage signal S11 and the first constant voltage signal S18 are substantially the same, so that the source of the PMOS transistor 242 The node is controlled at a constant voltage. In the PMOS transistor 252, the threshold voltage and the carrier mobility of the transistor change with changes in the ambient temperature, but the drain current thereof changes approximately linearly. At this time, since the current having the same value as the drain current of the PMOS transistor 242 flows through the resistance element 244, the second temperature signal S
The voltage signal of 10 changes approximately linearly.

[Explanation of temperature signal generation by temperature gradient signal calculation circuit]
The first temperature signal S12 and the second temperature signal S10 described with reference to FIGS. 5 and 6 have different temperature coefficients with respect to changes in the ambient temperature. The first temperature signal S12 has a negative polarity, and the second temperature signal S10 has a positive polarity.

The operational amplifier circuit 206 shown in FIG. 2 generates an output obtained by inverting the first temperature signal S12 as the first approximate primary signal S14 based on the second temperature signal S10.
For this reason, the first approximate primary signal S14 has a temperature gradient with a larger absolute value of the temperature coefficient than the first temperature signal S12 and the second temperature signal S10 with respect to a change in ambient temperature. It becomes.

  Similarly, the second approximate primary signal S16 is equal to a signal obtained by inverting the temperature coefficient of the first approximate primary signal S14 with reference to the first constant voltage signal S19 that is the reference voltage. The primary signal S16 is also a signal having a temperature gradient having a larger absolute value of the temperature coefficient than the first temperature signal S12 and the second temperature signal S10.

[Explanation of temperature signal selection circuit]
As shown in FIG. 7, the temperature signal selection circuit 300 includes, for example, two resistance dividing circuits and, for example, three nodes between the nodes of the first approximate primary signal S14 and the second approximate primary signal S16. It has selection circuit groups 330a, 330b, and 330c.
In the two resistance dividing circuits, a plurality of resistance elements 306a to 306n and 307a to 307n are inserted in parallel between nodes of the first approximate primary signal S14 and the second approximate primary signal S16.
Each of the selection circuit groups 330a, 330b, and 330c includes selection circuits 332a to 332n, 334a to 334n, and 336a to 336n.

The plurality of nodes that are resistance-divided by the resistance elements 306a to 306n are distributed and connected to each of the selection circuits 332a to 332n in the selection circuit group 330a.
The plurality of nodes divided by the resistance elements 307a to 307n are distributed and connected to each of the selection circuits 334a to 334n and 336a to 336n in the selection circuit groups 330b and 330c, that is, to the two selection circuit groups. ing.

In the selection circuit group 330a, one is selected by the selection circuits 332a to 332n selected by the memory circuit 305 to generate the approximate primary correction signal S20a.
Similarly, in the selection circuit group 330b, one is selected by the selection circuits 334a to 334n selected by the memory circuit 305 to generate the first gate voltage signal S20b.
In the selection circuit group 330c, one is selected by the selection circuits 336a to 336n selected by the memory circuit 305 to generate the second gate voltage signal S20c.

The approximate primary correction signal S20a is a base signal for correcting the temperature primary component for adjusting the oscillation frequency of an AT-cut quartz crystal resonator, which will be described later, to be constant with respect to a change in ambient temperature.
The first gate voltage signal S20b is a control signal for a temperature third-order component correction circuit that functions mainly when the ambient temperature is lower than room temperature and adjusts the oscillation frequency to be constant.
The second gate voltage signal S20c is a control signal for a temperature third-order component correction circuit that functions mainly when the ambient temperature is higher than room temperature and adjusts the oscillation frequency to be constant.

[Explanation of temperature correction signal generation circuit]
As shown in FIG. 2, the temperature correction signal generation circuit 270 includes a temperature third-order component correction signal generation circuit 2.
60 and a temperature primary component correction signal generation circuit 280.
The temperature third-order component correction signal generation circuit 260 includes PMOS transistors M1 and M2, NMOS transistors M3 and M4, and resistance elements 262 and 261.
The NMOS transistors M3 and M4 constitute a mirror circuit for turning back the current of the PMOS transistor M2. Resistive elements 262 and 261 are connected in series to the source nodes of the PMOS transistors M1 and M2, respectively, and are supplied with power from a constant voltage source. The constant voltage source is not particularly limited, but a general stabilized power circuit can be used.

A first gate voltage signal S20b is connected to the gate electrode of the PMOS transistor M1, and a second gate voltage signal S20c is connected to the gate electrode of the PMOS transistor M2.
The resistance element 262 has an exponential source-drain current increase characteristic due to a voltage drop caused by a current and a resistance value that flow through the PMOS transistor M1 in the process of the PMOS transistor M1 changing from a non-conductive state to a conductive state. It is changed in the direction that suppresses.
Further, the resistance element 262 suppresses the source-drain current with respect to the gate voltage of the PMOS transistor M1, so that the source-drain conductance is lowered.

  Similarly, the resistance element 261 has an exponential source-drain due to a voltage drop caused by the current and resistance value that flows through the PMOS transistor M2 in the process of the PMOS transistor M2 changing from the non-conductive state to the conductive state. While changing in the direction which suppresses the increase characteristic of an electric current, the source-drain electric current with respect to the gate voltage of PMOS transistor M2 is suppressed, and source-drain conductance is lowered.

  The temperature primary component correction signal generation circuit 280 is configured by a resistance element 283, for example. The approximate primary correction signal S20a is connected to one node of the resistance element 283, and the other node is an NMOS configured by a drain node of the PMOS transistor M1 and a mirror circuit obtained by folding the current of the PMOS transistor M2. The drain node of the transistor M3 is connected.

  The resistance element 283 is a PMOS transistor connected to the voltage of the approximate primary correction signal S20a, which is a correction voltage of a temperature primary component for adjusting the oscillation frequency of the AT-cut crystal resonator to be constant with respect to a change in ambient temperature. The exponential source-drain current generated in M1 and M2 is converted into a voltage by flowing in and out, and the correction voltage of the temperature third-order component is added to the correction signal S30.

  The resistance value of the resistance element 283 can control the voltage conversion value of the exponential source-drain current generated in the PMOS transistors M1 and M2. The current generated in the PMOS transistors M1 and M2 is controlled by the resistance element 283 and the temperature. The resistance value of the resistance element 283 is not reduced so as to affect the first approximate primary signal S14 and the second approximate primary signal S16 via the signal selection circuit 300.

  In addition, although the temperature primary component correction signal generation circuit 280 in FIG. 2 is configured by the resistance element 283, the present invention is not limited to this. For example, although not shown, as an adding circuit using a general operational amplifier, the approximate primary correction signal S20a is connected to the differential (+) side terminal and the temperature tertiary component correction signal is connected to the differential (−) side terminal. The generation circuit 260 may be connected, and the output of the operational amplifier may be used as the correction signal S30 and may be fed back to the differential (−) side terminal via a resistor.

  Note that the oscillation circuit 100, the frequency adjustment circuit 130, and the external frequency control voltage generation circuit 160 shown in FIG. 2 are well-known indirect compensation type oscillation circuits using CMOSs, and thus description thereof is omitted.

[Description of MIS variable capacitance element: FIGS. 3 and 4]
Next, the structure and characteristics of the first and second MIS variable capacitance elements 142a and 142b shown in FIGS. 1 and 2 will be described with reference to FIGS.
FIG. 3 schematically shows a cross-sectional view of the MIS variable capacitor. FIG. 3 shows a P-type semiconductor substrate 71 as an example, but the polarity of the MIS variable capacitance element is not particularly limited. In the case of FIG. 3, since the substrate is P-type, an N-type well 70 of opposite conductivity type is formed to electrically isolate the MIS variable capacitance element and the P-type semiconductor substrate 71.
The structure of the MIS variable capacitance element is a MOS structure (metal-oxide film-semiconductor). A gate oxide film 74 is formed on the surface of the N-type well 70, and a metal gate electrode 72 is formed. An N-type diffusion layer 76 is formed by selectively implanting a high concentration impurity on the surface of the N-type well 70.

FIG. 4 is a diagram schematically showing electrical characteristics of the MIS variable capacitor. The horizontal axis of the figure shows the Vg potential-Vw potential applied between the metal gate electrode 72 of the MIS type variable capacitance element and the N type well 70, and the vertical axis shows the effective capacitance value of the MIS type variable capacitance element. This is what is called a CV characteristic diagram.
The N-type well 70 is supplied with a potential (hereinafter referred to as Vw potential) from a terminal W connected to the N-type diffusion layer 76, and the metal gate electrode 72 is supplied with a potential (hereinafter referred to as Vg potential) from the terminal G. ) Is supplied.

  In the process in which the Vg potential of the metal gate electrode 72 changes to the negative side with respect to the Vw potential of the N-type well 70, the surface of the N-type diffusion region 78 immediately below the metal gate electrode 72 is changed from the accumulated state to the surface inversion layer. Until the formation, the capacitance value of the MIS variable capacitance element changes. When the change in the Vg potential or the Vw potential is controlled at a high frequency, a small number of the surface inversion layer is formed in the N type diffusion region 78. The carrier cannot be supplied instantaneously.

The right side in FIG. 4 is the accumulation state, and the Vg potential−Vw potential is set to the negative side toward the left side. As shown in FIG. 4, the capacitance value decreases as the Vg potential-Vw potential changes to the negative side, and when the Vg potential-Vw potential is controlled at a high frequency, the capacitance value becomes a certain fixed value. It is kept at a local minimum.
In the embodiment of the present invention, the oscillation frequency of the temperature compensated crystal oscillator is controlled using the CV characteristic shown in FIG.

[Explanation of Temperature Compensation Method: FIG. 1, FIG. 2, FIG. 5, FIG. 6, FIG. 8]
Next, the temperature compensation method for the oscillation frequency of the AT-cut crystal resonator will be described together with the circuit signal and circuit operation of the temperature compensated crystal oscillator of the present invention.
The first temperature detection circuit 208 shown in FIGS. 1 and 2 has the circuit configuration of FIG. 5 already described, and the first temperature signal S12 obtained here has the temperature dependence of the threshold voltage of the PMOS transistor 252. Since it is large, the temperature gradient coefficient is large. The threshold voltage changes primarily with respect to changes in the ambient temperature, and is very dominant in determining the voltage value with respect to the first temperature signal S12 because of the change in mobility which is approximately inversely related to temperature. The temperature signal S12 of 1 is a linear signal.

  The second temperature detection circuit 209 shown in FIGS. 1 and 2 has the circuit configuration shown in FIG. 6, and the second temperature signal S10 obtained here is generated by the voltage signal S11 and the resistance element 244. Therefore, the temperature gradient coefficient is large similarly to the first temperature signal S12, and the temperature coefficient has a polarity opposite to that of the first temperature signal S12.

The state of such a voltage signal is shown in FIG. In FIG. 8, the temperature increases toward the right side of the drawing with the horizontal axis as the temperature. The vertical axis represents the voltage value of each signal, indicating the gradient of the voltage signal due to temperature.
In FIG. 8, reference numeral 42 denotes an inflection point temperature of the temperature third-order component characteristic of the AT-cut crystal resonator, for example, 28 ° C.

The first temperature signal S12 is inverted with respect to the second temperature signal S10 by an inverting amplifier composed of the operational amplifier circuit 206 and the resistance elements 202 and 204 in the temperature gradient signal arithmetic circuit 200 shown in FIG. Thus, the first approximate primary signal S14 is generated.
The first approximate primary signal S14 has a larger temperature gradient than the first temperature signal S12 of the base signal.
That is, even if the signal amplification factor by the operational amplifier circuit 206 of the temperature gradient signal arithmetic circuit 200 is suppressed, a signal having a temperature gradient larger than that of the base signal can be generated.

The inverting amplifier composed of the operational amplifier circuit 216 and the resistance elements 212 and 214 generates a signal obtained by inverting the sign of the first approximate primary signal S14 with reference to the first constant voltage signal S19. It becomes the gate voltage source of the PMOS transistor that functions by correcting the temperature third order component.
The voltage signal S18 already shown in FIG. 6 is always a constant voltage with respect to changes in ambient temperature, and the first approximate primary signal S14 and the second approximate primary signal S16 as shown in FIG. Adjustment is made so as to intersect at the inflection point temperature 42 of the temperature third-order component characteristic of the AT-cut crystal resonator.

  In the temperature signal selection circuit 300, resistance division is performed by a resistance element inserted between the first approximate primary signal S14 and the second approximate primary signal S16, and the selection circuit groups 330a, 330b, and 330c. An approximate primary correction signal S20a, a first gate voltage signal S20b, and a second gate voltage signal S20c having a specific temperature gradient are selectively generated.

The generated first gate voltage signal S20b is generated by the PMOS transistor M1 in the temperature third-order component correction signal generation circuit 260 on the lower temperature side than the inflection point temperature 42 shown in FIG. Generate current.
Similarly, the second gate voltage signal S20c generates a source-drain current of an approximate secondary signal on the higher temperature side than the inflection point temperature 42 by the PMOS transistor M2 in the temperature third-order component correction signal generation circuit 260. .

The source-drain current generated on each of the low temperature side and the high temperature side is subjected to current-voltage conversion by the resistance element 283 in the temperature primary component correction signal generation circuit 280.
The approximate primary correction signal S20a is a base signal of the resistance element 283, and corrects the temperature primary component characteristics of the AT-cut crystal resonator. That is, the resistance element 283 is to add the approximate secondary signal generated by the PMOS transistors M1 and M2 and the approximate primary correction signal S20a, and an approximate cubic function like the correction signal S30 shown in FIG. A typical signal.

  An approximate cubic function correction signal S30 as shown in FIG. 8 is supplied to the frequency adjustment circuit 130 shown in FIGS. As described above, the frequency adjustment circuit 130 includes the DC cut capacitor 132a and the first MIS variable capacitor 142a connected in series, the DC cut capacitor 132b, and the first MIS variable capacitor 142b. And a DC cut capacitor element 150, and one electrode of each of the first and second MIS variable capacitor elements 142a and 142b is connected to the outside via resistor elements 135a and 135b. A frequency control voltage generation circuit 160 is connected, and a correction signal S30 is connected to the other electrode via a resistance element 138.

The external frequency control voltage generation circuit 160 generates a DC voltage for adjusting the voltage signal input from the external input terminal 50 to a predetermined frequency. On the other hand, the capacitance values of the first and second MIS variable capacitance elements 142a and 142b follow linearly.
The capacitance value according to the correction signal S30 linearly follows the voltage of the correction signal S30 and changes in an approximate cubic function with respect to changes in the ambient temperature. This change in the capacitance value cancels the frequency temperature characteristic of the AT-cut quartz resonator and keeps it at a constant frequency.

  The temperature compensated crystal oscillator of the present invention can improve the phase noise characteristics of the temperature compensated crystal oscillator by suppressing the signal amplification factor from the input side to the output side of the circuit system. Therefore, it is useful for an oscillator mounted on a communication device, and is particularly suitable as an oscillator mounted on a small portable device such as a mobile phone.

It is a block diagram explaining the structure of the temperature compensation type | mold crystal oscillator of this invention. It is a circuit diagram explaining the structure of the principal part of the temperature compensation type | mold crystal oscillator of this invention. It is a figure explaining the MIS type variable capacity element used for the temperature compensation type crystal oscillator of the present invention. It is a figure explaining the CV characteristic of the MIS type variable capacity element used for the temperature compensation type crystal oscillator of the present invention. It is a circuit diagram explaining the structure of the 1st temperature detection circuit used for the temperature compensation type | mold crystal oscillator of this invention. It is a circuit diagram explaining the structure of the 2nd temperature detection circuit used for the temperature compensation type | mold crystal oscillator of this invention. It is a circuit diagram explaining the structure of the temperature signal selection circuit used for the temperature compensation type | mold crystal oscillator of this invention. It is a figure explaining the temperature characteristic of the temperature compensation type | mold crystal oscillator of this invention. It is a block diagram explaining the prior art shown in patent document 1. FIG.

Explanation of symbols

DESCRIPTION OF SYMBOLS 50 External input terminal 100 Oscillation circuit 130 Frequency adjustment circuit 160 External frequency control voltage generation circuit 200 Temperature gradient signal calculation circuit 230 Temperature gradient signal generation circuit 270 Temperature correction signal generation circuit 208 1st temperature detection circuit 209 2nd temperature detection circuit 300 Temperature signal selection circuit 260 Temperature third-order component correction signal generation circuit 280 Temperature first-order component correction signal generation circuit 142a First MIS variable capacitor 142b Second MIS variable capacitor S12 First temperature signal S10 Second Temperature signal S14 first approximate primary signal S16 second approximate primary signal S20a approximate primary correction signal S20b first gate voltage signal S20c second gate voltage signal S30 correction signal

Claims (4)

An oscillation circuit unit including a crystal resonator having a temperature characteristic in which an oscillation frequency changes in a cubic function with respect to a temperature change;
A frequency adjustment circuit;
A temperature signal generation circuit;
A temperature third-order component correction signal generation circuit;
A temperature primary component correction signal generation circuit;
A temperature-compensated crystal oscillator comprising:
A plurality of temperature detection circuits for generating a temperature signal having a predetermined variation characteristic with respect to a temperature change;
A synthesis circuit that generates a temperature signal by synthesizing a potential difference between a plurality of temperature signals generated by each of the temperature detection circuits,
A temperature-compensated crystal characterized by driving the temperature third-order component correction signal generation circuit and the temperature first-order component correction signal generation circuit using the synthesized temperature signal to adjust the frequency of the oscillation circuit section. Oscillator. The temperature detection circuit includes:
A first temperature detection circuit for generating a first temperature signal that changes in a linear function with respect to a temperature change;
A second temperature detection circuit that changes in a linear function with respect to a temperature change and generates a second temperature signal different from the first temperature signal;
At least two of
The combining circuit combines the first temperature signal and the second temperature signal to generate a third temperature signal;
2. The frequency adjustment of the oscillation circuit unit is performed by driving the temperature third-order component correction signal generation circuit and the temperature first-order component correction signal generation circuit using the third temperature signal. A temperature-compensated crystal oscillator described in 1.   The first temperature signal has a positive or negative first-order coefficient with respect to a temperature change, and the second temperature signal has a polarity different from that of the first temperature signal with respect to a temperature change. The temperature-compensated crystal oscillator according to claim 2, having a first order coefficient.   4. The temperature compensated crystal oscillator according to claim 2, wherein the synthesis circuit generates a potential of a difference between the first temperature signal and the second temperature signal. 5.

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