JP2008515153A - Contactless multi-position switch using capacitive touch sensor - Google Patents

Abstract

  A touch switch device for detecting the presence of an object such as a human body part, wherein the touch pad, an electric field generated in the vicinity of the touch pad, and a controlled device connected to the touch pad, preferably Touch switch device with integrated local control circuit. Practical applications of touch switch devices include other structures, particularly those that use touch switch devices with multi-position contactless switches that emulate mechanical switches.

Description

(Cross-reference to related applications)
This application claims the benefit of US Provisional Patent Application No. 60 / 613,073 dated September 24, 2004, and US Patent Application No. 10 / 272,377 dated October 15, 2002. Is a continuation-in-part application. The latter application is US Provisional Patent Application No. 60 / 334,040, dated November 20, 2001, US Provisional Patent Application Nos. 60 / 341,350, No. 60, both dated December 18, 2001. / 341,550 and 60 / 341,551 and US Provisional Patent Application No. 60 / 388,245 dated June 13, 2002, and currently claims US Pat. No. 320,282, which is a continuation of US Patent Application No. 09 / 234,150 dated January 19, 1999 and is now US Pat. No. 6,713,897 This is a partially pending application of currently pending US patent application Ser. No. 10 / 027,884 dated Oct. 25. The disclosures of these references are hereby incorporated by reference.

(Field of Invention)
The present invention includes a touch switch (ie, a switch that operates by touching the touch pad or its periphery with a finger, for example, also referred to herein as a touch sensor or field effect sensor) and associated control circuitry; It relates to their practical application.

(Background of the Invention)
Mechanical switches have been used in all types of control devices, including home appliances, power tools, automobiles and related systems, as well as all types of other household tools and industrial equipment. Mechanical switches are generally mounted on a substrate and require some structure that penetrates the substrate. Such a penetrating structure, along with the penetrating structure within the switch itself, can cause dust, water and other contaminants to pass through the board and accumulate inside the switch, which can cause electrical shorts and other malfunctions. Connected.

  Touch switches are often used in place of conventional mechanical switches. Unlike mechanical switches, touch switches do not contain moving parts that are damaged or worn out. In addition, touch switches can be mounted or formed on a continuous substrate sheet, i.e., switch panel, without the need for openings in the substrate. The use of touch switches instead of mechanical switches is therefore particularly advantageous in environments where contaminants tend to be present. Touch switch panels are also easier to clean than mechanical switch panels because they can be made without the need for openings in the substrate that allow entry of contaminants.

  Existing touch switches typically include a touch pad having one or more electrodes. The touch pad communicates with control or interface circuitry, which is often complex and remote from the touch pad. Typically, a signal is supplied to one or more electrodes, including the touch pad, to generate an electric field around the affected electrode. The control / interface circuit detects disturbances to the electric field and generates a response for use by the controlled device.

  Touch switches solve many of the problems associated with mechanical switches, but existing touch switch designs are not perfect. For example, many existing touch switches malfunction when contaminants such as water or other liquids are present on the substrate. The contaminant acts as a conductor against the electric field generated around the touch pad, causing unintended switch drive. This is a problem in the areas of kitchens and certain factory environments where such contaminants are commonly found.

  Existing touch switch designs suffer from other problems related to crosstalk, ie, interference between electric fields around adjacent touch pads. Crosstalk drives the wrong touch switch or drives two switches simultaneously with a single touch close to one touch pad.

  Many existing touch switch designs can also cause unintended drive due to electrical noise and other mutual interference affecting the touch pad itself or leads extending from the touch pad to its associated control circuitry. is there. This problem is exacerbated in applications where the touch pad, such as found in conventional touch switch designs, is relatively far from the control circuitry.

  Existing touch switch designs typically require complex control circuitry to interface with the devices they control. These control circuits often include a large number of individual components that occupy a fairly large space on the circuit board. Due to their physical size, the control circuitry is generally located a long distance away from the touch pad itself. The physical size of the control / interface circuits and their distance from the touch pad further exacerbates the above-mentioned problems, such as crosstalk, electrical noise and mutual interference. Size and remoteness complicate the overall design of the touch switch panel, which further increases production costs and complexity.

  Some existing touch switch designs require a separate ground lead from the touch pad to the interface / control circuit or controlled device. Certain devices that employ conventional mechanical switches do not require such ground leads and are not easy to incorporate. In order to be able to use such a device with such a touch switch, a special grounding structure needs to be added, increasing the time, complexity and cost for design and production. The need for such a ground lead prevents the simple and direct replacement of a conventional mechanical switch panel with a touch switch panel.

  Recent advances in touch switch design include techniques that lower the input and output impedance of the touch switch itself to provide greater resistance to false drive by contaminants and external noise sources. US Pat. No. 5,594,222 describes a low impedance touch switch design that is less sensitive to malfunctioning in the presence of contaminants and electrical noise than many previous designs. Although this scheme has several advantages over the prior art, there are several attributes that limit its application. For example, the manufactured touch switch is sensitive to temperature changes. As long as the temperature change in the output is small compared to the true signal change and small compared to the signal change induced by transistor variations, a single transistor or other amplification device is sufficient. However, this technique requires additional circuitry for interfacing with the controlled device, thus increasing cost and complexity for the overall touch switch design. In applications such as when the dynamic range that allows compensation is not sufficient or when the temperature change is large compared to the true signal change, the different methods may better eliminate or reduce the temperature effect.

  In addition, while this technology's low-impedance scheme can distinguish between contaminants with a certain finite impedance and human contact with a certain finite impedance, this technology is extremely low impedance It is not enough to distinguish between levels. Such a situation appears when the entire touch switch (ie both the inner and outer electrodes) is covered with a large amount of contaminants. A similar, essentially zero impedance situation appears when a conductive material such as a metal pan totally covers the touch switch.

  US Pat. No. 6,310,611, assigned to the same assignee as the present application and incorporated herein by reference, is a differential measurement circuit that addresses many of the problems associated with common mode disturbances affecting touch switches. A touch switch device is disclosed. For example, a touch switch having a two-electrode touch pad can be configured to generate an electric field near each electrode. It is believed that common mode disturbances, such as contaminants that essentially cover both electrodes, have essentially the same effect on the electric field near each electrode. Each electrode supplies a signal proportional to the disturbance to the differential measurement circuit. Therefore, since the signals from the electrodes are assumed to be essentially equal, the differential measurement circuit does not sense the difference and does not respond to common mode disturbances. On the other hand, when only the electric field in the vicinity of one of the electrodes is disturbed, the signal supplied to the differential measurement circuit by that electrode is essentially the same as that supplied by the other, unaffected electrode. Should be different. The differential circuit can respond by providing an output based on the difference in the degree of stimulation at the first and second electrodes, thereby causing a switch drive based on the particular stimulation state of the electrode, or Information can be provided based on a number of stimulus states.

  While differential measurement circuitry can address many of the problems known in the prior art, it is relatively complex and expensive to design and manufacture. A differential measurement circuit generally includes more components than a conventional control circuit. The additional components occupy more space on the touch switch panel. Thus, the control circuit will be farther away from the touch pad than in the case of non-differential circuit designs, thus requiring long leads between the touch pad and its control circuit. This actually exacerbates the problems associated with electrical interference. In addition, when building a differential measurement circuit, component matching is important. Proper part alignment will create additional manufacturing burden and cost. Furthermore, when using differential sensing techniques, especially in low impedance applications, the resulting signal is relatively small compared to the dynamic range of the absolute signal change of the electrode. The resulting signal is thus affected by noise and other environmental effects. For proper buffering of the differential signal, additional components that build a switch or buffer are required. Furthermore, when a stimulus such as a pulse signal is supplied from a remote control circuit, the pulse signal is affected. Stimulus generation circuits, such as pulse generation circuits, generally require components and occupy physical space, but they may interfere with the sensing electrodes. Thus, signal generation circuits need to be physically away from the sensing electrodes if they occupy the physical space that adversely affects or biases the sensing electrodes and equivalently reduces the signal to noise ratio characteristics of the sensors. There is.

  Although the improvements described above reduce unintentional switch drive as a result of crosstalk between switches and electrical interference effects on their control circuits, they do not completely eliminate these problems. Furthermore, they do not address and solve the problems associated with the need for a separate ground circuit in a particular touch switch application. Furthermore, it is desirable that the above features can be realized using as small a physical structure as possible.

  In general, driving a field effect sensor does not require application of force or physical movement of the structure by the user, as in, for example, mechanical push buttons, toggles, or rotary switches. This is a desirable attribute for many applications, but in other applications the physical recognition that the switch has changed state, such as when the user applies force or physically moves the switch structure. It may be desirable to give to the user. In certain applications, it may be desirable to provide a switching mechanism that has the advantages provided by field effect sensors while preserving the mechanical feel of conventional mechanical switches.

(Summary of Invention)
The present invention provides a touch switch device that includes a touch pad and a control circuit located near the touch pad. The touch pad and control circuit are mounted on a dielectric substrate. The control circuit is small compared to the overall size of the device. In a preferred embodiment, the control circuit is essentially reduced to one or more integrated circuits. A physically compact embodiment with integrated control circuitry reduces the sensitivity of the touch switch to common mode interference and crosstalk and interference between adjacent touch switches. The integrated circuit scheme also provides better consistency and balance of control circuit components.

  The touch switch of the present invention can be configured as various preferred embodiments. In some embodiments, the touch switch can emulate a type of mechanical switch that retains conventional contact. In other embodiments, the touch switch can emulate a mechanical switch of the type that touches momentarily. Furthermore, in other embodiments, the touch switch can provide multiple outputs relative to sensing at the sensing electrodes.

  In a preferred embodiment, the touch pad has a first electrode and a second electrode proximate to the first electrode. At least one of the electrodes is electrically connected to a local control circuit. The first and second electrodes and the local control circuit are usually arranged on the same surface of the substrate, and the opposite surface of the substrate is used as a touch surface. However, they do not have to be coplanar and can be arranged on both sides of the substrate.

  In an alternative embodiment, the touch pad has a single electrode that is electrically connected to the local control circuit. In other alternative embodiments, the touch pad has more than two electrodes.

  In a preferred embodiment, the control circuit includes means for generating a signal and supplying it to the touch pad to generate an electric field near one or more electrodes including the touch pad. Alternatively, such a signal can be generated elsewhere and supplied to one or more electrodes to generate one or more electric fields in the vicinity thereof. The control circuit detects a disturbance to the electric field in response to a stimulus to the electric field, such as a user's fingertip touching or approaching a substrate adjacent to the touch switch. The control circuit responds selectively to such electric field disturbances by generating control signals for use by controlled devices, such as household appliances and industrial machinery.

  In a preferred embodiment, the control circuit is configured between the first electrode and the second electrode in response to a stimulus being applied in the vicinity of either the first electrode or the second electrode or both. Detect and respond to potential differences. Such differential measurement circuitry eliminates common mode signals such as temperature, electrical noise, power supply fluctuations and other inputs (ie signals that tend to affect almost equally on both electrodes). I will provide a. The differential measurement circuit also provides the ability to eliminate common mode signals caused by contaminants on the substrate adjacent to the touch switch.

  In a preferred embodiment, the signal is supplied to the first electrode and the second electrode. The signal is generated from within the control circuit or from elsewhere. An electric potential is generated at the location of each electrode, and thus an electric field is generated around each of the electrodes. In the differential measurement circuit, two matched transistors are arranged, the first transistor being connected to the first electrode and the second transistor being connected to the second electrode. The output of each transistor is connected to a peak detection circuit, and the output of each peak detection circuit is then supplied to a decision circuit.

  The output of each transistor changes when the electric field around the electrode to which it corresponds changes, that is, when a user touches or approaches the electrode. The peak detection circuit supplies a signal corresponding to the peak potential from the transistor to the determination circuit in response to a change in the output of the transistor. The decision circuit uses the peak potential in a predetermined manner to provide an output for use by other parts of the control circuit.

  In a preferred embodiment, the inner and outer electrodes function in conjunction with the input of the decision circuit and when the disturbance to the electric field around the first electrode is greater than the degree of disturbance to the electric field around the second electrode In addition, the decision circuit generates a high level output. Conversely, the decision circuit produces a low level output when the disturbance to the electric field around the second electrode is greater than the degree of disturbance to the electric field around the first electrode. When the electric field around both electrodes is disturbed to approximately the same level, the decision circuit produces a low level output.

  The first state appears, for example, when the fingertip essentially covers the first electrode but not the second electrode. The second state appears, for example, when fingertips or contaminants essentially cover the second electrode but not the first electrode. The third state appears when an object such as a contaminant or a metal pan covers both the first and second electrodes, for example.

  The decision circuit output is supplied to other circuit components such as an electrical latch, and the latter selectively outputs a control signal from the control circuit depending on the state of the decision circuit output. In a preferred embodiment, a high level output from the decision circuit will eventually cause a control signal to be output from the control circuit, but no control signal will be output when it is a low level output. In an alternative embodiment, a low level output from the decision circuit causes a control signal to be output from the control circuit, and no control signal is output when it is a high level output.

  Almost all functions that can be performed by mechanical switches such as device turn on and off, temperature adjustment or clock and timer settings can be performed using the touch switch device of the present invention. It can replace existing touch switches and solve problems associated with it. It can also be directly replaced by a mechanical or membrane switch. The touch switch device of the present invention can be used in an environment where the temperature change is extremely large, where there is a large amount of contaminants, or where a metal object is placed on the touch pad. Especially suitable.

  The present invention provides an input circuit section for more efficiently communicating signals between touch pad electrodes and logic and decision circuits. In a preferred embodiment, these inputs of the control circuit include various structures of peak detection circuits and active devices configured to convert high frequency transient pulses into DC signals. These embodiments do not require more complex AC processing circuitry and only use DC processing circuitry, thus reducing the size and cost of the integrated circuit of the touch switch assembly. Furthermore, these preferred embodiments can discharge the electric field associated with the peak detection circuit corresponding to the electric field at the location of the input electrode.

  In another preferred embodiment, the negative effect of stray capacitance caused by the bonding pad and wire bonding configuration is compensated by employing a swamping capacitance at the input of the control circuit described above. . Swamping according to these embodiments of the present invention can eliminate the imbalance introduced by the stray capacitance to the differential measurement circuit, thus allowing more consistent electrical information to be transmitted to the decision circuit.

  In another preferred embodiment, the protection of the control circuit from damage caused by stray currents and often electrostatic high potentials on the input electrodes of the touch pad provides an active blocking device structure at the input of the control circuit. Provided by.

  Other preferred embodiments provide statistical filtering and sampling in noisy and other environments. Furthermore, other preferred embodiments provide for linearization of the input signal sent to the decision circuit using differential measurement techniques.

  The present invention further provides a dual-connected latch circuit that facilitates the direct replacement of membrane and other mechanical switches with touch sensor switches. In a preferred embodiment, this latch circuit arrangement provides control of the touch switch assembly and isolation from the inherent leakage current path originating from the doped substrate used to make the integrated circuit. . One object of the present invention is to provide an analog output that takes advantage of the input configuration characteristics of the circuit utilized by the present invention. Another object of the present invention is to provide a method for sensing capacitive inputs.

  The present invention is also aimed at practical applications of touch switches. While the touch switches described herein are particularly suitable for use with many of the applications described herein, other touch switches and sensors, such as US Pat. No. 5,594, incorporated herein by reference. , 222 and 6,310,611 can be used in such applications as well as capacitive and field effect sensors.

  Various features, advantages and other uses of the present invention will become more apparent with reference to the following detailed description and drawings.

(Detailed description of the drawings)
U.S. Pat.Nos. 5,594,222, 5,856,646, 6,310,611, 6,320,282, 6,713,897 and 6,897,390, US patent application Ser. No. 10 / 271,933, filed Oct. 15, 2002 and entitled Intellectual Shelving System, assigned to the assignee of the present invention. See disclosure of 10 / 272,047 entitled Sensor (With Touch Sensor Integrated Integration), 10 / 850,272 entitled Integrated Touch Sensor and Light Apparatus (Integrated Touch Sensor and Light Apparatus). Therefore, it incorporated herein.

  The present invention relates to a touch switch device including a touch pad having one or a plurality of electrodes and one control circuit. To facilitate understanding of many of the drawings illustrating the control circuit, the circuit is drawn larger than the touch pad. However, in a typical application, the control circuit is small compared to the touch pad and is preferably in the form of one or more integrated circuit chips.

  FIG. 1 is a perspective view of one preferred embodiment of a touch switch device 20 of the present invention. The touch switch device 20 includes a touch pad 22, and a control circuit 24 including an integrated circuit (IC) chip 26 having eight output terminals PIN1-PIN8 and first and second resistors R1 and R2. In the illustrated embodiment, the touch pad 22 includes a first electrode E1 and a second electrode E2, although the touch pad may include more or less than two electrodes. . Although it is possible to manufacture the control circuit 24 using individual electronic components, it is desirable to embody the control circuit in an integrated circuit chip such as the IC chip 26.

  The control circuit 24 controls the first and second resistors R1 and R2, the first and second electrodes E1 and E2, and a remote device (not shown) via terminals PIN1-PIN8 of the IC chip 26. And / or connected to and in communication with input lines 30 configured to provide power signals. The control circuit 24 also communicates with a remote device (not shown) using the first output line 32. In some embodiments, a second output line 34 is also used for communication with a remote device (not shown).

  FIG. 2 is a partial cross-sectional view of an exemplary touch switch 20 of the present invention, in which components including the touch switch device 20 are mounted on a dielectric substrate 35 having a front surface 36 and an opposite back surface 37. . In the illustrated embodiment, the first and second electrodes E 1 and E 2 are mounted on the back surface 37 of the substrate 35. The IC chip 26 is also mounted on the back surface 37 of the substrate 35 close to the first and second electrodes E1 and E2. As can be seen from FIGS. 1 and 2, in the preferred embodiment, it is assumed that an IC chip 26 including a control circuit 24 is mounted in close proximity to the touch pad 22.

  The substrate 35 typically comprises a relatively strong dielectric material such as glass, plastic, ceramic or any suitable dielectric material. However, the substrate 35 can also include any other dielectric material including a flexible material. An example of a suitable flexible substrate is No. 1 from Consolidated Graphics. HS-500, Type 561, Level 2 0.127 mm (0.005 inch) thick polyester material. In embodiments in which the components of the touch switch device are mounted on a flexible substrate, it is often the case that the flexible carrier is later attached to another generally more robust substrate.

  In a preferred embodiment, the substrate 35 comprises glass having a uniform thickness of about 3 mm. In other embodiments, the thickness of the substrate 35 varies depending on the type of material used, its mechanical and electrical properties, and the physical strength and electrical sensitivity required for the particular application. . The maximum functional thickness for glass and plastic substrates is on the order of several centimeters (inches). However, in the most practical application, the thickness of the glass substrate is in the range of about 1.1 mm to about 5 mm, while the plastic substrate is much thinner.

  In a preferred embodiment, as shown in FIGS. 1 and 2, the second electrode E2 essentially surrounds the first electrode E1. A gap 28 is provided between the first electrode E1 and the second electrode E2. The first electrode E1 is sized to be “covered” by the user's fingertip or other human part when the user touches a corresponding portion of the front surface 36 of the substrate 35. In one preferred embodiment, the first electrode E1 has a rectangular shape, and the second electrode E2 has a rectangular pattern similar to the shape of the first electrode E1 and surrounding the periphery thereof.

  The shape of the touch pad shown in FIGS. 1 and 2 represents a preferred arrangement of the first and second electrodes E1 and E2, but this electrode shape is extensible to meet a wide variety of applications. It can be understood that it can be deformed. For example, the size, shape, and placement of the electrodes can be varied to accommodate the size of the site or other stimulus envisioned for driving the touch switch 20. For example, in special applications, a hand rather than a finger may be required to stimulate or drive the touch switch 20. In such an application, the first and second electrodes E1 and E2 are much larger and spaced apart.

  The first electrode E1 can take a plurality of arbitrary geometric shapes including, but not limited to, rectangular, trapezoidal, circular, elliptical, triangular, hexagonal and octagonal. Regardless of the shape of the first electrode E1, the second electrode E2 is configured to at least partially surround the first electrode E1 at an interval. However, in order to obtain the benefits of the invention, the second electrode E1 need not even partially surround the first electrode. For example, the first and second electrodes E1 and E2 can be adjacent to each other as shown in FIG. In an alternative embodiment, the second electrode E2 is omitted.

  Further, the electrode shapes need not be coplanar, but may be spheres, cubes, or three-dimensional shapes that conform to other geometric shapes. This design flexibility allows the invention to be used in a variety of applications with substrates of various shapes and components. In some applications, the touch pad 22 and control circuit 24 may not actually touch the touch substrate 35 located thereon or within. For example, FIG. 8 shows a touch switch device 20 in which first and second electrodes E 1 and E 2 are mounted on the outer surface 113 of the first frame 111 of the thermopane window 110. It can be driven by the user providing an appropriate stimulus 115 near the outer surface 114 of the frame 112 on the opposite side of the window.

  As mentioned above, the first and second electrodes E1 and E2 do not have to be on the same plane, and they can be mounted on both sides or both sides of the substrate, or together on different substrates. For example, FIG. 9 shows that the first electrode E1 is on the first surface 36 of the substrate 35 and the second electrode E2 and the IC chip 26 are on the second, opposite surface 37 of the substrate 35, respectively. The mounted touch switch device 20 is shown. In applications where the first and second electrodes E1 and E2 are mounted on the same side of one substrate, the IC chip 26 can be mounted on the same side of the substrate as the electrode or on another side of the substrate. If the first and second electrodes are mounted on different surfaces of one substrate or mounted together on different substrates, the IC chip 26 can be on the same surface as either of the electrodes or on different surfaces together Or it can mount on a board | substrate. However, it is desirable to mount the IC chip 26 close to the electrodes.

  The first electrode E1 is preferably a solid conductor. However, the first electrode E1 may have a plurality of openings or may have a mesh or grid pattern. In some embodiments, the second electrode E2 has a narrow ribbon shape that partially surrounds the first electrode E1. In other embodiments in which the first and second electrodes E1 and E2 are simply adjacent to each other, the second electrode E2 is also a solid conductor or has a mesh or grid pattern.

  The control circuit 24 can be designed in many different ways and can be used with a variety of power sources such as AC, periodically varying DC (eg, square wave), continuous DC, or the like. . FIGS. 4-7 illustrate a preferred control circuit design that can be easily adapted for use with a variety of power sources in a variety of operating modes. The embodiment of FIG. 4 uses square wave DC power in a differential input strobe mode of operation. The embodiment of FIG. 5 uses continuous DC power in a continuous DC mode with differential inputs. The embodiment of FIG. 6 uses square wave DC power in a single-ended input strobe mode. The embodiment of FIG. 7 uses continuous DC power in a single-ended input continuous DC mode.

  As is apparent from FIGS. 4-7, the control circuit 24 can be easily adapted to various different operating modes. The four modes described above are described in detail to demonstrate the design flexibility enabled by the present invention. However, it should be recognized that the invention is in no way limited to those four modes of operation. The special operating mode and power supply used for a particular application mainly depends on the requirements and specifications of the controlled device.

  Enclosed regions B1 and B2 in FIGS. 4-7 are on the IC chip 26, as are electrodes E1 and E2, resistors R1 and R2, controlled devices (not shown), and input line 30 and output line 32, respectively. A distinction is shown between a component that is supposed to be located and a component that is located off the IC chip 26. The portions other than the enclosed regions B1 and B2 in FIG. 4-7 are assumed to be located on the IC chip 26, and are the same for all of the four drawings and operation modes shown here. The enclosed area B6 includes the input part of the control circuit. Various configurations of the input unit included in the enclosed region B6 will be described below with reference to FIGS. 11A-18E.

  FIGS. 4-7 show the control circuit 24 including the start and bias section 40, the pulse generator and logic section 50, the decision circuit section 60 and the self-holding latch section 70, the functions of which are described below. explain. Each of the circuit sections 40, 50, 60 and 70 described above can be designed in a number of different ways as known to those skilled in the art of electronic integrated circuit design.

  Control circuit 24 also includes first, second and third transistors P1, P2 and P3. In the embodiment described herein, transistors P1-P3 are P-MOS devices, but N-MOS devices, bipolar devices, or other transistor types may be used. The control circuit 24 further includes an inverter I1, first, second and third diodes D1-D3, first and second capacitors C1 and C2, first, second, third and fourth transistor switches SW1. -Includes SW4 and third and fourth resistors R3 and R4. It will be appreciated that the third and fourth resistors R3 and R4 are replaced by current sources or active loads.

  In each embodiment shown in FIGS. 4-7, the source terminal 77 of the third transistor P3, the start-up and bias section 40, the pulse generator and logic section 50, the decision circuit 60 and the self-holding latch section 70. The power input terminals 41, 51, 61 and 71 are electrically connected to the terminal PIN 8 of the IC chip 26. The terminal PIN8 is then electrically connected to the power input line 30 of the control circuit 24, and the latter is then electrically connected to the power supply 25. Typically, the power source 25 is located at a controlled device (not shown).

  The bias output terminal 43 from the start and bias section 40 is electrically connected to the gate terminals G2 and G4 of the second and fourth transistor switches SW2 and SW4, respectively. In the preferred embodiment, the first through fourth transistor switches SW1-SW4 are N-MOS devices, as described herein with respect to FIGS. 4-7, but others as shown in FIGS. 11A-18E. These types of transistors and combinations can be used as well.

  The power-on reset output 44 from the start-up and bias section 40 is electrically connected to the power-on reset input 54 of the pulse generator and logic section 50. The power-on reset output 44 from the start-up and bias section 40 is also electrically connected to the gate terminals G1 and G3 of the first and third transistor switches SW1 and SW3.

  The internal ground reference output 42 from the start and bias section 40 is the low potential plates 102 and 103 of the first and second capacitors C1 and C2, and the source terminals S1, S2 of the first to fourth transistor switches SW1-SW4. , S3 and S4, the internal ground reference output 52 of the pulse generator and logic section 50, the internal ground reference output 62 of the decision circuit 60, the anode 98 of the third diode D3, the low of the third and fourth resistors R3 and R4 The potential ends 96 and 97 and the terminal PIN6 of the IC chip 26 are electrically connected. The nodes listed here will often be referred to hereinafter as the internal ground reference CHIP VSS.

  The pulse output 53 from the pulse generator and logic section output 50 is electrically connected to the source terminals 80 and 81 of the first and second transistors P1 and P2, respectively, and to the terminal PIN2 of the IC 26. The gate terminal 82 of the first transistor P1 is electrically connected to the terminal PIN1 of the IC 26. The gate terminal 83 of the second transistor P2 is electrically connected to the terminal PIN3 of the IC 26.

  The drain terminal 84 of the first transistor P1 is electrically connected to the anode 90 of the first diode D1 and the high potential end 94 of the third resistor R3. The drain terminal 85 of the second transistor P2 is electrically connected to the anode 91 of the second diode D2 and the high potential end 95 of the fourth resistor R4.

  The cathode 92 of the first diode D1 is connected to the PLUS input terminal 64 of the decision circuit 60, the drain terminals 86 and 87 of the first and second transistor switches SW1 and SW2, and the high potential plate 100 of the first capacitor C1. Is electrically connected. The cathode 93 of the second diode D2 is electrically connected to the MINUS input terminal 66 of the decision circuit 60, the drain terminals 88 and 89 of the third and fourth transistor switches SW3 and SW4, and the high potential plate 101 of the second capacitor C2. Connected.

  The logic output 63 of the decision circuit 60 is electrically connected to the input 75 of the inverter I1 and the latch trigger input 73 of the self-holding latch section 70. The output 72 of the self-holding latch section 70 is electrically connected to the terminal PIN4 of the IC 26.

  In the illustrated embodiment, decision circuit section 60 has its PLUS and MINUS inputs 64 and 66 each at an essentially equal potential, or MINUS input 66 is at an essentially higher potential than PLUS input 64. At some point, its output 63 is designed to be low potential. The output 63 of the decision circuit section 60 is high potential only when the PLUS input 64 is essentially at a higher potential than the MINUS input 66.

  The self-holding latch section 70 passes through the latch section 70 from the power supply 25 of the control circuit 24 to the internal ground reference CHIP VSS and the third diode D3 when the logic output 63 of the decision circuit section 60 is low. Designed so that no current flows through it. However, when the logic output 63 of the decision circuit 60 section is at a high potential, the latch trigger input 73 is at a high potential, triggering the latch circuit 70 and passing through the latch section 70 to supply power to the control circuit 24. From 25 to the internal ground reference CHIP VSS and through the third diode D3, current is caused to flow by the power input of latch 70 and by output terminals 71 and 72, respectively. Once the latch 70 is triggered, it remains triggered or contained until power is removed from the control circuit 24. The design and construction of latch sections operating in this manner are known to those skilled in the art and need not be described in detail here.

  The output terminal 76 of the inverter I1 is electrically connected to the gate terminal 78 of the third transistor P3. The drain terminal 79 of the third transistor P3 is electrically connected to the terminal PIN7 of the IC 26.

  A third diode D3 is provided to prevent reverse bias of the control circuit 24 when the touch switch device 20 is used in a multiplexed application. In applications where only a single touch pad 22 is used, or in applications where multiple touch pads 22 are used but not multiplexed, it can be omitted.

  The above description of the basic design of the control circuit 24 is the same for each of the four operation modes shown in FIGS. 4-7. The overall device configuration distinction between the four modes of operation resides primarily in the external terminal connections of the IC 26, as will be described in detail below. FIG. 4 shows a touch switch device 20 configured to operate in the differential input strobe mode described below. Control circuit 24 operating in this mode is generally configured as described above in connection with FIGS. 4-7. Terminal PIN2 of IC 26 is electrically connected to high potential ends 104 and 105 of first and second resistors R1 and R2, respectively. The terminal PIN1 of the IC 26 is electrically connected to both the low potential end 106 of the first resistor R1 and the first electrode E1. The terminal PIN3 of the IC 26 is electrically connected to both the low potential end 107 of the second resistor R2 and the second electrode E2.

  The circuit elements represented as C3 and C4 in FIGS. 4-7 are not individual electrical components. Rather, the reference symbols C3 and C4 represent the capacitance between ground of the first and second electrodes E1 and E2, respectively.

  The terminal PIN8 of the IC 26 is electrically connected to the input line 30. The latter is then electrically connected, for example, to a power signal source 25 in a controlled device (not shown). Terminal PIN4 of IC 26 is electrically connected to terminal PIN6 of IC 26, thereby electrically connecting output terminal 72 of latch 70 to anode 98 of third diode D3 and internal ground reference CHIP VSS. . In this embodiment, the terminal PIN7 of the IC chip 26 is not terminated externally. The terminal PIN5 of the IC 26 is electrically connected to the output line 32. The latter is then electrically connected to the high potential end 108 of the fifth resistor R5 and the output line 120, which is controlled directly or via a processor or other intermediate device (not shown). Connected to a device (not shown). The low potential end 109 of resistor R5 is electrically connected to system ground. In a typical application, resistor R5 is placed at a significant distance from other components including touch switch device 20. That is, in the preferred embodiment, resistor R5 is not assumed to be in the vicinity of touch pad 22 and control circuit 24.

  FIG. 5 illustrates an exemplary touch switch control circuit 24 configured to operate in a continuous DC mode with differential inputs, as described below. The overall control circuit and apparatus is the same as described above with respect to FIG. 4, but there are three different points. First, in the embodiment of FIG. 5, terminal PIN7 of IC 26 is electrically connected to high potential end 108 of resistor R5 and output line 120, the latter being either directly or a processor or other intermediate device ( The terminal PIN7 is not terminated externally in the embodiment of FIG. 4 although it is connected to a controlled device (not shown) via a not-shown). Second, in the embodiment of FIG. 5, the terminals PIN4 and PIN6 of the IC 26 are not electrically connected to each other or terminated externally, but in the embodiment of FIG. It has become. Third, in the embodiment of FIG. 5, the terminal PIN5 of the IC 26 is electrically connected to the low potential end 109 of the resistor R5. In the embodiment of FIG. 4, the terminal PIN5 of the IC 26 is 5 is electrically connected to a high potential end 109 of resistance and a controlled device (not shown). As in the embodiment of FIG. 4, the fifth resistor R5 is typically separated from other components including the touch switch device 20 by a considerable distance.

  FIG. 6 illustrates an exemplary touch switch control circuit configured to operate in a single-ended input strobe mode, as described below. The control circuit 24 is generally configured as described above with respect to FIGS. 4-7. Terminal PIN2 of IC 26 is electrically connected to high potential ends 104 and 105 of first and second resistors R1 and R2, respectively. The terminal PIN1 of the IC 26 is electrically connected to both the low potential end 106 of the first resistor R1 and the first electrode E1. The terminal PIN3 of the IC 26 is electrically connected to both the low potential end 107 of the second resistor R2 and the high potential end 110 of the sixth resistor electrode R6, and the second resistor R2 and the sixth resistor R6 In this way, a voltage divider is formed. The low potential end 111 of the sixth resistor R6 is typically electrically connected to the internal ground reference CHIP VSS at a point close to the terminal PIN5 of the IC 26. In FIG. 6, the electrical connection of the sixth resistor R6 to the internal ground reference CHIP VSS is represented by a dashed line “AA” for clarity.

  The terminal PIN8 of the IC 26 is electrically connected to the input line 30, and the latter is then electrically connected to the power signal source 25. Terminal PIN5 of IC 26 is electrically connected to output line 32, which in turn is electrically connected to high potential end 108 of fifth resistor R5 and output line 120. The output line 120 is electrically connected to a controlled device (not shown) directly or through a processor or other intermediate device. The terminal PIN4 of the IC 26 is electrically connected to the terminal PIN6 of the IC 26. The terminal PIN7 of the IC 26 is not terminated externally in this embodiment. In a typical application, the fifth resistor R5 is separated a significant distance from other components including the touch switch device 20.

  FIG. 7 illustrates an exemplary touch switch control circuit configured to operate in a single-ended input continuous DC mode, as described below. The control circuit 24 is generally configured as described above in connection with FIGS. 4-7. The overall control circuitry and apparatus is the same as described above with respect to FIG. 6, with three differences. First, in the embodiment of FIG. 7, terminal PIN7 of IC 26 is electrically connected to high potential end 108 of fifth resistor R5 and output line 120, which is then typically processor. And other connected controllers (not shown) via other controllers (not shown). The terminal PIN7 of the IC 26 is not terminated externally in the embodiment of FIG. Second, in the embodiment of FIG. 7, the terminals PIN4 and PIN6 of the IC 26 are neither electrically connected to each other nor terminated externally, but in the embodiment of FIG. It has become. Third, in the embodiment of FIG. 7, the terminal PIN5 of the IC 26 is electrically connected to the low potential end 109 of the resistor R5. However, in the embodiment of FIG. 5 is electrically connected to the high potential end 108 of the resistor 5 and the output line 120. In a typical application, the fifth resistor R5 is spaced a significant distance from other components including the touch switch device 20. In FIG. 7, the broken line “AA” represents the electrical connection of the sixth resistor R <b> 6 to the internal ground reference CHIP VSS.

  The touch switch device 20 configured to operate in the differential input strobe mode operates as follows. Referring to FIG. 4, a power / control signal 25 is applied to terminal PIN8 of IC 26, and then each of start and bias section 40, pulse generator and logic section 50, decision circuit section 60 and self-holding latch section 70, respectively. Electrically connected to power input terminals 41, 51, 61 and 71.

  After power-up and a suitable delay time for stabilization (approximately 25 microseconds is sufficient, but may be shorter or longer depending on the application), the start-up and bias section 40 is A power-on reset signal with a short pulse length is output from the output terminal 44 to the gate terminals G1 and G3 of the first transistor switch SW1 and the third transistor switch SW3, respectively, and the first and third transistor switches SW1 are output. And SW3 are turned on, thereby providing a current path from the high potential plates 100 and 101 of the first and second capacitors C1 and C2, respectively, to the internal ground reference CHIP VSS. The duration of the power-on reset signal is long enough that all charge present on the first and second capacitors C1 and C2 is essentially completely discharged to the internal ground reference CHIP VSS. In this way, the PLUS and MINUS inputs 64 and 66 to the decision circuit section 60 return to the initial low potential state.

  Essentially simultaneously, start-up and bias circuit 40 sends a power-on reset signal from output 44 to pulse generator and logic section 50 input 54 to initialize it. After a suitable delay for the pulse generator and logic section 50 to stabilize, the pulse generator and logic section 50 generates a pulse from the pulse output terminal 53 and the first and second resistors R1 and It outputs to the first and second electrodes E1 and E2 via R2, and also outputs to the source terminals 80 and 81 of the first and second transistors P1 and P2, respectively. The pulse may be any suitable waveform such as a square wave pulse.

  The start-up and bias circuit 40 also outputs a bias voltage from the bias output 43 to the gate terminals G2 and G4 of the second and fourth transistor switches SW2 and SW4, respectively. The bias voltage is different in phase from the pulse output for the first and second electrodes. That is, when the pulse output is high, the bias voltage output is low, and when the pulse output is low, the bias voltage output is high.

  The voltages at the gate terminals 82 and 83 of the first and second transistors P1 and P2 when a pulse is supplied to the first and second electrodes E1 and E2 via the first and second resistors R1 and R2, respectively. Initially biases the first and second transistors P1 and P2 to turn on because they are lower than the voltages at the source terminals 80 and 81 of the first and second transistors P1 and P2, respectively. When the first and second transistors P1 and P2 are turned on, current flows through the third and fourth resistors R3 and R4, and the anode terminals 90 and 91 of the first and second diodes D1 and D2 Produces a peak potential.

  When the peak potential at the anode terminals 90 and 91 of the first and second diodes D1 and D2 is higher than the potential across the first and second capacitors C1 and C2, the first and second diodes D1 and D2 are A peak current is generated to charge the first and second capacitors C1 and C2 and generate a peak potential at each of the PLUS and MINUS inputs 64 and 66 to the decision circuit section 60. This situation occurs, for example, following the first pulse after the control circuit 24 is initialized. This is because, as described above, the first and second capacitors C1 and C2 are charged when the power is turned on.

  As will be apparent to those skilled in the art, the bias of the first and second transistors P1 and P2, the current flowing through the third and fourth resistors R3 and R4, the anodes 90 and 91 of the first and second diodes D1 and D2 And the peak potential generated at each of the PLUS and MINUS inputs 64 and 66 to the decision circuit 60 are proportional to the state of the electric field at the first and second electrodes E1 and E2. The state of the electric field in the vicinity of the electrodes E1 and E2 changes according to the stimulus existing in the vicinity of the electrodes.

  As explained above, when the control circuit 24 is driven and there is no stimulus in the vicinity of any of the first and second electrodes E1 and E2, the PLUS and MINUS inputs 64 and 66 to the decision circuit 60 The potential at each of these will be referred to as a neutral state. In the neutral state, the potentials at each of the PLUS and MINUS inputs 64 and 66 are essentially equal. However, in order to prevent unintended driving, it is preferable to adjust the control circuit 24 so that the neutral state of the MINUS input 66 is somewhat higher than the neutral state of the PLUS input 64. This adjustment is performed by changing the configuration of the first and second electrodes E1 and E2 and the values of the first and second resistors R1 and R2 to achieve the desired neutral state potential. Regardless of the neutral potential, the output 63 of the decision circuit 60 is assumed to remain at a low potential until the PLUS input 64 is essentially higher.

  When the output 63 of the decision circuit 60 is at a low potential, the inverter I1 causes the potential of the gate terminal 78 of the third transistor P3 to be high, essentially equal to the potential of the source terminal 77. In this state, the third transistor P3 is not biased and remains turned off. However, in this embodiment, the terminal PIN7 of the IC 26 is not terminated. The drain terminal 79 of the third transistor P3 is therefore in an open circuit state, and the state of the third transistor P3 has no influence on the function of the device. In addition, if the decision circuit 60 output 63, and hence the latch trigger input 73, is low and the self-holding latch circuit 70 is not triggered, it will flow from the power supply 25 through the latch 70 to the internal ground reference CHIP VSS, and No current flows through the third diode D3.

  During the period determined by the pulse voltage, the values of the first and second resistors R1 and R2 and the capacitance of the first and second electrodes E1 and E2 to ground (represented in the drawing as virtual capacitors C3 and C4), The potential of the first and second electrodes E1 and E2 eventually rises until essentially equal to the pulse voltage, ie the voltage at the source terminals 80 and 81 of the first and second transistors P1 and P2. Thereby, the first and second transistors P1 and P2 are not biased. When this state is reached, the first and second transistors P1 and P2 are turned off, and the potentials at the anodes 90 and 91 of the first and second diodes D1 and D2 are internally grounded at an essentially equal rate. It starts to decrease towards the reference CHIP VSS level. Eventually, the anode potential of each of the first and second diodes D1 and D2 drops to a level lower than the corresponding cathode potential. At this point, the diodes D1 and D2 are in a reverse bias state, preventing the discharge of the first and second capacitors C1 and C2.

  When the output 53 pulse changes to a low state, the bias voltage output changes to a high state compared to the internal ground reference CHIP VSS, and the increased bias voltage is the gate terminal of the second and fourth transistor switches SW2 and SW4. Supplied to G2 and G4. In this state, the second and fourth transistor switches SW2 and SW4 are slightly biased to discharge the first and second capacitors C1 and C2 to the internal ground reference CHIP VSS in a controlled manner. Turn on for enough. When the pulse next changes to a high state, the bias voltage returns to a low state, the second and fourth transistor switches SW2 and SW4 are turned off, and the circuit responds as initially described. .

  When the pulse from the pulse generator and logic section 50 changes to a high potential, if there is a stimulus at or near the second electrode location, the first transistor P1 operates as described above. To do. That is, the first transistor P1 initially allows some current that is biased to flow through the third resistor R3, and generates a peak potential at the anode 90 of the first diode D1. The peak current flowing through the diode D1 is allowed, thereby charging the first capacitor C1 and generating a peak potential at the PLUS input 64 to the decision circuit 60. Once the voltage of the first electrode E1 is established in response to the input pulse, the first transistor P1 becomes unbiased and turns off.

  Except that the presence of a stimulus in the vicinity of the second electrode E2 changes the RC time constant of the circuit segment, and the time required for the potential of the second electrode E2 to stabilize becomes longer. The second transistor P2 operates in substantially the same manner. As a result, the second transistor P2 remains biased on for a longer time than the first transistor P1, and the peak current flowing through the fourth resistor R4 is greater than that flowing through the third resistor R3. A larger peak potential is generated at the anode 91 of the second diode D2 than the peak potential present at the anode 90 of the first diode D1. Accordingly, a peak current flows through the second diode D2 to charge the second capacitor C2, and eventually a peak potential greater than the peak potential at the PLUS input 64 to the decision circuit is applied to the decision circuit 60. Generate at MINUS input 66. Since the decision circuit 60 is configured such that when the potential at the MINUS input 66 is greater than or substantially equal to the potential at the PLUS input 64, its output is low, the output terminal of the decision circuit 60 63 becomes a low potential.

  If the output terminal 63 of the decision circuit 60 and, consequently, the latch / trigger input terminal 73 is at a low potential, the self-holding latch 70 is not triggered. Inverter I1 and third transistor P3 operate as previously described, but again the state of third transistor P3 is not important in this configuration.

  If contaminants or extraneous objects or other stimuli essentially cover or adhere to both the first and second electrodes E1 and E2, the system will not be both the first electrode and the second electrode. Respond as if no stimulus was present. However, when contaminants or extraneous objects are present in the vicinity of both electrodes E1 and E2, the RC time constant for those segments of the circuit changes, and the voltage on both the first and second electrodes E1 and E2 becomes a pulse voltage. The time required for the equalization becomes longer. Thus, both the first and second transistors P1 and P2 are turned on, and the third and fourth currents are greater than when neither the first nor second electrode E1 is stimulated. Flows through resistors R3 and R4. However, the first and second transistors P1 and P2 are essentially biased equally. Thus, essentially equal peak potentials occur at the anodes 90 and 91 of both the first and second diodes D1 and D2, and essentially equal peak currents are passed through the first and second diodes D1 and D2. Causing the first and second capacitors C1 and C2 to charge and establish a substantially equal peak potential at both the PLUS and MINUS inputs 64 and 66 to the decision circuit 60. In this state, the output terminal 63 of the decision circuit section 60 is at a low potential, the latch trigger input terminal 73 of the self-holding latch 70 is at a low potential, and the latch 70 remains untriggered. As previously mentioned, the state of inverter I1 and third transistor P3 is not important in this embodiment.

  In a situation where stimulation is applied in the vicinity of the first electrode E1, but not applied to the second electrode, the second transistor P2 is initially biased to turn on and turn on the fourth resistor. A current is passed through R4 to generate a peak potential at the anode terminal 90 of the second diode D2. A peak current flows through the second diode D2, charges the second capacitor C2, and establishes a peak potential at the MINUS input 66 of the decision circuit section 60. As the voltage at the gate terminal 81 of the second transistor P2 rises toward the level of the pulse voltage, the second transistor P2 becomes unbiased and turns off. The second D2 is then reverse biased to prevent the second capacitor C2 from discharging.

  As will be apparent to those skilled in the art, the presence of stimulation in the vicinity of the first electrode E1 extends the time required for the potential of the first electrode E1 to stabilize. As a result, the first transistor P1 stays biased for a longer time than the second transistor P2, flows a peak current larger than that flowing through the fourth resistor R4 through the third resistor R3, A peak potential higher than the potential at the anode 91 of the diode D2 is generated at the anode 90 of the first diode D1. Accordingly, a peak current having a larger amplitude and / or time length than that flowing through the second diode D2 flows through the first diode D1, charges the first capacitor C1, and finally the decision circuit 60. A peak potential that is substantially greater than the peak potential of the MINUS input 66 to the PLUS input 64 to the decision circuit 60 is produced. Since the decision circuit 60 is configured such that the output terminal 63 is in a state where the potential at the PLUS input 64 is higher than the potential at the MINUS input 66, the output 63 of the decision circuit 60 becomes a high potential.

  If the output 63 of the decision circuit 60 is at a high potential, the inverter I1 lowers the potential at the gate terminal 78 of the third transistor P3 compared to the potential at the source terminal 77, thereby biasing the third transistor P3. Turn it on. However, since the terminal PIN7 of the IC 26 is not terminated in this embodiment, the state of the third transistor P3 is not important.

  When the output terminal 63 of the decision circuit 60 is at a high potential, the trigger input terminal 73 of the self-holding latch circuit 70 is also at a high potential and triggers the latch 70. When the self-holding latch 70 is triggered, a current path is established from the power supply 25 to the internal ground reference CHIP VSS and through the third diode D3, the start-up and bias section 40, the pulse generator and logic section 50 and The rest of the control circuit 24 including the decision circuit section 60 is equivalently shorted. In this state, those sections of the control circuit 24 are essentially turned off and stop functioning.

  Once triggered, the self-holding latch 70 remains triggered regardless of how the stimulus subsequently approaches one or both of the electrodes E1 and E2. The latch 70 is reset when the power from the power supply 25 drops to near zero, such as when the square wave strobe signal from the power supply 25 in this example drops to zero.

  While the self-holding latch 70 is in the trigger state, a steady state signal is provided through the fifth resistor R5 and returned to the controlled device (not shown). In this way, the touch switch device 20 can emulate a change in state associated with a mechanical switch that retains contact.

  Referring now to FIG. 5, the touch switch device 20 configured for differential input continuous DC mode operates as follows. The control circuit 24 including up to the decision circuit 60 operates essentially the same as when configured for the differential input strobe mode of operation, as described above with reference to FIG. That is, when there is no stimulus close to both the first and second electrodes E1 and E2, when there is a stimulus close to both the electrodes E1 and E2, or in the vicinity of the second electrode E2. Exists, but does not exist in the vicinity of the first electrode E1, the output 63 of the decision circuit 60 is at a low potential. When the stimulus is in the vicinity of the first electrode E1 and not in the vicinity of the second electrode E2, the output 63 of the decision circuit 60 is at a high level.

  As can be readily seen from FIG. 5, the output 72 of the self-holding latch circuit 70 is not terminated in this embodiment, and therefore the self-holding latch 70 does not operate in the differential input DC mode. However, the drain terminal 79 of the third transistor P3 is electrically connected to the internal ground reference CHIP VSS and, in this embodiment, to the output line 32, so that it becomes the main part of the control circuit 24. When the output 63 of the decision circuit 60 is at a low potential, the inverter I1 sets the potential of the gate terminal 78 of the third transistor P3 to a high potential, and is therefore essentially equal to the potential source terminal 77. In this state, the third transistor P3 is not biased and does not turn on. When the output 63 of the decision circuit 60 is at a high potential, the inverter I1 sets the potential of the gate terminal 78 of the third transistor P3 to a potential lower than that of the potential / source terminal 77. In this state, the third transistor P3 is biased to turn on, and a current flows through the third transistor P3 and the fifth resistor R5. The output line resistor R5 limits the current through the third transistor P3 so that the rest of the control circuit 24 is not shorted and remains in operation.

  In the DC mode shown in FIG. 5, the control circuit 24 is also responsive to stimuli being removed from the vicinity of the first electrode E1. Each time the pulse goes high, unless the stimulus stays in the vicinity of the first electrode E1 and in the vicinity of the second electrode E2, the anode 90 of the first diode D1 is connected to the anode of the second diode D2. A peak potential higher than the peak potential of 91 is generated. Accordingly, the peak potential at the PLUS input 64 to the decision circuit 60 is higher than the peak potential at the MINUS input 66, and the control circuit 24 behaves as described above. However, when the stimulus is removed and there is no stimulus in the vicinity of either the first electrode E1 or the second electrode E2, the charge on the first capacitor C1 is biased to the second transistor switch SW2. Thanks to its function, it will eventually discharge to a neutral state. At this point, the potential at the PLUS input 64 of the decision circuit 60 is not higher or essentially higher than the potential at the MINUS input 66, so the output 63 of the decision circuit 60 returns to a low state.

  Thus, the touch switch device 20 operating in the DC mode of differential input emulates a mechanical switch of the type that touches momentarily and closes, releases and opens. It will be understood that with minor modifications, the control circuit can be configured to emulate a mechanical switch that is pushed open and released and closed.

  Referring now to FIG. 6, the touch switch device 20 configured for the single-ended input strobe mode of operation operates as follows. When a pulse is applied to the first electrode E1 and the first and second resistors R1 and R2, a current flows through the second resistor R2 and the sixth resistor R6. The second and sixth resistors R2 and R6 are configured as a voltage divider. That is, when the pulse output is high, the gate terminal 83 of the second transistor P2 has a lower potential than the source terminal 81 of the second transistor P2. Therefore, when the pulse output 53 is high, the second transistor P2 is continuously biased to supply a constant current flowing through the fourth resistor R4 and to the anode 91 of the second diode D2. Generate a potential. The reference potential at the anode 91 of the second diode D2 establishes a current through the second diode D2, charges the second capacitor C2, and generates a reference potential at the MINUS input 66 to the decision circuit 60. When the reference potential at the MINUS input 66 is essentially equal to the reference potential at the anode 91 of the second diode D2, the current flowing through the second diode D2 stops.

  At the same time, when there is no stimulus on the first electrode E1, the pulse supplied to the source terminal 80 of the first transistor P1 and the first electrode E1 is initially turned by biasing the first transistor P1.・ Turn it on. Thus, a current through the third resistor R3 is established, and a peak potential is generated at the anode 90 of the first diode D1. The peak potential establishes a peak current through the first diode D1 to charge the first capacitor C1 and generates a peak potential at the PLUS input 64 of the decision circuit. Resistors R1, R2, R3, R4 and R6 have a reference potential at the MINUS input 66 of the decision circuit 60 equal to the peak potential at the PLUS terminal 64 of the decision circuit 60 when no stimulus is present in the vicinity of the first electrode E1. Or larger than that.

  In this state, the output 63 of the decision circuit 60 becomes a low potential, and the self-holding latch 70 is not triggered. Further, the inverter I1 makes the third transistor P3 non-biased by turning the gate terminal 78 of the third transistor P3 to a high state, essentially equal to the potential of the source terminal 77, and is turned off. Stay on. However, in this embodiment, since the drain terminal 79 of the third transistor P3 is in an open circuit state, this is not important.

  This embodiment does not require a second electrode, but a two-electrode touch pad can also be adapted for use in this mode. When a two-electrode touch pad is adapted for use in this mode of operation, the presence or absence of a stimulus in the vicinity of the second electrode does not affect the operation of the circuit.

  When a stimulus is present in the vicinity of the first electrode E1, the operation of the second transistor P2 is the same as described above with respect to this embodiment. However, if there is a stimulus in the vicinity of the first electrode E1, the voltage at the gate terminal 82 of the first transistor P1 becomes equal to the potential of the source terminal 80 in the first transistor, so a longer time is required. Become. Thus, the first transistor P1 is turned on, allowing a larger current to flow through the third resistor R3 than the current that the second transistor P2 has flowed through the fourth resistor R4. . As a result, the peak potential at the anode 90 of the first diode D1 becomes larger than the reference potential at the anode 91 of the second diode D2. As a result, the peak potential of the PLUS input 64 of the decision circuit 60 is greater than the reference potential of the MINUS input 66 of the decision circuit 60, and the output 63 from the decision circuit 60 is accordingly high. In the state where the output 63 of the decision circuit 60 is high, the inverter I1 sets the potential of the gate terminal 78 of the third transistor P3 to a low state and turns on the transistor P3. However, this is not important since the drain terminal 79 of the third transistor P3 is not equivalently terminated.

  With the output 63 of the decision circuit 60 high, the latch trigger input 73 goes high and the self-holding latch 70 is triggered to trigger from the power supply 25 through the latch section 70 to the internal ground reference CHIP VSS, and the third. Current path through the diode D3, thereby effectively short-circuiting the rest of the control circuit 24. Self-holding latch 70 remains in this state until power to latch input terminal 71 is removed. Thus, a continuous digital control signal is output to the controlled device (not shown) until the latch 70 is reset. In this way, the touch switch device 20 can emulate a state change associated with a mechanical switch.

  Referring now to FIG. 7, the touch switch device 20 configured to operate in a continuous DC mode with single-ended input operates as follows. The operation and function of the control circuit 24 is essentially the same as in the single-ended input strobe mode described above with reference to FIG. However, in DC mode with a single-ended input, the self-holding latch output 72 is open circuit and therefore the self-holding latch 70 does not operate.

  When no stimulus is applied to the first electrode E1, the output 63 of the decision circuit 60 is at a low potential. Accordingly, the output 76 of the inverter I1 to the gate terminal 78 of the third transistor P3 is at a high potential. If the gate terminal 78 of the third transistor P3 has a high potential similar to the potential at the source terminal 77, the third transistor P3 is not biased and does not turn on. No current flows through the resistor R5.

  When a stimulus is present in the vicinity of the first electrode E1, the output 63 of the decision circuit 60 and thus the input 75 to the inverter I1 is in a high state. The inverter I1 changes the high level input to the low level output and supplies the output 76 to the potential of the gate terminal 78 of the third transistor P3. When the gate terminal 78 is at a lower potential than the source terminal 77, the third transistor is biased and it turns on and current flows through the third transistor P3 and the fifth resistor R5. . This raises the potential at the anode 108 of the fifth resistor R5, which is used via the output line 120 as an input to the controlled device (not shown).

  In the continuous DC mode of FIG. 7, the control circuit responds to the removal of the stimulus from the vicinity of the first electrode E1. As long as the stimulus remains in the vicinity of the first electrode E1, every time the pulse goes high, the anode 90 of the first diode D1 has a peak potential higher than the reference potential of the anode 91 of the second diode D2. appear. Therefore, the peak potential at the PLUS input 64 to the decision circuit 60 is higher than the reference potential at the MINUS input 66, and the control circuit 24 behaves as described above. When the stimulus is removed from the first electrode E1, the charge on the first capacitor C1 is finally discharged to a neutral state thanks to the bias function of the second transistor switch SW2. At this point, the peak potential at the PLUS input 64 of the decision circuit 60 is no longer higher or essentially higher than the reference potential of the MINUS input 66, and the output 63 of the decision circuit 60 returns to a low state.

  In this way, the touch switch device 20 operating in the DC mode with a single-ended input emulates a mechanical switch of a type that instantaneously touches. It will be understood that with minor modifications, the control circuit can be configured to emulate a mechanical switch that is pushed open and released and closed.

  Thus, this document has described the physical structure and operation of a single touch switch. Typical touch switch applications include multiple touch switches often used to control a device. FIG. 10 shows a switch panel including nine touch switches 20, and the nine touch switches 20 are arranged in a 3 × 3 matrix. Box B4 represents a touch panel component, while box B5 represents a controlled device component. Arbitrary multiple touch switches can theoretically be arranged in any way, but a matrix arrangement such as the one shown here can be easily multiplexed and required input / output from the controlled device. This is desirable because it reduces the number of lines.

  Box B6 of FIG. 4 shows the input of the touch switch control circuit, which includes active devices P1 and P2, diodes D1 and D2, resistors R3 and R4, and capacitors C1-C2. FIGS. 11A-18E above illustrate the present invention including providing low impedance buffering, reducing integrated circuit size and cost, linearizing input signals, stray capacitance swamping and blocking damaging current paths. Figure 7 illustrates another configuration of an input of a touch switch control circuit including an active device and a peak detector circuit that accomplishes some of the objectives described. The configuration shown in FIGS. 11A to 18E basically corresponds to the configuration of the box region B6 in FIG. 4 as understood by those skilled in the art of circuit design. Specifically, active devices M1 and M2 in FIG. 11A correspond to, for example, active devices P1 and P2 in FIG. 4, and active devices Q1 and Q2 in FIGS. 11A-18E correspond to diodes D1 and D2 in FIG. The resistors R7 and R8 in FIG. 11A correspond to, for example, the resistors R3 and R4 in FIG. 4, and the capacitors C9 and C10 in FIGS. 11A-18E correspond to the capacitors C1 and C2 in FIG. Furthermore, electrodes E1 and E2 and resistors R1 and R2 are the same as they appear in FIGS. 11A-18E in FIG. Pins OSCB, I_RNG and O_RNG appearing in FIGS. 11A-18E correspond to pins PIN2, PIN1 and PIN3 of FIG. The switches SW2 and SW4 in FIG. 4 correspond to, for example, the active devices M3 and M4 in FIG. 11A. The discharge signal DSCHGB of FIGS. 11A-18E corresponds to the current bias on trace 43 from the start-up and bias circuit 40 of FIG. Traces POS and NEG in FIGS. 11A-18E correspond to traces 64 and 66 in FIG. 4, respectively. Finally, trace OSCB of FIGS. 11A-18E corresponds to trace 53 from pulse generator and logic circuit 50 of FIG. 11A-18E is compatible with the circuit configuration described with reference to FIGS. 4-7.

  FIG. 11A shows the inner electrode E1 and the outer electrode E2 electrically connected to the oscillation signal generator OSCB through the pin OSCB and resistors R1 and R2, respectively. FIG. 11A also shows an interelectrode capacitance C6. Capacitances C7 and C8 are also shown which represent the intrinsic bond pad and wiring bond capacities when the electrical components are electrically connected to the integrated control circuit. Capacitors C7 and C8 also represent other capacities due to flip-chips that do not include bonding pad wires and other under-bump metal wiring, redistribution traces, etc., as known to those skilled in the art. Yes.

  In FIG. 11A, electrodes E1 and E2 are electrically connected to inputs of the touch switch control circuit at the gates of active devices M1 and M2, respectively, through pins I_RNG and O_RNG, respectively. In FIG. 11A, active devices M1 and M2 are shown as N-type MOSFET devices. The drains of the active devices M1 and M2 are electrically connected to the voltage source VDD via resistors R7 and R8, respectively, and their sources are connected to the oscillation signal OSCB.

  The drains of active devices M1 and M2 are also electrically connected to corresponding peak detection circuits including active devices M3, M4, Q1 and Q2 and capacitors C9 and C10, as described above, as shown in FIG. And includes components of switches SW2 and SW4, diodes D1 and D2 and capacitors C1 and C2, but the input active devices M1 and M2 are N-MOS active devices. 4, the active devices P1 and P2 are P-MOS devices, and the capacitors C9 and C10 and the sources of the active devices M1 and M2 are connected to the signal VDD instead of the voltage signal VSS through the resistors R7 and R8. . The peak detection circuit of FIG. 11A associated with active device M1 includes active device Q1, whose base is electrically connected to the source of active device M1 via trace SNEG and further to voltage signal VDD via resistor R7. And its emitter is electrically connected to the drain of the active device M3 and to the capacitor C9, and its collector is connected to the voltage signal VSS. The capacitor C9 has one terminal electrically connected to the voltage source VSS and the other end electrically connected to the emitter of the active device Q1 and the drain of the active device M3. The active device M3 has its drain electrically connected to the emitter of the active device Q1, its source connected to the voltage source VDD, and its base electrically connected to the discharge signal DCHGB. The configuration of the peak detection circuit associated with the active device M2 is similar, and includes active devices Q2 and M4 and a capacitor C10. In FIG. 11A, active devices Q1 and Q2 are P-type bipolar transistors, and active devices M3 and M4 are P-type MOSFET devices. The emitters of the active devices Q1 and Q2 are electrically connected as inputs to a decision circuit component (not shown) of the control circuit via traces NEG and POS, respectively. The operation of the decision circuit component is the same as described above with respect to FIGS. 4-7.

  In FIG. 11A, resistors R7 and R8 serve to convert drain current to voltage at the drains of active devices M1 and M2, respectively. These voltages are related to changes in the electric fields of electrodes E1 and E2 caused by contact and other stimuli. The voltage potential at the corresponding node of the drains of the active devices M1 and M2 is transmitted to the peak detector via traces SNEG and SPOS, respectively. The peak detector can convert the negative peak values of very fast transient pulses on traces SPOS and SNEG into DC signals on traces POS and NEG, respectively, which are easy for the decision circuit to process. Thus, FIG. 11A shows a dual input system with a negative pulse peak detection circuit. A similar positive pulse peak detection system is described in US Pat. No. 5,594,222 for single channel use. The sensing circuit that generates these negative pulses includes an N-type MOSFET device that can be pulled to a low level at high speed and a current source that can be pulled to a softer high level.

  Active devices M1 and M2 of FIG. 11A oscillate signal OSCB transmitted via both electrodes E1 and E2 and pins I_RNG and O_RNG to provide transient negative pulses on traces SNEG and SPOS, respectively. Turned on and off by. The maximum negative peak level of these pulses is proportional to the electric field strength at the input electrodes E1 and E2, which changes when the electrodes E1 and E2 are stimulated by contact or other causes.

  The signals on traces SNEG and SPOS are then sent to the respective bases of active devices Q1 and Q2 of the peak detection circuit corresponding to active devices M1 and M2. Low level signals transmitted to the bases of active devices Q1 and Q2 bias them on and provide the maximum negative voltage at the drains of active devices M1 and M2 to traces NEG and POS, respectively. Capacitors C9 and C10 are initially charged to VDD to slow down this voltage change on traces POS and NEG, thereby causing the transients of traces SPOS and SNEG to transition as shown in the timing diagram of FIG. 11A. Convert pulses to DC pulses on traces POS and NEG. Active devices Q1 and Q2 then disconnect capacitors C9 and C10 from charging as the transient signal ends. Active devices M3 and M4 are then controlled by discharge signal DCHGB to reset the initial charge VDD of capacitors C9 and C10, respectively.

  By advantageously utilizing short time length pulses, the touch sensor can maintain a low impedance. Furthermore, the average power consumed by the control circuit is small. For example, the peak current flowing through the capacitance of the input electrode is several milliamperes. This corresponds to a very low impedance during the period of peak current. Assuming that each pulse is active for 20 nanoseconds and sampled every 50 microseconds, a continuous average current is 0.8 microamperes for each channel and 1.6 microamperes for both channels. In addition, the input provides periodic sampling and statistical filtering of the sensing signal when the discharge signal DCHGB is not active.

  These low impedance and low average power consumption features can improve the stimulus interpretation performance of the touch sensor, as described in US Pat. No. 5,594,222, and can also provide mechanical switches, membranes This is advantageous when replacing type switches and equivalents with touch switch devices. Mechanical and other true switches cannot conduct current when open. Low-impedance and low-power solid-state switches directly replace mechanical switches by mimicking this property of true switches without risking unacceptable amounts of leakage current through “open” solid-state switches be able to. In addition, the low-impedance and low-power touch switch peak detector circuit uses amplifiers and operational amplifiers with relatively low gain and narrow bandwidth products for decision and other circuits, and for signal generation circuits. Compatible with using DC and relatively low gain, narrow bandwidth devices.

  FIG. 11B shows the inputs of the integrated control circuit, where active devices M1 and M2 are P-type MOSFET devices, active devices M3 and M4 are N-type MOSFETs, and active devices Q1 and Q2 are This is an N-type bipolar device. FIG. 11B shows that the resistors R7 and R8 and the sources of the active devices M3 and M4 are connected to the voltage signal VSS, and the collectors of the active devices Q1 and Q2 are connected to the voltage source VDD. The configuration is the same. Thus, FIG. 11B shows an embodiment using positive and transient transients as shown in the timing diagram of FIG. 11B. 11C and 11D show the input section, but the active devices M1 and M2 in FIG. 11A are replaced with active devices Q3 and Q4. They are N-type in FIG. 11C and P-type in FIG. 11D. FIG. 11C shows the peak detection circuit of FIG. 11A and includes P-type active devices Q1, Q2, M3 and M4, and FIG. 11D shows the peak detection circuit of FIG. Are all N-type devices. The operation of these input configurations is equivalent to that described above with respect to FIG. 11A and will be understood by those skilled in the art of circuit design.

11A-11D use resistors R7 and R8 that convert all drain or collector currents (either active devices M1 and M2, respectively, or Q3 and Q4, respectively) to a voltage proportional to the current at the drain or collector. is doing. That is, in FIGS. 11A-11D, this drain or collector voltage is equal to V− (I _r ) (R). Another way of providing this voltage conversion is shown in FIGS. 12A-15D. In these drawings, resistors R7 and R8 have been replaced with active devices.

Using an active device as a current to voltage converter can, for example, replace resistive components to allow high gain output and save integrated circuit space, as shown in FIGS. 12A-12D. 12A-12D generally correspond to FIGS. 11A-11D, respectively. 12A-12D, resistors R7 and R8 of FIGS. 11A-11B are replaced by MOSFET devices M5 and M6, and in FIGS. 12C-12D, resistors R7 and R8 of FIGS. 11C-11D are replaced by bipolar devices Q5 and Q6. Has been replaced. 13A-13D generally correspond to FIGS. 12A-12D, but the P-type active device current source of FIGS. 12A-12D has been replaced in FIGS. 13A-13D with an N-type active device current source (and in the same way 12A-12D is replaced with a P-type active device current source in FIGS. 13A-13D). Because the active load is the same type as the input devices of FIGS. 13A-13D, these active devices can be incorporated into an integrated circuit in the same manufacturing process. This gives better consistency. The output gain is determined by the voltage reference V _{ref used} and the device dimensions. V _ref can be set by a bias circuit that can mirror the current by scaling the gate width dimension when using a MOSFET device and scaling the emitter region when using a bipolar device.

  In the embodiment shown in FIGS. 12E-12H and 13E-13H, resistors R7 and R8 in FIGS. 11A-11D are connected to active devices M7 and M8 (FIGS. 12E-12F and 13E-13F) or Q7 and Q8 ( In addition to cascode connecting FIGS. 12G-12H and 13G-13H), active devices M5 and M6 (FIGS. 12E-12F and 13E-13F) or Q5 and Q6 (FIGS. 12G-12H and 13G-13H) Has been replaced. Such cascode bias helps to tolerate the control circuit against power and process variations.

  14A-14D show an embodiment using complementary devices. For example, in FIG. 14A, the square root extraction devices M9 and M10 are P-type MOSFET devices, and the input active devices M1 and M2 are N-type MOSFET devices. 14B-14D show an embodiment using complementary devices, which correspond to FIGS. 11B-11D. 14C-14D, active square root devices Q9 and Q10 are bipolar devices. The embodiments shown in FIGS. 14A-14D provide excellent stability even in the presence of variations in temperature, power supply, common mode noise and process variations during integrated circuit manufacturing. 15A-15D show an embodiment using an active square root device and the same type of input active device. That is, in FIG. 15A, the active square root devices M9 and M10 are N-type MOSFET devices like the input devices M1 and M2. Similar configurations are shown in FIGS. 15B (using N-type MOSFET devices), 15C (using N-type bipolar devices) and 15D (using P-type bipolar devices). Output linearity is maximized when matched MOSFET devices, i.e., MOSFET devices of the same type, are used for both input and active square root devices, as shown in FIGS. 15A-15B.

  11A-15D show the input capacitors C7 and C8 on the integrated circuit pin input connections I-RNG and O_RNG. These input capacities vary from part to part according to manufacturing tolerances and processes, but a compromise can be made between fluctuations and circuit performance. These variations tend to add to the electric field capacity of the electrodes and can cause variations and offsets in the performance of the control circuit. In typical applications, often the input detection circuit resolves very small variations in the electric field at the electrode when the input capacitance at the input node of the bonding pad is relatively large compared to the signal level of the input field effect capacitance. Therefore, it is advantageous to minimize the stray capacitances C7 and C8. One way to minimize the effect of this stray capacitance variation is to add “swamping” capacitors to the input circuit. This tends to reduce the sensitivity of the control circuit, but it does not accept input for variations due to input capacitance associated with bonding wires, under bump metallization, redistribution traces in flip-flop configurations, etc. Can be stabilized. The use of the swamping capacity is illustrated in FIG. This figure generally corresponds to FIG. 15A. In FIG. 16, the swamping capacitors C11 and C12 exist as equivalents to the stray capacitances C7 and C8, respectively, and are electrically connected to the voltage signal VSS. Obviously, the swamping capacitors C11 and C12 are compatible with all of the embodiments of the invention described herein and are not limited to use only with the embodiment shown in FIG.

  While the swamping capacitors C11 and C12 improve the performance of the control circuit, they require additional physical space. The embodiment shown in FIG. 17A shows the addition of a swamping capacitance resulting from the depletion capacitances of diodes D4-D7 to the control circuit inputs, here the gates of active devices M1 and M2. Space is saved. In FIG. 17A, diodes D4 and D6 are replaced with the swamping capacitor C12 of FIG. 16, and diodes D5 and D7 are replaced with the swamping capacitor C11 of FIG. The amount of capacitance per unit surface area is much larger for the type of diode configuration shown in FIG. 17A compared to the capacitance per unit area of a polysilicon or metal type capacitor. In addition, diodes D4-D7 are also used for protection of both positive and negative high voltage discharges. This protection is particularly advantageous for touch input applications. Human input devices such as keyboards, single input switches, etc., are exposed to electrostatic discharge transients and must include components such as MOSFETs and other devices to protect their sensitive input circuits. Can do. This problem is exacerbated when the sensing electrodes E1 and E2 are located very close to the input circuit IC, as shown in FIG. 17B.

  18A-18E show other possible configurations of input circuits for touch switches with integrated control circuits. 18A-18C show various examples of common mode stimulation for active devices M1 and M2. FIG. 18A generally illustrates the configuration of FIG. 17A and includes active devices M11-M14. In FIG. 18A, active devices M11-M14 are electrically connected to the sources of input active devices M1 and M2. The gates of the active devices M13 and M14 are connected to the oscillation signal OSCB, and their drains are connected to the gate of the active device M12. The gate of the active device M11 is connected to the current source bias signal CSBS, and its drain is connected to the source of the active device M12. The configuration shown in FIG. 18A provides negative feedback at the input stage to active devices M1 and M2.

  FIG. 18B shows an input circuit portion including active devices M15 and M16, which are here shown as N-type devices, whose sources are electrically connected to input pins I_RNG and O_RNG, respectively, Is electrically connected to the oscillation signal OSCB. The drains of the active devices M15 and M16 are connected to the sources of the active square root devices M9 and M10, respectively, and are connected to the bases of the active devices Q1 and Q2 of the peak detection circuit, respectively. In FIG. 18B, active devices M15 and M16 are stimulated by an oscillating signal OSCB through their gates and receive an input signal through their sources, but have previously been stimulated through their sources. , Replacing active devices M1 and M2, which were shown to receive input through their gates.

  FIG. 18C generally shows the configuration of FIG. 18B and includes swamping diodes D4-D7 as shown in FIG. 17A. The configuration of FIG. 18C can also be employed in a single input mode with only one electrode and can provide all the benefits of employing an input diode that provides a depletion mode swamping capacitance.

  FIG. 18D shows the configuration of FIG. 16, which includes swamping capacitors C11 and C12 that balance the inputs to active devices M1 and M2, but for single electrode mode, either outer electrode E2 or input pin O_RNG. There is no. 18E shows the configuration of FIG. 18D, but as also shown in FIG. 17A, the swamping capacitance is provided by diodes D4-D7 and is necessary to obtain the benefits of the swamping capacitance as described above. Space is minimized.

  FIG. 19 is a schematic circuit diagram of one possible configuration for the input circuit portion of the integrated circuit of the present invention, including the latch output LCH_O functioning as the self-holding latch 70 of FIGS. Various output functions and their configurations are shown. These output functions allow the touch cell to replicate the response of a conventional membrane or mechanical switch.

  Output pins NDB_O, NE_O and ND_O are outputs of touch cells and integrated circuit assemblies that pull the output electrically low through active devices. The integrated control circuit may cause the output to be electrically pulled to a low level through the active device when a stimulus is applied (eg, a human contact stimulus) or when no stimulus is present (eg, a human contact stimulus). It can be configured to pull the output electrically low through an active device (when there is no contact stimulus).

  As shown in FIG. 19, the output pin NDB_O is electrically connected to the drain of the active device M18, its source is the voltage signal VSS, its gate is the input of the inverter U2, the output of the inverter U2 , Connected to the gate of the active device M17 and the voltage signal TP_O. Output NE_O is electrically connected to the emitters of active devices Q13 and Q14, whose base is connected to the drain of active device M20 and whose collector is connected to voltage signal VSS. Active device M20 is then connected at its gate to the output of inverter U2 and at its source to voltage signal VSS. The output pin ND_O is electrically connected to the bases of the active devices Q13 and Q14 and the drain of the active device M20. Active device M20 acts as a negative pull down device for output NE_O and can bias the gates of active devices Q13 and Q14 for output ND_O on.

  Output pins PDS_O, PD_O and PE_O are outputs of touch cells and integrated circuit assemblies that pull the output electrically high through active devices. The integrated control circuit causes the output to be electrically pulled high through the active device when a stimulus is applied (eg, a human contact stimulus), or when no stimulus is present (eg, a human contact stimulus). It can be configured to pull the output electrically high through the active device (when there is no contact stimulus).

  In FIG. 19, the output pin PDS_O is electrically connected to the Schottky diode SD1, and the latter is then connected to the output pin PD_O. The output pin PD_O is electrically connected to the base of the active device Q12 and the drain of the active device M17, the latter source being connected to the voltage signal VDD and the gate being connected to the output of the inverter U1 and the input of the inverter U2. . The collector of the active device Q12 is connected to the emitter of the active device Q11, and the latter collector and base are both connected to the voltage signal VDD. As also shown in FIG. 19, the emitter of the active device Q12 is connected to the output pin PE_O.

  The integrated control circuit can be applied in conventional DC mode, DC matrix, pulsed DC matrix or latch matrix mode. FIG. 20A shows an application in which the integrated control circuit is applied to a touch cell configuration for DC mode. In all applications that utilize DC mode, each integrated control circuit is continuously connected to the system voltage signals VDD and VSS. In some cases, the outputs of several touch cells are connected in the form of electrical OR logic (eg, touch cells TC1-TC3 using PE_O output and TC7-TC9 using NE_O output). The rest of the touch cells (TC4-TC6 and TC10-TC13) show the use of various outputs: PD_S, PD_O, PD_E, NDB_O, NE_O and ND_O. For touch cells TC4-TC6, they can pull the output electrically high, but the output pin is connected to ground through a resistor, and for touch cells TC10-TC13, they are Although the output can be electrically pulled to a low level, the output pin is connected to the voltage signal VDD through a resistor.

FIG. 20B shows the case where the touch sensor is applied to the negative pulse DC matrix mode. The integrated control circuit of each touch cell has its voltage signal VDD connected to the system voltage source V _supply . It is also shown that the VSS of the integrated control circuit of each touch cell is connected to a row selection signal, ROW SELECT1 or ROW SELECT2. In FIG. 20B, the output pin NE_O of each touch cell integrated control circuit is connected to either a column return, ie, COLUMN RETURN1 (touch sensors TS1 and TS2) or COLUMN RETURN2 (touch sensors TS3 and TS4). ing. As can be seen from FIG. 20B, ROW SELECTS and COLUMN RETURNS can drive one touch sensor, one row of touch sensors, or one column of touch sensors. This is also shown in the timing diagram of FIG. 20B.

P-type active devices Q13 and Q14 shown in FIG. 19 pull NE_O low when an active stimulus is applied to the associated input. Inputs can also be configured such that those P-type active devices at the output pull NE_O low when the associated input is not stimulated. The emitter-base junction of active devices Q13 and Q14 prevents current from flowing through VSS to other devices in the matrix when any one device changes to an active low level. Whenever any one particular touch cell integrated control circuit pulls low, the output (in response to the forward bias voltage drop of the base-emitter junction of output active devices Q13 and Q14) (When measured from V _supply to NE_O). Depending on the application, one device can be used instead of the two active devices Q13 and Q14.

NDB_O or ND_O outputs employing MOSFET devices as shown in FIG. 19 can be used when it is desirable to avoid the V _be drop of the P-type device (s). The negative pulse DC matrix mode configuration of a touch sensor with ND_O output is shown in FIG. 20C, but is essentially similar to that shown in FIG. 10B. The voltage drop across the N-type MOSFET device M18 or M20 is relatively small at low current levels and depends on the RDSon resistance multiplied by the current flowing through the MOSFET device channel. Therefore, this current is set mainly by the external load resistance. At lower current levels, the voltage drop is smaller compared to the corresponding voltage drop for P-type bipolar transistors. On the other hand, at higher current levels, bipolar transistors cause a voltage drop (0.6 to 0.7 volts) of the base-emitter junction forward bias, while N-type MOSFET devices are RDSon's It has a tendency to increase the voltage drop according to a linear relationship to the drain current: V _drop = (RDSon) (I _drain ). That is, in typical logic circuits where lower current levels exist, N-type MOSFET outputs tend to cause less voltage drop than bipolar devices. This makes the MOSFET device generically suitable for other logic circuits. FIG. 20D shows a positive pulse DC matrix configuration with a touch sensor having a PD_O output using a P-type MOSFET device M17, as shown in FIG. It fits.

  However, MOSFET devices do not have the inherent blocking function like bipolar devices. FIG. 21A shows a cross-sectional view of a typical P-type substrate with doped N-type and P-type materials used in the construction of a typical CMOS circuit. FIG. 21B is a schematic diagram of an N-type MOSFET device N1, which is an output pull down device for output pin NBD_O (active device M18 in FIG. 19) or output pin ND_O (active device M20 in FIG. 19). Can be used as FIG. 21C is a schematic diagram of blocking device N2, which is connected in series with output device N1, and is a parasitic device associated with N1 that can occur as an unintended effect of the MOSFET device structure due to the depletion region surrounding the device. The leak current is prevented from being generated.

FIGS. 21A-21C show how the structure of an N-type MOSFET device produces a drain-to-source parasitic bipolar diode PD1 and prevents leakage current from VSS to the substrate. ing. A typical CMOS integrated circuit uses a P-type or N-type substrate. These substrates are typically electrically connected to the integrated circuit VSS or VDD. In the case of a P-type substrate, the substrate is connected to VSS, and in the case of an N-type substrate, the substrate is connected to VDD. Note that in FIG. 21B, the source of N-type MOSFET device N1 is coupled to voltage signal VSS, and the anode of parasitic diode PD1 is also coupled to the source node of device N1. The cathode of the parasitic diode PD1 is connected to the drain of the device N1. As a result, the integrated control circuit is incorporated into a negative pulse DC matrix mode with active electrical pull down using an N-type MOSFET device (with ND_O output as shown in FIG. 20C). There is a reverse current path through the parasitic diode PD1 and through the P-type substrate. When a strobe row pulse is supplied to the matrix and is higher than the potential at the output of ND_O, current flows from VSS to ND_O through the parasitic diode PD1. This current path affects the operation of the matrix and the power supply. This low current path provides a low impedance path that connects VSS to VDD through a strobe driver. A bipolar diode connected in series with an N-type MOSFET pull-down device blocks reverse current but also negates the characteristics of the N-type MOSFET pull-down device, ie, a low voltage drop at the output. Bipolar diodes tend to cause a base-emitter junction voltage drop V _be . In order to block this undesired current path, it is necessary to incorporate a blocking device, which is preferably compatible with conventional integrated circuit manufacturing and produces a minimum voltage drop. By properly connecting N-type MOSFET devices N1 and N2, the leakage current path can be blocked, and the P-type substrate and voltage signal VSS are decoupled from the leakage current path through ND_O device N1. At the same time, the voltage drop at the control circuit output is minimized.

Device N2 of FIG. 21A is a blocking device and is schematically illustrated in FIG. 21C. The drain and source of blocking device N2 are connected to VSS and VSS1, respectively, as shown in FIGS. 21A and 21C. The gate of blocking device N2 is connected to voltage signal VDD, the latter can be 3-5 volts for compatibility with many microprocessors, but this need not be the case. When the source of device N2 is at a low potential, such as ground, the channel resistance is very small as long as the gate voltage is slightly higher than the threshold voltage of the device. Since the gate of device N2 is at VDD which is on the order of 3 to 5 volts (V _supply ), during its active pulse period its source will be zero volts and its threshold voltage will be less than 1 volt, The channel resistance becomes very small and the device's channel voltage drop is also very small (ie smaller than a standard bipolar diode). When the source of device N2 is at a voltage equal to (or higher than) VDD, the gate-to-source voltage (V _GS ) will be less than the device threshold voltage. This greatly increases the channel resistance and essentially blocks current through the channel. Further, the voltage applied to the depletion junction from the source of the device N2 to the parasitic diode PD of the substrate PS is smaller than the barrier potential (about 0.6 to 0.7 volts) of the source / drain parasitic diode PD1. Thus, the parasitic diode PD1 essentially blocks current flowing through the substrate.

  Furthermore, blocking device N2 can also be used for reverse voltage protection in standard integrated circuit applications and can provide all of the benefits described above. When so used, blocking device N2 is connected to VSS of the integrated circuit as described above to protect the circuit from reverse current or voltage damage.

  FIGS. 21D-21F show blocking device BDP2 for an electrically high level pull device with forces PDS_O, PD_O and PE_O as shown in FIG. The device shown in FIGS. 21D-21F is complementary to the device shown in FIGS. 21A-21C and will be understood by those skilled in the art in light of the discussion with reference to FIGS. 21A-21C. In all the described DC mode configurations, there are three connections to each touch cell's integrated control circuitry. VDD and VSS for each touch cell integrated control circuit need to be connected to a power source for some time to process the input stimulus. The output of the integrated control circuit is found at PDS_O, PD_O, PE_O, NDB_O, ND_O and NE_O, depending on the desired configuration. These outputs constitute the third connection required for the integrated control circuit. However, in some cases it may be advantageous to use an integrated circuit that requires only two connections. For example, in an application that includes a membrane switch and has a touch sensing switch, typically only two connections are employed per switch, so an integrated control circuit that requires only two connections is a membrane type switch. Facilitates the direct replacement of switches with touch switches.

  A matrix of two-terminal membrane switches MS1-MS4 is schematically shown in FIG. FIG. 22 shows one way of addressing and reading the switches within the matrix. Of course, the matrix of FIG. 22 can be modified to include more rows, more columns, more switches, and other connections. In all cases, the interface to each switch typically includes two types of signal lines: ROW SELECT and COLUMN RETURN. Each ROW SELECT line is a potential source and when each switch MS1-MS4 is closed (in the case of a membrane switch, it is closed by pushing with a finger) through which current flows into the COLUMN RETURN line. Allow. Terminating resistors COLR1 and COLR2 on COLUMN RETURN lines 1 and 2 are used to generate voltages to be processed by the return logic circuit, respectively, to limit the current through the switch device. The sequence of strobe lines is such that only one row of switches (MS1 and MS3 or MS2 and MS4) is active at a given time. When a particular row is selected, the voltage generated through each termination resistor COLR indicates which switch in the selected row is electrically closed. The COLUMN RETURN line is generally processed simultaneously. The matrix scheme is efficient with respect to the number of interconnects used to process multiple switch inputs. For example, 64 switches can be read in an 8 × 8 matrix using 8 ROW SELECT lines and 8 COLUMN RETURN lines. Typically, certain logic devices can be connected to the strobe and return lines to determine the state of all switches in a short time. This is a typical matrix scheme for those skilled in the art to know how to implement. It can be used for controllers, computer keyboards, telephones and other devices widely used in the market.

  A solid-state sensing device that operates as a two-terminal switch that detects stimuli, without the addition of software, logic circuits and / or microprocessors that can cause resets and other failures, This is advantageous in that a reading circuit can be constructed. FIG. 23 shows an example of such a device arranged in a matrix and having only two integrated circuit connections. In this manner, the touch sensors TS1-TS4 in FIG. 23 can be replaced with the membrane switches MS1-MS4 in FIG. In FIG. 23, each touch sensor TS1-TS4 senses a difference in electric field potential. Depending on the presence or absence of an appropriate stimulus, the device (depending on the particular application) transitions from a high impedance state (equivalent to an open circuit) to a low impedance state (equivalent to a closed circuit), thereby causing a conventional Mimics a membrane or mechanical switch. The main feature of these devices is their ability to mimic the attributes of two-terminal switches.

  24A and 24B show possible circuits for the touch sensors TS1-TS4 of FIG. The circuit shown in FIGS. 24A and 24B is based on the latch circuit portion of the circuit shown in FIG. In FIG. 19, the latch circuit shown includes active devices M19 and Q15-Q19 in addition to resistor R9. Latch circuit output pin LCH_O is shown connected to the emitter of active device Q19. Active device Q19 is then connected at its base to the output of inverter U2, the drain of active device Q15 and the gate of active device M20, and its collector connected to the emitter of active device Q18. The latter base is connected to the voltage signal VDD, and its collector is connected to the resistor R9. The latter is then connected to the voltage signal VDD. The collector of active device Q18 is also shown connected to the bases of active devices Q15 and Q16 and active device Q17, whose emitters are connected to voltage signal VDD, the latter collector being connected to voltage signal VSS, The emitter is connected to the collector of the active device Q15. The collector of the active device Q18 is also connected to the drain of the active device M19, the latter gate is connected to the output pin INITB of the control circuit and its source is connected to the voltage signal VDD.

  24A and 24B show various embodiments of the latch circuit of FIG. Both of these embodiments omit the optional active devices Q16-Q18. FIG. 24A shows the incorporation of bipolar components Q15 and Q19 into the latch circuit as shown in FIG. 19, and FIG. 24B shows the incorporation of MOSFET components into the latch circuit. Other configurations can be implemented while keeping in mind the spirit and function of the two-terminal device.

  FIG. 24A shows a bipolar latch circuit that operates in conjunction with the control circuit. This provides the functions necessary to detect input stimuli, execute decisions and trigger bipolar latch circuits. The control circuit can also provide power-on reset functions, various internal blocks and function initialization and sequence settings. Inputs to the control circuit include those associated with the input sensing connections, i.e., OSCB, + (PLUS) and-(NEGATIVE), those associated with the power supply for the control circuit, i.e., voltage signals VDD and VSS and the latch circuit. The accompanying ones are included: INIT and TRIGGER. The latch output passes through the output pin LCH_O.

When there is a path on the ROW SELECT line through the active pull device P-type MOSFET for current to flow from system V _supply to GND, strobe line ROW SELECT in FIG. 24A becomes active. When power is supplied, the control circuit can operate. When the strobe pulse is first applied, the control circuit provides a gate signal via the INIT line to turn on the active device M19. This ensures that the base-emitter voltage of active device Q15 is essentially zero volts and does not conduct (except for leakage current). Since Q15 is off and no current flows through the base of Q19, Q19 is also off. If Q19 is off, the voltage at the base of Q15 is essentially at VDD even after the INIT signal is removed, and M19 is off. If the latch is essentially off (ie, no current flows), the control circuit can operate. In operation, the integrated control circuit simulates an open switch in a high impedance mode. The output voltage across the resistor R _column is equal to V _supply × R (integrated control circuit) / ([R (integrated control circuit) + R _column ]. The larger the equivalent resistance of the integrated control circuit is, the larger R _{column is.} The ratio of V _supply appearing at both ends becomes smaller, and the ratio of the drop appearing at both ends of the integrated control circuit becomes larger.

A perfect switch will exhibit infinite resistance and zero current when opened, so V _supply will appear across the switch during the strobe pulse, and since the current is zero, the voltage drop across R _column is zero. It becomes. Since the integrated circuit is not a switch, the integrated control circuit is designed to draw as little current as possible in order to more accurately replicate the characteristics of the open switch when V _supply is supplied by the strobe pulse. It is important to.

The input electrode can be configured such that the integrated control circuit remains in a high impedance mode when a stimulus is applied or when no stimulus is applied. When the integrated control circuit is in the high impedance mode, most of the V _supply is applied across the integrated control circuit. This allows the circuit to operate in a floating mode since the internal VDD and VSS are sufficient to operate the integrated circuit as a whole, and the internal control circuit is similar. The electrode configuration can also be such that when a stimulus is applied or when no stimulus is applied, the control circuit generates a trigger pulse to the latch circuit. When the control circuit generates a trigger pulse, the latch turns on. The trigger pulse in FIG. 24A is a positive pulse from VSS to VDD. This trigger pulse can turn M19 off after the INIT signal resets. This positive pulse forward biases the base-emitter junction of N-type bipolar device Q19, turning it on. The base current flows and the gain transfer of the active device Q19 causes a current to flow through the collector of the active device Q19 and thus the resistor R19. The current through resistor R9 generates a voltage potential that is sufficient to forward bias the emitter-base junction of active device Q15 toward VSS, turning it on toward the VSS. Pull down. The current gain of active device Q15 allows sufficient current to flow through the collector of active device Q15 and also causes the voltage at the base of active device Q19 to pass through the emitter-base junction of active device Q19 after the trigger pulse is removed. Pull it up enough to bias forward. The trigger pulse is removed enough to stop the operation of the control circuit due to the voltage drop across the control circuit. The latch current remains on after the trigger pulse is removed due to the positive current feedback loop between Q15 and Q19. The voltage drop of the latch is determined by the saturation voltage, the junction resistance, the gain of the active devices Q15 and Q19 and the resistance of R _column . Since it is important that the control circuit does not operate and the latch voltage drop is as small as possible within the current range, the latch circuit inside the integrated control circuit should remain on even after the trigger is removed. is there. In this low impedance mode, it is desirable to get as many of those attributes as possible to replicate a closed switch. A perfect closed switch will carry infinite current and the voltage drop will be zero volts at all current levels. In order to best replicate a perfect switch, i.e., one with a small voltage drop, the latch circuit expands the emitter region, bipolar transistor with low V _be drop and high W / L channel ratio, low threshold and It is desirable to utilize a MOSFET with a high gain device.

  FIG. 24B shows the latch circuit of FIG. 24A, where bipolar active devices Q15 and Q19 are replaced by MOSFET devices M21 and M22. The operation of the integrated control circuit in FIG. 24B is equivalent to the operation of the integrated control circuit in FIG. 24A. The operation of the latch unit shown in FIG. 24B will be described below.

When the INIT pulse is applied, the active device M19 turns on. This allows VDD to be applied to the gate of active device M21. In this state, the gate-source voltage of the active device M21 becomes smaller than the threshold voltage of the P-type MOSFET device M21, ie essentially zero volts, so that the active device M21 is turned off. Since the drain current of the active device M21 is essentially zero amperes (excluding leakage current), no voltage appears across resistor R10. Since the gate of active device M22 is essentially zero volts, its gate-source voltage is essentially less than the threshold voltage of the device. The drain current of active device M22 is essentially zero because its gate-source voltage is well below the threshold voltage. Since the current flowing through the resistor R9 becomes zero, the gate voltage of the active device M21 becomes VDD or very close to it, so that the gate-source voltage of the active device M21 remains after the INIT signal is removed. Essentially zero. This state places the latch circuit in a high impedance state. When a trigger pulse close to VDD is applied to the gate of active device M22, after the INIT pulse is removed, its gate-source voltage exceeds the threshold voltage of active device M22, turning M22 on. Let The drain current of the active device M22 increases and causes a voltage drop across the resistor R9. Due to the voltage drop across resistor R9, the gate-source voltage of active device M21 exceeds its threshold voltage, causing active device M21 to turn on. The drain current of active device M21 increases, causing the voltage drop across resistor R10 to increase above the threshold voltage of active device M22 even after the trigger pulse is removed. Thus, the latch goes to a low impedance state, but the voltage drop across it depends on the characteristics of the active devices M21 and M22, the values of resistors R9 and R10, and the resistance of R _column . The rest of the operation of the integrated control circuit of FIG. 24B is similar to that of the integrated control circuit of FIG. 24A. Both figures also show the blocking diodes of FIGS. 21A-21C, which are labeled D8 and D9, respectively, in FIGS. 24A and 24B.

  FIG. 25A shows the latch circuit portion of FIG. 19, showing the active devices Q15-Q19 in one possible configuration built on the substrate PS. FIG. 25B schematically shows the latch circuit portion. In FIG. 25A, active devices Q15 and Q16 share a P-doped well EMITTERQ15 / EMITTERQ16 as an emitter, and the collector of active device Q15 and the emitter of active device Q17 are the same P-doped connected to the gate of active device Q15. Well COLLECTORQ15 / EMITTERQ17. Active devices Q15, Q16 and Q17 also share the same N-doped well as their bases BASEQ15, BASEQ16 and BASEQ17, respectively. Substrate PS forms the collectors of active devices Q16 and Q17, COLLECTORQ16 and COLLECTORQ17, respectively. The active device Q19 is shown in a separate N-doped well in the substrate PS, COLLECTORQ19, the collector of its N-doped well, is connected to resistor R9, and the base of its P-doped well, BASEQ19 Is connected to COLLECTORQ15 / EMITTERQ17 which is a P-doped well, and EMITTERQ19 which is the emitter of its N-doped well is connected to the voltage signal VSS at the anode of the diode D10. In FIG. 25A, the active device M19 is connected in parallel to the resistor R9. The operation of the configuration shown in FIGS. 25A and 25B will be understood by those skilled in the art of active device and circuit design from the discussion of the latch circuit associated with FIG. 24A. Active devices Q16-Q18 amplify the signal delivered to output LCH_O. The configuration shown in FIG. 25A has a reduced latch-on voltage compared to the voltage drop associated with a standard latch, thanks to the dynamic impedance of the active device Q17 and the branching of the VSS current through the substrate PS. It is advantageous in terms of descent. A diode D10 having a cathode connected to the output LCH_O and an anode connected to the emitter of the active device Q19 and the voltage signal VSS can prevent feedback to the latch portion of the integrated circuit shown in FIG. 25B. FIG. 25C shows diode D10 having its anode connected to voltage signal VSS and the collectors of active devices Q17 and Q16, and its cathode connected to the emitter of active device Q19 and output LCH_O. The configuration of FIG. 25C changes the voltage signal on the emitter of active device Q19, which can thus be biased on by output TRIG, from VSS in FIG. 25B to VSS1. This latch circuit configuration can advantageously reduce the voltage drop in this case because the voltage drop across diode D10 is not in series with the base-emitter voltage of active device Q19. The optional active device Q18 of FIGS. 25B and 25C is useful for increasing the reverse breakdown voltage of the latch circuit.

The integrated circuit of the present invention can respond to capacitive inputs that vary in various ways. For example, FIGS. 26A-26C illustrate a capacitive input sensing device that is compatible with the integrated circuit of the present invention, where the capacitive inputs are the electrodes GE and SE that make up the capacitive C _sense shown schematically in FIG. 26D. It changes when the distance d between is changed. The capacitance C _sense is a function of the electrode capacitance constant E _o , the relative permittivity E _r , the electrode surface area s, and the distance d between them. The apparatus shown in FIG. 26A having an integrated control circuit ICC and a sensor electrode SE on one side 143 of the substrate 144 and a grounded electrode GE configured as a button 122 forming a cavity 121 on the opposite side 145. FIGS. 26B and 26C show the separated layers of the device shown in FIG. 26A. The cavity 121 of FIG. 26A allows the button 122 to be pressed by, for example, a human finger or other probe, thereby changing the distance between the electrodes GE and SE. The control circuit shown in FIG. 26D can respond to changes in capacitance resulting from changes in distance d. The control circuit in FIG. 26D corresponds to the control circuit shown in FIG. 18D, but the capacitor C3 in FIG. 18D is renamed C _{sense in} FIG. 26D.

  Thus far, this specification has described various preferred embodiments of touch sensors (or field effect sensors) in general according to the present invention. In the following, various practical applications relating to such sensors will be described. While it is desirable to implement these applications using the touch sensors described above, other types of touch sensors, such as US Pat. Nos. 5,594,222 and 6,310,611, may be used. It is also generally possible to implement using the sensors described in, conventional capacitive sensors, and other types of sensors known to those skilled in the art.

27A-27D show a capacitive input liquid level sensing device compatible with the integrated circuit of the present invention. Here, the capacitive input changes as the dielectric constant _Er between the two electrodes changes. This change occurs, for example, when the liquid replaces the air between the two electrodes GE and SE1 that form the capacitance C _sense . Thus, in FIG. 27A, the grounded electrode GE on the substrate 123 is separated from the sensor electrode SE1 by a gap that can be filled with the liquid 125. FIG. 27B shows a configuration in which the liquid 125 fills a gap between the substrate 124 forming a storage container for the liquid 125 and the grounded electrode GE and the sensor electrode SE1 when the liquid 125 reaches a certain level. The substrate 123 is shown. FIGS. 27C and 27D show one possible possible configuration of the grounded electrode GE and the sensor electrode SE1 connected to the integrated control circuit ICC. In both FIGS. 27C and 27D, the electrodes GE and SE1 are long and horizontally arranged. That is, their long axes are parallel to the surface of the liquid 125, and when the level of the liquid 125 is slightly increased, the capacitance C _sense is greatly changed as schematically shown in FIG. 27D. The control circuit shown in FIG. 27E is the same as that shown in FIG. 26D, which is equally compatible with the device shown in FIGS. 27A-27D.

28A-28B show a capacitive input sensing device compatible with the integrated circuit of the present invention. Here, the capacitive input changes as the surface area s _s3 of the sensor electrode SE3 changes. In FIG. 28A, the substrate 126 carries a grounded electrode GE, and the movable substrate 127 carries two sensor electrodes SE2 and SE3 connected to the integrated control circuit ICC. The sensor electrode SE3 has a surface area s _s2 that varies along the direction in which the substrate 127 is moved. Thus, FIG. 28B shows the substrate 127 moved upward relative to the position of FIG. 28A. Accordingly, the surface area s _s3 of the sensor electrode SE3 as seen from the grounded electrode GE decreases. This change in surface area corresponds to the change in capacitance C _sense3 schematically shown in FIG. 28C. The control circuit shown in FIG. 28C is similar to the circuit shown in FIG. 18E, but has the dual electrode structure shown in FIG. 11A. Here, the electrodes E1 and E2 are renamed as sensor electrodes SE2 and SE3, and the capacitor C6 is renamed as a capacitor C23. The operation of this circuit will be understood by those skilled in the art from the previous discussion with respect to FIGS. 11A and 18E.

29A-29D show a capacitive input sensing dial device compatible with the integrated circuit of the present invention. Here, the input pulse width and sequence determine the response of the integrated control circuit. FIGS. 29A-29D show sensor electrode SE4 connected to integrated control circuit ICC on substrate 128 and grounded electrodes GE1 and GE2 on rotating disk 129. FIG. In FIGS. 29A-29D, the grounded electrodes GE1 and GE2 (including the space between them) together occupy about half of the area of the rotating disk 129 and are separated from each other. This and other similar configurations allow the control circuit to distinguish between clockwise and counterclockwise rotation of the dial device. FIGS. 29B-29C show the motion of the rotating disk 129 relative to the stopped substrate 128. FIGS. 29E and 29F show the output pulses of the dial device shown in FIGS. 29A-29D, which can generate a response at the input of the integrated control circuit, as shown in FIG. 29G. FIG. 29E shows the relatively widely spaced input pulses that occur when the rotating disk 129 is rotated counterclockwise at one speed, and FIG. 29F rotates the rotating disk 129 clockwise at a higher speed. FIG. 2 shows an input pulse that is relatively narrow and close when it is rotated to the right. A change in the capacitance C _sense formed between the electrode SE4 and the GE1 or GE2 and schematically shown in FIG. 29G (similar to the configuration shown in FIG. 27E) is the implementation of the integrated control circuit of the present invention. It can be detected by morphology.

  30A-30E illustrate another capacitive sensing dial device that is compatible with the integrated circuit of the present invention. Here, the connection to ground is provided by the user. FIG. 30A shows a rotating disk 130 having transfer electrodes TE1-TE8 of various sizes. This corresponds to various sizes of input pulse widths when they are connected to ground. FIG. 30B shows the transfer electrodes TE1-TE8 of the rotating disk 130 connected to the coupling electrode CE mounted on the cylinder 131. FIG. 30C shows a cylinder 132 having sensor electrodes SE5 and SE6 connected to the integrated control circuit ICC and configured to fit inside the cylinder 131 of FIG. 30B. FIG. 30D shows the components shown in FIGS. 30A-30C that are combined together to form a rotary capacitive input device. FIG. 30E shows a hand 133 that grips the cylinder 131. The hand 133 connects the coupling electrode CE and the transfer electrodes TE1-TE8 to a virtual ground. As shown in FIG. 30C, each sensor electrode SE5 and SE6 is configured to receive a capacitive input from one transfer electrode at a time. As shown in FIGS. 30F-30H, two input pulses can be sent to the integrated control circuit at a time. The direction of rotation and arc length by the user of the dial including the rotating disk 130 and cylinder 131 can be determined from the inputs shown in FIGS. 30F and 30G. FIG. 30F shows a pulse train generated as a result of rotating the dial device twice in the counterclockwise direction, while FIG. 30G shows a pulse train generated by rotating twice in the clockwise direction. FIG. 30H shows a schematic diagram of the dial device of FIG. 30E, including a hand 133 to ground and a coupling electrode CE connected to the transfer electrode TE, which are connected to resistors RIN1 and RIN2, respectively. A capacitance is formed together with SE5 and SE6. The integrated control circuit ICC supplies the oscillation signal OSC to the sensor electrodes SE5 and SE6 via the resistors RIN1 and RIN2, respectively, and supplies the outputs OUT1 and OUT2 to the decision circuit (not shown). The various parts of the dialing device, including the rotating disk 130 and cylinders 131 and 132, are described as “Molded / Integrated Touch Switch / Control Panel Assembly and Method for. It can be formed according to the invention described in US Pat. No. 6,897,390 entitled “Making Same)” or other methods.

  31A-31F illustrate the structure and isolation layer of a touch switch assembly having an integrated control circuit according to the present invention. 31A-31E show the individual layers that make up the assembled touch switch shown in FIG. 31F. FIG. 31A shows the back side of the substrate 133 including the opaque region 135 and the window region 136. The opaque area 135 may be a decorative frit, decorative epoxy, UV curable ink, or any other decorative layered material. FIG. 31B shows the touch switch electrode 134 mounted in the window region 136 on the back side of the substrate 133. The electrode 134 is shown overlying the opaque region 135 and can include a transparent conductive material including indium tin oxide or other suitable material. FIG. 31C shows the bottom conductive layer of the touch switch assembly as viewed from the back side, with circuit trace 138 including silver-containing frit, silver epoxy, copper epoxy, electroplated conductors and similar materials and combinations thereof. Is included. FIG. 31D shows a dielectric layer of a touch switch having a dielectric layer region 140, which may be an insulated ceramic frit, UV ink, epoxy, or the like. FIG. 31E shows the crossover layer of the touch switch assembly from the back side and includes a crossover conductor 137 comprising the material described with reference to FIG. 31C. FIG. 31F shows the separation layer shown in FIGS. 31A-31E together assembling the completed touch switch assembly. FIG. 31F also shows the assembly as viewed from the back side.

  Although the embodiment shown above has been described as a DC mode, the integrated control circuit of the present invention is also compatible with the AC input and can therefore operate in the AC mode. The case of AC is shown in FIG. FIG. 32 shows a touch switch with integrated control circuitry adapted to receive AC input. In FIG. 32, AC signal AC is connected to rectifier bridge RB including diodes D11-D14 via resistors R10 and RLOAD. The rectifier bridge RB diodes D11 to D14 are connected in parallel with the Zener diode Z1 and the capacitor C15. The AC signal AC can simulate a touch switch with an integrated control circuit that includes the latch portion shown in FIG. 24A except for the diode D8. This configuration is advantageous in that the integrated circuit can be designed to draw a relatively small current, and provides a floating circuit that is less dependent on ground because the current is characterized by a small sensing impedance.

  Although the embodiments of the present invention described above have been described as providing digital outputs, many of the benefits of touch switches with the integrated control circuitry described above are provided by the integrated control circuitry. It can be enjoyed even when supplying analog output. In the case of a digital output, the output reflects only two states of information, stimulated or unstimulated, provided by input to the electrodes. In some applications, it is desirable to be able to accommodate more than two conditions. For example, it is desirable to be able to provide an output reflecting many states corresponding to many liquid levels rather than two states in a liquid sensing application similar to the situation described in connection with FIGS. 27A-27D. Analog output can accommodate many input states. FIG. 33A shows a possible circuit for an analog electric field sensor with an integrated control circuit. The circuit configuration of FIG. 33A corresponds to the circuit shown in FIG. 4, and includes a start-up and bias circuit 40 that supplies a current bias to the gates of the switches SW2 and SW4, and a power-on reset signal POR to the gates of the switches SW1 and SW3. A pulse generator and a logic circuit. The configuration of FIG. 33A also includes an input having active devices M1, M2, M5 and M6, similar to the input described with reference to FIG. 12A. The drains of active devices M1 and M2 are connected to traces INPUT1 and INPUT2, and are connected to traces PKOUT1 and PKOUT2 via diodes D1 and D2. The latter supplies an input to the differential amplifier circuit 160. The operation of this circuit can be understood from the description given in connection with FIGS. 4-7. The configuration shown in FIG. 33A can provide the benefits of the configuration shown in FIGS. 4-7, including buffering of sensor electrodes and strobe signals, common mode rejection of electrical interference in electrodes and circuits, temperature stability, etc. . 33B and 33C show timing diagrams for the circuit shown in FIG. 33A. 33B and 33C show the oscillation signal OSC and the signals supplied to the traces IN1, IN2, INPUT1 and INPUT2. FIG. 33B shows these signals as a function of time in microseconds, and FIG. 33C shows these signals as a function of time in nanoseconds.

  FIG. 34 shows a 2 × 2 matrix of the electric field sensor of FIG. 33A that accepts an analog input and provides an analog output. The multiplexing system of FIG. 34 is the same as that shown in FIG. Trace ROW SELECT1, which has a signal supplied by control circuit 141, is high only during periods when analog switches ATS1 and ATS3 are powered. The analog output AOUT of analog switches ATS1 and ATS3 is provided to trace COLUMN RETURN1 and sent to analog interface circuit 142, but provides an output proportional to the stimulus applied to the electrodes of analog switches ATS1 and ATS3. These outputs are stable over temperature and exhibit good signal-to-noise performance characteristics due to the low impedance of the circuit, as well as common mode rejection characteristics. The analog signal can be processed in a similar manner as described in US Pat. No. 5,594,222 or using other analog processing techniques understood by those skilled in the art of electrical circuit design.

  FIGS. 35A-35B illustrate an embodiment 1100 of the present invention in which a field effect sensor is used with other structures to emulate a mechanical push button switch. Embodiment 1100 includes a dielectric substrate 1102, which can be implemented in any suitable shape. It is desirable that the substrate 1102 be essentially hard. For example, the substrate 1102 may be a conventional printed wiring board or panel, or a part such as, for example, an automobile door panel or dashboard, a refrigerator interior panel, or part of a larger assembly. Alternatively, the substrate 1102 may be a flexible circuit carrier. In such embodiments, it is desirable to affix a flexible circuit carrier to an essentially rigid secondary substrate (not shown). The substrate 1102 can have any other suitable shape as will be appreciated by those skilled in the art.

  The substrate 1102 defines an opening 1104. The field effect sensor 1106 </ b> A is disposed in the vicinity of the opening 1104 on the substrate 1102. Field effect sensor 1106A is shown in FIG. 35A as being disposed on one side of substrate 1102. FIG. On the contrary, the field effect sensor 1106A can be arranged on the opposite side of the substrate 1102. Further, in embodiments where the field effect sensor 1106A includes two or more electrodes, one or more such electrodes are disposed on one side of the substrate 1102, and the other electrode or electrodes are on the opposite side of the substrate 1102. It can also be arranged. In other embodiments, the field effect sensor 1106A is shown in FIG. 35D with respect to the field effect sensor 1106E, and can also be contained within the substrate 1102, as discussed further below.

  The shaft 1108 is inserted so as to slide into the opening 1104. A sleeve, bushing, or the like (not shown) may be provided along with the opening 1104 to allow the shaft 1108 to slide within the opening 1104 without swaying. The shaft 1108 preferably includes a knob 1110. In the illustrated embodiment, the shaft 1108 is a threaded plastic bolt having a knob 1110 at its head. In other embodiments, the shaft 1108 can have any suitable shape, as will be recognized by those skilled in the art, and can include any suitable material. The shaft 1108 preferably includes a non-conductive material, such as plastic or resin.

  An electric field stimulator 1112 is secured to the shaft 1108 at a predetermined location. The electric field stimulator 1112 includes a material that can easily stimulate or disturb the electric field, as described above. The electric field stimulator 1112 preferably includes a metal or other conductive material, but other materials are equally suitable, as is known to those skilled in the art. In the embodiment of FIG. 35A, the electric field stimulator 1112 is a metal washer fixed to the shaft 1108 and a threaded plastic washer 1116 is provided at both ends of the electric field stimulator 1112. In other embodiments, the electric field stimulator 1112 has other shapes, includes other materials, and can be secured to the shaft 1108 by any suitable means, as is known to those skilled in the art.

  The plastic washer 1116 provided between the substrate 1102 and the electric field stimulator 1112 is desirably sufficiently thick so that the electric field stimulator 1112 does not contact the electrode (s) of the field effect sensor 1106A. Alternatively, other structures (not shown) may be provided to prevent the field stimulator 1112 from contacting the electrode (s) of the field effect sensor 1106A, as is known to those skilled in the art.

  FIG. 35A shows a state in which the electric field stimulator 1112 is arranged on the same side of the substrate 1102 as the field effect sensor 1106A and on the opposite side of the head 1101 of the substrate 1102. In contrast, the field stimulator 1112 and the field effect sensor 1106A may be disposed on the opposite side of the substrate 1102, and the field stimulator 1112 and the head 1110 may be disposed on the same side of the substrate 1102.

  The shaft 1108 is longitudinally biased so that the field stimulator 1112 is typically placed at a predetermined position relative to the field effect sensor 1106A. Accordingly, the shaft 1108 and the electric field stimulator 1112 can be moved from their normal positions by applying an appropriate force to the head 1110. In the embodiment of FIG. 35A, the field stimulator 1112 is typically biased by a coil spring 1114 mounted around a shaft 1108 between the knob 1110 and the corresponding substrate 1102 surface. 1106A vicinity. The field stimulator 1112 moves away from the field effect sensor 1106A when a longitudinal force is applied to the shaft 1108. In another embodiment, the shaft 1108 is typically remote from the field effect sensor 1106A and near the field effect sensor 1106A when an appropriate force is applied to the shaft 1108, as will be appreciated by those skilled in the art. Can also be biased to move. In yet another embodiment, the coil spring 1114 can be replaced with a suitable structure that biases the shaft 1108. For example, a layer of flexible and / or resilient material (not shown) is placed around the opening 1104 on the substrate 1102 or when the substrate 1102 itself is pressed against the knob 1110 It can be made of a flexible and / or resilient material that, when deformed and released, returns to its origin and returns the knob 1110, shaft 1108 and field stimulator 1112 to their respective origin positions. Any number of other structures can be used to bias the shaft 1108, as is known to those skilled in the art.

  In operation, as described above, an electric field is generated around the field effect sensor 1106A. The electric field stimulator 1112 couples to this electric field when the shaft 1108 is in the normal position shown in FIG. 35A. As noted above, a detection circuit (not shown) associated with field effect sensor 1106A detects this coupling. For example, when the shaft 1108 moves longitudinally in response to the user depressing the knob 1110, the field stimulator 1112 moves away from the field effect sensor 1106A and couples with the electric field around the field effect sensor 1106A. Cuts out. In response, a detection circuit (not shown) detects this decoupling and provides a signal to the control circuit, which in turn provides a control signal to the controlled device as described above. Thus, embodiment 1100 emulates a mechanical push button switch.

  FIG. 35C illustrates another embodiment 1140 of the present invention that emulates a mechanical pull switch. Embodiment 1140 is structurally similar to embodiment 1100, except that shaft 108 is biased such that field stimulator 1112 is typically located at a predetermined distance from field effect sensor 1106A. is there. Thus, the field stimulator 1112 is normally decoupled from the field effect around the field effect sensor 1106A. To activate field effect sensor 1106A, the user takes knob 1110 and pulls field stimulator 1112 closer to field effect sensor 1106A so that field stimulator 1112 couples to the electric field around field effect sensor 1106A. . It is desirable to provide a mechanical stop, for example, a mechanical stop 1119 that is fixed to the shaft 1108 at a predetermined position, to limit the range of movement of the shaft 1108 by the coil spring 1114 or other biasing means.

  FIG. 35D shows another embodiment 1160 of the present invention emulating a mechanical pushbutton switch. Embodiment 1160 includes a post 1102 attached to a substrate 1102. The field effect sensor 1106E is enclosed in the substrate 1102 near the post 1118. In other embodiments, the field effect sensor 1106E is placed on either surface of the substrate 1102, similar to the field effect sensor 1106A, as shown and described with respect to FIG. 35A.

  A push button 1120 having a seating surface 1122 is combined to slide with the post 1118. The electric field stimulator 1112 is attached to the lower side of the push button 1120 closest to the substrate 1102. Although the push button 1120 and the electric field stimulator 1112 may be separated, it is not necessary to do so. Indeed, the push button 1120 and the electric field stimulator 1112 can be embodied as one monolithic structure.

  Coil spring 1114 biases push button 1120 so that field stimulator 1112 is typically located at a predetermined distance from field effect sensor 1106E. The field stimulator 1112 is moved toward the field effect sensor 1106E by applying an appropriate force to the seating surface 1122. Field effect sensor 1106E and associated detection circuitry respond as described above. In particular, this embodiment 1160 does not include an opening in the substrate 1102. Accordingly, the embodiment 1160 is particularly suitable for use in applications where it is desirable to eliminate liquid or contaminant intrusion into the substrate 1102. Embodiment 1160 can be readily modified to function as a pull switch, as will be appreciated by those skilled in the art.

  More than one field effect sensor can be used for any of the previous embodiments. FIG. 35E shows an embodiment in which four field effect sensors 1106A to 1106D are arranged around the opening 1104 of embodiments 1100 and 1140. Embodiment 1160 can be similarly modified. Other embodiments can use more or less than four field effect sensors.

  In an embodiment using multiple field effect sensors 1106A-1106n, the detection and control circuitry associated with the sensors is essentially as the field stimulator 1112 moves toward or away from the field effect sensors 1106A-1106n. At the same time, the field stimulator 1112 may be configured to couple to or break the field (s) around the individual field effect sensor 1106i. Alternatively, such an embodiment may be such that when the field stimulator 1112 reaches different points in the range of travel in the direction toward or away from the field effect sensors 1106A-1106n, the field stimulator 1112 may It can be configured to couple with the electric field (s) around sensor 1106i or to break the coupling (eg, through the shape of the sensor and / or stimulator).

  Other modifications can be made to the above-described embodiment. For example, the biasing means may be omitted from any of the above-described embodiments so that the shaft 1108 or push button 1120 remains in the final position installed by the user. Further, although shaft 1108 and post 1118 are shown as being essentially perpendicular to substrate 1102, shaft 1108 and post 118 are at different angles with respect to substrate 1102, as is known to those skilled in the art. It can also comprise so that it may become.

  36A-36B illustrate another embodiment 1200 of the present invention that emulates a mechanical toggle switch. Embodiment 1200 includes a substrate 1202 that defines an opening 1204. The field effect sensor 1206A is disposed on the substrate 1202 in the vicinity of the opening 1204. The shaft 1208 is coupled to the substrate 1202 so as to rotate therethrough at the opening 1204. The shaft 1208 is not required, but can include a knob 1210. A bearing (not shown), such as a round or spherical bearing, or other means (not shown) is provided in the opening 1204 to support and / or limit the degree and direction of movement of the shaft 1208. You can also For example, in an embodiment intended to be used as a simple on / off switch, the shaft 1208 may be limited so that it moves only one side, eg, left and right in the embodiment of FIG. 36A. It is desirable to be able to do so.

  Electric field stimulator 1212 is secured to shaft 1208 at a predetermined position as described above with respect to embodiment 1100. In the embodiment of FIGS. 36A-36B, a coil spring 1214 is inserted between the head 1210 and the substrate 1202 to bias the shaft 1208 to a central position where the shaft 1208 is essentially perpendicular to the substrate 1202. To do. In other embodiments, other means may be used to bias the shaft 1208 to a central position or another desired position, as will be appreciated by those skilled in the art. Alternatively, such biasing means can be omitted so that the shaft 1208 normally rests in the final position to which it has been moved.

  In operation, as described above, an electric field is generated around the field effect sensor 1206A. If the shaft 1208 is in the center position, the electric field stimulator 1212 is sufficiently far from the electric field so that the electric field stimulator 1212 does not disturb the electric field. When the shaft 1208 is moved, for example, by a user applying a force perpendicular to the shaft 1208, the field stimulator 1212 moves so that at least a portion of the field stimulator 1212 approaches the field effect sensor 1206A. The electric field in the vicinity of the field effect sensor 1206A is disturbed. A detection circuit associated with the field effect sensor 1206A detects this disturbance and then sends an output signal to the corresponding control circuit as described above.

  Embodiment 1200 may be toggled by adjusting the coupling of shaft 1208 and opening 1204 such that shaft 1208 can toggle around and slide within opening 1204 as will be appreciated by those skilled in the art. It can be easily modified to realize a push button combination embodiment (not shown).

  FIG. 36C shows another embodiment that includes four field effect sensors 1206A-1206D that are positioned near the opening 1204 and spaced apart from each other and spaced 90 degrees around the opening 1204. FIG. Each field effect sensor 1206A-1206D includes a corresponding field generation and detection circuit. Certain field effect sensors 1206i operate when the field stimulator 1212 is sufficiently close to such a field effect sensor 1206i to disturb the electric field near the field effect sensor 1206i in response to a toggle of the shaft 1208. Typically, only one field effect sensor 1206i operates at a time. However, the field effect sensors 1206A-1206D (and their corresponding electric field generation and detection circuitry) may be coupled to two (or more) adjacent field effect sensors 1206i when the field stimulator 1212 is located near them. Can be adapted to operate simultaneously. For example, in the embodiment of FIG. 36C, when the shaft 1208 toggles to position at least a portion of the field stimulator 1212 between the field effect sensors 1206A and 1206B, the field stimulator 1212 may It can be combined with both 1206B. In other embodiments, as will be appreciated by those skilled in the art, more or less than four field effect sensors may be disposed in the vicinity of the opening 1204 on the substrate 1202 in any desired configuration.

  FIG. 36D shows another embodiment 1240 of the present invention that emulates a mechanical toggle switch. Embodiment 1240 includes a shaft 1208 connected to substrate 1202 at pivot point 1224. In this embodiment, the shaft 208 does not penetrate the substrate 1202. The electric field stimulator 1212 is fixed to the shaft 1208 at a predetermined distance from the pivot point 1224. Biasing means (not shown) can be provided to bias the shaft 1208 to any desired position.

  FIGS. 37A-37D show another embodiment 1300 of the present invention that emulates a mechanical rotary switch. The substrate 1302 defines an opening 1304. Inner field effect sensor 1306A and outer field effect sensor 1306B are placed on one surface of substrate 1302 at first and second predetermined distances from opening 1304, respectively. The shaft 1308 is inserted into the opening 1304 and freely rotates therein. Bushings, bearings or other means (not shown) may be provided to help the shaft 1308 rotate within the opening 1304 and prevent the shaft 1308 from sliding out of the opening 1304. It is desirable to provide a knob 1310 on the shaft 1308 so that the user can easily grasp and rotate the shaft 1308.

  The electric field stimulator mount plate 1330 is secured to the shaft 1308 by any suitable means at a predetermined distance from the substrate 1302, as is known to those skilled in the art. The inner electric field stimulator 1332 is mounted on the electric field stimulator mount plate 1330 in a ring shape at a predetermined distance from the center of the electric field stimulator mount plate 1330. This predetermined distance corresponds to and is preferably equal to a predetermined distance from the center of the opening 1304 to the inner field effect sensor 1306A. Similarly, the outer field stimulator 1334 is annularly mounted on the field stimulator mount plate 1330 at a predetermined distance from the center of the electric field stimulator mount plate 1330, and this predetermined distance is , Corresponding to and equal to a predetermined distance from the center of the opening 1304 to the inner field effect sensor 1306B. Desirably, the angular spacing between adjacent inner field stimulators 1332 is equal. Similarly, the angular spacing between adjacent outer field stimulators 1334 is desirably equal.

  In operation, the user turns knob 1310, which in turn rotates shaft 1308 and electric field stimulator mount plate 1330. As the field stimulator mount plate 1330 rotates, each inner field stimulator 1332 alternately repeats coupling and decoupling with the electric field around the inner field effect sensor 1306A. Similarly, each outer field stimulator 1334 also repeats coupling and decoupling with the electric field around the outer field effect sensor 1306B alternately. A detection circuit associated with the field effect sensors 1306A and 1306B detects the coupling and decoupling, and supplies an output signal corresponding to the coupling and decoupling to a control circuit (not shown). The control circuit can be adapted to know the angle and speed of rotation of the knob 1310 based on those signals.

  It is desirable that the inner electric field stimulators 1332 are not aligned in the radial direction nor positioned in the center between the electric field stimulators 1334 that are angularly adjacent. Thus, the inner field stimulator 1332 couples and decouples with the electric field around the inner field effect sensor 1306A at a specific angular displacement of the knob 1310, and the outer field stimulator 1334 differs from the knob 1310. In the angular displacement, coupling and decoupling are performed with the electric field around the outer field effect sensor 1306B. FIG. 37E shows a typical flow of output signals from the detection circuitry associated with the field effect sensors 1306A, 1306B when the knob 1310 is turned in a particular direction. As will be appreciated by those skilled in the art, based on these signals, the microprocessor determines whether knob 1310 is being turned clockwise or counterclockwise.

  In another embodiment, one of the inner field effect sensor 1306A and the outer field effect sensor 1306B can be omitted. In such embodiments, it may be desirable to also omit the corresponding inner field stimulator 1332 or outer field stimulator 1334.

  In another alternative embodiment, the shaft 1308 is adapted to be able to rotate within the opening 1304 and slide longitudinally, to emulate a mechanical pushbutton switch. As described above in connection with this, means for biasing the shaft 1308 in the longitudinal direction can be provided, thereby realizing an embodiment emulating a rotary / push button and / or a pull switch. Such embodiments, as will be appreciated by those skilled in the art, may include one or more additional field effect sensors and / or field stings to facilitate such push button and / or pull switch functions. May include a simulator.

  FIG. 37F shows an embodiment 1350 of the present invention emulating another rotary switch. Embodiment 1350 includes a second substrate 1340 that has a predetermined spatial relationship with substrate 1302. On the second substrate 1340, second inner and outer field effect sensors 1306C, 1306D are installed. Second inner and outer electric field stimulators 1342 and 1344 are attached to the second surface of electric field stimulator mounting plane 1330 opposite to the surface where inner and outer electric field stimulators 1332 and 1334 are located. It is done. The shaft 1308 rotates inside the opening 1304 and slides freely.

  FIG. 37F shows the field stimulator mounting plane 1330 in the first position, where the inner and outer field stimulators 1332, 1334 are the substrates 1302 (and thus the inner and outer field effect sensors). 1306A and 1306B are relatively close to each other), and the second inner and outer electric field stimulators 1342 and 1344 are relatively far from the second substrate 1340. When the knob 1310 is turned in this position, the inner and outer field stimulators 1332 and 1334 alternate between coupling and decoupling between the electric field around the inner and outer field effect sensors 1306A and 1306B, respectively. In this position, the second inner and outer field stimulators 1332, 1334 are sufficiently far away from the second inner and outer field effect sensors 1306C, 1306D, so the second inner and outer field stimulators 1342 , 1344 do not perform coupling and decoupling with electric fields around the corresponding field effect sensors 1306C, 1306D, respectively.

  Pushing the knob 1310 allows the user to move the electric field stimulator mount plate 1330 to a second position, where the inner and outer electric field stimulators 1342, 1344 are relatively far from the substrate 1302. And the second inner and outer field stimulators 1332, 1334 compared to the second substrate 1340 (and thus the ring in which the second inner and outer field effect sensors 1306C, 1306D are located). Approach. When the knob 1310 is rotated in this position, the second inner and outer field stimulators 1342 and 1344 are coupled to the electric field around the second inner and outer field effect sensors 1306C and 1306D, respectively. Repeat decoupling alternately. In this position, the inner and outer field stimulators 1332, 1334 remain sufficiently far away from the inner and outer field effect sensors 1306A, 1306B, so that the inner and outer field effect sensors 1306A, 1306B correspond to each other. Neither coupling nor decoupling is performed with the electric field around the field effect sensors 1306A and 1306B.

  As shown in FIG. 37F, a coil spring 1314 is provided to bias the electric field stimulator mount plate 1330 to the “normal” position. In other embodiments, the electric field stimulator mount plate 1330 can be biased to a different “normal” position. In yet another embodiment, the coil spring 1314 can be omitted so that the electric field stimulator mount plate 1330 remains in any desired position between the substrate 1302 and the second substrate 1340. Further, the embodiment 1350 includes inner and outer field stimulators 1332, 1334 when the field stimulator mount plate 1330 is located essentially midway between the substrate 1302 and the second substrate 1340. Both sets of 1342, 1344 can be adapted to couple with corresponding field effect sensors 1306A, 1306B, 1306C, 1306D. Alternatively, embodiment 1350 can be adapted so that when a field stimulator mount plate is in a position, there is no field stimulator coupled to the corresponding field effect sensor.

  All of the above embodiments are suitable for use with analog or digital detection and control circuitry, as will be appreciated by those skilled in the art. FIG. 37G illustrates another embodiment 1360 of the present invention that emulates a mechanical rotary switch. This is particularly suitable for use in analog detection and control circuits. Embodiment 1360 includes a substrate 1302 that defines an opening 1304. The field effect sensor 1306 is installed in the vicinity of the opening 1304 on the substrate 1302. The shaft 1308 is inserted into the opening 1304 and freely rotates therein. In the illustrated embodiment, the shaft 1308 is fixed in the longitudinal direction. In other embodiments, the shaft 1308 can be adapted to slide within the opening 1304. The shaft 1308 preferably includes a knob 1310 at one end. The electric field stimulator 1328 is fixed to the shaft 1308 at a predetermined distance from the substrate 1302. The field stimulator 1328 is preferably tapered like a wing of a propeller so that the distance between the field effect sensor 1306 and the field stimulator 1328 changes with the rotation of the knob 1310 and the shaft 1308. To do. Alternatively, the electric field stimulator 1328 is essentially flat, parallel to the substrate 1302, and has a width or thickness that varies with distance from the shaft 1308, as shown in FIG. 37I. Accordingly, the degree of coupling between the field stimulator 1328 and the electric field around the field effect sensor 1306 depends on the rotation of the knob 1310, the distance between the field stimulator 1328 and the field effect sensor 1306, and / or the field effect sensor 1306. It varies as a function of the effective area of the adjacent electric field stimulator. Through the use of appropriate analog detection and control circuitry, embodiment 1360 can emulate a potentiometer, for example.

  FIG. 37H illustrates another alternative embodiment 1380 of the present invention that is particularly suitable for use with analog detection and control circuitry. Embodiment 1380 includes a substrate 1302 defining an opening 1304 having a screw 1305 therein. The field effect sensor 1306 is disposed on the substrate 1302 near the opening 1304. A threaded shaft 1308 having a knob 1310 at one end is screwed into the opening 1304. The electric field stimulator 1312 is fixed to the shaft 1308 at a predetermined position. When the knob 1310 is turned clockwise, the electric field stimulator 1312 moves in a direction away from the field effect sensor 1306. Conversely, when the knob 1310 is turned counterclockwise, the electric field stimulator 1312 moves in a direction approaching the field effect sensor 1306. Accordingly, the rotation of the knob 1310 is changed to intend a coupling between the field stimulator 1312 and the field effect sensor 1306. Such coupling changes can be easily detected and processed by analog detection and control circuitry, as is known to those skilled in the art.

  38A-38D illustrate yet another embodiment 1400 of the present invention that emulates a rotary switch. Embodiment 1400 includes a substrate 1402. Inner and outer knobs 1450, 1452 are secured to the substrate 1402 by any suitable means so that they can rotate about an axis that is essentially perpendicular to the substrate 1402, as will be appreciated by those skilled in the art. Yes. One or more electric field stimulators 1412 are installed at the base of each of the inner and outer knobs 1450, 1452. The inner and outer field effect sensors 1406A, 1406B are installed essentially in line with the electric field stimulators 1412 installed on the corresponding inner and outer knobs 1450, 1452, respectively, so that each electric field stimulator 1412 As the knobs 1450 and 1452 rotate, coupling and decoupling are alternately repeated between the electric fields around the respective field effect sensors 1406A and 1406B. The electric field stimulator 1412 can be embodied by various methods. For example, each electric field stimulator 1412 may be a conductive object 1413, for example, a ball bearing set at the bottom of the corresponding knob 1450, 1452. Alternatively, each electric field stimulator 1412 may be a bump 1417 in a ring 1415 inserted in the bottom of the corresponding knob 1450, 1452, as shown in FIG. 38D. In the preferred embodiment, the ring 1415 comprises beryllium copper with stamped bumps 1517.

  39A-39B show an alternative embodiment 1500 of the present invention that emulates a rotary switch. This embodiment is particularly suitable for applications that sense angular position. Those embodiments use a single field effect sensor with multiple sensing electrodes. This embodiment includes a substrate 1502 on which is disposed a string of sensing electrodes 1505A-1505H incorporated between a generally annularly configured detection 1503 and resistors R1-R7. In an alternative embodiment, the detection circuit 1503 is located remotely and more or fewer sensing electrodes and resistors than those shown are used.

  The knob 1510 is connected to the substrate 1502 so that the knob 1510 can be rotated about an axis that is essentially perpendicular to the substrate 1502. In the embodiment of FIG. 39A, the shaft 1508 is inserted into an opening 1504 defined by the substrate 1502 so that it can rotate freely, and a knob 1510 is secured to the shaft 1508. In other embodiments, the shaft 1508 is secured to the substrate 1502 so that the knob 1510 can rotate about the shaft 1508. The electric field stimulator 1512 is embedded in or attached to the knob 1510, along the arc corresponding to the electrodes 1505 </ b> A to 1505 </ b> H and the resistors R <b> 1 to R <b> 7 arranged annularly on the substrate 1502 by the knob 1510. The emulator 1512 can be rotated.

  In operation, when the user turns the knob 1510, the electric field stimulator 1512 repeats coupling and decoupling alternately with the electric field around the corresponding electrodes 1505A-1505H. The analog detection circuit can be adapted to determine the range, speed and direction of rotation of the knob 1510, as will be appreciated by those skilled in the art. In a preferred embodiment, the detection circuit 1503 can take the form shown in FIG. 33A, and the electrodes 1505A and 1505H in the embodiment of FIG. 39A are respectively the electrodes E1 and E2 shown in FIG. 33A. Occupies the place. The signal strength at the (+) and (−) inputs, and thus the output of the adder 160, is unique and predetermined for each position of the electric field stimulator 1512 relative to the electrodes 1505A-1505H, as will be appreciated by those skilled in the art. Have a value. (A detection circuit of the form shown in FIG. 33A can also be used to detect changes in the distance between two conductive sheets, as will be appreciated by those skilled in the art. Thus, the detection circuit of FIG. Can be used with a pair of conductive sheets configured as vibration sensors, sound pressure sensors, air pressure sensors, position sensors, and the like, and in certain embodiments, one conductive material between the conductive sheets. A layer of foam material can be placed.)

  As shown in FIG. 39B, a conductor 1507 having an impedance that varies with length can be used in place of the electrode-resistor string shown in FIG. 39A. The continuously changing impedance of the conductor 1507 continuously changes the output of the detection circuit 1503 as the electric field stimulator 1512 changes its position in response to the rotation of the knob 1510. Therefore, for example, it is desirable to use the conductor 1507 when it is desired to increase the resolution of the angular position.

  The embodiment of FIGS. 39A-39B can be readily adapted for use as an angular position sensor, as will be appreciated by those skilled in the art. The principle of the embodiment of FIGS. 39A-39B is facilitated, for example, by placing field effect sensors 1505A-1505H and resistors R1-R7 in a straight line and replacing knob 1510 with a slide, as shown in FIG. 42A. It can be adapted to provide a slide switch or slide potentiometer. These principles can be further extended to detect the location of the stimulus in the xy array by forming an array of detection circuits and electrode-resistor strings, as shown in FIG. 42E.

  FIG. 40 shows yet another embodiment 1600 of the present invention that emulates a rotary switch. Embodiment 1600 includes a shaft 1608 having an outer knob 1610 and an inner knob 1611 and a substrate 1602. The substrate 1602 is formed by, for example, a molding method so as to capture the inner knob 1611 and enclose the field effect sensor 1606. A field coupling element 1612 is contained or embedded in the outer knob 1610. Alternatively, the electric field coupling element 1612 is enclosed or embedded in the inner knob 1611. A light emitting device 1621 can be enclosed in the substrate 1602. The light emitting device 1621 is used to selectively illuminate at least a portion of the outer knob 1610 by using a transparent or translucent material for at least a portion of the substrate 1602, inner and outer knobs 1610, 1611 and shaft 1608. Can do.

  FIG. 41A shows an embodiment 1700 of the present invention that emulates a mechanical rocker switch. The embodiment 1700 includes a substrate 1702 and two field effect sensors 1706A and 1706B mounted on the surface of the substrate 1702, and includes electric field stimulators 1713A and 1713B in the form of a rocker 1713 fixed to the substrate 1702. In the illustrated embodiment, the rocker 1713 is a piece of bent spring steel secured to the substrate 1702 and the electric field stimulators 1713A, 1713B are monolithic portions of the rocker 1713. In alternative embodiments, the rocker 1713 includes other materials and can take other forms, and the field stimulators 1713A, 1713B are embedded within the rocker 1713, as will be appreciated by those skilled in the art. Another element may be a ball bearing, for example.

  In operation, the user pushes either the electric field stimulator 1713 A corresponding to the left side of the rocker 1713 or the electric field stimulator 1713 B corresponding to the right side of the rocker 1713 toward the substrate 1702. As field stimulators 1713A, 1713B approach or contact substrate 1702, field stimulators 1713A, 1713B couple with corresponding field effect sensors 1706A, 1706B. In the illustrated embodiment, both field stimulators 1713A, 1713B can be moved toward the substrate 1702 simultaneously. It is desirable to configure the rocker 1713 to always move only one of the electric field stimulators 1713A, 1713B.

  FIG. 41B shows another embodiment 1750 of the present invention that emulates a mechanical rocker switch. Embodiment 1750 is similar to embodiment 1700 except that embodiment 1750 uses a hard rocker 1713. In certain embodiments, the rocker 1713 includes a material that does not fully bond with the field effect sensors 1706A, 1706B when pressed. In such embodiments, as will be appreciated by those skilled in the art, a conductive object 1715 can be embedded at an appropriate location on the rocker 1713 to enhance such coupling. In another embodiment, the rocker 1713 is configured such that when a user pushes a corresponding portion of the rocker 1713 toward the substrate 1702, the user's finger over the rocker 1713 couples with the electric field around the field effect sensors 1706A, 1706B. It is produced in such a size and shape.

  Biasing means may be provided to bias the rocker 1713 to a predetermined “normal” position. In FIG. 41B, the biasing means is implemented as a pair of plastic tabs 1725 secured to the substrate 1702. The plastic tab 1725 is flexible enough to bend when the rocker 1713 is pushed and elastic enough to return the rocker 1713 to the “normal” position when the rocker 1713 is released. Any other suitable biasing means may be used, as will be appreciated by those skilled in the art.

  Alternatively, as shown in FIG. 41C, the rocker 1713 and tab can be adapted to lock the rocker 1713 in a special location until moved by the user. In such an embodiment, the tab 1725 desirably includes a protrusion 1725 that protrudes toward the end of the rocker 1713, and the rocker 1713 includes a recess 1727 for receiving the protrusion 1725. desirable.

  FIG. 42A shows an embodiment 1800 of the present invention emulating a mechanical slide switch. Embodiment 1800 includes a substrate 1802. One or more field effect sensors 1806 are provided on the substrate 1802. An electric field stimulator 1812, such as a conductive cylinder or ball bearing, is secured to the slide 1811. The slide 1811 meshes with a rail 1803 fixed to the substrate 1802. In operation, the user slides the slide 1811 back and forth along the substrate 1802. As the field stimulator 1812 approaches a particular field effect sensor 1806, the field stimulator 1812 couples with the electric field around that field effect sensor 1806. Similarly, when the field stimulator 1812 moves away from a particular field effect sensor 1806, the field stimulator 1812 breaks coupling with the electric field around the field effect sensor 1806.

  In an alternative embodiment, the slide 1811 can be replaced with a slide 1817 having a notch 1819 designed to accommodate the user's finger. In this embodiment, the user's finger functions as a dielectric field stimulator 1812. In another alternative embodiment, the slide 1811 can be omitted entirely. The same principle can be applied to rotary switch emulation by placing a field effect sensor 1806 around a cylinder (not shown) or around a frustum of a cone 1807, as shown in FIG. 42D. .

  In certain embodiments, a portion of slide 1811 can be omitted. Such an embodiment includes a light pipe 1821 and includes a light source (not shown) for illuminating the light pipe 1821 along with the substrate 1802, and a light channel 1823 installed on the slide 1811 includes a light pipe. It is desirable to receive light from 1821. In other embodiments, it may be desirable to illuminate slide 1811 or a portion thereof using other means.

  FIG. 42B shows an embodiment 1850 of the present invention emulating a slide switch. The embodiment 1850 is similar to the embodiment 1800 except that the embodiment does not include the slider 1811 at all. A flexible sheet 1827 under the rail 1803 so as to cover the substrate 1802. Sheet 1827 is preferably easily replaced, and may include, for example, graphics that display the position of field effect sensors (not shown) installed on substrate 1802 under sheet 1827. Usually, there is a gap between the sheet 1827 and a field effect sensor (not shown) installed on the substrate 1802 below the sheet 1827. When the user contacts the sheet 1827 to activate the field effect sensor, air is excluded from the air gap, allowing the user's finger and the electric field around the field effect sensor to be coupled and strengthened.

  FIG. 42C shows another embodiment 1860 of the present invention that emulates a slide switch. A field effect sensor 1806 is provided on the substrate 1802. The substrate 1802 includes rails 1803. The slide 1811 meshes with the substrate 1802 so as to slide through the rail 1803. The electric field stimulator 1812 is preferably a conductive object placed on the slide 1811. In the embodiment of FIG. 41C, the cross-sectional area of the electric field stimulator 1812 changes from one end of the slide 1811 to the other end. When the slide 1811 is in the position shown in FIG. 42C, the field stimulator 1812 is away from the field effect sensor 1806 and does not couple with the electric field around the field effect sensor 1806. For example, when the slide 1811 is moved to the right by the user's finger, the field stimulator eventually moves sufficiently close to the field effect sensor 1806 and couples with the electric field around the field effect sensor 1806. Initially, such coupling is small due to the small area of the field effect sensor 1806 and hence the field stimulator 1812 in the vicinity of the corresponding electric field. As the slide 1811 moves further to the right, a larger portion of the field stimulator 1812 approaches the field effect sensor 1806 and thus the corresponding electric field, increasing the coupling between the field stimulator 1812 and the electric field. The analog detection circuit recognizes the changing coupling state and supplies a corresponding analog output to the corresponding control circuit. Biasing means, such as a coil spring 1814, may be provided to hold the slide 1811 in such a “normal” position when there is no force to move the slide from the “normal” position.

  FIG. 43 shows an embodiment 1900 of the present invention that emulates a mechanical spherical switch or track ball. Embodiment 1900 includes a substrate 1902 that constitutes a container 1962 of a ball 1960. One or more field effect sensors 1906 are embedded within the substrate 1902 or are disposed on the substrate surface. Around the ball 1960 is an electric field stimulator 1912 arranged in a unique non-repeating pattern. In operation, the field stimulator couples and decouples the electric field around the field effect sensor 1906 as the ball 1960 rotates within the container 1962. The detection and control circuitry associated with field effect sensor 1906 can be adapted to determine the direction and magnitude of rotation of ball 1960, as will be appreciated by those skilled in the art. In an alternative embodiment, the ball 1960 can be fixed and the substrate 1902 and container 1962 can be rotated or moved around the container 1962. This embodiment can be used, for example, to detect tilt or vibration.

  FIG. 44 shows an application specific embodiment 2000 that emulates a mechanical switch according to the present invention, in particular a snowmobile or personal watercraft throttle. The field effect sensor 2006 is embedded or installed inside the handle 2002. An electric field stimulator 2012 in the form of a conductive object is mounted on the throttle lever 2016. When the user presses or releases the throttle lever 2016, the electric field stimulator 2012 approaches or moves away from the field effect sensor 2006, respectively. Using analog detection and control, the throttle position can be determined based on the signal received from the field effect sensor 2006. In a preferred embodiment, additional field effect sensors 2031, 2033 and 2035 can be attached to the handle 2002. These additional sensors include, for example, a spare sensor 2031 for throttle control, and a hand position sensor 2033 that stops the throttle until it detects that the rider's hand is on the handle 2002, such as a watercraft. A water sensor 2035 is included that stops the throttle when immersed in water.

  45A-45B show an application specific embodiment of a tire pressure sensor 2100 according to the present invention. In a preferred embodiment, a compressible, preferably conductive, foamable substrate 2104 is attached to the surface of the substrate 2102. A plurality of field effect sensors 2106 are arranged in a matrix on the opposite surface of the substrate 2102. In operation, for example, an automobile (not shown) tire is placed on the foam substrate 2104, thereby compressing the portion of the foam substrate 2104 that contacts the tire 2108. The compressed portion of the foamable substrate 2104 combines with the electric field around the corresponding field effect sensor 2106 to activate those sensors, as will be appreciated by those skilled in the art. A microprocessor (not shown) with a programmed load weight on the tire 2108 may be configured based on the signal it receives from the field effect sensor 2106 corresponding to the area of compressed foam material of the tire 2108. Air pressure can be determined. In other embodiments, the foamable substrate 2104 can be omitted and the tire 2108 itself can be coupled to the field effect sensor 2106.

  FIG. 46 shows an automobile seat 2202 having a seat portion 2202A and a backrest portion 2202B. The seat 2202 is preferably packed with a compressible foam material 2204, which includes a plurality of field effect sensors 2206A that detect the weight applied to the seat 2202, and the physicality of a person sitting on the seat 2202 or a placed object. A plurality of field effect sensors 2206B that sense the dimensions are embedded. The field stimulator 2212 embedded as a seat support post in the illustrated embodiment is placed in a predetermined spatial relationship with respect to the field effect sensor 2206A.

  When seat 2202 is empty, field effect sensor 2206A is at a predetermined distance from field stimulator 2212, and field effect sensor 2206A is not driven. When a load, such as a person or luggage, is placed on the seat 2202, the foam material of the seat portion 2202A is compressed and moves the field effect sensor 2206A closer to the electric field stimulator 2212 so that the electric field stimulator 2212 The electric field around the effect sensor 2206A is disturbed. The heavier the load placed on the seat 2202, the greater the compression of the foam material 2204 of the seat portion 2202B and thus the corresponding displacement of the field effect sensor 2206A. An analog detection and control circuit (not shown) that receives the output signals from the field effect sensor 2206A, based on those signals, determines the displacement of the field effect sensor 2206A in response to the load being placed on the seat 2202. Can be determined. Based on this displacement data and the compression characteristics of the foam material 2204, the control circuit can determine the weight of an object placed on the seat 2202 or a seated person.

  Further, when the seat 2202 is empty, there is no stimulus coupled to the electric field around the field effect sensor 2206B. When a person sits on the seat 2202 or a load is placed, the person or portion of the load that approaches any field effect sensor 2206B couples with the electric field around those sensors. Analog or digital detection and control circuitry that receives the output signal from the field effect sensor 2206B can determine the physical contour of the load (human or luggage) on the seat 2202. The control circuit can use this data along with the weight derived from the signal received from the field effect sensor 2206A as described above to determine whether the load on the seat 2202 is a person or a load. If the load determined by the control circuit is not a person but a load, the passenger airbag will be deactivated. If the load determined by the control circuit is a person rather than a load, the airbag deployment speed will be adjusted to the size and weight of the person occupying the seat.

  While several embodiments of the present invention have been shown, it will be apparent to those skilled in the art that many modifications can be made without departing from the spirit of the claims set forth below.

1 is a perspective view of components of a preferred embodiment of a touch switch of the present invention. FIG. 1 is a cross-sectional view of a two-electrode touch pad and integrated circuit chip of the present invention. The top view of embodiment of the touch switch apparatus of this invention. FIG. 2 is a schematic electrical circuit diagram of a touch switch control circuit configured for a preferred mode of operation. FIG. 6 is a schematic electrical schematic of a touch switch control circuit configured for an alternative preferred mode of operation. FIG. 6 is a schematic electrical schematic of a touch switch control circuit configured for another alternative preferred mode of operation. FIG. 6 is a schematic electrical schematic of a touch switch control circuit configured for yet another alternative preferred mode of operation. FIG. 6 is a cross-sectional view of an alternative embodiment of a touch pad of the present invention. FIG. 6 is a cross-sectional view of another alternative embodiment of the touch pad of the present invention. The diagram of an embodiment of a touch switch panel using a plurality of touch switches in a matrix. 8 is a schematic electrical circuit diagram of an input circuit for a touch switch control circuit that is compatible with the circuit shown in FIGS. 4-7. FIG. 11A is a schematic electrical circuit diagram of the input circuit for the touch switch control circuit of FIGS. 11A-11D in which the active device functions as a current source. FIG. 12A is a schematic electrical circuit diagram of the input circuit for the touch switch control circuit of FIGS. 12A-12H including different combinations of active devices. FIG. 11A is a schematic electrical circuit diagram of the input circuit for the touch switch control circuit of FIGS. 11A-11D having an active square root device. FIG. 14A is a schematic electrical circuit diagram of the input circuit for the touch switch control circuit of FIGS. 14A-14D having different active square root devices. FIG. FIG. 15B is a schematic electrical circuit diagram of the input circuit for the touch switch control circuit of FIG. 15A having a swamping capacitance provided by a capacitor. FIG. 17 is a schematic electrical circuit diagram of the input circuit for the touch switch control circuit of FIG. 16 in which the swamping capacity is provided by the depletion capacity of the input diode. FIG. 6 is a structural diagram of a touch switch assembly showing one possible configuration for bringing electrodes close to an integrated circuit. FIG. 3 is a schematic electric circuit diagram showing a configuration for providing negative feedback directly to an input circuit. FIG. 4 shows a common gate structure with a swamping capacitor at the front end and how the input structure differs from the common source configuration shown in all previous drawings. The figure which shows the structure of FIG. 18B provided with the depletion diode. FIG. 18B shows the configuration of FIG. 18B using two swamping capacitors in a single electrode format and shows cost-effective integrated circuit matching. The figure which shows the structure of FIG. 18D provided with the depletion diode. FIG. 2 is a schematic electric circuit diagram of an output circuit for an integrated circuit of a touch switch control circuit. Schematic diagram of a touch cell matrix for use in various modes of operation. Schematic diagram of a MOSFET blocking device. Schematic diagram of one method of constructing a membrane or other mechanical switch matrix and addressing and timing it. FIG. 23 is a schematic diagram of FIG. 22 where the switch is a touch switch assembly having two connections to the matrixed address lines. FIG. 10 is a schematic electrical circuit diagram illustrating certain functions of the output circuit shown in FIG. 9 in communication with a touch switch control circuit. FIG. 4 shows a possible configuration of an active device for constructing a latch circuit according to the present invention. 1 is a schematic diagram of a latch circuit according to the present invention. FIG. 27D shows a capacitive switch device for use with the integrated circuit of the present invention, the circuit shown in FIG. 26D being responsive to the capacitance between them changing in response to changes in the distance between the two electrodes. . FIG. 26 shows a circuit according to the invention for use in the application described with reference to FIGS. 26A-26C. FIG. 28E illustrates a capacitive sensing device that senses liquid for use with the integrated circuit of the present invention, and the circuit shown in FIG. 27E is responsive to changes in the relative dielectric constant of the electrodes. FIG. 28 shows a circuit according to the invention for use in the application described with reference to FIGS. 27A-27D. FIG. 28C illustrates a capacitive switch device for use with an integrated circuit of the present invention, wherein the circuit of FIG. 28C is responsive to capacitance between two electrodes that changes due to effective changes in the surface area of one electrode. FIG. 29 shows a circuit according to the invention for use in the application described with reference to FIGS. 28A-28B. FIGS. 29A-29D show the electrode configuration of the device in various input stages, and FIGS. 29E-29F show the device configuration of the device that functions as a dialing device for use with the integrated circuit of the present invention. FIG. 29G shows a pulse output of one type of rotation, and FIG. 29G shows a possible integrated circuit configuration for use with the device shown in FIGS. 29A-29D). FIG. 5 shows another type of capacitive switch dial device for use with the integrated circuit of the present invention, with the electrode being grounded by the user. The figure which shows the pulse output of two rotations of an apparatus. FIGS. 30A-30E are schematic diagrams illustrating input connections between the device of FIGS. 30A-30E and an integrated circuit used with the device. FIGS. The figure which shows the structure and isolation | separation layer of a touch switch provided with the integrated control circuit assembled in the 2x2 matrix form on the board | substrate. 1 shows an embodiment of an integrated circuit of the present invention that uses an AC input and low current. FIG. FIG. 4 shows the input and other parts of an embodiment of an integrated circuit of the present invention for use in an electric field sensing application having an analog output. FIG. 33B is a timing diagram of the integrated circuit shown in FIG. 33A. The figure which shows the matrix of an analog output sensor. FIG. 3 is a side view of an embodiment of push button switch emulation according to the present invention. FIG. 5 is a bottom view of an embodiment of push button switch emulation according to the present invention. FIG. 5 is a side view of an alternative embodiment of push button switch emulation according to the present invention. FIG. 6 is a side view of another alternative embodiment of push button switch emulation according to the present invention. FIG. 6 is a side view of yet another alternative embodiment of push button switch emulation according to the present invention. FIG. 3 is a side view of an embodiment of toggle switch emulation according to the present invention. FIG. 3 is a side view of an embodiment of toggle switch emulation according to the present invention. FIG. 6 is a bottom view of an embodiment of toggle switch emulation according to the present invention. FIG. 6 is a side view of an alternative embodiment of toggle switch emulation according to the present invention. FIG. 3 is a side view of a rotary switch emulation according to the present invention. FIG. 3 is a bottom view of some embodiments of rotary switch emulation according to the present invention. FIG. 6 is a bottom view of another portion of an embodiment of a rotary switch emulation according to the present invention. FIG. 6 is a bottom view of another portion of an embodiment of a rotary switch emulation according to the present invention. FIG. 6 is a timing diagram of an embodiment of a rotary switch emulation according to the present invention. FIG. 6 is a side view of an alternative embodiment of rotary switch emulation according to the present invention. FIG. 6 is a side view of another alternative embodiment of rotary switch emulation according to the present invention. FIG. 6 is a side view of yet another alternative embodiment of rotary switch emulation according to the present invention. FIG. 6 is a top view of a portion of yet another alternative embodiment of a rotary switch emulation according to the present invention. FIG. 6 is a side view of yet another alternative embodiment of rotary switch emulation according to the present invention. FIG. 6 is a top view of yet another alternative embodiment of rotary switch emulation according to the present invention. FIG. 6 is a bottom view of a portion of yet another alternative embodiment of a rotary switch emulation according to the present invention. FIG. 9 is a partial cross-sectional view of a portion of yet another alternative embodiment of a rotary switch emulation according to the present invention. FIG. 4 is a bottom view of an embodiment of a rotary switch emulation and / or angular position sensor according to the present invention, and a schematic diagram illustrating an electrode structure used in connection therewith. FIG. 39B is a schematic diagram of an alternative electrode structure for use with the embodiment illustrated in FIG. 39A. FIG. 6 is a side view of yet another alternative embodiment of rotary switch emulation according to the present invention. FIG. 3 is a side view of an embodiment of a rocker switch emulation according to the present invention. FIG. 6 is a side view of an alternative embodiment of rocker switch emulation according to the present invention. FIG. 6 is a side view of another alternative embodiment of rocker switch emulation according to the present invention. 1 is a side view of an embodiment of a slide switch emulation according to the present invention. FIG. FIG. 6 is a side view of an alternative embodiment of slide switch emulation according to the present invention. FIG. 6 is a side view of another alternative embodiment of slide switch emulation according to the present invention. FIG. 6 is a perspective view of an alternative embodiment of rotary switch emulation according to the present invention. FIG. 3 is a top view of an xy position sensor according to the present invention. FIG. 3 is a side view of an embodiment of ball switch emulation according to the present invention. The figure which shows the throttle control and related sensor which embody the principle of this invention. 1 is a perspective view of a tire pressure sensing device according to the present invention. 1 is a side view of a tire pressure sensing device according to the present invention. 1 is a side view of an automobile seat including a weight and position sensor according to the present invention. FIG.

Claims (3)

A device that emulates a mechanical switch,
A substrate,
A first field effect sensor installed on the substrate;
A first electric field stimulator, wherein the first electric field stimulator is movable along a predetermined path, and the predetermined path is relatively close to the first field effect sensor. A first field stimulator having at least a first point and at least a second point relatively far from the first field effect sensor;
Including said device.   2. The apparatus according to claim 1, further comprising a second field effect sensor installed on the substrate, wherein the first electric field stimulator is movable along a predetermined path. The apparatus, wherein the predetermined path has at least a third point relatively close to the second field effect sensor and at least a fourth point relatively far from the second field effect sensor. . The apparatus of claim 1, further comprising:
A second field effect sensor installed on the substrate;
A second electric field stimulator, wherein the second electric field stimulator is movable along a predetermined path, and the predetermined path is relatively close to the second field effect sensor. A second field stimulator having at least a third point and at least a fourth point relatively far from the second field effect sensor;
Including said device.

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