JP2008236641A - Transmitter - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a transmitter having a function for optimizing a noise level of transmitter noise or the like while taking into account the newly enacted standards of unwanted radiation of a radio unit. <P>SOLUTION: A digital wireless transmitter is equipped with a non-linear distortion compensation circuit using a Cartesian negative feedback linearizer, wherein variable attenuators 8, 21 are provided, respectively in a part of a forward circuit and a feedback circuit of transmission output. The digital wireless transmitter is further equipped with a means 25 for detecting spurious power in a predetermined frequency band of the transmission output and a control means 24 for adjusting a forward direction and/or feedback direction gain by means of the variable attenuators on the basis of a result of detection performed by the means, thereby reducing an increase in noise in the predetermined frequency band of the transmission output. <P>COPYRIGHT: (C)2009,JPO&INPIT

Description

  The present invention relates to a transmission apparatus for performing wireless transmission and wired transmission using a linear modulation method, and particularly relates to a transmitter.

  In a wireless system using a linear digital modulation scheme, such as 16QAM or π / 4 shift QPSK, it is essential to perform nonlinear distortion compensation of a power amplifier, and therefore various nonlinear distortion compensation schemes (linearizers) are used. ing. Among them, in particular, the Cartesian loop negative feedback linearizer has been used for a long time.

  For example, according to Patent Document 1 below, a Cartesian loop negative feedback system that stabilizes a negative feedback amplifier by adjusting a phase shift amount of a local signal, and detects spurious power and inputs it to a phase control unit. However, it is already known that the phase shift shift amount of the phase shifter is adjusted so as to reduce the power of the spurious.

  Further, for example, according to Patent Document 2 below, in a multiplex radio communication system using common modulation, nonlinear distortion compensation of a power amplifier is performed by performing a negative feedback loop at an IF frequency and inverse transformation is performed, and each common modulation is performed. It has already been known that a local transmission signal generator is provided for each channel to suppress the occurrence of carrier leakage, thereby maintaining the performance of adjacent channel leakage power characteristics and spurious characteristics stably.

  Further, according to Patent Document 3 below, in a digital radio having a power amplifier distortion compensation method using a Cartesian loop, the variable attenuators installed on the forward direction and the feedback side are simultaneously operated gently so that the characteristics are reversed. It is already known that the Cartesian loop is kept stable by making the loop gain constant, and at the same time, the regulation regarding the leakage power at the time of carrier off by the TDMA operation is satisfied.

  In addition, according to Patent Document 4 below, in a transmitter that uses a Cartesian loop system to remove distortion generated in the power amplifier, the third-order distortion level ratio of the power amplifier is more than a certain level by reducing the power. Thus, it is already known that when the distortion compensation by the Cartesian loop method is no longer necessary, the power supply to the feedback direction section is made in the middle stage to suppress the loop oscillation and keep the loop gain constant.

JP 2002-344555 A JP 2001-44859 A JP 2002-171178 A JP 2003-198390 A

  In the prior art, for example, as described above, in the negative feedback by the Cartesian loop method, the I component and Q component of the input signal and the I signal of the feedback signal are added on the input side of the adder and the adder in order to stabilize the system. It is necessary that the phase of the component and the Q component be opposite to each other, and therefore, the phase adjustment is generally performed by a phase shifter.

  By the way, according to the newly established standard for unnecessary radiation of radios, it has become necessary to suppress the noise level such as transmitter noise, which has not been a problem in the past, to a low level. .

  Here, a newly established standard for unnecessary radiation of a radio is described. Although there was no distinction between the conventional unnecessary emission areas, according to the new standard, it was divided into two areas, “out-of-band area” and “spurious area”, and a prescribed value was provided for each. In addition, a new reference bandwidth for spurious measurement is specified.

  Here, the vicinity band 1 (carrier wave ± 62.5 kHz to 1 MHz) of the TELEC measurement method will be described with an example. Conventionally, in this region, the frequency resolution setting of the spectrum analyzer is 10 kHz, and the spurious value at this setting has been measured. On the other hand, in this revision, the reference bandwidth is defined as 100 kHz, and the specified value similar to the conventional value must be satisfied even at a frequency resolution of 100 kHz. In other words, in spurious that does not have a band, there is no change in the spurious value due to the difference in frequency resolution, but in unnecessary components having a band such as noise, the band is added as much as the frequency resolution is increased, Spurious tube surface display of spectrum analyzer becomes high. Therefore, it is necessary to suppress the noise level such as transmitter noise to a low level.

  However, in the above-described prior art, the noise level such as transmitter noise is not sufficiently considered, that is, there is no function for optimizing the noise level such as transmitter noise. There was a problem that the specification of the new spurious standard could not be achieved.

  Therefore, in view of the problems in the conventional technology, specifically, the present invention has an object to provide a transmitter having a function of optimizing a noise level such as transmitter noise.

  To achieve the above object, according to the present invention, there is provided a digital radio transmitter including a nonlinear distortion compensation circuit using a Cartesian negative feedback linearizer, and at least a forward circuit and a feedback direction circuit for transmission output. Each of which includes a variable attenuator, and further, means for detecting spurious power in a predetermined frequency band of the transmission output, and the variable attenuator based on a detection result by the means Thus, a transmitter is provided that includes control means for adjusting the forward and / or feedback gain, thereby reducing an increase in noise in a predetermined frequency band of the transmission output.

  In the present invention, in the transmitter described above, the predetermined frequency band preferably includes a carrier ± 62.5 kHz to 1 MHz, and in the present invention, spurious power may be detected from an IF signal. .

  According to the present invention described above, the noise in a predetermined neighborhood band is monitored, and the signal level in the forward direction and the feedback direction is controlled so as to minimize the level. It is possible to achieve a practically excellent technical effect that it is possible to reliably provide a digital wireless transmitter that achieves the standard.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the accompanying drawings.

  First, a basic configuration of a transmitter according to an embodiment (first embodiment) of the present invention will be described with reference to FIG. FIG. 1 is a block diagram showing a configuration of a transmission unit of a digital radio using a Cartesian negative feedback linearizer.

  As shown in the figure, the baseband signal generator 1 is connected to the adder 2 and the adder 3, and the adder 2 and the adder 3 are connected to the quadrature modulator 4. The quadrature modulator 4 is connected to a mixer 6 via a band pass filter 5. The mixer 6 is connected to a directional coupler 10 via a bandpass filter 7, a variable attenuator 8, and a power amplifier 9. The directional coupler 10 is connected to an antenna 13 through an isolator 11 and a low-pass filter 12, and is connected to a power distributor 22. The power distributor 22 is connected to the quadrature demodulator 18 via the variable attenuator 21, the mixer 20, and the band pass filter 19, and is also connected to the mixer 23. The mixer 23 is connected to the control unit 24 via the band pass filter 26 and the level detector 25. The control unit 24 is connected to the variable attenuator 8 and the variable attenuator 21, respectively. The quadrature demodulator 18 is connected to the adder 2 and the adder 3, respectively. The reference signal generator 14 is connected to a frequency synthesizer 15 and a frequency synthesizer 17. The frequency synthesizer 15 is connected to the quadrature modulator 4 and the quadrature demodulator 18 via the phase shifter 16, and the frequency synthesizer 17 is connected to the mixer 6, the mixer 20, and the mixer 23.

  That is, the baseband signal generator 1 outputs the in-phase component (I component) and the quadrature component (Q component) of the baseband signal. Of these outputs, the I component is added to the feedback signal by the adder 2 and then output, while the Q component is added to the feedback signal by the adder 3 and then output. 4 is input.

  On the other hand, the reference frequency signal generated by the reference signal generator 14 is input as a reference signal for the frequency synthesizer 15 and the frequency synthesizer 17. The frequency synthesizer 15 generates a first local oscillation signal (LO signal) based on the input reference signal, and inputs the generated first LO signal to the quadrature modulator 4 and the phase shifter 16. The phase of the first LO signal input to the phase shifter 16 is adjusted by the phase shifter 16 and is supplied to the quadrature demodulator 18. Further, the frequency synthesizer 17 generates a second LO signal based on the reference signal, and supplies the second LO signal to the mixer 6 and the mixer 20.

  In the quadrature modulator 4 described above, the I component and the Q component of the input baseband signal are modulated into a signal in the intermediate frequency band (IF band) by the first LO signal. The modulated modulated wave signal has its unnecessary components removed by the band pass filter 5 and is input to the mixer 6. The modulated wave signal input to the mixer 6 is converted to a desired frequency by the second LO signal output from the frequency synthesizer 17 and further input to the bandpass filter 7. The level of the signal from which unnecessary spurious components have been removed by the bandpass filter 7 is adjusted by the variable attenuator 8 and input to the power amplifier 9. The signal input to the power amplifier 9 is amplified to a specified output level and input to the low-pass filter 12 via the directional coupler 10 and the isolator 11. Thereafter, the input low-pass signal is transmitted from the antenna 13 after the harmonic component generated in the power amplifier 9 is cut in the low-pass filter 12.

  In this way, in the transmitter according to the present invention, the negative feedback amplifier has a negative feedback linearizer configuration based on a Cartesian loop, so that a part of the output signal of the power amplifier 9 is fed back by the directional coupler 10. Are input to the variable attenuator 21. The level of the feedback signal input to the variable attenuator 21 is adjusted to an appropriate value and input to the mixer 20. Further, the feedback signal frequency-converted to the IF band by the mixer 20 is input to the quadrature demodulator 18 after unnecessary components are removed by the band-pass filter 19. The signal input to the quadrature demodulator 18 is demodulated into an I component and a Q component by the first LO signal input from the phase shifter 16. That is, the I component is input to the adder 2 and the Q component is input to the adder 3 as a feedback signal.

  In the present invention, as described above, in the negative feedback amplifier taking the configuration of the negative feedback linearizer by the Cartesian loop, a part of the output signal of the power amplifier 9 is fed back by the directional coupler 10, The signal input to the variable attenuator 21 via the power distributor 22 and the signal input to the power distributor 22 are input to the mixer 23 and frequency-converted to the IF band. A spurious component is extracted from the signal converted into the IF band by the band-pass filter 26, and the spurious level is detected by the level detector 25. Then, the control unit 24 adjusts the gains of the variable attenuator 8 and the variable attenuator 21 so as to minimize the detected spurious level.

  In the negative feedback described above, in order to stabilize the system, the I component and the Q component of the input signal on the input side of the adder 2 and the adder 3, and the phases of the I component and the Q component of the feedback signal are determined. These phase adjustments are performed by the phase shifter 14, although it is necessary that the phases are opposite to each other.

  Subsequently, the behavior of noise in the near band 1 with respect to the loop gain in the above-described Cartesian loop method will be described below with reference to FIGS.

  As described above, since this transmitter has a negative feedback linearizer (Cartesian loop) configuration, it is configured to suppress unnecessary components in the forward direction (nonlinear distortion, nearby noise, etc.) by the amount of the loop gain. . Here, the loop gain is a gain obtained by adding the gains in the forward direction and the feedback direction.

  When this loop gain is small, as shown in FIG. 2, the non-linear distortion or noise in the vicinity is not sufficiently suppressed. Therefore, in some cases, the regulation according to the new standard can be achieved. There may be no cases.

  A case where the loop gain is too large will be described with reference to FIG. Since this transmitter has a negative feedback configuration, the IQ output of the baseband signal generator 1 and the feedback signal have the adders 2 and 3 as described above, and the phases are opposite in the signal band. Need to be. FIG. 3 is a graph (Board diagram) showing a loop gain vs. frequency characteristic and a phase difference vs. frequency characteristic of an input signal and a feedback signal. In the figure, the signal band is indicated by diagonal lines.

  As is clear from the figure, as the frequency goes away from the signal band, the loop gain decreases and the phase difference starts from 180 °. When the loop gain disappears (0dB), the margin for the phase difference in phase (0 °) is called the phase margin, and the loop gain (minus (-)) at the point where the phase difference becomes the same phase is the gain margin. To tell. Then, by eliminating each of these margins, a loop oscillation occurs. That is, when the phase difference is in phase in the positive loop gain region, the loop oscillates.

  In the present invention, a case where the loop gain becomes large is considered based on this fact. That is, as shown in FIG. 4, when the loop gain increases (solid line → broken line), both the phase margin and gain margin decrease. Further, when the loop gain is increased, the phase margin and gain margin are reduced, leading to loop oscillation. When the oscillation tends to occur, a spectrum waveform is obtained as shown in FIG. 5, that is, noise rises (rises) at the point where the phase difference is in phase. In some cases, this excitement may be one factor that prevents the above-mentioned regulations from being achieved.

  From the above, when the loop gain is small, the neighborhood noise is (see FIG. 2). On the other hand, when the loop gain is large, the noise at the point where the loop gain becomes “0”, “noise at the in-phase point” ( As shown in FIG. 3, it can be seen that this is a factor that makes it impossible to achieve the regulation according to the new standard.

  The present invention has been achieved based on the above-mentioned recognition by the inventor. Specifically, in the Cartesian loop method, the loop gain is optimized in order to optimize the noise in the predetermined neighborhood band 1 described below. Execute the conversion. Details of the method will be described below.

  That is, according to the configuration shown in FIG. 1 (first embodiment), the signal extracted by the power distributor 22 is further converted to the IF band by the mixer 23 based on the LO signal from the frequency synthesizer 17. Convert frequency. Thereafter, the adjacent band 1 (carrier wave ± 62.5 kHz to 1 MHz) is extracted from the converted signal by the band-pass filter 26, and the level detector 25 detects the total power of the noise. That is, the control unit 24 controls the variable attenuator 8 for adjusting the level of the signal from which unnecessary spurious components have been removed so that the level of the detected nearby band 1 is minimized, and further the level of the feedback signal. The setting of the variable attenuator 21 for adjustment is controlled (that is, set to an optimum value).

  The optimum value may be automatically acquired before shipment of the product, for example, or may follow a change in temperature or the like in the operating environment even during the operation (operation). However, it may be controlled at all times.

  Next, another embodiment (second embodiment) of the present invention will be described with reference to FIG. The negative feedback configuration is the same as that in the above-described embodiment, and the description thereof is omitted. However, also in other embodiments, the same reference numerals as those in FIG. 1 indicate the same components.

  In this embodiment, unlike the configuration of FIG. 1, a spurious level detection is performed in the IF band. More specifically, the band-pass filter 5 in the IF band is connected to the power distributor 27. A part of the signal is extracted, and the spurious is extracted by the band-pass filter 26 for extracting the neighboring band 1. . In this case, the variable attenuator 8 and the variable attenuator 21 are controlled so as to minimize the value as in the above-described embodiment.

  According to such a configuration, since it is a loop system, the noise level of the output from the transmitter can be detected in the IF band provided in the middle of the forward path.

  In addition, still another embodiment (third embodiment) will be described with reference to FIG. In the other embodiments as well, the same reference numerals as those in FIG. 1 indicate the same constituent elements.

  As is apparent from the figure, in this further embodiment, the signal distributed by the power distributor 22 is input to the mixer 23 together with the level detector 31. Then, the level detector 31 detects the level of transmission power, and the control unit 30 controls the variable attenuator 21 in order to make this constant.

  According to such a configuration, in the negative feedback circuit, the output power level is determined by the gain in the feedback direction (provided that the forward gain is large). That is, the variable attenuator 21 is used for controlling transmission power. Further, the signal input to the mixer 23 is frequency-converted to the IF band as in the above-described embodiment, and further, the level detector 25 passes through the band-pass filter 26 for extracting the neighboring band 1. Measure the spurious level. Information obtained by the measurement is sent to the control unit 24. After that, the control unit 24 that has received the information adjusts the variable attenuator 8 and the variable attenuator 28 to optimize the above-described neighborhood noise level in the same manner as described above.

  According to this other embodiment (fourth embodiment), the variable attenuator 21 constituting the feedback path is also used for adjusting the output power level of the transmission signal, so that the spurious is optimized. However, it is not possible to adjust this, however, the feedback gain is adjusted by changing the Cartesian loop input. This mechanism will be described below.

  Normally, when the loop input is increased, the output level is increased accordingly. However, in this case, since output control (APC) is performed by controlling the variable attenuator 21, the output level is kept constant. That is, the fact that the output level does not change (is constant) even though the loop input is increased is nothing but that the feedback gain is increased by APC. As described above, in this further embodiment, it is possible to adjust the loop gain by controlling the Cartesian loop input while the APC is functioning.

  Further, FIG. 8 attached herewith shows a configuration of a transmitter according to still another embodiment of the present invention. It should be noted that this other embodiment is also an example in which the loop gain is adjusted by controlling the loop input and the spurious level is optimized by controlling the loop input, and the spurious is detected. This is an example in which the location to be detected is detected in the IF band. However, since the operation is the same as described above, the description thereof is omitted.

1 is a block diagram illustrating a basic configuration of a transmitter (a transmission unit of a digital radio) according to a first embodiment of the present invention. It is a figure which shows an example of the spectrum analyzer display in case a loop gain is small explaining the behavior of the noise in the near band in a Cartesian loop system. It is a graph (Board diagram) showing a loop gain vs. frequency characteristic and a phase difference vs. frequency characteristic of an input signal and a feedback signal when the loop gain is too large in the Cartesian loop method. It is a graph (Board diagram) showing a loop gain vs. frequency characteristic according to the present invention and a phase difference vs. frequency characteristic of an input signal and a feedback signal when the loop gain is too large in the Cartesian loop method. It is a figure which shows an example of the spectrum analyzer display in case a loop gain is large explaining the behavior of the noise in the near band in a Cartesian loop system. It is a block diagram which shows the basic composition of the transmitting apparatus (transmitting part of a digital radio) which becomes the 2nd Embodiment of this invention. It is a block diagram which shows the basic composition of the transmitting apparatus (transmitting part of a digital radio) which becomes the 3rd Embodiment of this invention. It is a block diagram which shows the basic composition of the transmitting apparatus (transmitting part of a digital radio) which becomes the 4th Embodiment of this invention.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 ... Baseband signal generator 2, 3 ... Adder 4 ... Quadrature modulator 5, 7, 19, 26 ... Band pass filter 6, 20, 23 ... Mixer 8, 21, 28, 29 ... Variable attenuator 9 ... Electric power Amplifier 10 ... Directional coupler 11 ... Isolator 12 ... Low pass filter 13 ... Antenna 14 ... Reference signal generator 15, 17 ... Frequency synthesizer 16 ... Phase shifter 18 ... Quadrature demodulator 22, 27 ... Power divider 24, 30 ... Control unit 25, 31 ... Level detector

Claims (2)

  A digital radio transmitter equipped with a nonlinear distortion compensation circuit using a Cartesian negative feedback linearizer, wherein at least a part of the forward circuit and the feedback circuit of the transmission output are each equipped with a variable attenuator Further, in order to adjust the forward direction gain and / or the feedback direction gain by the variable attenuator based on the detection result of the spurious power in the predetermined frequency band of the transmission output and the detection result by the means. And a control means for reducing an increase in noise in a predetermined frequency band of the transmission output.   2. The transmitter according to claim 1, wherein the predetermined frequency band includes a carrier wave ± 62.5 kHz to 1 MHz.

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