JP2008193222A - Estimating method and estimating device using the same, and reception device and reception system - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To improve estimation precision of the amount of a component corresponding to noise. <P>SOLUTION: An OFDM symbol comprising a guard interval prescribed with the same pattern with a rear portion of an effective symbol and an effective symbol disposed behind the guard interval is input to an estimating unit 54. The estimating unit 54 derives the difference between a part of the effective interval which should correspond to the guard interval and the guard interval for the input OFDM symbol to estimate the amount of the component corresponding to the noise. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to an estimation technique, and more particularly to an estimation method for estimating the quality of an OFDM symbol, and an estimation apparatus, reception apparatus, and reception system using the estimation method.

A signal received by a receiving device in a digital wireless communication system is affected by a wireless transmission path. Therefore, the strength of the received signal varies greatly depending on the state of the wireless transmission path and the distance between the transmitting device and the receiving device. In general, a receiving apparatus converts a received signal into a digital signal by an AD converter, and then performs a receiving process on the digital signal. Since the dynamic range of the AD converter is finite, it is desirable that the magnitude of the signal input to the AD converter is adjusted according to the dynamic range. For this reason, an AGC (Automatic Gain Control) is generally arranged in front of the AD converter. (For example, refer to Patent Document 1).
JP 2005-117341 A

  In wireless communication, effective use of limited frequency resources is generally desired. One of the technologies for effectively using frequency resources is the adaptive array antenna technology. Adaptive array antenna technology forms the directivity pattern of an antenna by controlling the amplitude and phase of signals transmitted and received by a plurality of antennas. That is, an apparatus provided with an adaptive array antenna changes the amplitude and phase of signals received by a plurality of antennas, and adds the changed reception signals. This corresponds to receiving a signal equivalent to a signal received by an antenna having a directivity pattern corresponding to the amount of change between the amplitude and phase (hereinafter referred to as “weight”). In addition, a signal is transmitted by an antenna directivity pattern corresponding to the weight.

  In the adaptive array antenna technique, an example of a process for calculating a weight is a method based on a minimum mean square error (MMSE) method. In the MMSE method, a Wiener solution is known as a condition for giving an optimum weight value, and a recurrence formula with a smaller amount of calculation than directly solving the Wiener solution is also known. As the recurrence formula, for example, an adaptive algorithm such as an RLS (Recursive Least Squares) algorithm or an LMS (Least Mean Squares) algorithm is used.

  When deriving weights in adaptive array antenna technology, information on signal strength received by a plurality of antennas is generally required. That is, it is required to maintain the intensity ratio among a plurality of received signals even after being converted into a digital signal by the AD converter. Therefore, AGC performs amplification with respect to a plurality of received signals with a common amplification factor. With such an amplification factor setting, there may be a reception signal that is not sufficiently amplified among the plurality of reception signals. As a result, the characteristic improvement by the adaptive array antenna technology is not sufficient.

  Under such circumstances, the present inventor has come to recognize the following problems. In order to perform sufficient amplification for each of the plurality of received signals, it is desirable to use an amplification factor corresponding to each of the plurality of received signals. On the other hand, in such a case, in order to maintain the intensity ratio between the plurality of received signals, after amplification, each of the plurality of received signals is corrected according to the magnitude of the noise component. Good. For this reason, the amount of improvement in characteristics by the adaptive array antenna technology depends on the estimation accuracy of the amount of components corresponding to noise.

  The present invention has been made in view of such circumstances, and an object thereof is to provide an estimation technique and a reception technique that improve the estimation accuracy of the amount of a component corresponding to noise.

  In order to solve the above-described problem, an estimation apparatus according to an aspect of the present invention provides an OFDM formed by a guard interval defined in the same pattern as the rear part of an effective symbol, and an effective symbol arranged at the subsequent stage of the guard interval. By deriving the difference between the card interval and the portion of the effective symbol that should correspond to the guard interval with respect to the input unit that inputs the symbol, and the OFDM symbol that is input to the input unit, the component corresponding to the noise An estimation unit for estimating the quantity.

  According to this aspect, the portion of the effective symbol that should correspond to the guard interval and the card interval are defined to be the same pattern, and the difference between the two is derived, so the amount of the component corresponding to the noise The estimation accuracy of can be improved.

  The input unit may continuously input a plurality of OFDM symbols, and the estimation unit may derive a difference between a rear part of the guard interval and a part of the effective symbol that should correspond to the rear part of the guard interval. In this case, since the rear part of the guard interval is used when the difference is derived, the influence of intersymbol interference can be reduced, and the estimation accuracy of the amount of the component corresponding to noise can be improved.

  You may further provide the derivation | leading-out part which derives | leads-out the magnitude | size of the delay component of the OFDM symbol input into the input part. The estimation unit may adjust the length of the rear part of the guard interval to be used for deriving the difference based on the size of the delay component derived by the deriving unit. In this case, since the length of the rear part of the guard interval to be used for deriving the difference is adjusted according to the magnitude of the delay component, the estimation accuracy can be improved according to the radio transmission path environment.

  A plurality of OFDM symbols input to the input unit constitute a packet signal, an OFDM symbol arranged in front of the packet signal has a known value, and an OFDM symbol following an OFDM symbol of a known value is unknown. The estimation unit estimates the amount of the component corresponding to the noise in the period of the OFDM symbol having a known value, and supports the noise in the period of the OFDM symbol having an unknown value. The amount of the added component may be updated. In this case, since the estimation is performed not only in the period of the OFDM symbol having a known value but also in the period of the OFDM symbol having an unknown value, the estimation accuracy of the amount of the component corresponding to noise can be improved.

  Among the packet signals input to the input unit, an OFDM symbol having a known value includes a plurality of effective symbols in succession, and the estimation unit performs guarding in the period of the OFDM symbol having a known value. The amount of the component corresponding to the noise may be estimated by deriving a difference between the interval and the effective symbol and deriving a difference between a plurality of effective symbols. In this case, since the difference between the guard interval and the effective symbol is derived and also the difference between a plurality of effective symbols is derived, it is possible to improve the estimation accuracy of the amount of the component corresponding to the noise.

  Another aspect of the present invention is a receiving device. The apparatus includes a plurality of input units corresponding to the plurality of antennas, a plurality of correction units corresponding to the plurality of input units, and a reception processing unit connected to the plurality of correction units. The signal input at each of the plurality of input units includes an OFDM symbol formed by a guard interval defined in the same pattern as the rear part of the effective symbol and an effective symbol arranged at the subsequent stage of the guard interval. Each of the input OFDM symbols is subjected to analog-digital conversion after the OFDM symbol received at the corresponding antenna is amplified with an amplification factor independent of the OFDM symbols received at the other antennas. Each of the plurality of correction units eliminates the noise by deriving the difference between the card interval and the portion of the effective symbol that should correspond to the guard interval with respect to the OFDM symbol input at the corresponding input unit. Estimate the amount of the corresponding component Performs correction by component corresponding to the noise, the reception processing unit, the OFDM symbols corrected in each of the plurality of the correction unit executes the reception process.

  According to this aspect, the signal subjected to the analog-digital conversion after being amplified by the independent amplification factor is corrected by the component corresponding to the noise, so that a plurality of signals are processed. However, the analog-digital conversion can be executed efficiently.

  An OFDM symbol input in each of a plurality of input units is composed of a plurality of sequences, and each of the plurality of correction units estimates an amount of a component corresponding to noise after being separated into OFDM symbols in units of sequences. May be. In this case, since a plurality of sequences are separated, a component corresponding to noise can be estimated even when a signal formed by a plurality of sequences is a processing target.

  Yet another embodiment of the present invention is also a receiving device. The apparatus includes a plurality of amplification units corresponding to each of the plurality of antennas, a plurality of conversion units corresponding to each of the plurality of amplification units, a plurality of correction units corresponding to each of the plurality of conversion units, and a plurality of A reception processing unit connected to the correction unit. Each of the plurality of amplifying units is formed by a guard interval that is an OFDM symbol received at a corresponding antenna and is defined in the same pattern as the rear part of the effective symbol, and an effective symbol that is arranged after the guard interval. The amplified symbol is amplified by an amplification factor independent of the amplification factor set by the other amplification units, and each of the plurality of conversion units is applied to the OFDM symbol amplified by the corresponding amplification unit. The analog-to-digital conversion is performed, and each of the plurality of correction units is different from the card symbol interval in the portion corresponding to the guard interval in the effective symbols with respect to the OFDM symbol converted in the corresponding conversion unit. Is used to estimate the amount of the component corresponding to the noise and Run the correction based component corresponding to the reception processing unit, the OFDM symbols corrected in each of the plurality of the correction unit executes the reception process.

  According to this aspect, after the amplification by the independent amplification factor, the correction by the component corresponding to the noise is performed on the analog-digital converted signal, so even when a plurality of signals are to be processed, Analog-to-digital conversion can be executed efficiently.

  Yet another embodiment of the present invention is a receiving system. The reception system includes a plurality of RF processing devices corresponding to the plurality of antennas, and a baseband processing device connected to each of the plurality of RF processing devices via a cable. Each of the plurality of RF processing devices includes an OFDM symbol received at a corresponding antenna and defined by the same pattern as the rear part of the effective symbol, and an effective symbol arranged at the subsequent stage of the guard interval. For the formed OFDM symbol, after performing amplification with an amplification factor independent of the amplification factor set by another RF processing device, analog-digital conversion is performed, and the baseband processing device For each OFDM symbol converted in each of the RF processing devices, by deriving the difference between the portion of the effective symbol that should correspond to the guard interval and the card interval, the amount of the component corresponding to noise is estimated. After performing correction with the component corresponding to noise, Respect, executes reception processing.

  According to this aspect, after the amplification by the independent amplification factor, the correction by the component corresponding to the noise is performed on the analog-digital converted signal, so even when a plurality of signals are to be processed, Analog-to-digital conversion can be executed efficiently.

  Yet another embodiment of the present invention is an estimation method. This method inputs an OFDM symbol formed by a guard interval defined in the same pattern as the rear part of the effective symbol and an effective symbol arranged at the subsequent stage of the guard interval, and is effective for the input OFDM symbol. The amount of the component corresponding to the noise is estimated by deriving the difference between the portion of the symbol that should correspond to the guard interval and the card interval.

  After a plurality of OFDM symbols are continuously input, a difference between the rear part of the guard interval and the part of the effective symbol that should correspond to the rear part of the guard interval may be derived. The size of the delay component of the input OFDM symbol may be derived, and the length of the rear portion of the guard interval to be used for deriving the difference may be adjusted based on the size of the derived delay component. A plurality of input OFDM symbols constitute a packet signal, an OFDM symbol arranged in front of the packet signal has a known value, and an OFDM symbol following an OFDM symbol of a known value has an unknown value. In the period of the OFDM symbol having a known value, the amount of the component corresponding to noise is estimated, and in the period of the OFDM symbol having an unknown value, the amount of the component corresponding to noise is updated. May be. Among the input packet signals, the OFDM symbol having a known value includes a plurality of effective symbols in succession, and the difference between the guard interval and the effective symbol in the period of the OFDM symbol having the known value. And the amount of the component corresponding to the noise may be estimated by deriving a difference between a plurality of effective symbols.

  It should be noted that any combination of the above-described constituent elements and a conversion of the expression of the present invention between a method, an apparatus, a system, a recording medium, a computer program, etc. are also effective as an aspect of the present invention.

  According to the present invention, it is possible to improve the estimation accuracy of the amount of components corresponding to noise.

  Before describing the present invention in detail, an outline will be described. Embodiments of the present invention relate to a receiving apparatus in a communication system using an OFDM (Orthogonal Frequency Division Multiplexing) modulation scheme such as a wireless LAN (Local Area Network) compliant with IEEE802.11a / g. Here, in the OFDM modulation scheme, continuous OFDM symbols are used, and each OFDM symbol is configured by a combination of a guard interval and an effective symbol. Note that the guard interval is defined to have the same value as the rear part of the effective symbol. The receiving apparatus according to the embodiment of the present invention includes a plurality of antennas, but is not configured integrally, and is configured by a plurality of apparatuses. That is, the receiving device is configured by a plurality of RF processing devices and one baseband processing device, and each of the plurality of RF processing devices is connected to the baseband processing device via a cable.

  The RF processing apparatus is connected to an antenna, and performs frequency conversion from a radio frequency to a baseband frequency and also performs analog to digital AD conversion on a signal received by the antenna. The baseband processing device receives baseband and digital received signals from each of the plurality of RF processing devices, and performs adaptive array signal processing on them. In such a configuration, in order to efficiently perform AD conversion, the embodiment of the present invention next executes processing.

  The RF processing apparatus performs AD conversion after performing amplification by AGC. Here, since each RF processing apparatus performs an operation | movement independently, the gain in AGC is set independently for every RF processing apparatus. For this reason, amplification with an amplification factor suitable for each of a plurality of received signals is executed, so that efficient AD conversion is executed. When the baseband processing apparatus receives a plurality of received signals, the baseband processing apparatus estimates components corresponding to the noise included in each of the received signals. In order to perform the estimation, the baseband processor calculates the difference between the rear part of the effective symbol and the guard interval in each OFDM symbol. Since such a calculation can be performed over the entire packet signal, the estimation accuracy can be improved. Further, the baseband processing device corrects the received signal with a component corresponding to the estimated noise, and performs adaptive array signal processing on the corrected received signal.

  FIG. 1 shows a spectrum of a multicarrier signal according to an embodiment of the present invention. In particular, FIG. 1 shows the spectrum of a signal in the OFDM modulation scheme. One of a plurality of carriers in the OFDM modulation system is generally called a subcarrier, but here, one subcarrier is designated by a “subcarrier number”. For example, in a communication system compliant with the IEEE 802.11n standard (hereinafter referred to as “MIMO system”), 56 subcarriers from subcarrier numbers “−28” to “28” are defined. The subcarrier number “0” is set to null in order to reduce the influence of the DC component in the baseband signal. On the other hand, in a system that does not support the MIMO system (hereinafter referred to as “conventional system”), 52 subcarriers from subcarrier numbers “−26” to “26” are defined. An example of a conventional system is a wireless LAN compliant with the IEEE 802.11a / g standard.

  Further, one signal unit composed of a plurality of subcarriers and one signal unit in the time domain is referred to as an “OFDM symbol”. Each subcarrier is modulated by a variably set modulation scheme. Any one of BPSK (Binary Phase Shift Keying), QPSK, 16QAM (Quadrature Amplitude Modulation), and 64QAM is used as the modulation method. Here, among the plurality of subcarriers, pilot signals are arranged on four subcarriers of subcarrier numbers “−21”, “−7”, “7”, and “21”. In addition, the pilot signal arranged in one subcarrier has a pattern that has the same value every 4 OFDM symbols.

  FIGS. 2A to 2C show the format of a packet signal according to the embodiment of the present invention. Since the present invention is directed to the conventional system, FIG. 2A corresponds to the packet signal format of the conventional system. In the packet signal, “L-SIG” and “data” are arranged after “L-STF” and “L-LTF” as preamble signals or training signals. “L-STF”, “L-LTF”, and “L-SIG” respectively correspond to a known signal for AGC setting, a known signal for channel estimation, and a control signal corresponding to the conventional system. FIG. 2B shows the configuration of “L-LTF” in FIG. “L-LTF” is configured by “GI2”, “T1”, and “T2”.

  “T1” and “T2” have the same signal pattern, and each has a length of a signal output by one-time IFFT (Inverse Fast Fourier Transform). “GI2” is a guard interval, which is formed by duplicating the rear part of “T2”. FIG. 2C shows the configuration of “data” in FIG. “Data” is composed of a plurality of “OFDM symbols” and is illustrated as (i−1) OFDM symbol, iOFDM symbol, (i + 1) OFDM symbol. Each OFDM symbol is formed by “GI” and “effective symbol”, and “effective symbol” corresponds to a signal obtained by converting the spectrum of FIG. 1 into the time domain. The “GI” is defined in the same pattern as the rear part of the “effective symbol”, and is arranged in front of the “effective symbol”. Here, “GI” is defined by half the length of “GI2”, and “T1” and “T2” described above correspond to “effective symbols”.

  FIG. 3 shows a configuration of the communication system 100 according to the embodiment of the present invention. The communication system 100 includes a transmission device 10, a first RF processing device 12a, a second RF processing device 12b, an NRF processing device 12n, and a baseband processing device 34, which are collectively referred to as an RF processing device 12. Here, the RF processing device 12 and the baseband processing device 34 correspond to a receiving device. The transmission device 10 includes a baseband unit 26, a modulation unit 28, a radio unit 30, and a transmission antenna 16. The first RF processing device 12a includes a first receiving antenna 14a, the second RF processing device 12b includes a second receiving antenna 14b, and the NRF processing device 12n includes an Nth receiving antenna 14n.

  Here, the first receiving antenna 14a, the second receiving antenna 14b, and the Nth receiving antenna 14n are collectively referred to as the receiving antenna 14. The baseband processing device 34 includes a first correction unit 50a, a second correction unit 50b, an Nth correction unit 50n, a signal processing unit 18, a demodulation unit 20, an IF unit 22, and a control unit 24, which are collectively referred to as a correction unit 50. . In addition, as signals, the first amplified received signal 320a, the second amplified received signal 320b, the Nth amplified received signal 320n, which are collectively referred to as the amplified received signal 320, the first digital received signal 300a, which is collectively referred to as the digital received signal 300, 2 digital reception signal 300b, Nth digital reception signal 300n, and composite signal 304 are included.

  The transmission device 10 is connected to the reception device and executes communication with the reception device. The baseband unit 26 is an interface with a PC connected to the transmission device 10 and an application inside the transmission device 10, and performs transmission processing of an information signal to be transmitted in the communication system 100. Further, error correction and automatic retransmission processing may be performed, but the description thereof is omitted here. The modulation unit 28 generates a transmission signal by executing the above-described mapping to BPSK or the like, IFFT, or quadrature modulation.

  Here, the signal output from the modulation unit 28 in the transmission process forms a multicarrier signal. In addition, the multicarrier signal constitutes a packet signal. Such a multi-carrier signal is shown as in FIGS. 1 and 2 (a)-(c). The radio unit 30 executes frequency conversion processing. The radio unit 30 performs amplification processing, DA conversion processing, and the like. The radio unit 30 transmits a radio frequency signal to the radio apparatus via the transmission antenna 16.

  The RF processing device 12 is connected to the receiving antenna 14 at one end, and is connected to the baseband processing device 34 at the other end. Here, one RF processing device 12 is connected to one receiving antenna 14. That is, the plurality of RF processing devices 12 correspond to the plurality of reception antennas 14, respectively. The RF processing device 12 performs frequency conversion on a radio frequency multicarrier signal received by the receiving antenna 14 to derive a baseband signal. As described above, the multicarrier signal constitutes a packet signal, and the training signal is continuously included in the head portion of the packet signal. The multicarrier signal includes a pilot signal in a predetermined subcarrier, and the pilot signal is formed by repeating a predetermined pattern.

  Each of the plurality of RF processing devices 12 includes a local oscillator, and the RF processing device 12 frequency-converts each of the plurality of multicarrier signals by a local signal output from the local oscillator. In addition, each of the plurality of RF processing devices 12 performed amplification with a gain independent of the gain set in the other RF processing device 12 with respect to the signal received by the corresponding receiving antenna 14. Later, analog-to-digital conversion is performed. The RF processing device 12 outputs the amplified baseband signal to the baseband processing device 34 as an amplified received signal 320. In general, baseband signals are formed by in-phase and quadrature components, so they should be transmitted by two signal lines. Here, for clarity of illustration, only one signal line is used. Shall be shown.

  The baseband processing device 34 is connected to each of the plurality of RF processing devices 12 via a cable. The correction unit 50 is provided so as to correspond to each of the RF processing devices 12, and receives the amplified reception signal 320 from the RF processing device 12. In other words, the signal input at each of the correction units 50 is amplified with an amplification factor independent of the signals received at the other receiving antennas 14 with respect to the signals received at the corresponding receiving antennas 14. Later, analog-to-digital conversion was performed. Each of the plurality of correction units 50 estimates a component corresponding to the noise and performs correction using the component corresponding to the noise on the amplified reception signal 320. Details of the component estimation method corresponding to the noise and the correction method using the component corresponding to the noise in the correction unit 50 will be described later. Further, the correction unit 50 outputs the corrected signal as a digital reception signal 300.

  The signal processing unit 18 is connected to the plurality of correction units 50, and performs reception processing on the digital reception signals 300 from the plurality of correction units 50. Here, the signal processing unit 18 converts each of the plurality of digital reception signals 300 into the frequency domain, and performs adaptive array signal processing on the frequency domain signal. The signal processing unit 18 outputs the result of adaptive array signal processing as a synthesized signal 304. Here, it is assumed that composite signal 304, which is a frequency domain signal, includes a plurality of subcarrier components as shown in FIG. For the sake of clarity, it is assumed that the frequency domain signals are arranged in the order of subcarrier numbers to form a serial signal.

  FIG. 4 shows the configuration of a signal in the frequency domain. Here, one combination of subcarrier numbers “−28” to “28” shown in FIG. 1 is referred to as an “OFDM symbol”. Here, one signal unit in the frequency domain is also referred to as an “OFDM symbol”. In the “i” th OFDM symbol, subcarrier components are arranged in the order of subcarrier numbers “1” to “28” and subcarrier numbers “−28” to “−1”. Also, the “i−1” th OFDM symbol is arranged before the “i” th OFDM symbol, and the “i + 1” th OFDM symbol is arranged after the “i” th OFDM symbol. And In the conventional system, a combination of subcarrier numbers “−26” to “26” is used for one “OFDM symbol”. Returning to FIG.

  The reception process in the signal processing unit 18 will be described in more detail. The signal processing unit 18 derives reception weight vectors for the plurality of digital reception signals 300 over the training signal period of the packet signal. The reception weight vector is derived based on the transmission path characteristics. The transmission path characteristic indicates the amount of signal attenuation and the amount of phase rotation of the signal in the transmission path between the transmitting antenna 16 and the receiving antenna 14. Therefore, the transmission path characteristic has a number of components corresponding to the multiplication result of the number of combinations of the transmitting antenna 16 and the receiving antenna 14 and the number of subcarriers.

  For example, in the case of FIG. 3, since there is one transmitting antenna 16 and N receiving antennas 14, the transmission path characteristic for one subcarrier includes N components. Also, the reception weight vector derived from the transmission path characteristics has the same number of components as the transmission path characteristics. That is, the reception weight vector for one subcarrier has a component corresponding to each of the reception antennas 14. After the training signal period ends, the signal processing unit 18 performs array synthesis on the digital received signal 300 while using the received weight vector.

  The demodulator 20 performs demodulation and deinterleaving. Note that demodulation is performed in units of subcarriers. The demodulator 20 outputs the demodulated signal to the IF unit 22. The IF unit 22 is an interface with a network (not shown). The control unit 24 controls the timing of the baseband processing device 34 and the like.

  This configuration can be realized in terms of hardware by a CPU, memory, or other LSI of any computer, and in terms of software, it is realized by a program having a communication function loaded in the memory. Describes functional blocks realized by collaboration. Accordingly, those skilled in the art will understand that these functional blocks can be realized in various forms by hardware only, software only, or a combination thereof.

  FIG. 5 shows the configuration of the first RF processing apparatus 12a. The first RF processing device 12a includes a frequency conversion unit 146, an AGC 148, a quadrature detection unit 150, an AD conversion unit 152, and a local oscillation unit 166. Further, the other RF processing devices 12 are similarly configured.

  The frequency conversion unit 146 performs frequency conversion between the radio frequency and the intermediate frequency on the received signal. The AGC 148 automatically controls the gain so that the amplitude of the received signal becomes an amplitude within the dynamic range of the AD conversion unit 152. Since a known technique may be used as the configuration of the AGC 148, the description thereof is omitted here. However, the AGC 148 included in the first RF processing apparatus 12a operates independently of the AGC 148 included in another RF processing apparatus 12 (not shown). That is, the AGC 148 performs amplification at an amplification factor independent of the amplification factor set by the AGC 148 included in the other RF processing device 12.

  The quadrature detection unit 150 performs quadrature detection on the intermediate frequency signal to generate a baseband analog signal. The local oscillation unit 166 supplies a local signal having a predetermined frequency to the quadrature detection unit 150. The AD conversion unit 152 converts the baseband analog signal into a digital signal, and outputs the converted result as the first amplified received signal 320a.

  FIG. 6 shows the configuration of the AD conversion unit 152. The AD converter 152 includes a resistor R, a first comparator 80a, a second comparator 80b, a third comparator 80c, a fourth comparator 80d, a fifth comparator 80e, a sixth comparator 80f, a seventh comparator 80g, which are collectively referred to as a comparator 80, A ninth comparator 80h and an encoder 82 are included.

  R and R / 2 are connected in series as shown, and receive reference voltages “+ Vref” and “−Vref”. R and R / 2 generate a plurality of reference signals by changing the voltages of the reference voltages “+ Vref” and “−Vref”. The comparator 80 receives the signal from the quadrature detection unit 150 of FIG. 5 as “Vin” at one end and the reference signal from R at the other end. The comparator 80 outputs a comparison result between “Vin” and the reference signal to the encoder 82.

  The encoder 82 encodes the comparison result from the comparator 80 and generates a first amplified received signal 320a. If “Vin” is large as a whole, many bits are used, so that the accuracy of the digital signal becomes high. On the other hand, if “Vin” is small as a whole, only a few bits are used, and the accuracy of the digital signal is deteriorated. Here, a flash AD converter is shown as the AD conversion unit 152, but the present invention is not limited to this. For example, a sequential conversion AD converter or a pipeline AD converter may be used. Also good.

  FIG. 7 shows a configuration of the first correction unit 50a. The first correction unit 50 a includes an execution unit 52 and an estimation unit 54. The other correction units 50 are configured in the same manner.

  The first amplified received signal 320a is input to the first correction unit 50a. The first amplified received signal 320a is configured as shown in FIGS. 2A to 2C, and is a packet signal formed by a plurality of OFDM symbols. The OFDM symbol arranged in front of the packet signal has known values such as “L-STF” and “L-LTF”, and the subsequent OFDM symbols are “L-SIG” and “data”. It has an unknown value.

  The estimation unit 54 estimates the amount of the component corresponding to noise by deriving the difference between the card symbol interval and the portion of the effective symbol that should correspond to the guard interval with respect to the OFDM symbol. Here, the process of the estimation part 54 is demonstrated, using FIG. FIG. 8 shows an outline of processing in the estimation unit 54. FIG. 8 shows one OFDM symbol and corresponds to one of the plurality of OFDM symbols shown in FIG. Further, in FIG. 8, for convenience of explanation, predetermined timings are shown as “G1” to “G3” and “S1” to “S3”. “G1” is the GI start timing, and “G3” is the GI end timing. “S1” is the timing of the portion corresponding to the head of the GI in the effective symbol, and “S3” is the end timing of the effective symbol. It can be said that “S3” is the timing of the portion corresponding to the end of the GI among the effective symbols. “G2” and “S2” will be described later.

  That is, in the transmitting apparatus 10, the portion defined by the period from “S1” to “S3” (hereinafter referred to as “corresponding portion”) of the effective symbols is defined to correspond to the GI. The estimation unit 54 associates the corresponding part with the GI. Specifically, the correspondence is made so that the timing of the GI starting from “G1” is equal to the timing of the corresponding portion starting from “S1”. Further, the estimation unit 54 calculates the difference between the associated GI and the corresponding part. For example, the sum of squares of the difference between the two in-phase components and the quadrature component is calculated at a predetermined timing, and such calculation is executed over a corresponding period. Subsequently, the estimation unit 54 derives a difference in the OFDM symbol by integrating the calculation results. As described above, the associated GI and the corresponding part are defined as the same value on the transmission side. Therefore, the calculated difference corresponds to a component corresponding to noise.

  Moreover, the estimation part 54 may accumulate | store the difference in a some OFDM symbol. That is, the estimation unit 54 estimates the amount of the component corresponding to the noise during the training signal period. Furthermore, the estimation unit 54 updates the amount of the component corresponding to the noise during the data period. For example, the update is performed by adding the newly derived difference to the previously derived difference. At that time, division by a predetermined coefficient may be performed.

  In the packet signal, a plurality of OFDM symbols are continuously arranged as shown in FIG. Further, in the wireless transmission path from the transmission device 10 to the RF processing device 12 in FIG. As a result, the front part of the GI is affected by the preceding OFDM symbol, and intersymbol interference occurs in that part. Such intersymbol interference affects the calculation results described above. Therefore, the estimation unit 54 derives a difference between the rear part of the GI and the part that should correspond to the rear part of the GI among the corresponding parts. In FIG. 8, the rear part of the GI corresponds to a part defined by the period from “G2” to “S3”. “S2” is the timing of the portion corresponding to “G2” in the effective symbol, and the portion of the corresponding portion that should correspond to the rear portion of the GI is defined by the period from “S2” to “S3”. It corresponds to the part made. Here, in order to simplify the description, it is assumed that the timings of “G2” and “S2” are defined in advance. Since the calculation for deriving the difference is as described above, the description is omitted here. Returning to FIG.

  The execution unit 52 corrects the amplified received signal 320 with the noise derived by the estimation unit 54. The correction is made, for example, by dividing the amplified received signal 320 by the square root of the component corresponding to the noise.

  FIG. 9 shows the configuration of the signal processing unit 18. The signal processing unit 18 includes a first FFT unit 40 a, a second FFT unit 40 b, an NFFT unit 40 n, a synthesis unit 60, a reception weight vector calculation unit 68, and a reference signal storage unit 70 that are collectively referred to as the FFT unit 40. The synthesizing unit 60 includes a first multiplying unit 62a, a second multiplying unit 62b, an Nth multiplying unit 62n, and an adding unit 64, which are collectively referred to as a multiplying unit 62. The signal includes a reference signal 306, a first reception weight vector signal 312a, a second reception weight vector signal 312b, and an Nth reception weight vector signal 312n, collectively referred to as a reception weight vector signal 312.

  The FFT unit 40 performs FFT on the input digital reception signal 300. That is, the FFT unit 40 converts a time domain signal into a frequency domain signal. Here, the signal converted into the frequency domain is also referred to as a digital received signal 300. The digital received signal 300 converted to the frequency domain is configured as shown in FIG. Here, the digital reception signal 300 is a plurality of multicarrier signals corresponding to each of the plurality of reception antennas 14 and a plurality of multicarrier signals in which pilot signals are arranged on at least one subcarrier.

  The synthesizer 60 weights the digital reception signal 300 by the reception weight vector signal 312 in the reception antenna vector unit 312 and the subcarrier unit in the multiplication unit 62, adds the results by the addition unit 64, and adds the combined signal 304. Is output. Note that multiplication in one multiplication unit 62 is performed for each subcarrier, and the combined signal 304 is configured as shown in FIG. The reference signal storage unit 70 outputs a known training signal stored in advance during the training signal period as a reference signal 306.

  The reception weight vector calculation unit 68 derives transmission path characteristics from the digital reception signal 300 and the reference signal 306 over the training signal period. Since a known technique may be used for deriving the transmission path characteristics, description thereof is omitted here. As described above, the transmission path characteristics have components corresponding to each of the plurality of receiving antennas 14 and the plurality of subcarriers, and each component has an in-phase component and a quadrature component. The reception weight vector calculation unit 68 derives the reception weight vector signal 312 from the transmission path characteristics. Since a known technique may be used for deriving the reception weight vector signal 312, the description is omitted here. The reception weight vector signal 312 is formed by the same number of components as the transmission path characteristics.

  Operations of the RF processing device 12 and the baseband processing device 34 configured as described above will be described. Each RF processing device 12 performs amplification with a unique amplification factor in the AGC 148 on the received signal, and then performs analog-digital conversion in the AD conversion unit 152. Each RF processing device 12 outputs an amplified received signal 320 to the baseband processing device 34. Each correction unit 50 in the baseband processing device 34 estimates a component corresponding to the noise included in the amplified received signal 320 during the training signal period, and corrects the amplified received signal 320 with the component corresponding to the estimated noise. . The signal processing unit 18 performs adaptive array signal processing on the corrected amplified reception signal 320, that is, the digital reception signal 300.

  Hereinafter, modifications according to the present invention will be described. In the embodiment, as shown in FIG. 8, the start timings “G2” and “S2” for calculating the error are defined in advance. In order to reduce the influence of intersymbol interference, it is desirable that “G2” and “S2” are behind. That is, it is desirable that the period from “G1” to “G2” and the period from “S1” to “S2” are longer. On the other hand, in order to improve the averaging effect, it is desirable that “G2” and “S2” are in front. That is, it is desirable that the period from “G2” to “G3” and the period from “S2” to “S3” are longer. Therefore, the timings of “G2” and “S2” should be defined in consideration of the influence of intersymbol interference and the effect of averaging. In the wireless transmission path, the length of the period in which intersymbol interference occurs varies. Therefore, in order to improve the estimation accuracy of the component corresponding to noise, the timings of “G2” and “S2” should be adjusted according to the wireless transmission path.

  The communication system 100 according to the modification is the same type as that in FIG. 3, and the first correction unit 50a is the same type as that in FIG. The estimation unit 54 derives the magnitude of the delay component of the amplified received signal 320. The estimation unit 54 includes a matched filter, and stores an “L-LTF” pattern in the matched filter in advance. Further, the estimation unit 54 derives a delay component by performing a correlation process between “L-LTF” in the amplified received signal 320 and stored “L-LTF” in the matched filter. Further, the estimation unit 54 calculates a period from “G2” to “G3” and a period from “S2” to “S3” to be used for deriving the difference based on the size of the derived delay component. Adjust. For example, referring to a table in which the amount of the delay component and the length of the period are associated with each other, the estimation unit 54 determines that the period is shortened if the delay component is long. The period is determined to be longer. In addition, in the estimation unit 54 included in each of the plurality of correction units 50, an independent period may be determined, or a common period may be determined. In the latter case, for example, the shortest period among the plurality of periods is selected.

  Hereinafter, another modification will be described. So far, processing applicable to any part of the format of the packet signal shown in FIG. 2A has been described. On the other hand, as shown in FIG. 2B, in the L-LTF, known signals are continuously arranged. In another modification, known signals arranged in succession are used to improve the estimation accuracy of the component corresponding to noise. The communication system 100 according to another modified example is the same type as that in FIG. 3, and the first correction unit 50a is the same type as that in FIG.

  In the L-LTF among the amplified reception signals 320 input to the estimation unit 54, as shown in FIG. 2B, a plurality of effective symbols, that is, two effective symbols “T1” and “T2” are placed after “GI2”. "Is included in succession. The estimation unit 54 derives the difference between the GI2 and the effective symbol in the “L-LTF” period, similarly to the processing so far. The estimation unit 54 also derives a difference between “T1” and “T2”. Furthermore, the estimation part 54 estimates the quantity of the component corresponding to noise by adding the absolute value of both difference.

  FIG. 10 shows an outline of processing in the estimation unit 54 according to another modification of the present invention. FIG. 10 shows the configuration of “L-LTF” as in FIG. In FIG. 10, timing “a1” is the start timing of GI2, timing “a2” is the start timing of T1, and timing “a3” is the end timing of T1. The timing “b1” is the timing of the portion of T1 corresponding to “a1” of GI2, the timing “b2” is the start timing of T2, and the timing “b3” is the end timing of T2. is there. Here, the difference between the portion from timing “a1” to “a2” and the portion from timing “b1” to “b2” is derived in the same manner as in the previous embodiment. Further, the difference between the portion from timing “a2” to “a3” and the portion from timing “b2” to “b3”, that is, the difference between T1 and T2 is also derived. Furthermore, the difference in L-LTF is derived | led-out by adding the absolute value of both difference. By taking the latter into account, the effect of averaging increases, and the estimation accuracy of the component corresponding to noise is improved.

  Hereinafter, still another modification will be described. So far, the communication system 100 is assumed to be a conventional system. However, in still another modification of the present invention, the communication system 100 is assumed to be a MIMO system, not a conventional system. Since the packet signal in the MIMO system is composed of a plurality of sequences, the transmission apparatus 10 includes a plurality of transmission antennas 16, a plurality of radio units 30, and a plurality of modulation units 28 in order to cope with it. The baseband processing device 34 includes a plurality of signal processing units 18 and a plurality of demodulation units 20. In such a configuration, the transmission device 10 and the baseband processing device 34 process a plurality of sequences in parallel. First, a packet signal when the MIMO system is applied will be described.

  FIGS. 11A to 11C show the format of a packet signal according to still another modification of the present invention. FIG. 11A corresponds to the case where the number of series is “4”, FIG. 11B corresponds to the case where the number of series is “3”, and FIG. This corresponds to the case where the number of is “2”. In FIG. 11A, it is assumed that data included in the four sequences is to be transmitted, and packet formats corresponding to the first to fourth sequences are shown in order from the top to the bottom.

  In the packet signal corresponding to the first stream, “L-STF”, “HT-LTF” and the like are arranged as a preamble signal or a training signal. “HT-SIG” corresponds to a control signal corresponding to the MIMO system. The control signal corresponding to the MIMO system includes, for example, information on the number of sequences and the destination of the data signal. “HT-STF” and “HT-LTF” correspond to a known signal for AGC setting and a known signal for channel estimation corresponding to the MIMO system. In the first stream, HT-LTFs are arranged in the order of “HT-LTF”, “−HT-LTF”, “HT-LFT”, and “−HT-LTF” from the top. Here, these are sequentially referred to as “first component”, “second component”, “third component”, and “fourth component” in all series. “Data 1” is a data signal. Note that L-LTF and HT-LTF are used not only for AGC setting but also for timing estimation.

  Also, in the packet signal corresponding to the second stream, “L-STF (−50 ns)”, “HT-LTF (−400 ns)” and the like are arranged as preamble signals. Further, in the packet signal corresponding to the third stream, “L-STF (−100 ns)”, “HT-LTF (−200 ns)”, and the like are arranged as preamble signals. In the packet signal corresponding to the fourth stream, “L-STF (−150 ns)”, “HT-LTF (−600 ns)”, and the like are arranged as preamble signals.

  Here, “−400 ns” or the like indicates a timing shift amount in CDD (Cyclic Delay Diversity). CDD is a process in which a waveform in the time domain is shifted backward by a shift amount in a predetermined period, and the waveform pushed out from the last part of the predetermined period is cyclically arranged at the head part of the predetermined period. That is, “L-STF (−50 ns)” is cyclically shifted with a delay amount of −50 ns with respect to “L-STF”. Note that L-STF and HT-STF are configured by repetition of a period of 800 ns, and other HT-LTFs and the like are configured by repetition of a period of 3.2 μs. Here, “data 1” to “data 4” are also CDDed, and the timing shift amount is the same value as the timing shift amount in the HT-LTF arranged in the preceding stage.

  In the part from “L-LTF” to “HT-SIG” and the like, “52” subcarriers are used as in the conventional system. Of the “52” subcarriers, “4” subcarriers correspond to pilot signals. On the other hand, “56” subcarriers are used in the subsequent parts such as “HT-LTF”.

  In FIG. 11A, the sign of “HT-LTF” is defined as follows. The codes are arranged in the order of “+”, “−”, “+”, “−” in order from the top of the first sequence, and the codes are “+”, “+”, “+” in order from the top of the second sequence. Arranged in the order of “+” and “+”, the codes are arranged in the order of “+”, “−”, “−” and “+” in order from the top of the third series, and from the top of the fourth series. In order, the codes are arranged in the order of “+”, “+”, “−”, and “−”. However, the code | symbol may be prescribed | regulated as follows. The codes are arranged in the order of “+”, “−”, “+”, “+” in order from the top of the first sequence, and the codes are “+”, “+”, “+” in order from the top of the second sequence. Arranged in the order of “−” and “+”, the codes are arranged in the order of “+”, “+”, “+”, “−” in order from the top of the third series, and from the top of the fourth series. In order, the symbols are arranged in the order of “−”, “+”, “+”, “+”. Even such a code corresponds to a combination of codes of predetermined components having an orthogonal relationship between sequences.

  FIG. 11B corresponds to the first to third series in FIG. FIG. 11C is similar to the first stream and the second stream in the packet format shown in FIG. Here, the arrangement of “HT-LTF” in FIG. 11C is different from the arrangement of “HT-LTF” in FIG. That is, the HT-LTF includes only the first component and the second component. In the first sequence, HT-LTFs are arranged in the order of “HT-LTF” and “HT-LTF” from the top, and in the second sequence, HT-LTFs are arranged from the top to “HT-LTF”, “− They are arranged in the order of “HT-LTF”. These can also be said to be orthogonal as described above.

  Since the configuration of the baseband processing device 34 according to another modification is the same type as that in FIG. 3, the description thereof is omitted here. FIG. 12 shows a configuration of a first correction unit 50a according to still another modification of the present invention. The first correction unit 50 a includes an execution unit 52, an estimation unit 54, a separation unit 56, and an average unit 58. The other correction units 50 are configured in the same manner.

  The first amplified received signal 320a is input to the first correction unit 50a, and the first amplified received signal 320a is composed of a plurality of sequences. The separation unit 56 separates the first amplified received signal 320a into a sequence unit signal. The separation process in the separation unit 56 is executed as follows. The correction unit 50 extracts the first series of components by performing the calculation of the first component−the second component + the third component−the fourth component with respect to the reception signal in FIG. Further, the correction unit 50 extracts the second series of components by performing the calculation of the first component + the second component + the third component + the fourth component with respect to the received signal. Further, the correction unit 50 extracts the third series of components by calculating the first component−the second component−the third component + the fourth component with respect to the received signal. Further, the correction unit 50 extracts the fourth series of components by performing the calculation of the first component + the second component−the third component−the fourth component with respect to the received signal. These correspond to the combination of codes of predetermined components having an orthogonal relationship between sequences. The addition / subtraction process is executed by vector calculation.

  Moreover, the correction | amendment part 50 extracts the component of a 1st series by calculating the 1st component + 2nd component with respect to the received signal of FIG.11 (c). Moreover, the correction | amendment part 50 extracts the component of a 2nd series by calculating a 1st component-a 2nd component with respect to a received signal. As described above, the separating unit 56 outputs each extracted component to the averaging unit 58. The average unit 58 calculates the average value of each received component and outputs the average value to the estimation unit 54.

  FIG. 13 shows a configuration of a signal processing unit 18 according to still another modification of the present invention. The signal processing unit 18 includes a first FFT unit 40a, a second FFT unit 40b, an NFFT unit 40n, which are collectively referred to as the FFT unit 40, a first synthesis unit 220a, a second synthesis unit 220b, which are collectively referred to as a synthesis unit 220, and an Mth synthesis. Section 220m, reception weight vector calculation section 222, and reference signal storage section 70.

  The signal processing unit 18 receives the digital reception signal 300 corresponding to each of the plurality of reception antennas 14 and formed by a plurality of series as shown in FIGS. input. The FFT unit 40 corresponds to the FFT unit 40 in FIG. The synthesizer 220 performs array synthesis. Further, M synthesis units 220 are provided in accordance with the series. One combining unit 220 performs weighting on the digital reception signal 300 in units of the receiving antenna 14 and subcarriers, and performs combining in units of subcarriers. The weighting and composition processing in one composition unit 220 is performed in the same manner as the composition unit 60 in FIG.

  The reception weight vector calculation unit 222 generates a reception weight vector used in the weighting in the synthesis unit 220. Since M combining sections 220 correspond to M sequences, reception weight vector calculation section 222 derives reception weight vectors respectively corresponding to M sequences. Further, the reception weight vector corresponding to one combining unit 220 has components corresponding to the number of reception antennas 14 and the number of subcarriers, similarly to the reception weight vector signal 312 described so far.

  According to the embodiment of the present invention, the portion of the effective symbol that should correspond to the GI and the GI are defined so as to have the same pattern, and the difference between the two is derived. The accuracy of estimating the amount of can be improved. Further, since the rear part of the GI is used when the difference is derived, the influence of intersymbol interference can be reduced, and the estimation accuracy of the amount of the component corresponding to noise can be improved. Further, since the length of the rear part of the GI to be used for deriving the difference is adjusted according to the magnitude of the delay component, the estimation accuracy can be improved according to the radio transmission path environment. In addition, since estimation is performed not only in the training signal period but also in the data period, the estimation accuracy of the amount of the component corresponding to noise can be improved. Further, since the difference between the GI and the effective symbol is derived and the difference between the plurality of effective symbols is also derived, the estimation accuracy of the amount of the component corresponding to the noise can be improved.

  In addition, since the analog-to-digital converted signal is amplified by an independent amplification factor and corrected by a component corresponding to noise, the efficiency is improved even when a plurality of signals are processed. Can often perform analog-to-digital conversion. In addition, since the signal that has been amplified by an independent amplification factor and then subjected to analog-to-digital conversion is corrected by a component corresponding to noise, even if multiple signals are processed, Can be used.

  Further, even when an RF processing apparatus in which an independent AGC has been incorporated is used, the same performance as when an AGC system that sets a common amplification factor is used can be realized. In addition, since a plurality of sequences are separated, a component corresponding to noise can be estimated even when a signal formed by a plurality of sequences is a processing target. In addition, since the components corresponding to the noise between sequences are averaged, the estimation accuracy of the component corresponding to the noise can be improved.

  In addition, the effective bits of the AD converter can be used to the maximum. Further, when the AGC is commonly controlled, the bit width of the AD conversion unit is not effectively used. However, according to the present embodiment, the effective bits of the AD conversion unit can be used to the maximum extent. In addition, even when AGC is controlled in common, even if the same current is set to VGA, if the amplification factor is slightly different due to the difference in characteristics of LNA, or if the temperature of each RF processing device is different, the amount of noise that appears in the first place Differently, noise of different magnitudes may be added depending on the characteristics of the analog circuit depending on the characteristics of the analog circuit. However, according to this embodiment, the components corresponding to the noise included in each system are corrected in the digital domain. The effect of maximum ratio synthesis and array synthesis can be increased.

  In the above, this invention was demonstrated based on the Example. This embodiment is an exemplification, and it will be understood by those skilled in the art that various modifications can be made to the combination of each component and each processing process, and such modifications are also within the scope of the present invention. .

  In the embodiment of the present invention, the receiving device includes a plurality of RF processing devices 12 and a baseband processing device 34. However, the present invention is not limited to this. For example, the receiving device may be integrally configured. At that time, a plurality of AGCs corresponding to each of the plurality of receiving antennas 14, a plurality of AD conversion units corresponding to each of the plurality of AGCs, a plurality of correction units 50 corresponding to each of the plurality of AD conversion units, A signal processing unit 18 connected to the correction unit 50 is included. Each of the plurality of AGCs is an OFDM symbol received at a corresponding antenna, and is formed by a guard interval defined in the same pattern as the rear part of the effective symbol, and an effective symbol arranged at the subsequent stage of the guard interval. For the OFDM symbol, amplification is performed with an amplification factor independent of the amplification factors set by other AGCs. Each of the plurality of AD conversion units performs analog-digital conversion on the signal amplified in the corresponding AGC. Each of the plurality of correction units 50 copes with noise by deriving a difference between a card interval and a portion of an effective symbol that should correspond to a guard interval with respect to an OFDM symbol converted by a corresponding conversion unit. The amount of the corrected component is estimated, and correction by the component corresponding to the noise is executed. The signal processing unit 18 performs reception processing on the OFDM symbols corrected in each of the plurality of correction units 50. According to this modification, the signal is subjected to analog-to-digital conversion after being amplified with an independent amplification factor, and correction is performed using a component corresponding to noise. However, the analog-digital conversion can be executed efficiently.

  In the embodiment of the present invention, the communication system 100 is applied to the communication system 100 based on CSMA. However, the present invention is not limited thereto, and for example, the receiving apparatus may be applied to a communication system other than CSMA. For example, TDI (Time Division Multiple Access), CDMA (Code Division Multiple Access), SDMA (Space Division Multiple Access), etc. are used. May be. According to this modification, the present invention can be applied to various communication systems. That is, any receiving device that receives a signal from the transmitting device 10 may be used.

  In the embodiment of the present invention, the estimation unit 54 is described as being included in the correction unit 50. However, the present invention is not limited to this. For example, the estimation unit 54 may not be included in the correction unit 50. In that case, the estimation part 54 is comprised as an apparatus for measuring the component corresponding to noise. According to this modification, the present invention can be applied to various devices.

It is a figure which shows the spectrum of the multicarrier signal which concerns on the Example of this invention. FIGS. 2A to 2C are diagrams illustrating a format of a packet signal according to the embodiment of the present invention. It is a figure which shows the structure of the communication system which concerns on the Example of this invention. It is a figure which shows the structure of the signal of the frequency domain in FIG. It is a figure which shows the structure of the 1st RF processing apparatus of FIG. It is a figure which shows the structure of the AD conversion part of FIG. It is a figure which shows the structure of the 1st correction | amendment part of FIG. It is a figure which shows the outline | summary of the process in the estimation part of FIG. It is a figure which shows the structure of the signal processing part of FIG. It is a figure which shows the outline | summary of the process in the estimation part which concerns on another modification of this invention. FIGS. 11A to 11C are diagrams showing a format of a packet signal according to still another modified example of the present invention. It is a figure which shows the structure of the 1st correction | amendment part which concerns on another modification of this invention. It is a figure which shows the structure of the signal processing part which concerns on another modification of this invention.

Explanation of symbols

  12 RF processing device, 14 receiving antenna, 18 signal processing unit, 20 demodulation unit, 22 IF unit, 24 control unit, 34 baseband processing device, 100 communication system.

Claims (10)

An input unit for inputting an OFDM symbol formed by a guard interval defined in the same pattern as the rear part of the effective symbol, and an effective symbol arranged at the subsequent stage of the guard interval;
An estimation unit that estimates the amount of a component corresponding to noise by deriving a difference between a card interval and a portion that should correspond to a guard interval of effective symbols with respect to the OFDM symbol input to the input unit; ,
An estimation apparatus comprising: The input unit continuously inputs a plurality of OFDM symbols,
The estimation apparatus according to claim 1, wherein the estimation unit derives a difference between a rear part of the guard interval and a part of the effective symbol that should correspond to the rear part of the guard interval. A derivation unit for deriving the magnitude of the delay component of the OFDM symbol input to the input unit;
The said estimation part adjusts the length of the rear part of the guard interval which should be used in order to derive a difference based on the magnitude | size of the delay component derived | led-out in the said derivation | leading-out part. Estimating device. The plurality of OFDM symbols input to the input unit constitute a packet signal, the OFDM symbol arranged in front of the packet signal has a known value, and the OFDM symbol following the OFDM symbol of the known value is unknown. Has the value of
The estimation unit estimates an amount of a component corresponding to noise in an OFDM symbol period having a known value, and updates an amount of a component corresponding to noise in an OFDM symbol period having an unknown value. The estimation apparatus according to any one of claims 1 to 3, wherein Among the packet signals input to the input unit, the OFDM symbol having a known value includes a plurality of effective symbols continuously,
The estimation unit derives the difference between the guard interval and the effective symbol in the period of the OFDM symbol having a known value, and also derives the difference between the plurality of effective symbols, so that the amount of the component corresponding to the noise The estimation apparatus according to claim 4, wherein: A plurality of input units corresponding to each of a plurality of antennas;
A plurality of correction units corresponding to each of the plurality of input units;
A reception processing unit connected to the plurality of correction units,
A signal input at each of the plurality of input units includes an OFDM symbol formed by a guard interval defined in the same pattern as the rear part of the effective symbol and an effective symbol arranged at the subsequent stage of the guard interval. In each of the input OFDM symbols, the OFDM symbol received at the corresponding antenna is amplified by an amplification factor independent of the OFDM symbol received at the other antenna, and then analog-digital Conversion is done,
Each of the plurality of correction units copes with noise by deriving a difference between a card interval and a portion of a valid symbol that should correspond to a guard interval with respect to an OFDM symbol input at a corresponding input unit. The amount of the corrected component is estimated, and the correction by the component corresponding to the noise is executed.
The reception apparatus, wherein the reception processing unit performs reception processing on the OFDM symbol corrected in each of the plurality of correction units. The OFDM symbol input at each of the plurality of input units is composed of a plurality of sequences,
The receiving apparatus according to claim 6, wherein each of the plurality of correction units estimates an amount of a component corresponding to noise after being separated into OFDM symbols in units of sequences. A plurality of amplifiers corresponding to each of a plurality of antennas;
A plurality of conversion units corresponding to each of the plurality of amplification units;
A plurality of correction units corresponding to each of the plurality of conversion units;
A reception processing unit connected to the plurality of correction units,
Each of the plurality of amplifying units is an OFDM symbol received at a corresponding antenna and defined in the same pattern as the rear part of the effective symbol, and an effective symbol arranged in the subsequent stage of the guard interval, For the OFDM symbol formed by the above, the amplification by the amplification factor independent of the amplification factor set in the other amplification unit is executed,
Each of the plurality of conversion units performs analog-to-digital conversion on the OFDM symbol amplified in the corresponding amplification unit,
Each of the plurality of correction units copes with noise by deriving a difference between a card interval and a portion of the effective symbol that should correspond to the guard interval with respect to the OFDM symbol converted by the corresponding conversion unit. The amount of the corrected component is estimated, and the correction by the component corresponding to the noise is executed.
The reception apparatus, wherein the reception processing unit performs reception processing on the OFDM symbol corrected in each of the plurality of correction units. A plurality of RF processing devices corresponding to each of a plurality of antennas;
A baseband processing device connected to each of the plurality of RF processing devices via a cable;
Each of the plurality of RF processing devices is an OFDM symbol received at a corresponding antenna and defined by the same pattern as the rear part of the effective symbol, and an effective symbol arranged at the subsequent stage of the guard interval After performing the amplification with the amplification factor independent of the amplification factor set by the other RF processing apparatus, the analog-digital conversion is performed on the OFDM symbol formed by
The baseband processing device derives a difference between a card interval and a portion of an effective symbol that should correspond to a guard interval with respect to an OFDM symbol converted in each of the plurality of RF processing devices. A reception system that estimates the amount of a component corresponding to the signal and performs a correction process using the component corresponding to noise, and then performs a reception process on the corrected signal.   An OFDM symbol formed by a guard interval defined in the same pattern as the rear part of the effective symbol and an effective symbol arranged at a subsequent stage of the guard interval is input. An estimation method characterized in that the amount of a component corresponding to noise is estimated by deriving a difference between a card interval and a portion that should correspond to the guard interval.

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