JP2008076290A - Radar device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a radar device capable of expanding the dynamic range of a self-receiver against jamming other than a desired wave. <P>SOLUTION: When a desired wave or a reflected wave is received that is reflection of thetransmitted wave emitted at a self device 10 from a target, timing of jamming wave phase data detected at a signal processing section 20 is adjusted through a phase regulating apparatus 28 in the signal processing section 20 and moreover two phase regulators of the phases of 0-π and π/2-3π/2 is switched at an I/Q signal changeover switch 42 and an orthogonal transformer 48, and if a plurality of radar beams exists, a level of the maximum jamming wave emitted at other surrounding radar devices is suppressed to expand dynamic range of self-receiver against jamming other than a desired wave. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to a radar apparatus, for example, a radar apparatus that expands a dynamic range of a receiving system for an interference wave other than a desired wave in a vehicle-mounted radar.

  In recent years, for example, as described in Patent Document 1, an in-vehicle radar system mounted on a moving body such as an automobile has been developed. In such a radar system reception system, the reception dynamic range of the front end including the reception antenna to the analog-digital converter affects the performance of the radar system such as distance measurement and relative speed detection.

An in-vehicle radar traveling on a road needs to receive a desired wave (a reflected wave of its own device transmission wave) from a distance measuring target vehicle in a radio wave environment in which nearby interference waves existed. Here, assuming that the maximum reception level of the jamming wave is Pu_max, the minimum reception level of the desired wave is Pd_min, the reception dynamic range of the front end unit is Ga, and under the condition that satisfies the following equation (1), the required CNR of the reception system There is a problem that, for example, distance measurement becomes impossible even though (Carrier to Noise Ratio) is within an allowable range.
Pu_max−Pd_min> = Ga (1)
Japanese Patent Laid-Open No. 2002-22826

  In order to solve such problems, it is necessary to increase the dynamic range by increasing the linearity of the amplifiers at each stage of the front end unit or increasing the number of bits of the analog / digital conversion unit. Thus, there is a problem that the cost is increased and the power consumption of the radar apparatus is increased.

  It is an object of the present invention to provide a radar apparatus that can eliminate the drawbacks of the prior art and expand the dynamic range of the receiver for interference waves other than the desired wave.

  In order to solve the above-described problems, the present invention provides a radar apparatus using a two-phase phase modulation method, in which the ratio between the level of a received interference wave and the level of a desired reception wave is a predetermined radio wave environment. Switching means for switching between the first phase angle in the two-phase phase modulation of the disturbing wave and the second phase angle obtained by shifting the first phase angle by 90 degrees, and the first phase Receiving means for receiving a two-phase phase modulated wave by the angle and a two-phase phase modulated wave by the second phase angle.

  According to the present invention, the first and second phase angles are switched between the first phase angle in the two-phase phase modulation of the interference wave and the second phase angle obtained by shifting the first phase angle by 90 degrees. When the ratio of the level of the disturbing wave to the level of the desired received wave is a value indicating a predetermined radio wave environment by receiving the two-phase modulated wave by the receiver, the receiver for the disturbing wave other than the desired wave The dynamic range of the radar apparatus can be expanded, and the increase in the scale and cost of the radar apparatus can be suppressed.

  Next, an embodiment of a radar apparatus according to the present invention will be described in detail with reference to the accompanying drawings. Referring to FIG. 1, an embodiment of a radar apparatus to which the present invention is applied is shown. The radar apparatus is an on-vehicle radar apparatus mounted on a moving body such as an automobile. In the following description, portions that are not directly related to the present invention are not shown and described, and reference numerals of signals are represented by reference numerals of connection lines that appear.

  As shown in the figure, the radar apparatus 10 includes a signal processing unit 20 that inputs the transmission data 12 and the synchronization signal pattern signal 14, adjusts the phase, outputs the transmission data to the output 16, and generates the IQ switching signal 18. Yes. The signal processing unit 20 includes a correlation detector 26 that detects a correlation of received data and generates a desired wave radar analysis signal 22 and an interference wave phase synchronization signal 24. The signal processing unit 20 includes a phase adjuster 28 that adjusts the phase of the transmission data 16 based on the generated interference wave phase synchronization signal 24. The output 16 of the phase adjuster 28 forms the output of the signal processor 20 and is connected to a digital / analog converter (DAC) 30. The DAC 30 converts the transmission data into an analog transmission signal and outputs it to the output 32. . An output 32 of the DAC 30 is connected to a low pass filter 34.

  The low-pass filter 34 is a filter that passes a low band of the transmission signal converted into an analog signal by the DAC 30, and an output 36 thereof is connected to the baseband amplifier 38. The baseband amplifier 38 amplifies the input transmission signal and outputs the amplified transmission signal to the output 40. The output 40 is connected to the I / Q changeover switch 42.

  The I / Q switch 42 switches the in-phase (I) and quadrature (Q) phases according to the IQ switch signal 18 supplied from the signal processing unit 20, outputs the in-phase signal to the output 44, and outputs the quadrature signal. Output to output 46. Outputs 44 and 46 of the I / Q changeover switch 42 are connected to an I / Q quadrature modulator 48. The I / Q quadrature modulator 48 phase angle modulates the IQ signal by the input carrier frequency signal 50. The I / Q quadrature modulator 48 has a function of phase angle modulating the transmission signal with the phase 0-π of the first priority and a function of phase angle modulating with the phase π / 2-3π / 2 of the second priority. Have.

  The I / Q quadrature modulator 48 outputs the modulated transmission signal to the output 52, and this output 52 is connected to the intermediate frequency band-pass filter 54. An output 56 of the intermediate frequency band bandpass filter 54 is connected to an upconversion mixer 58. The up-conversion mixer 58 converts the frequency of the input signal by the carrier frequency signal 60 and outputs it to the output 62. The output 62 is connected to the transmission bandpass filter 66 through the amplifier 64, and the output of the transmission bandpass filter 66 is connected to the transmission antenna 68. These low-pass filter 34, baseband amplifier 38, I / Q changeover switch 42, I / Q quadrature modulator 48, intermediate frequency band bandpass filter 54, up-conversion mixer 58, amplifier 64, filter 66 and transmission antenna 68 are 2 A front end portion on the transmission side for transmitting the phase-phase modulated wave is formed.

  Next, the reception-side front end unit that receives the two-phase modulated wave will be described. The reception antenna 80 on the reception side is connected to the low-noise amplifier 84 via the reception bandpass filter 82, and the low-noise amplifier 84 is The amplified received signal is output to the output 86. The output 86 of the low noise amplifier 84 is connected to the down conversion mixer 88.

  The down-conversion mixer 88 converts the frequency of the reception signal 86 by the carrier signal 92 supplied from the first local unit (1st-LO) 90 and outputs the converted reception signal to the output 94. The output 94 is connected to the reception band pass filter 96. An output 98 of the reception bandpass filter 96 is connected to the I / Q quadrature demodulator 100. The first local unit 90 generates a carrier wave signal 60, and its output 60 is connected to the above-described up-conversion mixer 58.

  The I / Q quadrature demodulator 100 demodulates the received signal 98 based on the carrier signal 104 supplied from the second local unit (2nd-LO) 102, and outputs the demodulated in-phase and quadrature phase received signals to the outputs 110 and 112, respectively. Output. Outputs 110 and 112 are connected to baseband amplifiers 114 and 116, respectively. Baseband amplifiers 114 and 116 amplify the received signals and output to outputs 118 and 120, respectively. Outputs 118 and 120 are connected to low-pass filters 122 and 124, respectively, and the low-pass received signal is connected to analog / digital converters (ADCs) 130 and 132 via outputs 126 and 128, respectively.

  Each of ADCs 130 and 132 converts an input received signal into a digital signal and outputs the converted received data to outputs 134 and 136, respectively. These outputs 134 and 136 are connected to the correlation detector 26 in the signal processing unit 20, respectively.

  The output 98 of the reception bandpass filter 96 is connected to a received electric field strength signal (RSSI) detector 140. The RSSI detector 140 inputs the received signal 98 and outputs a received electric field strength signal 142 corresponding to the electric field strength. Output.

  With the above configuration, the interference wave phase data detected by the signal processing unit 20 is adjusted in data timing by the phase adjuster 28 in the signal processing unit 20, and the I / Q signal changeover switch 42 and the orthogonal transformer 48 are further adjusted. , Two phase modulations of phase 0-π and phase π / 2-3π / 2 are switched. When there are multiple radar waves, the maximum disturbing wave level emitted from other surrounding radar devices is suppressed when receiving the desired wave that is the reflected wave reflected from the target by the transmitted wave emitted from the own device 10 Then, the dynamic range of the receiver for the interference wave other than the desired wave is expanded, and the distance to the target and the relative speed are determined.

  Next, the operation of the radar apparatus 20 in the present embodiment will be described with reference to the flowchart shown in FIG. 2 with reference to FIG. When ranging is started, in step S200, the front end units on the transmission side and reception side of the radar apparatus 20 are set to the first priority phase modulation mode. Here, the phase 0-π mode is set temporarily. The radio wave environment where Equation (1) does not hold is when the desired wave has a sufficiently higher reception level than the interference wave.In this case, the receiving side's radar processing unit uses the desired wave to determine the distance to the target vehicle, etc. Information can be obtained, and the radar wave reception operation is performed in the first priority phase modulation mode.

  On the other hand, if there is no ranging detection signal of the desired wave in the processing unit on the receiving side in step S202, and the RSSI detection unit 140 detects the RSSI signal in step S204, Expression (1) is established. In this case, the reception environment is an environment where the interference wave level is high, and the interference wave is regarded as phase modulation (0-π), the data transmission timing of the transmission wave is adjusted in step S208, and then the process proceeds to step S210. Then, the transmission-side phase modulation of the apparatus 10 is set to the second priority phase modulation (π / 2-3π / 2) mode. At the same time, the receiving side similarly sets the phase demodulator 100 to the π / 2-3π / 2 phase modulation mode. Thereafter, the process proceeds to step S212, and distance measurement processing is executed.

  Examples of characteristic improvement according to the present embodiment are shown in FIGS. FIG. 3 shows the suppression level ratio of the interference wave in the quadrature demodulator 96 when the interference wave is input to the own apparatus 10. A characteristic curve 300 shows the interference wave level and the desired wave reception level, and a characteristic curve 302 shows the interference wave level after the interference wave suppression. Thus, it can be seen that the difference between the interference wave level before application of the present embodiment and the interference wave level after application is the suppression ratio 304, which is greatly improved. The simulation data was calculated under the conditions including the phase noise of the radio and the amplitude error and phase error of the quadrature demodulator 96 in consideration of the performance degradation amount in the analog circuit when the device 10 was mounted on the actual device. It is a value, and suppression of 25 [db] or more is possible. In this characteristic example, the phase noise at a frequency of 100 [kHz] is 80 [db], the amplitude error is 2 [db], and the phase error is 2 percent.

  FIG. 4 shows Eb / No versus bit error rate (BER) characteristic data obtained by improving the suppression ratio shown in FIG. As shown in the figure, a characteristic curve 400 shows a theoretical value of BPSK (Binary Phase Shift Keying), and a characteristic curve 402 is an example of the characteristic according to this embodiment. A case where the present embodiment is not applied is indicated by a characteristic curve 404. Thus, it is clear from the graph that the BER characteristics are improved by this embodiment. As a result, an increase in the scale and cost of the radar apparatus 10 can be suppressed.

It is a block diagram which shows the Example of the radar apparatus to which this invention was applied. It is a flowchart which shows operation | movement of the radar apparatus in the Example shown in FIG. It is a graph which shows the example of an effect of the interference wave suppression ratio by a radar apparatus. It is a graph which shows the example of improvement of the bit error rate by a radar apparatus.

Explanation of symbols

10 Radar equipment
34 Low-pass filter
38 Baseband amplifier
42 I / Q selector switch
48 I / Q quadrature modulator
54 Intermediate frequency bandpass filter
58 Upconversion mixer
64 amplifier
66 Filter
68 Transmit antenna
80 Receive antenna
82 Receive bandpass filter
84 Low noise amplifier
88 Downconversion mixer
90 1st local part (1st-LO)
96 Receive bandpass filter
100 I / Q quadrature demodulator
102 2nd local part (2nd-LO)
114, 116 Baseband amplifier
122, 124 Low-pass filter

Claims (2)

In a radar apparatus using a two-phase phase modulation method, the apparatus includes:
When the ratio between the level of the received jamming wave and the desired level of the received wave is a value indicating a predetermined radio wave environment, the first phase angle in the two-phase phase modulation of the jamming wave and the first Switching means for switching the second phase angle obtained by shifting the phase angle of the second phase angle by 90 degrees;
A radar apparatus comprising: a receiving unit configured to receive a two-phase phase-modulated wave based on the first phase angle and a two-phase phase-modulated wave based on the second phase angle.   2. The apparatus according to claim 1, wherein the predetermined radio wave environment is a radio wave environment in which a ratio between the level of the interference wave and the level of the desired reception wave exceeds a reception dynamic range of the apparatus. Radar device.

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