JP2008067582A - Motor controller, motor control method and actuator - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a motor controller which can control a motor to generate a larger torque even during saturation, and to provide a motor control method. <P>SOLUTION: The motor controller 20 for controlling the drive current of a motor by determining the current values id and iq of d and q phases through dq conversion for converting into the dq rectangular coordinates and performing current loop processing of d and q phases is provided with a means 25 for controlling the vector angle θv of the composition vector of voltage command values Vd and Vq of d and q phases for the q axis. Consequently, a larger torque can be generated from the motor even during saturation. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to a motor control device, a motor control method, and an actuator.

  As this type of motor control device, the d-axis in the direction of magnetic flux created by the magnetic field from the current value of any two-phase current flowing in the three-phase winding and the electrical angle of the rotor and the q-axis orthogonal to the d-axis Convert to dq coordinates, calculate d-phase and q-phase current values, set d-phase current target value to 0, calculate q-phase current target value, perform current loop processing for each phase, and perform d-phase voltage The command value and the q-phase voltage command are calculated, the d-phase voltage command value and the q-phase voltage command value are converted into a three-phase voltage command value, and the motor is PWM (Pulse) based on the three-phase voltage command value. What controls (Width Modulation) is known (for example, refer patent document 1).

  The process in the current loop takes a deviation between the d-phase and q-phase current values, the d-phase current target value, and the q-phase current target value, and based on this deviation, the d-phase voltage based on the proportional-integral control law. When the command value and the q-phase voltage command value are calculated and the combined vector of the d-phase voltage command value and the q-phase voltage command value reaches the saturation voltage, that is, the d-phase voltage command value and the q-phase voltage command value When the combined vector is saturated, a voltage higher than the power supply voltage cannot be applied between the terminals of each winding, so that the absolute value of the integral value in the current loop increases, and the motor is operated normally. It becomes a state that cannot be controlled.

Therefore, in the conventional motor control device, when the combined vector of the d-phase voltage command value and the q-phase voltage command value reaches the saturation voltage, the integral value is saturated in the current loop, and the motor is controlled. A stable torque is generated.
Japanese Patent Application Laid-Open No. 6-1553569

  However, in the conventional motor control device, when the combined vector of the d-phase voltage command value and the q-phase voltage command value reaches the saturation voltage, the angle of the combined vector of the d-phase voltage and the q-phase voltage with respect to the q axis is forcibly fixed. Therefore, a large torque cannot be generated by the motor at the time of saturation.

  In other words, assuming that the q-phase takes a positive value when the motor is rotated forward, the q-phase current may take the maximum value at the time of saturation, but in the state where the q-phase current does not take the maximum value. Since the angle of the combined vector of the d-phase voltage and the q-phase voltage with respect to the q-axis is kept constant, the q-phase current cannot be controlled to the maximum value.

  Further, the q-phase current during saturation changes depending on the angle of the combined vector of the d-phase voltage command value and the q-phase voltage command value with respect to the q-axis, and further changes depending on the electrical angular velocity.

  Therefore, the present invention was devised to improve the above-described problems, and the object of the present invention is to provide a motor control device and a motor control method capable of generating a large torque with a motor even when saturated. Furthermore, the present invention also provides an actuator capable of generating a larger torque by a motor and outputting a larger thrust even when saturated.

  In order to achieve the above-described object, the motor control device in the problem solving means of the present invention obtains current values of the d-phase and the q-phase by dq conversion for converting to dq orthogonal coordinates, and performs a current loop process for each dq phase. A motor control apparatus for controlling a motor drive current includes a vector angle control means for controlling a vector angle of a combined vector of voltage command values for each phase of dq.

  The actuator in the problem solving means of the present invention includes a motion conversion mechanism including a rotating member and a linear motion member that exhibits linear motion by the rotating motion of the rotating member, and a motor that is connected to the rotating member and applies torque to the rotating member. And the motor control device.

  Further, in the motor control method in the problem solving means of the present invention, the d-phase and q-phase current values are obtained by dq conversion to be converted into dq orthogonal coordinates, and a current loop process is performed for each dq phase to control the motor drive current. And a vector angle control step of controlling a vector angle of a combined vector of the voltage command values for each phase of dq.

  According to the motor control device of the present invention, it is possible to control the angle of the combined vector of the voltage command value of each phase of dq with respect to the q axis, and when the combined vector of the voltage command value of each phase of dq is saturated. However, it is possible to appropriately control the torque generated by the motor, and it is possible to generate a larger torque by the motor as compared with the conventional motor control device.

  In addition, since a larger torque can be generated by the motor, the torque control range is increased, and the motor mounted on various devices can be reduced in size.

  The present invention will be described below based on the embodiments shown in the drawings. FIG. 1 is a conceptual diagram of an actuator to which a motor control device according to an embodiment is applied. FIG. 2 is a diagram showing a change in the vector angle of the combined vector of the voltage command values of each phase of dq that maximizes the q-axis current with respect to the electrical angular velocity. FIG. 3A is a graph showing a change in the vector angle of the combined vector of the voltage command values for each phase of dq with respect to the stroke displacement of the actuator when the motor is controlled so that the d-phase current is zero and the integration is stopped when saturated. It is. FIG. 3B is a graph showing the change in the vector angle of the combined vector of the voltage command values of dq each phase with respect to the stroke displacement of the actuator when the motor is controlled by performing the field weakening control and the integration is stopped at the time of saturation. FIG. 4 is a system diagram of the motor control device according to the embodiment. FIG. 5 is a diagram showing a PWM circuit. FIG. 6A is a diagram showing a vector angle final value map created using the electrical angular velocity used as a parameter when the motor generates torque in the reverse rotation direction. FIG. 6B is a diagram showing a vector angle final value map created by using the electrical angular velocity used as a parameter when the motor generates torque in the forward rotation direction. FIG. 7 is a table for determining the rotation direction with respect to the q-axis of the combined vector of the voltage command values for each phase of dq. FIG. 8 is a diagram for explaining the rotation direction of the combined vector when the q-phase current target value is positive. FIG. 9 is a diagram illustrating the rotation direction of the combined vector in the dq orthogonal coordinates when the q-phase current target value is positive. FIG. 10 is a diagram illustrating the rotation direction of the combined vector when the q-phase current target value is negative. FIG. 11 is a diagram for explaining the rotation direction of the combined vector in the dq orthogonal coordinates when the q-phase current target value is negative. FIG. 12 is a diagram showing a map of limit values for limiting the q-phase current target value created using the electrical angular velocity as a parameter. FIG. 13 is a flowchart illustrating a processing procedure in the motor control device according to the embodiment. FIG. 14 is a system diagram of a motor control device according to another embodiment. FIG. 15 is a system diagram of the switch controller in the selection means. FIG. 16 is a flowchart illustrating a processing procedure in a motor control device according to another embodiment.

  As shown in FIG. 1, the actuator to which the motor control device in one embodiment is applied has a pinion gear 1 that is a rotating member and a rack shaft 2 that is a linear motion member, and the rotational movement of the pinion gear 1 is a rack shaft. The motion conversion mechanism H which converts into 2 linear motions, and the motor M which has the rotor R connected with the pinion gear 1 are comprised.

  Specifically, the pinion gear 1 is meshed with the rack teeth 2a of the rack shaft 2, and when the motor M is driven to rotationally drive the pinion gear 1, the rack shaft 2 can be linearly moved. .

  In the above description, the motion conversion mechanism H is a rack and pinion mechanism composed of a pinion gear and a rack shaft 2. However, in addition to this, a rotary member and a friction wheel are used as a shaft and a linear motion member is used as a shaft. Alternatively, a feed screw mechanism in which one of the ball screw nut and the screw shaft is a rotating member and the other is a linear motion member may be used. Further, a reduction gear constituted by a gear mechanism or the like may be interposed between the pinion gear 1 and the rotor R, and the rotational motion of the rotor R may be reduced and transmitted to the pinion gear 1.

  The actuator rotates the pinion gear 1 by the motor M as described above, and further moves the rack shaft 2 directly by the rotation of the pinion gear 1, and functions as a linear actuator. When the pinion gear 1 and the rack shaft 2 exhibit a linear relative motion in the axial direction, the pinion gear 1 that is a rotating member exhibits a rotational motion, and the rotational motion of the pinion gear 1 is transmitted to the rotor R of the motor M. Therefore, the rack shaft 2 can be braked by suppressing the rotational movement of the pinion gear 1 with the torque generated by the motor M.

  Subsequently, the motor M includes a cylindrical frame 10, a stator S that is an armature provided on the inner peripheral side of the frame 10, and a rotor R that is rotatably supported by the frame 10. Specifically, the stator S includes a ring-shaped stator core 11 having a plurality of teeth, and windings 12 in U, V, and W phases wound around the teeth. The other rotor R includes a shaft 13 connected to one end of the screw shaft 1 and a driving magnet 14 attached to the outer periphery of the intermediate portion of the shaft 13.

  The drive magnet 14 is formed into a block so that a predetermined number of poles can be realized and bonded to the outer periphery of the shaft 13 or formed in an annular shape and divided and magnetized so as to have an outer periphery of the shaft 13. To be fitted.

  The motor M is equipped with a rotation angle sensor 15 for detecting the rotation angle (electrical angle) θ of the rotor R. Specifically, for example, the rotation angle sensor 15 is provided on the shaft 13. The resolver core and the resolver stator provided on the frame 10 are opposed to the resolver core, and the electrical angular velocity ω can be obtained from the electrical angle θ. In addition, as long as a calculation unit for calculating the electrical angular velocity ω from the electrical angle θ is separately provided, an optical encoder may be employed, or in the case where a sensing magnet is provided in the rotor R, a Hall element is used. A magnetic sensor such as an MR element or the like may be provided on the frame 10.

On the other hand, the motor control device 20 basically performs proportional-integral control of the motor M. Specifically, the motor control device 20 performs two-phase conversion for converting the current flowing through the three-phase windings U, V, and W of the motor M into the current of each of the dq coordinates in the dq coordinates in the proportional-integral control. The deviations εd and εq between the current values id and iq of each phase of dq and the current target values id ^* and iq ^{* of} each phase of dq are obtained, and the proportional gain KP is multiplied by the deviations εd and εq for each phase of dq. The value and the value obtained by integrating the deviations εd and εq and multiplied by the integral gain KI are added to calculate the voltage command values Vd and Vq for each phase of dq, and the voltage command values Vd and Vq are calculated as the actual three-phase voltages. It converts into command value Vu, Vv, Vw, applies a voltage to the three-phase winding 12 with PWM opening degree (duty ratio) according to voltage command value Vu, Vv, Vw, and controls the drive current of the motor M .

  Here, when the combined vector of the voltage command values Vd and Vq for each phase of dq is saturated, that is, when the length of the combined vector of the voltage command values Vd and Vq for each phase of dq reaches the saturation voltage Vs, the d-phase current id. And the q-phase current iq interfere with each other and the influence of the induced electromotive force proportional to the electrical angular velocity ω, the current values id and iq of each phase are controlled in accordance with the current target values id * and iq * of each phase. And the motor M cannot generate the torque as intended.

Specifically, the d-phase and the q-phase by the dq conversion are converted into orthogonal two-phase windings equivalent to the three-phase windings 12 of U, V, and W in the actual motor M. The value of the combined vector length of the voltage and the q-phase voltage cannot take a value exceeding the saturation voltage Vs, but the length of the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq (Vd ² + Vq ² ) ^1/2 is calculated based on the respective current target values id *, iq * and the deviations εd, εq of the d-phase and q-phase current values id, iq regardless of the above limitation. The limit may be exceeded. Here, the saturation voltage Vs is a value corresponding to the voltage of the power source E in the dq coordinates.

  Therefore, when the combined vector length of the d-phase voltage command value Vd and the q-phase voltage command value Vq exceeds the saturation voltage Vs, that is, the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq is saturated. If the d-phase voltage command value Vd and the q-phase voltage command value Vq are converted to the voltage command values Vu, Vv, Vw to be applied to the three-phase windings 12, respectively, Since the three-phase windings 12 cannot be applied to the values Vu, Vv, and Vw, the current values id and iq cannot follow the current target values id * and iq *, and the current target values id *. , Iq * and the current values id, iq, the absolute values of deviations εd, εq become large.

  Then, the absolute values of the d-phase voltage command value Vd and the q-phase voltage command value Vq increase even though the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq is saturated. As a result, ripples are generated in the torque of the motor M and the controllability is deteriorated.

  On the other hand, when the motor M generates torque that rotates the rotor R in the negative speed direction, the sign that the q-axis current value iq has a positive value is used, and the d-axis is the vertical axis and q In the dq orthogonal coordinate with the axis as the horizontal axis, the d-phase when the q-phase current value iq takes the maximum value when the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq is saturated. The angle (vector angle) of the combined vector of the voltage command value Vd and the q-phase voltage command value Vq with respect to the q axis changes as the electrical angular velocity ω changes as shown in FIG. For convenience of explanation and easy understanding, the vector angle of the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq is the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq. Will be described as the angle formed with respect to the q axis in the dq orthogonal coordinates, but the intent of the present invention is to control the vector angle of the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq. The phase current is appropriately controlled, and an angle formed by a combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq with respect to an arbitrary straight line extending from the origin of the dq orthogonal coordinates is used as a reference. It may be a vector angle, for example, the straight line may be the d-axis in dq orthogonal coordinates.

  As shown by the solid line in FIG. 2, when the rotor R is reversely rotated at a negative speed and the motor M generates torque that assists the reverse rotation of the rotor R (power running state), The angle of the combined vector at which the q-phase current value iq becomes the maximum value approaches 0 to 90 degrees clockwise from 0 degrees as the electrical angular velocity ω decreases (increases to the minus side), and finally becomes -90 degrees. On the other hand, when the rotor R is rotating forward at a positive speed and the motor M is generating torque that suppresses the forward rotation of the rotor R (braking state), the current value of the q phase The angle of the combined vector at which iq becomes the maximum value approaches 90 degrees counterclockwise from 0 degrees as the electrical angular velocity ω increases, and finally becomes 90 degrees.

  In addition, as shown by the broken line in FIG. 2, when the rotor R is rotating forward with a positive speed and the motor M is generating torque that assists in the forward rotation of the rotor R (power running state). Is that the angle of the combined vector at which the q-phase current value iq becomes the minimum value (maximum value on the minus side) approaches −90 degrees counterclockwise from −180 degrees as the electrical angular velocity ω increases. In contrast, when the rotor R is reversely rotated at a negative speed and the motor M generates a torque that suppresses the reverse rotation of the rotor R (braking state), q The angle of the combined vector at which the phase current value iq becomes the minimum value (maximum value on the minus side) approaches 180 degrees from 90 degrees clockwise due to the decrease (increase on the minus side) of the electrical angular velocity ω, and finally 90 degrees.

  Since the q-phase current is a current that contributes to the torque output by the motor M, the greater the absolute value of the q-phase current value iq, the greater the generated torque of the motor M.

  From this, when the combined vector of the voltage command values Vd and Vq of each phase of dq is saturated, if the angle of the combined vector with respect to the q axis is controlled, the q that contributes to the torque generated by the motor M even in the saturated state. The phase current can be controlled, and a large torque can be generated by the motor M.

  On the other hand, when the motor M performs two-phase conversion of dq to control the d-phase current id to zero and the combined vector of the d-phase and q-phase voltage command values Vd and Vq is saturated, the integral calculation of the deviations εd and εq If the combined vector of the d-phase and q-phase voltage command values Vd and Vq is saturated, as shown in FIG. 3A, in the powering state, the d-phase voltage command value Vd and the q-phase The angle of the combined vector of the voltage command value Vq with respect to the q-axis changes between about 0 degrees and −50 degrees, and changes between about 80 degrees and about 120 degrees in the braking state.

  Further, when field weakening control is performed and the motor M is controlled by stopping the integral calculation of the deviations εd and εq when the combined vector of the d-phase and q-phase voltage command values Vd and Vq is saturated, FIG. As shown in FIG. 4, in the power running state, the angle of the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq with respect to the q axis changes between about −20 degrees and −50 degrees, and in the braking state Changes between 30 degrees and 70 degrees.

  3A and 3B are results when the motor M is controlled by a conventional control method. Specifically, each graph shows sinusoidal vibration on the rack shaft 2. The d-phase voltage command value Vd and the q-phase voltage when the rack shaft 2 is reciprocated so that the combined vector of the d-phase and q-phase voltage command values Vd and Vq is saturated during the stroke. The angle with respect to the q-axis of the combined vector of the command value Vq is shown, and a torque that reversely rotates the rotor R is applied to the motor M. Further, since the rack shaft 2 is stroked by applying sinusoidal vibration, the stroke speed becomes low near both ends and the center of each graph, so that the combined vector of the d-phase and q-phase voltage command values Vd and Vq is saturated. Therefore, there is a change in the angle formed with respect to the q axis of the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq. Furthermore, the absolute value of the electrical angular velocity ω is greatest at the 1/4 and 3/4 portions of the horizontal axis.

  As can be understood from the above description, if the motor M is controlled in the same manner as in the past, even if the resultant vector of the d-phase and q-phase voltage command values Vd and Vq is saturated, the vector angle is as in the present invention. Since the control is not performed, the absolute value of the angle of the combined vector with respect to the q axis decreases with the increase of the absolute value of the electrical angular velocity ω, and the absolute value of the q-phase current value iq is maximized. It has not been.

  Therefore, the motor control device 20 according to the embodiment of the present invention includes vector angle control means capable of controlling the angle of the combined vector of the voltage command values Vd and Vq of each phase of dq with respect to the q axis. ing.

  When the vector angle control unit controls the angle of the combined vector of the voltage command values Vd and Vq for each phase of dq with respect to the q axis, the combined vector of the voltage command values Vd and Vq for each phase of dq is saturated. Even in this case, it is possible to appropriately control the torque generated by the motor M, and it is possible to cause the motor M to generate a larger torque than that of the conventional motor control device.

  Further, in this motor control device 20, since a larger torque can be generated by the motor M, the torque control range is increased, and the necessary thrust of the actuator can be obtained even if a smaller motor is used. Can be secured. In other words, a smaller motor can be used, the cost of the actuator can be reduced, and the mountability of the actuator in a device using the actuator can be improved.

  Next, the motor control device 20 will be described more specifically. Specifically, as shown in FIG. 4, the motor control device 20 includes a current target value calculation unit 21 that calculates each current target value id *, iq *, and a q-phase current target value iq * that has a limit value. A current limiter 22 serving as a q-phase current limiter that limits this to a limit value Ilim, and a current flowing in two phases of the three phases of the winding 12 by dq conversion to obtain a d-phase current value id and a q-phase A two-phase current calculation unit 23 for calculating a current value iq, and a d-phase voltage command value Vd and a q-phase voltage command based on each current target value id *, iq * and the d-phase and q-phase current values id, iq A proportional-integral control unit 24 that calculates a value Vq, a vector angle control unit 25 that is a vector angle control unit that controls the angle of the combined vector of the d-phase and q-phase voltage command values Vd and Vq with respect to the q-axis, and d-phase And q-phase voltage command values Vd, Vq Saturation determination unit 26, which is a selection unit that determines whether the vector is saturated and selects either proportional integral control or vector angle control, and d-phase voltage command value Vd and q-phase voltage command value Vq are set to U , V, W of the three-phase voltage command values Vu, Vv, Vw, and the voltage command values Vu, Vv, Vw output from the three-phase conversion calculator 27 When the PWM opening is fully open, that is, when the PWM duty ratio is greater than or equal to the maximum value, the limiter 28 that limits the voltage command values Vu, Vv, and Vw to values that maximize the PWM duty ratio, and the U of the motor M , V, and W, a current detector 29 that detects a current value flowing in two phases iu and iv, and each winding 12 of U, V, and W with a predetermined PWM opening according to the voltage command values Vu, Vv, and Vw And a PWM circuit 30 for applying voltage It has been.

  The motor control device 20 basically includes the d-phase and q-phase current target values id * and iq * determined by the current target value calculation unit 21 and the calculation result of the two-phase current calculation unit 23. The motor M is subjected to proportional-integral control based on the deviations εd and εq from the d-phase and q-phase current values id and iq obtained as follows. Note that proportional differential integration control may be performed by adding an element obtained by differentiating the deviations εd and εq.

  Here, the current target value calculation unit 21 outputs the d-phase and q-phase current target values id * and iq * to the proportional-plus-integral control unit 24 according to a predetermined control law. For the side, one suitable for the device in which the actuator is used is adopted.

  Further, the current target value calculation unit 21 basically calculates the q-phase current target value iq * by setting the d-phase current target value id * to 0, but the electrical angular velocity ω of the rotor is large. In this case, of course, the field weakening control may be performed by inducing the d-phase current target value id * to a negative value.

  The current detector 29 may be a non-contact type using a Hall element, a winding, or the like, or a current sensor that obtains a current value from a voltage drop of a resistor connected in series with any two of the three-phase windings 12. May be used.

  The current detector 29 only needs to detect a current value flowing in two phases of the U, V, and W phases. If the current values of the two phases are known, this will be described later from the electrical angle θ of the rotor R. This is because it can be converted into d-phase and q-phase current values using Equation (1).

  Further, as shown in FIG. 5, the PWM circuit 30 includes a power source E, six switching elements 30a for supplying current to the windings 12 of the three phases of the motor M, and a PWM pulse signal to each switching element 30a. The PWM circuit 31 includes a pulse generator (not shown) such as a multivibrator to be applied. The PWM circuit 31 is configured to change the phase to a predetermined PWM opening based on each voltage command value output from the proportional integration control unit 21. Supply current.

比例積分制御部２４は、各電流目標値ｉｄ*，ｉｑ*とｄ相およびｑ相の電流値ｉｄ，ｉｑの各偏差εｄ，εｑを求め、上記各偏差εｄ，εｑをそれぞれ積分した値に積分ゲインＫＩを乗じるとともに、各偏差εｄ，εｑに比例ゲインＫＰを乗じることで得られる二つの値を加算して、ｄ相電圧指令値Ｖｄおよびｑ相電圧指令値Ｖｑを演算する。 Then, using the electrical angle θ, the two-phase current calculation unit 23 converts the current values iv and iu into d-phase and q-phase current values id and iq, as shown in the following formula (1). The converted d-phase and q-phase current values id and iq are output to the proportional-plus-integral control unit 24.

The proportional-integral control unit 24 calculates each current target value id *, iq * and each of the d-phase and q-phase current values id, iq, and εd, εq, and integrates the values obtained by integrating the deviations εd, εq. The gain KI is multiplied, and two values obtained by multiplying the deviations εd and εq by the proportional gain KP are added to calculate the d-phase voltage command value Vd and the q-phase voltage command value Vq.

  Specifically, the deviations εd and εq for each phase are calculated by the calculation formulas of εd = id * -id and εq = iq * -iq, and the integration of the deviations εd and εq is calculated for each phase. The integral values fd and fq are calculated by adding the phase deviations εd and εq corresponding to the d-phase and q-phase integral values fdpre and fqpre that were calculated at the previous control, respectively. Calculated. That is, the d-phase integration value fd is calculated by fd = fdpre + εd, and the q-phase integration value fq is calculated by fq = fqpre + εq.

  Therefore, the d-phase voltage command value Vd is calculated by Vd = KI · fd + KP · εd, the q-phase voltage command value Vq is calculated by Vq = KI · fq + KP · εq, and the proportional-integral control unit 24 described above is The voltage command values Vd and Vq calculated as described above are output.

  That is, in the case of the motor control device 20, the current loop is a current feedback loop formed by the two-phase current calculation unit 23, the proportional integration control unit 24, and the motor M to be controlled.

以上によって、モータ制御装置２０は、基本的には、電流ループ処理を行って比例積分制御によってモータＭを駆動するようになっている。 Further, the d-phase voltage command value Vd and the q-phase voltage command value Vq are input to the three-phase conversion calculation unit 27 that converts the voltage command values of the U, V, and W phases as described above. The phase conversion calculation unit 27 converts the d-phase voltage command value Vd and the q-phase voltage command value Vq to the actual voltage command values Vu, Vv, Vw of the U, V, and W phases by the calculation of the following equation (2). The converted voltage command values Vu, Vv, and Vw are output to the PWM circuit 30. The limiter 28 is provided to limit the voltage command values Vu, Vv, and Vw so that the PWM opening in the PWM circuit 30 is within a range of values from the maximum to the minimum.

As described above, the motor control device 20 basically performs current loop processing and drives the motor M by proportional-integral control.

  Subsequently, in the motor control device 20 of the present invention in the present embodiment, the vector angle control unit 25 is controlled so that the angle of the combined vector of the voltage command values Vd and Vq of each phase of dq with respect to the q axis can be controlled. In addition, by controlling the vector angle, the torque generated by the motor M can be appropriately controlled even when the combined vector of the voltage command values Vd and Vq of each phase of dq is saturated. It has become.

The vector angle control unit 25 obtains a vector angle target value theta _dq, from this vector angle target value theta _dq synthetic vector length ^{(Vd 2 + Vq 2) 1/2} , Vd voltage command value Vd of the d-phase = (Vd ² + Vq ² ) ^1/2 · sin θ _dq , the q-phase voltage command value Vq is calculated by Vq = (Vd ² + Vq ² ) ^1/2 · cos θ _dq , and the voltage of each phase of these dq The command values Vd and Vq are output to the three-phase conversion calculation unit 27.

In calculating sin θ _dq and cos θ _dq , the vector angle control unit 25 actually uses a sin function map. This sin function map is originally used when the three-phase conversion calculation unit 27 performs the dq-UVW conversion. Therefore, by using the sin function map also in the vector angle control unit 25, the sin function map is provided to the motor control device 20. There is no need to keep duplicates.

Returning in detail, the vector angle control unit 25 first determines that the q-phase current iq has the maximum value based on the electrical angular velocity ω of the rotor R when the q-phase current target value iq * is positive. When the q-phase current target value iq * is negative, the vector angle at which the q-phase current iq takes the minimum value is set as the vector angle final value θ _dq *.

Specifically, since the vector angle final value θ _dq * varies depending on the electrical angular velocity ω, as shown in FIGS. 6A and 6B, a vector that varies in advance using the electrical angular velocity ω as a parameter in advance. The final angle value θ _dq * is mapped, the electric angle velocity ω is monitored by the vector angle control unit 25, and the final vector angle value θ _dq * is calculated by referring to the map.

By performing the map calculation in this way, the final vector angle value θ _dq * necessary for controlling the vector angle can be easily obtained, thereby reducing the calculation time and improving the control response. In addition, it is possible to avoid a situation where a high-performance CPU, which is expensive, must be used.

  The map shown in FIG. 6A includes a map in the case where the motor M shown in FIG. 6A generates a torque that reversely rotates the rotor R, and a torque that causes the motor M shown in FIG. There are two types of maps for generating

Actually, the final vector angle value θ _dq * continuously changes with respect to the change in the electrical angular velocity ω. However, when creating the map, the discontinuity is obtained when the electrical angular velocity ω is changed by an arbitrary value. such as a vector angle final value theta _{dq *,} if not actually present in the vector angle final value theta _{dq *} is on a map corresponding to the monitored electrical angular velocity ω, the operation vector angle final value theta _{dq *} by linear interpolation Just do it.

つづいて、ベクトル角度制御部２５は、前回制御時の電圧指令値Ｖｄ，Ｖｑの合成ベクトルを取り込み、前回制御時の電圧指令値Ｖｄ，Ｖｑの合成ベクトルのベクトル角度θｖを今回制御に当たっての現在のベクトル角度として、ベクトル角度最終値θ_ｄｑ*とベクトル角度θｖの角度制御偏差たるベクトル角度制御偏差εθ_ｄｑと、ｑ相電流目標値ｉｑ*の符号、および、ｑ相電流目標値ｉｑ*とｑ相電流値ｉｑの偏差である電流制御偏差ε_ｑ（ε_ｑ＝ｉｑ*−ｉｑ）とから、図７に示すような判断を行って、合成ベクトルのｑ軸に対する回転方向を決定し、この回転方向からベクトル角度目標値θ_ｄｑを設定するようにする。 Further, instead of performing the map calculation, θ _dq * is approximately obtained by the equation (3), and when the motor M is caused to generate a torque for rotating the rotor R in the reverse direction, θ _d is in a range of −90 degrees to 90 degrees. _{When dq} ^* is obtained and torque for causing the motor M to rotate the rotor R forward is generated, θ _dq ^* may be obtained in the range of −180 degrees to −90 degrees and 90 degrees to 180 degrees. In equation (3), ω represents an electrical angular velocity, and a is an arbitrary positive small constant.

Subsequently, the vector angle control unit 25 takes in the combined vector of the voltage command values Vd and Vq at the previous control, and obtains the vector angle θv of the combined vector of the voltage command values Vd and Vq at the previous control as the current control. As the vector angle, the vector angle final value θ _dq * and the vector angle control deviation εθ _{dq which} is the angle control deviation of the vector angle θv, the sign of the q-phase current target value iq *, and the q-phase current target value iq * and the q-phase Based on the current control deviation ε _q (ε _q = iq * −iq), which is a deviation of the current value iq, the determination as shown in FIG. 7 is performed to determine the rotation direction of the combined vector with respect to the q axis. From this, the vector angle target value θ _dq is set.

The vector angle θv is an angle of the combined vector of the voltage command values Vd and Vq with respect to the q axis, and the combined vector changes from 0 degrees to 180 degrees counterclockwise with respect to the q axis. The composite vector is assumed to change clockwise from 0 degrees to −180 degrees with respect to the axis, and the vector angle target value θ _dq is the same.

The vector angle control deviation εθ _dq is an amount obtained based on the final vector angle value θ _dq * and the simple deviation eθ of the vector angle θv (eθ = θ _dq * −θv, the unit is an angle). When the deviation eθ is in the range of −180 or more and less than 180 degrees, the simple deviation eθ is directly used as the vector angle control deviation εθ _dq . When the deviation eθ is 180 degrees or more, the vector angle control deviation εθ _dq is εθ _dq = When calculated by eθ-360 and the simple deviation eθ is less than −180 degrees, the vector angle control deviation εθ _dq is calculated by εθ _dq = eθ + 360. Therefore, the vector angle control deviation εθ _dq is always adjusted to take a value within the range of −180 degrees or more and less than 180 degrees.

In the above determination, when the sign of the q-phase current target value iq * is positive, that is, when the motor M generates a torque that reversely rotates the rotor R, the current control deviation ε _q takes a positive value. When the vector angle control deviation εθ _dq is not less than −180 degrees and less than 0 degrees, the vector angle θv is rotated clockwise. Conversely, when the vector angle control deviation εθ _dq is not less than 0 degrees and less than 180 degrees, the vector angle Rotate θv counterclockwise.

On the other hand, even when the sign of the q-phase current target value iq * is positive, when the current control deviation ε _q takes a negative value, the vector angle control deviation εθ _dq is −180 degrees or more and less than 0 degrees. The vector angle θv is rotated counterclockwise. Conversely, when the vector angle control deviation εθ _dq is not less than 0 degrees and less than 180 degrees, the vector angle θv is rotated clockwise.

That is, on the assumption that the sign of the q-phase current target value iq * is positive, if the q-phase current value iq is smaller than the q-phase current target value iq *, that is, the current control deviation ε _q is a positive value. If the vector angle control deviation εθ _dq is not less than −180 degrees and less than 0 degrees, as shown in FIG. 8, the vector angle θv during the previous control is the final vector angle that maximizes the q-phase current iq. It exists in the area | region (a) between the line L1 of value (theta) _dq *, and the line L2 from which this phase L1 differs in phase by 180 degrees. Further, since the q-phase current value iq is smaller than the q-phase current target value iq *, in this case, the vector angle θv at the previous control is brought closer to the line L1 of the vector angle final value θ _dq * that maximizes the q-phase current iq. Thus, the q-phase current iq can be approximated to the q-phase current target value iq *, so that the combined vector is moved downward in FIG. 8 with respect to the line L1, that is, the combined vector is considered in terms of the dq axis orthogonal coordinates. When rotated clockwise, the rotation angle of the combined vector can be made 180 degrees or less at the maximum, and the q-phase current iq can follow the q-phase current target value iq * more quickly.

Explaining the above in more detail, for example, when the vector angle final value θ _dq * is −90 degrees, when considering the orthogonal coordinates of the dq axis, as shown in FIG. When the angle θv is in a range greater than −90 degrees and less than or equal to 90 degrees, and there is a combined vector in such a range, the vector angle θv that is an angle formed by the combined vector with respect to the q axis is determined as the vector angle final. It will be understood that the rotation angle is smaller when the synthesized vector is rotated clockwise in FIG. 9 in order to obtain the value θ _dq *.

In this way, by making the above determination, in order to cause the q-phase current iq to follow the q-phase current target value iq * more quickly, the resultant vector is counterclockwise and counterclockwise with respect to the vector angle final value θ _dq *. It is possible to determine which direction should be rotated.

Similarly, on the assumption that the sign of the q-phase current target value iq * is positive, when the q-phase current value iq is smaller than the q-phase current target value iq *, that is, the current control deviation ε _q is a positive value. When the vector angle control deviation εθ _dq is not less than 0 degrees and less than 180 degrees, as shown in FIG. 8, the vector angle θv in the previous control is the vector angle final value that maximizes the q-phase current iq. It exists in the area | region (b) between the line L1 of value (theta) _dq *, and the line L2 from which this phase L180 differs in phase. Further, since the q-phase current value iq is smaller than the q-phase current target value iq *, in this case, the vector angle θv at the previous control is brought closer to the line L1 of the vector angle final value θ _dq * that maximizes the q-phase current iq. Thus, the q-phase current iq can be approximated to the q-phase current target value iq *, so that the combined vector is moved upward in FIG. 8 relative to the line L1, that is, the combined vector is considered in terms of the orthogonal coordinates of the dq axis. When rotated counterclockwise, the rotation angle of the combined vector can be made 180 degrees or less at maximum.

Furthermore, on the assumption that the sign of q-phase current target value iq * is positive, when q-phase current value iq is larger than q-phase current target value iq *, that is, current control deviation ε _q is a negative value. If the vector angle control deviation εθ _dq is not less than −180 degrees and less than 0 degrees, as shown in FIG. 8, the vector angle θv during the previous control is the final vector angle that maximizes the q-phase current iq. It exists in the region (a) between the line L1 of the value θ _dq * and the line L2 that is 180 degrees out of phase with the line L1, but this time the q-phase current target value iq Since the q-phase current value iq is larger than *, in this case, it is necessary to keep the vector angle θv at the previous control away from the line L1 of the final vector angle value θ _dq * that maximizes the q-phase current iq.

Therefore, when the composite vector is moved upward in FIG. 8 with respect to the line L1, that is, when the composite vector is rotated counterclockwise in terms of the orthogonal coordinates of the dq axis, the composite vector is set to the vector angle final value θ _dq *. The q-phase current iq can be made to follow the q-phase current target value iq * more quickly by reliably moving away from the line L1. Here, if the synthesized vector is rotated clockwise, the q-phase current iq increases to the maximum value and then follows the q-phase current target value iq *, so that the controllability deteriorates. Therefore, it becomes possible to prevent deterioration of controllability by determining the rotation direction based on the above-described determination.

That is, on the assumption that the sign of the q-phase current target value iq * is positive, when the q-phase current value iq is larger than the q-phase current target value iq *, that is, the current control deviation ε _q is negative. If the vector angle control deviation εθ _dq is not less than 0 degrees and less than 180 degrees, the vector angle θv in the previous control is the vector angle that maximizes the q-phase current iq, as shown in FIG. Although it exists in the area | region (b) between the line L1 of final value (theta) _dq *, and the line L2 which differs in phase 180 degree | times with respect to this line L1, it is q phase current target similarly to the above. Since the q-phase current value iq is larger than the value iq *, in this case, it is necessary to keep the vector angle θv at the previous control away from the line L1 of the vector angle final value θ _dq * that maximizes the q-phase current iq.

Therefore, when the composite vector is moved downward in FIG. 8 with respect to the line L1, that is, when the composite vector is rotated clockwise in view of the orthogonal coordinates of the dq axis, the composite vector is the line of the vector angle final value θ _dq *. The q-phase current iq can be made to follow the q-phase current target value iq * more quickly by reliably moving away from L1.

Subsequently, when the sign of the q-phase current target value iq * is negative, that is, when the motor M is caused to generate torque that causes the rotor R to rotate in the forward direction, the current control deviation ε _q takes a positive value. When the vector angle control deviation εθ _dq is not less than −180 degrees and less than 0 degrees, the vector angle θv is rotated counterclockwise. Conversely, when the vector angle control deviation εθ _dq is not less than 0 degrees and less than 180 degrees, the vector angle θv Rotate clockwise.

On the other hand, even when the sign of the q-phase current target value iq * is negative, in the situation where the current control deviation ε _q takes a negative value, the vector angle control deviation εθ _dq is −180 degrees or more and less than 0 degrees. The vector angle θv is rotated clockwise. Conversely, when the vector angle control deviation εθ _dq is not less than 0 degrees and less than 180 degrees, the vector angle θv is rotated counterclockwise.

That is, on the assumption that the sign of the q-phase current target value iq * is negative, if the q-phase current value iq is smaller than the q-phase current target value iq *, that is, the current control deviation ε _q is a positive value. If the vector angle control deviation εθ _dq is not less than −180 degrees and less than 0 degrees, as shown in FIG. 10, the vector angle θv at the previous control is the vector angle final value that minimizes the q-phase current iq. It exists in the area | region (c) between the line L3 of value (theta) _dq *, and the line L4 from which a phase differs 180 degree | times with respect to this line L3.

Further, since the q-phase current value iq is smaller than the q-phase current target value iq *, in this case, the vector angle θv at the previous control is moved away from the line L3 of the vector angle final value θ _dq * that minimizes the q-phase current iq. As a result, the q-phase current iq can be made closer to the q-phase current target value iq *. Therefore, the combined vector is moved upward in FIG. 10 with respect to the line L3. When rotated counterclockwise, the combined vector can be surely moved away from the line L3 of the vector angle final value θ _dq *, and the q-phase current iq can follow the q-phase current target value iq * more quickly. Here, if the combined vector is rotated clockwise, the q-phase current iq once decreases to the minimum value that is the negative maximum value and then follows the q-phase current target value iq *. Since the controllability is deteriorated, it is possible to prevent the deterioration of the controllability by determining the rotation direction based on the above-described determination.

To explain the above in more detail, for example, when the vector angle final value θ _dq * is 90 degrees, considering the orthogonal coordinates of the dq axis, as shown in FIG. 11, the vector angle of the combined vector at the previous control time When θv is in the range of greater than 90 degrees and less than 180 degrees and in the range of less than −90 degrees and greater than −180 degrees, and there is a composite vector in such a range, the composite vector is in relation to the q axis. It will be understood that it is reasonable to rotate the resultant vector counterclockwise in FIG. 11 in order to keep the vector angle θv, which is the angle formed, away from the final vector angle value θ _dq *.

Similarly, on the assumption that the sign of the q-phase current target value iq * is negative, when the q-phase current value iq is smaller than the q-phase current target value iq *, that is, the current control deviation ε _q is a positive value. When the vector angle control deviation εθ _dq is not less than 0 degrees and less than 180 degrees, as shown in FIG. 10, the vector angle θv at the previous control is the vector angle final value that minimizes the q-phase current iq. It exists in the area | region (d) between the line L3 of value (theta) _dq *, and the line L4 from which a phase differs 180 degrees with respect to this line L3.

Further, since the q-phase current value iq is smaller than the q-phase current target value iq *, in this case, the vector angle θv at the previous control is moved away from the line L3 of the vector angle final value θ _dq * that minimizes the q-phase current iq. As a result, the q-phase current iq can be brought close to the q-phase current target value iq *. Therefore, the combined vector is moved downward in FIG. 10 with respect to the line L3. When rotated clockwise, the rotation angle of the combined vector can be made 180 degrees or less at maximum.

Furthermore, on the assumption that the sign of q-phase current target value iq * is negative, if q-phase current value iq is larger than q-phase current target value iq *, that is, current control deviation ε _q is a negative value. If the vector angle control deviation εθ _dq is not less than −180 degrees and less than 0 degrees, as shown in FIG. 10, the vector angle θv at the previous control is the vector angle final value that minimizes the q-phase current iq. It exists in the region (c) between the line L3 of the value θ _dq * and the line L4 that is 180 degrees out of phase with the line L3, but this time the q-phase current target value iq Since the q-phase current value iq is larger than *, in this case, it is necessary to bring the vector angle θv at the previous control closer to the line L3 of the vector angle final value θ _dq * that minimizes the q-phase current iq.

Therefore, when the composite vector is moved downward in FIG. 10 with respect to the line L3, that is, when the composite vector is rotated clockwise in view of the orthogonal coordinates of the dq axis, the composite vector is the line of the vector angle final value θ _dq *. The Q phase current iq can be made to follow the q phase current target value iq * more quickly.

  That is, considering the orthogonal coordinates of the dq axes, if the resultant vector is rotated clockwise, the rotation angle of the resultant vector can be reduced to 180 degrees or less at the maximum, and the q-phase current iq can be converted to the q-phase current target value iq earlier. * Can be followed.

Further, on the assumption that the sign of the q-phase current target value iq * is negative, if the q-phase current value iq is larger than the q-phase current target value iq *, that is, the current control deviation ε _q is a negative value. If the vector angle control deviation εθ _dq is not less than 0 degrees and less than 180 degrees, as shown in FIG. 10, the vector angle θv at the previous control is the final vector angle value that minimizes the q-phase current iq. It exists in the region (d) between the line L3 of θ _dq * and the line L4 having a phase difference of 180 degrees with respect to the line L3, but the q-phase current target value iq is similar to the above. Since the q-phase current value iq is larger than *, in this case, it is necessary to bring the vector angle θv at the previous control closer to the line L3 of the vector angle final value θ _dq * that maximizes the q-phase current iq.

Therefore, when the composite vector is moved upward in FIG. 10 with respect to the line L3, that is, when the composite vector is rotated counterclockwise in terms of the orthogonal coordinates of the dq axis, the composite vector is set to the vector angle final value θ _dq *. The q-phase current iq can be made to follow the q-phase current target value iq * sooner by approaching the line L3.

That is, by determining the rotation direction of the vector angle θv by the above determination, the q-phase current is changed to the q-phase current target value iq * by making the vector angle θv a perspective with reference to the vector angle final value θ _dq *. It is possible to follow.

When the rotation direction of the combined vector with respect to the q-axis is determined from the above determination, the vector angle target value θ _dq is set from the rotation direction. This vector angle target value θ _dq is, for example, the current vector angle The angle is set to θv by adding a predetermined α degree in the rotational direction. Therefore, the vector angle target value θ _dq is calculated as θ _dq = θv ± α. Note that the sign before α is determined as described above according to the rotation direction of the combined vector.

  The angle α added to the vector angle θv is calculated by multiplying the q-phase current deviation εq by the control gain. In this case, since the q-phase current is increased or decreased by the sin function with respect to the vector angle θv, the control gain for calculating α can be expressed in a non-linear manner that can be expressed by the sin function according to this. It is possible to ensure the same responsiveness even in a difficult state.

Thus, the vector angle control unit 25 calculates the d-phase voltage command value Vd and the q-phase voltage command value Vq from the vector angle target value θ _dq as described above, and these voltage command values Vd and Vq are The motor M is driven by the PWM circuit 30 after being input to the three-phase conversion calculation unit 27 that converts the voltage command values Vv, Vu, Vw of each phase of U, V, W.

  In this way, the vector angle control unit 25 controls the vector angle θv of the combined vector of the voltage command values Vd and Vq for each phase of the dq, so that the torque of the motor M can be increased to the limit even in the saturated state. It is possible to output torque as close as possible to demand.

A vector angle capable of realizing the q-phase current target value iq * is obtained from the electrical angular velocity ω of the rotor R, and this vector angle may be directly controlled as the vector angle target value θ _dq . Thus, by obtaining the rotation direction and setting the vector angle target value θ _dq , it is possible to prevent the voltage command values Vd and Vq of each phase of dq from fluctuating rapidly, and the torque fluctuation increases. Can be prevented. Further, if the vector angle target value θ _dq is set to a vector angle that can realize the q-phase current target value iq *, the rotation direction of the vector angle cannot be uniquely determined, but the rotation direction of the vector angle is determined by the above determination. Therefore, the operation amount of the vector angle can be reduced.

As described above, the vector angle target value θ _dq can be obtained, and thereby the torque control range can be expanded. However, when the rotation direction is determined by the above determination, the vector angle final value θ _dq * is set to Since the rotation direction is reversed as a reference, the vector angle θv may be vibrated near the final value of the vector angle θ _dq * or may change abruptly. Therefore, a current limiting unit 22 is provided that limits the absolute value of the q-phase current target value iq * to the maximum value that can be taken.

  The current limiting unit 22 limits the q-phase current target value iq * to the limit value Ilim that is the maximum value that the q-phase current can take when the motor M generates torque so as to reversely rotate the rotor R. On the other hand, when the motor M is generating torque so as to rotate the rotor R forward, the q-phase current target value iq * is limited to the limit value Ilim that is the minimum value that the q-phase current can take. .

  Specifically, the limit value Ilim is a state in which the combined vector of the voltage command values Vd and Vq of each phase of dq has reached the saturation voltage Vs when the motor M generates a torque that reversely rotates the rotor R. In this case, the maximum value that can be taken by the q-phase current value iq when the resultant vector is rotated 360 degrees around the origin in the dq coordinate, while the motor M generates a torque for rotating the rotor R forward. The q-phase current value iq can be taken when the combined vector of the voltage command values Vd and Vq for each phase reaches the saturation voltage Vs and the combined vector is rotated 360 degrees around the origin in the dq coordinates. It is the minimum value.

Therefore, when the motor M generates torque for rotating the rotor R in the reverse direction, when the upper limit of the q-phase current target value iq * is saturated in the combined vector of the d-phase and q-phase voltage command values Vd and Vq, When the q-phase current value iq is limited to the maximum value that can be taken and the motor M generates a torque that causes the rotor R to rotate forward, the lower limit of the q-phase current target value iq * is the d-phase and q-phase voltage command. When the combined vector of the values Vd and Vq is saturated, the q-phase current value iq is limited to the minimum value that can be taken. Therefore, even if the combined vector is saturated, the q-phase current target value iq * and the q-phase current iq The absolute value of the deviation εq does not increase, and during the vector angle control described above, the vector angle θv becomes a small value near the vector angle final value θ _dq *, and the vector angle θv becomes the vector angle final value θ _dq *. Become biased Since εq is zero, so that a trouble such as vector angle θv is oscillatory around _* vector angle final value theta _dq is eliminated.

  The maximum absolute value of the q-phase current value iq approaches zero due to the influence of the mutual interference between the d-phase and the q-phase and the induced electromotive force as the absolute value of the electrical angular velocity ω of the rotor R increases. It becomes smaller and becomes a value depending on the electrical angular velocity ω.

  Therefore, the limit value Ilim also needs to be changed depending on the electrical angular velocity ω. However, as shown in FIG. 12, the limit value Ilim that varies with the electrical angular velocity ω as a parameter is mapped in advance, and the current limiter The electrical angular velocity ω may be monitored at 22 and the limit value Ilim may be selected with reference to the map. The reason why the limit value Ilim when the electrical angular velocity ω takes a value close to 0 is constant with respect to the change in the electrical angular velocity ω is that it is limited by the maximum energization capability of the motor control device 20. The difference in limit value Ilim between powering and braking when generating torque in the direction is due to the influence of the direction of generation of induced electromotive force.

  By performing the map calculation in this way, it is possible to easily obtain the limit value Ilim necessary for limiting the q-phase current target value iq *, thereby reducing the calculation time and improving the control response. Furthermore, it is possible to avoid a situation in which a high-performance CPU, which is expensive, must be used.

  The map may be created by plotting the limit value Ilim when the electrical angular velocity ω changes from a positive practical maximum value to a negative practical minimum value. In practice, the limit value Ilim is the electrical angular velocity. Although it changes continuously with respect to the change of ω, when creating a map, it corresponds to the actually monitored electrical angle ω as the discontinuous limit value Ilim when the electrical angular velocity ω is changed by an arbitrary value. If the limit value Ilim does not exist on the map, the limit value Ilim may be calculated by linear interpolation.

  In this way, the q-phase current target value iq * is limited to the limit value Ilim, but the limit value Ilim is the maximum value of the q-phase current iq when the combined vector of the d-phase and q-phase voltage command values Vd and Vq is saturated. In the case of the minimum value, the maximum value and the minimum value of the q-phase current value iq depend on the voltage of the power supply E. Therefore, the limit value Ilim may be corrected according to the voltage change of the power supply E.

  Note that the maximum value and the minimum value of the q-phase current iq when the combined vector of the d-phase and q-phase voltage command values Vd and Vq is saturated change substantially in proportion to the power supply voltage. The limit value Ilim may be proportionally corrected with respect to the change. In this case, the limit value Ilim in the map of the limit value Ilim created using the electrical angular velocity ω as a parameter is set in advance as the power source at the time of map creation. A correction map is created by dividing by the voltage of E, and the limit value Ilim may be corrected by multiplying the limit value Ilim obtained by referring to this map by the actual power supply voltage.

  Subsequently, in the motor control device 20, whether or not the combined vector of the voltage command values Vd and Vq for the d phase and the q phase is saturated, that is, the combined voltage command values Vd and Vq for each phase of the dq. A saturation judgment unit 26 is provided as a selection means for judging whether or not the length of the vector is larger than the threshold value and selecting either the above-described proportional-integral control or vector angle control.

  The saturation determination unit 26 takes in the voltage command values Vd and Vq of each phase of dq output from the proportional integration control unit 24 that performs proportional integration control, and the length of the combined vector of the voltage command values Vd and Vq is greater than the threshold value. When the combined vector length of the voltage command values Vd and Vq is larger than the threshold value, the vector angle control unit 25 outputs the switch 26a to select the vector angle control. When the voltage command values Vd and Vq are operated so as to be input to the three-phase conversion calculation unit 27, and when the combined vector of the voltage command values Vd and Vq is equal to or less than the threshold value, the switch 26a is connected to the proportional integration control unit 24. Is operated so that the voltage command values Vd and Vq of each phase dq output from the input are input to the three-phase conversion calculation unit 27.

Note that, regardless of which control is selected, the proportional-integral control unit 24 continues the processing in the current loop in which the current target values id ^* and iq ^* of the dq phases are input, The voltage command values Vd and Vq are output to the saturation judgment unit 26.

Further, when the combined vector of the voltage command values Vd and Vq is saturated, the saturation judgment unit 26 performs proportional-integral control even though the current values id and iq of each phase of dq cannot follow the current target values id ^* and iq ^*. Since the absolute values of the integration values fd and fq of the integration path of the unit 24 will increase, the integration calculation in the integration path is stopped.

  As described above, when the combined vector of the voltage command values Vd and Vq is saturated, the vector angle control is selected and the voltage command values Vd and Vq for each phase dq output from the vector angle control unit 25 become valid. Thereafter, when the combined vector of the voltage command values Vd and Vq of the respective dq phases output from the proportional-integral control unit 24 is not saturated, the switch 26a is switched and the dq-phases output from the proportional-integral control unit 24 are switched. The voltage command values Vd and Vq are made valid.

  At the time of such switching, that is, when proportional integral control is selected and transition is made to proportional integral control, the saturation determination unit 26 outputs an integral value rewriting command for rewriting the integral values fd and fq to the proportional integral control unit 24. On the other hand, the proportional integral control unit 24 monitors the voltage command values Vd and Vq of each phase of dq output from the vector angle control unit 25, and receives the above integral value rewrite command from the saturation judgment unit 26 by switching the switch 26a. Then, the values of the integrated values fd and fq in the integration path are rewritten so that the voltage command values Vd and Vq output from the vector angle control unit 25 are the same. In this rewriting, the vector angle control unit 25 takes into account the current target values id * and iq * of the respective dq phases input to the proportional-integral control unit 24 and the deviations εd and εq of the current values id and iq of the respective dq phases. Is obtained by subtracting the values εd · KP and εq · KP in the proportional path calculated based on the deviations εd and εq from the voltage command values Vd and Vq output from The integration values fd and fq are calculated by reverse calculation and the integration values fd and fq are rewritten.

  That is, immediately after the switch 26a is switched, the proportional-integral control unit 24 has a voltage having the same value as the voltage command values Vd and Vq of the dq phases output from the vector angle control unit 25 before the switch 26a is switched. The command values Vd and Vq are output.

  Then, as described above, when the saturation determination unit 26 shifts from vector angle control to proportional integral control, a situation in which the value of the voltage command values Vd and Vq of each phase of dq changes suddenly is prevented. A situation in which a ripple occurs in the output torque of M is avoided.

Furthermore, also in the vector angle control, as described above, the vector angle target value θ _dq is set so that the voltage command values Vd and Vq of each phase of dq do not change abruptly. Since the command values Vd and Vq are not suddenly changed, the output torque is not suddenly changed even if the switch 26a is switched to any one of the proportional integral control and the vector angle control. There is no need for in-fade out, and there is no problem that the control algorithm becomes complicated.

  Further, when the threshold value is set to the saturation voltage Vs in the determination of the saturation determination unit 26, it is proportional until the combined vector of the voltage command values Vd and Vq of each phase of dq is saturated, that is, normal (non-saturated). Proportional integral control is performed in the integral control unit 24, and vector angle control with a high calculation load needs to be performed only during saturation, which is advantageous in terms of control responsiveness.

  In the determination by the saturation determination unit 26, when the threshold value is the saturation voltage Vs, the threshold value can improve the voltage between the terminals of the U, V, and W phase windings 12 to the voltage of the power source E. In the case where the control is carried out, it is sufficient to set 1 / √2 to a value obtained by multiplying the initial setting voltage of the power source E, and a sine wave voltage is applied to the U, V, W phase windings 12. In the case of control, a value obtained by multiplying √6 / 4 by the set voltage of the power source E may be used.

  In turn, as hardware resources in each part other than the current detector 29, the rotation angle sensor 15 and the PWM circuit 30 of the motor control device 20 configured as described above, specifically, for example, the current detector 32 and the rotation An amplifier for amplifying each signal output from the angle sensor 15, a converter for converting an analog signal into a digital signal, a storage device such as a CPU (Central Processing Unit), a ROM (Read Only Memory), and the CPU It is configured as a well-known computer system (not shown) having a RAM (Random Access Memory) that provides a storage area, a crystal oscillator, and a bus line that connects these, and the voltage command values Vu, Vv, Vw can be output There. In addition, as hardware, each part other than the current detector 32, the rotation angle sensor 15, and the PWM circuit 30 of the motor control device 20 may be integrated into a controller or the like mounted on a device to which the motor M is applied.

  In this case, the current target value calculation unit 21, current limiting unit 22, two-phase current calculation unit 23, proportional integral control unit 24, vector angle control unit 25, saturation determination unit 26, three-phase conversion calculation unit 27, and limiter The processing procedure in 28 is stored in advance in a ROM or other storage device as a program, and each of the above units is realized by the CPU reading the program and executing the above-described arithmetic processing.

  Here, the processing procedure in the motor control apparatus 20 will be described based on the flowchart shown in FIG.

  First, in step S1, the motor control device 20 determines the current values iu and iv of any two phases of the three-phase windings 12, for example, the U phase and the V phase, the electrical angle θ of the motor M, and the electrical angular velocity. Read ω.

  Subsequently, the process proceeds to step S2, and the motor control device 20 calculates each current target value id *, iq *.

  Furthermore, in step S3, a map for calculating the limit value Ilim from the current target value iq * is selected, and a process for limiting the current target value iq * based on the electrical angular velocity ω is performed.

  Subsequently, in step S4, the motor control device 20 uses the current values iu and iv and the electrical angle θ to perform an operation for converting the current values iv and iu into d-phase and q-phase current values id and iq. Go to step S5.

In step S5, the length of the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq was greater than or equal to the threshold during the previous control, that is, the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq. Whether the length (Vd ² + Vq ² ) ^1/2 is equal to or greater than the threshold is determined by referring to the selection flag, and the length of the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq at the previous control is If it is not equal to or greater than the threshold, the process proceeds to step S6. If the combined vector length of the d-phase voltage command value Vd and the q-phase voltage command value Vq is equal to or greater than the threshold during the previous control, the process proceeds to step S10. The selection flag is set in steps S14 and S17 described later. For example, when the selection flag is 0, the length of the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq at the previous control time. Is less than the threshold and the selection flag is 1, it indicates that the combined vector length of the d-phase voltage command value Vd and the q-phase voltage command value Vq was equal to or greater than the threshold during the previous control.

  Subsequently, the process proceeds to step S6, where the motor control device 20 determines whether or not the proportional integral control was performed at the previous control with reference to the control transition flag, and the proportional integral control is performed at the previous control. If yes, the process proceeds to step S7. If not, the process proceeds to step S16. The control transition flag is set in steps S9 and S15 described later. For example, when the control transition flag is 0, proportional-integral control is performed during the previous control, and the control transition flag is 1. In this case, it indicates that the vector angle control was performed during the previous control.

  In step S7, since the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq is not equal to or greater than the threshold value during the previous control and the normal proportional-integral control was performed in the previous time, the normal control is performed. The motor control device 20 calculates the deviations εd and εq between the current target values id * and iq * and the d-phase and q-phase current values id and iq, and calculates the integrated values fd and fq, and step S8. Migrate to In step S7, since normal proportional integral control is performed, the integral values fd and fq are calculated by calculating fd = fdpre + εd and fq = fqpre + εq, respectively.

  In step S8, Vd = KI · fd + KP · εd is calculated to calculate the d-phase voltage command value Vd, and Vq = KI · fq + KP · εq is calculated to calculate the q-phase voltage command value Vq. Migrate to

  In step S9, proportional integral control is performed in this control, so the control transition flag is set to 0 and the routine proceeds to step S17.

On the other hand, in step S10, the motor control device 20 performs vector angle control because the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq is greater than or equal to the threshold value during the previous control. A vector angle θv is obtained from the output voltage command values Vq and Vd of each phase of dq. In calculating the vector angle θv, tan ⁻¹ (Vd / Vq) may be calculated.

Then, in step S11, select the map for obtaining the vector angle final value theta _{dq *} from the sign of the q-phase current target value iq *, obtains the vector angle final value theta _{dq *} based on the electrical angular velocity omega.

Subsequently, in step S12, the sign of the q-phase current target value iq *, the vector angle control deviation εθ _dq, and the current control deviation ε _q which is the deviation between the q-phase current target value iq * and the q-phase current value iq are obtained. Then, the rotation direction of the combined vector is determined, and the vector angle target value θ _dq is obtained from the vector angle θv, the aforementioned α and the rotation direction of the combined vector by θ _dq = θv + α or θ _dq = θv−α.

Then, the process proceeds to step S13, and the voltage command values Vd and Vq of each phase of _{dq are obtained} from the vector angle target value θdq. That is, the d-phase voltage command value Vd is calculated by Vd = (Vd ² + Vq ² ) ^1/2 · sin θ _dq , and the q-phase voltage command value Vq is calculated as Vq = (Vd ² + Vq ² ) ^1/2 · cos θ _dq. Calculate with.

Further, the process proceeds to step S14, where only the integration calculation of the integration path in the proportional integration control is stopped to calculate the voltage command values Vd and Vq for each phase of dq, and the dq phase voltage command value calculated by stopping the integration. If the length of the combined vector of Vd and Vq is greater than or equal to the threshold value, the selection flag is set to 1, otherwise the selection flag is set to 0. In this determination, if an operation for comparing the combined vector length (Vd ² + Vq ² ) ^1/2 of the d-phase voltage command value Vd and the q-phase voltage command value Vq with a threshold value is necessary, a route operation is required and the operation is performed. Since the time becomes longer, it is preferable to compare the square value (Vd ² + Vq ² ) of the combined vector length of the d-phase voltage command value Vd and the q-phase voltage command value Vq with the square value of the threshold value. .

  Subsequently, in step S15, since vector angle control is performed in the current control, the control shift flag is set to 1 and the process proceeds to step S18.

  In step S16, the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq is not greater than or equal to the threshold during the previous control, but the vector angle control is performed during the previous control. In order to shift to the normal control, the motor control device 20 integrates the integration in the integration path of each phase of dq from the voltage command values Vd, Vq of each phase of dq and the deviations εd, εq at the previous control when vector angle control was performed. The values fd and fq are rewritten to calculate the voltage command values Vd and Vq for each phase of dq, and the process proceeds to step S9.

  Returning, step S17 is a step that moves after the processing of step S8 and step S16 described above. In step S17, the motor control device 20 performs the dq phase voltage command value Vd calculated in step S8 or step S16. , Vq, the selection flag is set to 1 when the length of the combined vector is greater than or equal to the threshold, and the selection flag is set to 0 otherwise.

  In step S18, the motor control device 20 performs an operation for converting the d-phase voltage command value Vd and the q-phase voltage command value Vq into three-phase voltage command values Vu, Vv, and Vw, and proceeds to step S19. To do.

  In step S19, the motor control device 20 performs the limiting process when the process of limiting the voltage command values Vu, Vv, and Vw to a value that can be taken is necessary, and performs the limiting process when the process is not necessary. Without moving to step S20.

  In step S20, the motor control device 20 outputs the voltage command values Vu, Vv, Vw to the PWM circuit 30, and ends these series of processes.

  The motor control device 20 controls the motor M by repeatedly performing the above steps S1 to S20.

  Therefore, when the control device 20 executes the above-described series of processing, the current target value calculation unit 21, the current limiting unit 22, the two-phase current calculation unit 23, the proportional integration control unit 24, the vector angle control unit 25, Processing of each part of the saturation determination unit 26, the three-phase conversion calculation unit 27, and the limiter 28 is realized, and thereby the angle of the combined vector of the voltage command values Vd and Vq of each phase of dq with respect to the q axis can be controlled. , Dq Even when the combined vector of the voltage command values Vd and Vq of each phase is saturated, the torque generated by the motor M can be appropriately controlled, and the motor M can be controlled by conventional motor control. It is possible to generate a larger torque compared to the device.

  Further, in the motor control device 20, since a larger torque can be generated by the motor M, the torque control range can be increased, and the motors mounted on various devices can be reduced in size. Even if a smaller motor is used, the necessary thrust of the actuator can be ensured. In other words, a smaller motor can be used, the cost of the actuator can be reduced, and the mountability of the actuator in a device using the actuator can be improved.

  Next, a motor control device 50 according to another embodiment will be described. The motor control device 50 in the other embodiment is different from the motor control device 20 in the one embodiment in that it does not directly control the vector angle of the combined vector of the voltage command values Vd and Vq of each phase of dq. The vector angle is controlled by independently adjusting the d-phase voltage command value Vd and the q-phase voltage command value Vq.

Hereinafter, this different point will be described in detail. Specifically, as shown in FIG. 14, the motor control device 50 according to the other embodiment includes a current target value calculation unit 21 that calculates each current target value id *, iq *, and one embodiment. As with the current limiting unit 22, in addition to limiting the limit to the limit value Ilim when the q-phase current target value iq * exceeds the limit value, the lower limit of the d-phase current target value id ^* is the value of −Φ / Ld A current limiter 22a that limits the current flowing in two phases of the three phases of the winding 12, and a two-phase current calculator 23 that calculates a d-phase current value id and a q-phase current value iq; A proportional-integral control unit 51 for calculating a d-phase voltage command value Vd and a q-phase voltage command value Vq based on each current target value id *, iq * and the d-phase and q-phase current values id, iq; Q axis of the combined vector of the voltage command values Vd and Vq of the q and q phases A vector angle control unit 52 serving as a vector angle control means for controlling the angle with respect to the signal, and determining whether or not the resultant vector of the d-phase and q-phase voltage command values Vd and Vq is saturated, proportional integral control and vector angle Saturation determination unit 53, which is a selection means for selecting one of the controls, and d-phase voltage command value Vd and q-phase voltage command value Vq as voltage command values Vu, Vv, Vw for each of the three phases U, V, and W. Among the three-phase conversion calculation unit 27 for conversion to the above and the voltage command values Vu, Vv, Vw output from the three-phase conversion calculation unit 27, the PWM opening is fully opened, that is, the PWM duty ratio is greater than or equal to the maximum value. In this case, the limiter 28 that limits the voltage command values Vu, Vv, and Vw to values that maximize and minimize the PWM duty ratio, and the current value that flows through the two phases iu and iv of the motors U, V, and W are detected. Current detection 29, is constituted by a PWM circuit 30 for applying U, V, each winding 12 of the W at a predetermined PWM opening in accordance with the voltage command values Vu, Vv, Vw.

  The proportional-plus-integral control unit 51, like the proportional-plus-integral control unit 24 of the motor control device 20 in the above-described embodiment, sets the current target values id *, iq * and the d-phase and q-phase current values id, iq. Each deviation εd, εq is obtained, and the values obtained by integrating the deviations εd, εq are multiplied by an integral gain KI, and two values obtained by multiplying the deviations εd, εq by a proportional gain KP are added, d-phase voltage command value Vd and q-phase voltage command value Vq are calculated.

  Specifically, the proportional-plus-integral control unit 51 includes addition / subtraction units 61 and 62 for calculating the deviations εd and εq of the phases, multiplication units 63 and 64 for multiplying the deviations εd and εq by the proportional gain KP, and the deviations εd. , Εq integrating units 65, 66, multipliers 67, 68 for multiplying the integral values fd, fq, which are the calculation results of the integrators 65, 66, and the integral gain KI, and the proportional gain calculated by the multiplier 63, respectively. An addition unit 69 for adding the deviation εd after multiplication by KP and the integral gain fd after multiplication by the integral gain KI calculated by the multiplication unit 67; and multiplying the deviation εq by multiplication by the proportional gain KP calculated by the multiplication unit 64 And an adding unit 70 that adds the integrated value fq after multiplication by the integral gain KI calculated by the unit 68.

  That is, in the case of the motor control device 50, the current loop is a current feedback loop formed by the two-phase current calculation unit 23, the proportional-integral control unit 51, and the motor M to be controlled.

  Subsequently, in the motor control device 50 of the invention in this other embodiment, the vector angle control unit 52 calculates the integration value fq in the q-phase current loop, that is, the integration unit 66 of the proportional integration control unit 51. The integrated value fq is corrected by feeding back the integrated value fq, and the q-phase voltage command value Vq is adjusted to control the vector angle of the combined vector of the voltage command values Vd and Vq for each phase of dq.

Hereinafter, the vector angle control in the other embodiment will be described in detail. The vector angle control unit 52 is an integration that is a deviation between the integration value fq calculated by the integration unit 66 and the q-phase integration value target value fq ^*. _{Addition /} subtraction unit 71 for calculating the value deviation ε _fq , multiplication unit 72 for multiplying the integral value deviation ε _fq by the gain G, output G · ε _fq of the multiplication unit 72 and addition / subtraction of the proportional integration control unit 51 described above The switch 73 includes a switch 73 that selectively inputs one of the deviations εq calculated by the unit 62 to the integrating unit 66 and a switch controller 75 that controls switching of the switch 73.

Here, when the switch 73 inputs the deviation εq calculated by the addition / subtraction unit 62 to the integration unit 66, the vector angle control by the vector angle control unit 52 is not performed, but the normal proportional integration control is performed. When the switch 73 inputs G · ε _fq that is the output of the multiplication unit 72 to the integration unit 66, the proportional integration control is switched to the vector angle control by the vector angle control unit 52, and the vector angle control unit 52 The integral value fq, which is the calculation result of the integrator 66, is gradually guided to the integral value target value fq ^* by a loop that feeds back the integral value fq, and the angle of the combined vector of the voltage command values Vd and Vq of each phase of dq with respect to the q axis. To control.

Then, the vector angle final value θ _dq * obtained from the electrical angular velocity ω is obtained using the map shown in FIG. 6, and the length (Vd ² + Vq ²⁾ of the combined vector of the voltage command values Vd and Vq of each phase of dq is obtained. ) By setting the multiplication result of ^1/2 and cos θ _dq * to the integral value target value fq ^* , the vector angle of the combined vector can be controlled in the same manner as in the embodiment.

That is, since the q-phase current target value iq * is limited by the current limiting unit 22, the proportional path in the proportional-integral control unit 51 is small because the q-phase current value iq follows the q-phase current target value iq *. Since the value of the integration path in which the vector angle control is performed now dominates the q-phase voltage command value Vq, the q-phase voltage command value Vq is finally integrated. will be induced to a value desired value fq ^*, it can be controlled to a value targeting q-phase voltage command value Vq.

Actually, since it takes time to multiply (Vd ² + Vq ² ) ^1/2 · cos θ _dq *, the electrical angular velocity ω is used as a parameter (Vd ² + Vq ² ) ^1/2 · cos θ _dq *. Map calculation may be performed using the map. Therefore, for example, in the above power running state, if the electrical angular velocity ω is a large negative value, the vector angle is set to −90 degrees, so if the integral value target value fq ^* is set to 0, Good.

  Further, when the motor M generates torque that assists the rotation of the rotor R (powering state), field weakening control is performed in which the d-phase current target value id * output from the current target value calculation unit 21 is a negative value. carry out.

  When performing vector angle control in the above power running state, it is necessary to guide the vector angle θv of the combined vector to a negative angle. However, when control is performed to set the d-phase current target value id * to 0, actually There is a possibility that the deviation εd becomes positive due to the influence of the d-phase current id taking a negative value, the d-phase voltage command value Vd is induced in the positive direction, and the vector angle cannot be controlled as intended. Therefore, as described above, particularly in the powering state, the d-phase voltage command value Vd is obtained by performing field-weakening control in which the d-phase current target value id * output from the current target value calculation unit 21 is a negative value. A negative value can be set, and the vector angle θv of the combined vector can be controlled as intended.

Note that when the field weakening control is performed to induce the d-phase voltage command value Vd to be negative, the current target value calculation unit 21 sets the d-phase current target value id * to a negative value. If this value is negative and excessively large, the absolute value of the q-phase current iq cannot be maximized. Therefore, the d-phase current id that maximizes the q-phase current iq is a value −Φ / Ld obtained by dividing the maximum number of flux linkages Φ of the motor M by the d-phase inductance Ld. The output d-phase current target value id * is limited to -Φ / Ld by the current limiting unit 22a. Therefore, in the above power running state, the d-phase current target value id * is limited to -Φ / Ld, and this prevents a situation in which the absolute value of the q-phase current iq cannot take the maximum value. . Here, current limiting unit 22a limits q-phase current target value iq ^{* in the same} manner as in the above-described embodiment.

  In the braking state, when the current target value calculation unit 21 originally controls the d-phase current target value id * to be 0, the d-phase current value in a state where the combined vector of each dq phase is saturated. Since id takes a negative value, the d-phase deviation εd has a positive value, and the d-phase voltage command value Vd becomes a positive value. Therefore, particularly in such a braking state, the current target value calculation is performed. The unit 21 may not perform the field weakening control for setting the d-phase current target value id * to a negative value.

In addition, when the vector angle control is actually performed, when the absolute value of the electrical angular velocity ω is small, that is, when the rotational speed of the rotor R is small, the combined vector of the voltage command values Vd and Vq for each phase of dq is saturated. Since there are not many cases to do so, setting the integral target value fq ^* to 0 instead of making it variable as described above has no problem in practical use, and the calculation required for control becomes simple. The calculation cycle can be shortened.

Incidentally, when the integral value target value fq ^* is set to 0, the angle formed by the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq with respect to the q axis is 90 degrees and −90 degrees. Therefore, it is not necessary to describe the q-phase integral value target value fq ^* on the program for realizing the processing procedure of the motor control device 20, and the program can be simplified. .

In the above description, the integrated value fq is fed back, but the value finally output by the integration path, that is, the multiplication unit 68 outputs by the loop that feeds back the value fq · KI multiplied by the integral gain KI. Of course, the value fq · KI to be derived may be derived by multiplying the integral value target value fq ^* by the integral gain KI, and the multiplier 68 in the q-phase integral path is used to feed back the integral value. It also includes feeding back the value output by.

As described above, in the motor control device 50, the vector angle control unit 52 corrects the integral value fq by feeding back the integral value fq. Therefore, the integral value fq is a feedback loop of a first-order lag system. The integral value fq ^* is gradually changed depending on the time constant, and only the integral value fq is corrected without performing the correction process for the proportional path. Using the filter characteristics of the integration unit 66 in the current loop, the q-phase voltage command value Vq is not changed suddenly, the generated torque of the motor M is not changed suddenly, and the stroke of the actuator is There is no sudden change in speed.

  Further, since the q-phase voltage command value Vq does not change suddenly when the switch 73 is switched, there is no need to fade in and fade out the correction control when correcting the integral value fq, and the control algorithm is complicated. There is no problem that becomes.

  In other words, although it is possible to directly correct the q-phase voltage command value Vq and the value of the proportional path, it is necessary to fade in and fade out in the control. Since the change of the q-phase voltage command value Vq becomes abrupt, the control becomes complicated, but the motor control device 50 of the present embodiment does not have such a problem.

  Subsequently, switching of the switch 73 will be described. In the present embodiment, switching of the switch 73, that is, switching between proportional integral control and vector angle control is performed under the condition that the combined vector of the voltage command values Vd and Vq for each phase of dq is saturated.

  That is, until the combined vector of the voltage command values Vd and Vq for each phase of dq saturates, it is sufficient to perform proportional integral control. Therefore, even if the combined vector of the voltage command values Vd and Vq for each phase of dq is saturated. Since vector angle control may be performed to generate a large torque in the motor M, switching from proportional integral control to vector angle control is performed on condition that the combined vector of the voltage command values Vd and Vq of each phase of dq is saturated.

From the above, when normal proportional-integral control is possible, the switch 73 is set so that the deviation εq calculated by the adder / subtractor 62 is input to the integrator 66, and the voltage command value Vd for the d-phase and q-phase is supplied. , Vq satisfying the condition that the combined vector is saturated, the switch 73 may be switched so that G · ε _fq, which is the output of the multiplier 72, is input to the integrator 66.

  For this reason, the motor control device 50 according to the present embodiment includes a saturation determination unit 53 that determines whether or not the combined vector of the voltage command values Vd and Vq of each phase of dq is saturated. If the condition is satisfied as a result of the determination, the switch 73 is switched so that the switch controller 75 selects the vector angle control and switches from the proportional integral control to the vector angle control.

More specifically, the saturation determination unit 53 compares the square value of the combined vector length of the d-phase voltage command value Vd and the q-phase voltage command value Vq (Vd ² + Vq ² ) with the square value of the threshold value described above. Thus, it is determined whether the combined vector length of the d-phase voltage command value Vd and the q-phase voltage command value Vq exceeds a threshold value. In this case, as described above, by setting the threshold value to the saturation voltage Vs, it is determined whether the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq is saturated based on the above determination. Judgment can be made.

Thus, the square value of the combined vector length of the d-phase voltage command value Vd and the q-phase voltage command value Vq, that is, the square value of the d-phase voltage command value Vd and the square value of the q-phase voltage command value Vq. Is added to the sum (Vd ² + Vq ² ) and the square of the threshold, and the resultant vector length (Vd ² + Vq ² ) ^{1/2 of the} d-phase voltage command value Vd and the q-phase voltage command value Vq Therefore, it is not necessary to perform the root calculation for the calculation necessary for this determination, which can contribute to shortening the calculation time.

  In addition, instead of the above-described determination of saturation, when the q-phase current target value iq * exceeds the limit value Ilim, the q-phase current target value iq * is a value that cannot be taken by the q-phase current iq. Therefore, since the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq is saturated, the q-phase current target value iq * exceeds the limit value Ilim in this way. The saturation determination unit 53 may determine that the combined vector is saturated.

  The saturation determination unit 53 outputs 0 to the switch controller 75 when the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq is saturated, and 1 when it is not saturated. Output to the switch controller 75.

  On the other hand, as shown in FIG. 15, the switch controller 75 includes a negative operation unit 76 that negates the value output from the saturation determination unit 53, and the switch controller 75 directly uses the value output from the saturation determination unit 53 as q Multiplication unit 77 provided between addition / subtraction unit 62 for calculating deviation εq between phase current target value iq * and q phase current iq and switch 73 and multiplication provided before integration unit 65 in the d-phase integration loop. And outputs the value output from the negative operation unit 76 to the multiplication unit 78 provided between the switch 73 and the multiplication unit 72 in the middle of the feedback loop for correcting the integral value fq. The value output from the saturation determination unit 53 is output to the switch 73 as it is.

In this case, when the switch 73 receives a value less than 1 from the switch controller 75, G · ε _fq that is the output of the multiplication unit 72 is input to the integration unit 39 in order to validate the vector angle control. When a value of 1 or more is received from the switch controller 75, the deviation εq calculated by the adder / subtractor 62 is switched so as to be input to the integrator 66.

  When the saturation determination unit 53 determines that the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq is saturated and outputs 0, the switch controller 75 outputs 0 to the switch 73. Is output to the multipliers 77 and 80 and 1 is output to the multiplier 78.

Then, in order to select vector angle control, the switch 73 switches so that G · ε _fq that is the output of the multiplication unit 72 is input to the integration unit 66, and G · ε _fq that is the output of the multiplication unit 72 and Since 1 output from the switch controller 75 is input to the multiplication unit 78, G · ε _fq is input to the integration unit 66 as it is. In addition, since 0 and the deviation εd output from the switch controller 75 are input to the multiplication unit 80, 0 is input to the integration unit 65.

That is, in this case, since the vector angle control is performed, control for correcting the integral value fq is performed, and 0 is output to the multiplication unit 80. Therefore, the integral calculation in the d-phase current loop is stopped. Will be. The current target value calculation unit 21 calculates the d-phase current target value id ^* regardless of whether or not the saturation determination unit 53 saturates the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq. The field weakening control is performed to a negative value.

  On the other hand, when the saturation determination unit 53 determines that the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq is not saturated, the switch controller 75 outputs 1 to the switch 73. 1 is output to the multipliers 77 and 80, and 0 is output to the multiplier 78.

  Then, the switch 73 switches so that the deviation εq calculated by the addition / subtraction unit 62 is input to the integration unit 66 in order to select the proportional integration control, and the multiplication unit 77 outputs the deviation 1 from the switch controller 75. Since εq is input, the deviation εq is input to the integrating unit 66 as it is. Furthermore, since 1 and the deviation εd output from the switch controller 75 are input to the multiplication unit 80, the deviation εd is input to the integration unit 65 as it is.

  That is, in this case, there is no need to perform vector angle control by correcting the integral value fq, so that normal proportional-integral control is performed.

Further, in the vector angle control, the integration calculation in the d-phase current loop is stopped, and the q-phase integration value fq is induced to a small integration value target value fq ^*. The absolute values of the values fd and fq do not have large values, and the control response is not deteriorated when the proportional-integral control is restarted. Further, in the vector angle control in the present embodiment, the integral calculation in the d-phase current loop is stopped and the q-phase integral value fq is corrected. Therefore, the vector angle control is performed when switching to the proportional integral control. Without fading out or rewriting the integrated values fd and fq of each phase of dq, a large step does not occur in the voltage command values Vd and Vq of each phase of dq, and a large fluctuation occurs in the torque generated by the motor M. Is prevented.

  Thus, as described above, the saturation determination unit 53 determines whether the motor control device 50 should perform normal proportional-integral control or vector-angle control, and either the proportional-integral control or the vector-angle control. Since such control is selected, the motor M can be appropriately controlled.

  Further, by configuring the switch controller 75 as described above, it is possible to control the switch 73 by using only the logical operation of the value output from the saturation determination unit 53 without requiring high-level arithmetic processing, and to perform proportional integral control and vector processing. One of the angle controls can be selected.

  The motor control device 50 in the other embodiment also controls the angle of the combined vector of the voltage command values Vd and Vq for each phase of dq with respect to the q axis, similarly to the motor control device 20 in the one embodiment described above. Even when the combined vector of the voltage command values Vd and Vq of each phase of dq is saturated, the torque generated by the motor M can be appropriately controlled, and the motor M In addition, it is possible to generate a larger torque compared to the conventional motor control device.

  Further, even in this motor control device 50, since a larger torque can be generated by the motor M, the torque control range is increased, and the necessary thrust of the actuator can be obtained even if a smaller motor is used. Can be secured. In other words, a smaller motor can be used, the cost of the actuator can be reduced, and the mountability of the actuator in a device using the actuator can be improved.

  Further, in the motor control apparatus of this other embodiment, in the vector angle control, the d-phase and q-phase voltage command values Vd and Vq are independently adjusted without using the vector angle calculation which requires time for calculation. This makes it possible to control the vector angle of the combined vector of the voltage command values Vd and Vq for each phase of dq, so that the calculation cycle is shortened, the calculation load is reduced, and the practicality of the motor control device 50 is improved. Will improve.

  Subsequently, the hardware resources in each part other than the current detector 29, the rotation angle sensor 15, and the PWM circuit 30 of the motor control device 50 configured as described above are the same as those of the motor control device 20 described above, and the current target. Value calculation unit 21, current limiting unit 22, two-phase current calculation unit 23, proportional integral control unit 51, vector angle control unit 52, saturation determination unit 53, three-phase conversion calculation unit 27, limiter 28, switch 73 and switch controller 75 The processing procedure is stored in advance in a ROM or other storage device as a program, and these units are realized by the CPU reading the program and executing the above-described arithmetic processing.

  Here, a processing procedure in the motor control device 50 described above will be described based on a flowchart shown in FIG.

  First, in step S31, the motor control device 50 determines the current values iu and iv of any two phases of the three-phase windings 12, for example, the U phase and the V phase, the electrical angle θ and the electrical angular velocity ω of the motor M. Is read.

  Subsequently, the process proceeds to step S32, and the motor control device 50 calculates each current target value id *, iq *. If necessary, the current target value iq * is limited based on the electrical angular velocity ω. Further, the d-phase current target value id * is calculated by applying field-weakening control.

  Furthermore, in step S33, the motor control device 50 performs an operation for converting the current values iv and iu into d-phase and q-phase current values id and iq using the current values iu and iv and the electrical angle θ. Then, the process proceeds to step S34.

In step S34, the length of the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq was greater than or equal to the threshold value during the previous control, that is, the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq. It is determined with reference to the selection flag whether the length (Vd ² + Vq ² ) ^1/2 is equal to or greater than the threshold value. This threshold value is the same as that in the above-described embodiment. When the selection flag is 1, the combined vector length of the d-phase voltage command value Vd and the q-phase voltage command value Vq is equal to or greater than the threshold value, and when the selection flag is 0, the d-phase voltage command value Vd and It is determined that the combined vector length of q-phase voltage command value Vq is less than the threshold value. If the combined vector length of the d-phase voltage command value Vd and the q-phase voltage command value Vq is not greater than or equal to the threshold during the previous control, the process proceeds to step S35, and the d-phase voltage command value Vd and the q-phase voltage command are performed during the previous control. If the length of the combined vector of the value Vq is greater than or equal to the threshold value, the process proceeds to step S37.

  Subsequently, the process proceeds to step S35, and the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq is not equal to or greater than the threshold value during the previous control. The device 50 calculates the deviations εd and εq between the current target values id * and iq * and the d-phase and q-phase current values id and iq, calculates the integral values fd and fq, and proceeds to step S36. . In step S35, since normal proportional integration control is performed, the integral values fd and fq are calculated by calculating fd = fdpre + εd and fq = fqpre + εq, respectively. Here, as described above, fdpre and fqpre are respectively integrated values of the d-phase and the q-phase calculated during the previous control.

  In step S36, Vd = KI · fd + KP · εd is calculated to calculate the d-phase voltage command value Vd, and Vq = KI · fq + KP · εq is calculated to calculate the q-phase voltage command value Vq. Migrate to

On the other hand, in step S37, the motor control device 50 performs vector angle control in the current control because the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq is greater than or equal to the threshold value during the previous control. The integral value target value fq ^* is calculated.

Then, the process proceeds to step S38 to perform a process of correcting the integral value fq of the q-phase integration loop, and the integration calculation in the d-phase integration loop is stopped. Specifically, the integral value fq at the time of the current control is obtained by calculating fq = fqpre + G · _εfq , and the d-phase integrated value fd is obtained by calculating fd = fdpre.

  Then, the process proceeds to step S39, where dq phase voltage command values Vd and Vq are calculated, and the process proceeds to step S40. Specifically, Vd = KI · fd + KP · εd is calculated to calculate the d-phase voltage command value Vd, and Vq = KI · fq + KP · εq is calculated to calculate the q-phase voltage command value Vq.

  In step S40, the selection flag is set to 1 if the length of the combined vector of the voltage command values Vd and Vq for the dq phase is equal to or greater than the threshold value, and the selection flag is set to 0 otherwise. When the threshold value is the saturation voltage Vs, the combined vector of the d-phase voltage command value Vd and the q-phase voltage command value Vq is determined depending on whether or not the q-phase current target value iq * exceeds the limit value Ilim. It is possible to determine whether or not it is saturated, and the step S40 for setting the selection flag is provided between the step S32 and the step S33, and the voltage command value Vd for each phase of the dq at the previous control time. Instead of selecting either proportional-integral control or vector-angle control depending on whether the combined vector of Vq is saturated, one of proportional-integral control and vector-angle control is selected in this control routine. May be.

  In step S41, the motor control device 50 performs an operation of converting the d-phase voltage command value Vd and the q-phase voltage command value Vq into three-phase voltage command values Vu, Vv, and Vw, and proceeds to step S43. To do.

  In step S42, the motor control device 50 performs the restriction process when a process for restricting the voltage command values Vu, Vv, and Vw to a value that can be taken is necessary, and performs the restriction process when the process is not necessary. Without moving to step S43.

  In step S43, the motor control device 50 outputs the voltage command values Vu, Vv, Vw to the PWM circuit 30 and ends these series of processes.

  The motor controller 50 controls the motor M by repeatedly performing the above steps S31 to S43.

  Therefore, when the motor control device 50 executes the above-described series of processes, the current target value calculation unit 21, the current limiting unit 22, the two-phase current calculation unit 23, the proportional integration control unit 51, and the vector angle control unit 52 described above. , The saturation determination unit 53, the three-phase conversion calculation unit 27, the limiter 28, the switch 73, and the switch controller 75 are processed, whereby the dq respective phase voltage command values Vd and Vq are combined with respect to the q-axis. The angle can be controlled, and even when the combined vector of the voltage command values Vd and Vq of each phase of dq is saturated, the torque generated by the motor M can be controlled appropriately, and Further, it is possible to generate a larger torque in the motor M than in the conventional motor control device.

  Further, in the vector angle control of the motor control device 50 in this other embodiment, the input numerical value in the current loop of each phase of dq of the proportional integration control unit 51 is changed. At the time of switching between control and vector angle control, the voltage command values Vd and Vq of each phase of dq do not change abruptly without performing special processing, and there is no problem that the control becomes complicated.

  This is the end of the description of the embodiment of the present invention, but the scope of the present invention is of course not limited to the details shown or described.

It is a key map of an actuator to which a motor control device in one embodiment is applied. It is a figure which shows the change of the vector angle of the synthetic | combination vector of the voltage command value of dq each phase which makes q-axis current the largest with respect to electrical angular velocity. (A) is a graph which shows the change of the vector angle of the synthetic | combination vector of the voltage command value of dq each phase with respect to the stroke displacement of an actuator at the time of setting a d-phase electric current to zero and controlling a motor so that integration is stopped at the time of saturation. . (B) is a graph showing a change in the vector angle of the combined vector of the voltage command values of each phase of dq with respect to the stroke displacement of the actuator when the motor is controlled by performing the field weakening control and the integration is stopped at the time of saturation. It is a system diagram of a motor control device in one embodiment. It is a figure which shows a PWM circuit. (A) is a figure which shows the map of the vector angle final value created using the electrical angular velocity used as a parameter when generating the torque of a reverse rotation direction in a motor. (B) is a figure which shows the map of the vector angle final value produced using the electrical angular velocity used as a parameter when generating the torque of a normal rotation direction in a motor. It is a table | surface for determining the rotation direction with respect to the q-axis of the synthetic | combination vector of the voltage command value of dq each phase. It is a figure explaining the rotation direction of the synthetic | combination vector in case q phase current target value is positive. It is a figure explaining the rotation direction of the synthetic | combination vector in dq orthogonal coordinate when q phase current target value is positive. It is a figure explaining the rotation direction of the synthetic | combination vector in case q phase current target value is negative. It is a figure explaining the rotation direction of the synthetic | combination vector in dq orthogonal coordinates when q phase current target value is negative. It is a figure which shows the map of the limit value which restrict | limits the q phase current target value produced using electrical angular velocity as a parameter. It is a flowchart which shows the process sequence in the motor control apparatus of one Embodiment. It is a system diagram of the motor control device in other embodiments. It is a system diagram of the switch controller in the selection means. It is a flowchart which shows the process sequence in the motor control apparatus of other embodiment.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Pinion gear which is a rotating member 2 Rack shaft 10 which is a linear motion member Frame 11 Stator core 12 Winding 13 Shaft 14 Driving magnet 15 Rotation angle sensor 20, 50 Motor control device 21 Current target value calculating unit 22, 22a Current limiting unit 23 Phase current calculation unit 24, 51 Proportional integral control unit 25, 52 Vector angle control unit 26, 53 Saturation determination unit 27 Three-phase conversion calculation unit 28 Limiter 29 Current detector 30 PWM circuit 30a Switching element 61, 62, 71 Addition / subtraction unit 63 , 64, 67, 68, 72, 77, 78 Multiplying unit 65, 66 Integration unit 69, 70 Addition unit 73 Switch 75 Switch controller 76 Negative operation unit 80 Multiplying unit E Power supply H Motion conversion mechanism M Motor R Rotor S Stator

Claims (41)

In a motor control device that obtains d-phase and q-phase current values by dq conversion to be converted into dq orthogonal coordinates and performs current loop processing for each dq phase to control the drive current of the motor, A motor control device comprising vector angle control means for controlling a vector angle of a combined vector. 2. The motor control device according to claim 1, further comprising a selection unit that selects either proportional-integral control based on a current control deviation of each phase of dq or vector angle control, and performs control selected by the selection unit. 3. The motor control apparatus according to claim 2, wherein the selection means switches to vector angle control when the combined vector length of the voltage command value of each of the dq phases is larger than a threshold value. 4. The motor control device according to claim 1, wherein the vector angle control means obtains a voltage command value for each phase of dq from the vector angle target value and the combined vector length. The vector angle control means determines the rotation direction of the combined vector from the q-phase current control deviation and the angle control deviation of the vector angle, and sets the vector angle target value from the rotation direction. 4. The motor control device according to any one of 4. 6. The motor control device according to claim 1, wherein the angle control deviation is determined based on a vector angle final value that maximizes an absolute value of the q-phase current and a deviation of the vector angle. 7. The motor control device according to claim 6, wherein the vector angle control means obtains a final vector angle value with reference to a vector angle map created in advance using the electrical angular velocity as a parameter. 4. The motor control device according to claim 1, wherein the vector angle control unit corrects an integral value in the q-phase current loop to control a vector angle of the combined vector. 5. 9. The motor control device according to claim 8, wherein the vector angle control means feeds back an integral value in the q-phase current loop to correct the integral value. 10. The motor according to claim 8, wherein the vector angle control means obtains a q-phase integral value target value from the electrical angular velocity, and corrects the q-phase integral value to become the q-phase integral value target value. Control device. The vector angle control means obtains the q-phase integral value target value from the electrical angular velocity, and obtains a value obtained by multiplying the integral difference between the q-phase integral value target value and the q-phase integral value by the correction gain. 11. The motor control device according to claim 8, wherein the motor control device corrects the q-phase integral value by feedback to the input. The vector angle control means sets the d-phase current target value to a negative value when the rotor is rotating forward and the motor is generating torque that assists the rotor in rotating forward. The motor control device according to any one of 11. The motor control device according to any one of claims 1 to 12, further comprising q-phase current limiting means for limiting the q-phase current target value to a limit value depending on an electrical angular velocity. The motor control device according to claim 13, wherein the limit value is set to a maximum value that can be taken by an absolute value of the q-phase current. 15. The d-phase current limiting means for limiting the absolute value of the d-phase current target value to a value obtained by dividing the maximum number of interlinkage magnetic fluxes by the d-phase inductance is provided. Motor control device. The motor control device according to claim 3, wherein the threshold value is set to a value obtained by multiplying the power supply voltage by 1 / √2. The motor control device according to claim 3, wherein the threshold value is set to a value obtained by multiplying the power supply voltage by √6 / 4. The motor control device according to claim 3, wherein the threshold value is set to a value that does not saturate the PWM opening degree. After switching from the control by the vector angle control means to the control by the proportional integration control means, the voltage command value of each dq phase output for the first time by the proportional integration control means is the voltage command value of each dq phase output by the vector angle control means before switching. The motor control device according to claim 1, wherein the motor control device has the same value as the value. The integral value of the dq phase of the integral loop in the proportional integral control means is the voltage command value of each dq phase that the proportional integral control means outputs for the first time after switching from the control by the vector angle control means to the control by the proportional integral control means. The motor control device according to any one of claims 1 to 18, wherein the control means is set to have the same value as the voltage command value of each phase of dq outputted before switching. 21. A motion conversion mechanism comprising a rotating member and a linear motion member that exhibits a linear motion by the rotating motion of the rotating member, a motor that is coupled to the rotating member and applies torque to the rotating member, and Actuator with a motor control device. In a motor control method in which current values of d and q phases are obtained by dq conversion to be converted into dq orthogonal coordinates and current loop processing is performed for each dq phase to control the motor drive current, the voltage command value of each phase of dq A motor control method comprising a vector angle control step of controlling a vector angle of a composite vector. 23. The motor control method according to claim 22, comprising a selection step of selecting either proportional-integral control or vector angle control based on a current control deviation of each of the dq phases, and performing the selected control. 24. The motor control method according to claim 23, wherein, in the selection step, the vector angle control step is selected when the length of the voltage command value combined vector of each phase of dq is equal to or greater than a threshold. 25. The motor control method according to claim 22, wherein in the vector angle control step, a voltage command value for each phase of dq is obtained from the vector angle target value and the combined vector length. 23. The vector angle control step includes determining a rotation direction of the combined vector from the q-phase current control deviation and the vector angle angle control deviation, and setting a vector angle target value from the rotation direction. The motor control method according to claim 25. 27. The motor control method according to claim 22, wherein the angle control deviation is determined based on a vector angle final value that maximizes an absolute value of the q-phase current and a vector angle deviation. 28. The motor control method according to claim 27, wherein, in the vector angle control step, the final value of the vector angle is obtained by referring to a vector angle map created in advance using the electrical angular velocity as a parameter. 25. The motor control method according to claim 22, wherein in the vector angle control step, the vector angle of the combined vector is controlled by correcting the integral value in the q-phase current loop. 30. The motor control method according to claim 29, wherein in the vector angle control step, the integral value in the q-phase current loop is fed back to correct the integral value. 31. The motor according to claim 29 or 30, wherein, in the vector angle control step, a q-phase integral value target value is obtained from an electrical angular velocity, and the q-phase integral value is corrected to become a q-phase integral value target value. Control device. In the vector angle control step, the q-phase integral value target value is obtained from the electrical angular velocity, and the value obtained by multiplying the q-phase integral value target value and the integral value deviation of the q-phase integral value by the correction gain is set in the integrator in the q-phase current loop. 32. The motor control method according to claim 29, wherein feedback is made to the input to correct the q-phase integral value. The motor control method according to any one of claims 29 to 32, wherein the d-phase current target value is set to a negative value in the vector angle control step. 34. The motor control method according to claim 22, further comprising a q-phase current limiting step of limiting the q-phase current target value to a limit value depending on the electrical angular velocity. The motor control method according to claim 34, wherein the limit value is set to a maximum value that can be taken by an absolute value of the q-phase current. 36. The motor according to claim 29, further comprising a d-phase current limiting step for limiting the absolute value of the d-phase current target value to a value obtained by dividing the maximum number of flux linkages by the d-phase inductance. Control method. 37. The motor control method according to claim 24, wherein the threshold value is set to a value obtained by multiplying the power supply voltage by 1 / √2. 37. The motor control method according to claim 24, wherein the threshold value is set to a value obtained by multiplying the power supply voltage by √6 / 4. The motor control method according to any one of claims 24 to 38, wherein the threshold value is set to a value at which the PWM opening is not saturated. After switching from vector angle control to proportional-integral control, the voltage command value for each phase of dq output in the proportional-integral control step becomes the same value as the voltage command value for each phase of dq output before switching in the vector angle control step. The motor control method according to any one of claims 22 to 39, further comprising an adjustment step of adjusting the frequency to the above. In the adjustment step, the voltage command value of each phase of dq that the proportional integration control step outputs for the first time after switching from vector angle control to proportional integration control is the same as the voltage command value of each phase of dq that is output before switching by the vector angle control step. 41. The motor control device according to claim 40, wherein an integration value of the dq phase of the integration loop in the proportional integration control step is set so as to be a value.

Images