JP2007325408A - Electrically powered motor controller and electrically powered steering system using the same - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide an electrically powered controller which secures stability while keeping torque constant by suppressing a phase lag and attenuation of a harmonic component in a current control system. <P>SOLUTION: The electrically powered controller includes a motor control means 20 having the current control system which controls an electric motor 12 based on a current command value and a motor current. The controller includes a motor angular velocity detecting means 34 to detect motor angular velocity of the electric motor 12, and the motor control means 20 is constituted so that it may calculate the current command value which keeps the torque constant in consideration of characteristics of the current control system, based on the motor angular velocity detected by the motor angular velocity detecting means. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to an electric motor control device including motor control means having a current control system for controlling an electric motor based on a current command value and a motor current, and an electric power steering device using the same.

As an electric motor control device that suppresses torque fluctuation, a command value of compensation current for suppressing electrical ripple caused by distortion of the induced electromotive force waveform of the motor is shown as a relationship between compensation current per unit torque and electrical angle. Based on the compensation current map and the frequency characteristic map indicating the frequency characteristics of the current control system of the motor, the current compensation values Δi _q0 and Δi per unit torque based on the motor electrical angle θmre corrected to compensate for the phase delay in the current control system. _The current compensation values Δi _q1 and Δi _d1 are calculated by determining _d0 and multiplying these compensation values by a coefficient corresponding to the q-axis basic current command value i ^* _q0 so that the amplitude of the compensation current is proportional to the motor load. Further, by multiplying the compensation value by the correction rate Rm to compensate for the gain reduction in the current control system, the q-axis and d-axis currents are used as compensation current command values. .Delta.i _q, electric power steering apparatus to calculate the .delta.i _d has been proposed (e.g., see Patent Document 1).

The detected motor current is converted into an n-th order dq-axis signal and an m-th order dq-axis signal by a coordinate conversion unit, and then the direct current components are extracted by a low-pass filter, and the deviation between the direct current components and the command value is extracted. A motor control device has also been proposed in which the motor current control is performed by converting the signal into a three-phase AC signal again (see, for example, Patent Document 2).
JP 2004-328814 A JP 2005-328691 A

However, in the conventional example described in Patent Document 1, each phase parameter is converted to the dq coordinate on the assumption that the parameters of each phase are balanced, so that there are resistance fluctuations and variations in each phase of the motor. There is an unsolved problem that torque ripple occurs and steering feeling deteriorates.
Moreover, in the conventional example described in Patent Document 2, since the dq conversion of each order component of the harmonic is performed, the calculation amount is very large, the load on the arithmetic processing unit is increased, and the low-pass filter is used. Since the DC component of each secondary component is extracted using a low-pass filter, the response characteristic of the control system is reduced due to a response delay caused by the low-pass filter, and stability can be maintained when the electric motor is operated at a low speed. For example, when an electric power steering apparatus has a wide operating range from a low speed region to a high speed region, there is an unsolved problem that stability cannot be guaranteed.

In addition, amplitude reduction and phase delay due to responsiveness of the current control system that performs PI control or the like based on the current command value and the motor current detection value cannot be avoided. Has an unresolved problem that it attenuates and causes output reduction and torque fluctuation.
Therefore, the present invention has been made paying attention to the unsolved problems of the above-described conventional example, and suppresses phase delay and harmonic component attenuation in the current control system to ensure stability while keeping the torque constant. It is an object of the present invention to provide an electric motor control device that can be used and an electric power steering device using the same.

  In order to achieve the above object, an electric motor control device according to claim 1 is an electric motor control device including motor control means having a current control system for controlling an electric motor based on a current command value and a motor current. Motor angular velocity detection means for detecting the motor angular velocity of the electric motor, and the motor control means is configured to maintain constant torque in consideration of the characteristics of the current control system based on the motor angular speed detected by the motor angular velocity detection means. It is comprised so that the electric current command value which becomes may be calculated.

  According to a second aspect of the present invention, there is provided an electric motor control device comprising: a motor control means having a current control system for controlling a multiphase electric motor having an induced voltage including high-order harmonics based on a current command value and a motor current. The electric motor control device is provided, wherein the motor control means sets the advance amount and the amplitude compensation value of the current command value of each harmonic component so that the torque is constant in consideration of the characteristics of the current control system. Current command value compensation means is provided for calculating and compensating the current command value of the multiphase electric motor based on the calculated advance angle amount and amplitude compensation value.

  In the invention according to claim 2, the current command value compensation means calculates the advance amount and the amplitude compensation value of the current command value of each harmonic component so that the torque is constant in consideration of the characteristics of the current control system. Since the current command value of the multiphase electric motor is compensated based on the calculated advance amount and amplitude compensation value, the output can be controlled as designed, and torque fluctuations and operating noise can be reduced.

  Furthermore, the electric motor control device according to claim 3 includes a motor control means having a current control system for controlling a multiphase electric motor having an induced voltage including high-order harmonics based on the current command value and the motor current. An electric motor control device comprising: motor angular velocity detection means for detecting a motor angular velocity of the electric motor, wherein the motor control means is configured on the dq coordinate rotating at a frequency corresponding to the motor angular velocity. The advance angle of the current command value in each harmonic component of the dq coordinate of the multiphase motor so that the torque is constant based on the motor angular speed detected by the motor angular speed detecting means in consideration of the characteristics of the current control system. A current command value compensation means for computing the amount and amplitude compensation value and compensating the current command value of the multiphase electric motor based on the computed advance angle amount and amplitude compensation value; It is characterized.

  In the invention according to claim 3, the current command value compensation means obtains the characteristics of the current control system based on the motor angular speed detected by the motor angular speed detection means on the dq coordinate rotating at a frequency corresponding to the motor angular speed. The advance amount and amplitude compensation value of the current command value in each harmonic component of the dq coordinate of the polyphase motor are calculated so that the torque is constant in consideration, and the calculated advance angle amount and amplitude compensation value are calculated. Since the current command value compensation means for compensating the current command value of the multiphase electric motor is provided based on the dq coordinate, the current is compensated on the dq coordinate, the output is controlled according to the design value, torque fluctuation and The operating noise can be reduced.

Furthermore, the electric motor control device according to a fourth aspect is the invention according to the second or third aspect, wherein the current command value compensation means includes a motor angular velocity and an advance angle that suppress attenuation of harmonic components in the current control system. A control map representing a relationship between the amount and the amplitude compensation gain, wherein the advance angle amount and the amplitude compensation gain are calculated with reference to the control map based on the motor angular velocity. Yes.
Still further, an electric power steering apparatus according to claim 5 controls an electric motor that generates a steering assist force for the steering mechanism based on the steering torque by the electric motor control apparatus according to any one of claims 1 to 4. It is characterized by doing so.

  According to the first aspect of the present invention, the current command value for constant torque is calculated based on the motor angular speed in consideration of the characteristics of the current control system, so that the actual motor current and the ideal current actually supplied to the electric motor are Can be expected to achieve the expected output and reduce torque fluctuation and operating noise. For example, when applied to an electric power steering device, good steering performance and steering feeling can be obtained. The effect that it can be obtained is acquired.

According to the invention of claim 2, the current command value compensation means calculates the advance amount and amplitude compensation value of the current command value of each harmonic component so that the torque is constant, and the calculated advance angle Since the current command value of the multiphase electric motor is compensated based on the amount and the amplitude compensation value, the output can be controlled as designed, and torque fluctuation and operating noise can be reduced.
Further, according to the invention of claim 3, the current command value compensating means makes the torque constant on the dq coordinate rotating at the frequency corresponding to the motor angular speed based on the motor angular speed detected by the motor angular speed detecting means. The advance amount and amplitude compensation value of the current command value in each harmonic component of the dq coordinate of the multiphase motor are calculated so that the multiphase electric motor is based on the calculated advance amount and amplitude compensation value. Since current command value compensation means for compensating the current command value of the motor is provided, current compensation is performed on the dq coordinate to control the output according to the design value, and to reduce torque fluctuation and operating noise. Can do.

Furthermore, according to the fourth aspect of the present invention, the advance amount and the amplitude compensation gain are calculated with reference to the control map based on the motor angular velocity. Therefore, the advance amount and amplitude in consideration of the characteristics of the current control system. The compensation gain can be calculated accurately and easily, and good current compensation control can be performed.
Furthermore, according to the fifth aspect of the invention, since it is applied as an electric motor control device of an electric power steering device, it is possible to generate an expected steering assist force and to reduce torque fluctuations and operating noise. And good steering performance and steering feeling can be obtained.

DESCRIPTION OF THE PREFERRED EMBODIMENTS Embodiments of the present invention applied to an electric power steering device will be described below with reference to the drawings.
FIG. 1 is an overall configuration diagram showing an embodiment in which the present invention is applied to an electric power steering apparatus. In the figure, reference numeral 1 denotes a steering wheel, which is applied to the steering wheel 1 from a driver. A steering force is transmitted to a steering shaft 2 having an input shaft 2a and an output shaft 2b. The steering shaft 2 has one end of the input shaft 2a connected to the steering wheel 1 and the other end connected to one end of the output shaft 2b via a steering torque sensor 3 as steering torque detecting means.

  The steering force transmitted to the output shaft 2 b is transmitted to the lower shaft 5 via the universal joint 4 and further transmitted to the pinion shaft 7 via the universal joint 6. The steering force transmitted to the pinion shaft 7 is transmitted to the tie rod 9 via the steering gear 8 and steers steered wheels (not shown). Here, the steering gear 8 is configured in a rack and pinion type having a pinion 8a connected to the pinion shaft 7 and a rack 8b meshing with the pinion 8a, and the rotational motion transmitted to the pinion 8a is linearly moved by the rack 8b. It has been converted to movement.

A steering assist mechanism 10 for transmitting a steering assist force to the output shaft 2b is connected to the output shaft 2b of the steering shaft 2. The steering assist mechanism 10 includes a reduction gear 11 connected to the output shaft 2 b and a three-phase brushless motor 12 that generates a steering assist force connected to the reduction gear 11.
The steering torque sensor 3 detects the steering torque applied to the steering wheel 1 and transmitted to the input shaft 2a. For example, the steering torque sensor 3 is a torsion bar (not shown) in which the steering torque is interposed between the input shaft 2a and the output shaft 2b. The torsional angular displacement is converted into a torsional angular displacement, and the torsional angular displacement is converted into a resistance change or a magnetic change to be detected.

  Further, as shown in FIG. 2, the three-phase brushless motor 12 is connected to one end of the U-phase coil Lu, the V-phase coil Lv and the W-phase coil Lw to form a star connection, and each of the coils Lu, Lv and Lw The other end is connected to a steering assist control device 20 as motor control means, and motor drive currents Iu, Iv and Iw are individually supplied. In addition, the three-phase brushless motor 12 includes a rotor position detection circuit 13 including a resolver, an encoder, and the like that detect the rotational position of the rotor.

  As shown in FIG. 2, the steering assist control device 20 receives the steering torque T detected by the steering torque sensor 3 and the vehicle speed detection value Vs detected by the vehicle speed sensor 21 and is detected by the rotor position detection circuit 13. Is output from the motor current detection circuit 22 that detects the motor drive currents Iu, Iv, and Iw supplied to the phase coils Lu, Lv, and Lw of the three-phase brushless motor 12. Drive current detection values Iud, Ivd, and Iwd are input.

  The steering assist control device 20 calculates a steering assist target current value based on the steering torque T, the vehicle speed detection value Vs, the motor current detection values Iud, Ivd and Iwd, and the rotor rotation angle θ, and the motor voltage command value Vu. , Vv and Vw, a control arithmetic unit 23 for outputting the motor, a motor drive circuit 24 composed of a field effect transistor (FET) for driving the three-phase brushless motor 12, and a phase voltage command value Vu output from the control arithmetic unit 23 , And an FET gate drive circuit 25 for controlling the gate current of the field effect transistor of the motor drive circuit 24 based on Vv and Vw.

As shown in FIG. 3, the control arithmetic unit 23 uses a current control system, that is, a current control unit to drive the three-phase brushless motor 12 so as not to cause torque fluctuations by utilizing the excellent characteristics of vector control. In consideration of the attenuation of each harmonic component at 40, vector control d of each harmonic component including the fundamental wave component, advance of the q component and amplitude compensation are performed, and the target current value i _d ′, i After determining _q ′, these target current values i _d ′, i _q ′ are converted into respective phase target current values Iu ^* , Iv ^* and Iw ^* corresponding to the respective excitation coils Lu to Lw, and are output. Unit 30, current feedback processing using the phase current command values Iu ^* , Iv ^* and Iw ^* output from the target current setting unit 30 and the motor current detection values Iud, Ivd and Iwd detected by the motor current detection circuit 22 To go phase Pressure command value Vu, the Vv and Vw and a current control unit 40 outputs to the FET gate driving circuit 25.

The target current setting unit 30 is configured as shown in FIG. That is, the target current setting unit 30 receives the steering torque T detected by the steering torque sensor 3 and the vehicle speed detection value Vs detected by the vehicle speed sensor 21, and calculates the steering assist current command value I _ref based on these inputs. the assist current command value calculating section 31, and the electrical angle converting portion 32 for converting the rotor rotating angle theta detected by the rotor rotation angle detecting circuit 13 to an electrical angle theta _e, the electrical angle theta output from the electrical angle converting portion 32 a differentiating circuit 33 which calculates the electrical angular speed omega _e by differentiating the _e, as the motor angular velocity detection means for calculating a motor angular speed omega _m by dividing the electrical angular velocity omega _e outputted from the differentiation circuit 33 in the motor pole pairs p The three-phase brushless motor 12 based on the steering auxiliary current command value I _ref and the motor angular velocity ω _m output from the motor angular velocity conversion unit 34 and the steering auxiliary current command value calculation unit 31. In a rotating dq-axis orthogonal coordinate system in which the direction of the exciting current component of the flowing current is set to the d-axis and the direction of the torque current component is set to the q-axis orthogonal to the d-axis, the d-axis current command value DC component I _dDC , 6n-order current command value component amplitudes i _{d6 to} i _d6n , d-axis current command value I _d , q-axis current command value DC component I _qDC , 6n-order current command value component amplitudes i _q6 to a current command value limiting unit 35 for calculating a d-axis current command value I _q composed of i _q6n , and a d-axis current command value I _d and a q-axis current command value I output from the current command value limiting unit 35 Compensation d-axis current command value i _d ′, compensated q-axis current by performing advance angle and amplitude compensation of each harmonic component including fundamental wave component based on _q , motor angular velocity ω _m and electrical angle θ _e Current command to calculate command value i _q ′ and electrical angle θ _e ′ after advance Each harmonic component advance angle / amplitude compensation calculation unit 36 serving as a value compensation means, and a post-advanced d-axis current command value based on the post-advance electrical angle θ _e ′ calculated by the advance angle / amplitude compensation calculation unit 36 A two-phase / three-phase converter 37 that converts i _d ′ and the compensated q-axis current command value i _q ′ into three-phase current command values Iu ^* , Iv ^*, and Iw ^* is provided.

The steering assist current command value calculator 31 described above calculates the steering assist current command value I _ref with reference to the steering assist current command value calculation map shown in FIG. 4 based on the steering torque T and the vehicle speed detection value Vs. Here, as shown in FIG. 4, the steering assist current command value calculation map takes the steering torque T on the horizontal axis, the steering assist current command value I _ref on the vertical axis, and the vehicle speed detection value Vs as a parameter. It consists of a characteristic diagram represented by a parabolic curve. The steering assist current command value I _ref maintains “0” when the steering torque T is from “0” to the set value Ts1 in the vicinity thereof. When the steering torque T exceeds the set value Ts1, the steering assist current is initially set. Although the command value I _ref increases relatively gradually as the steering torque T increases, the steering assist current command value I _ref is set to increase steeply as the steering torque T further increases. A plurality of characteristic curves are set so that the inclination becomes smaller as the vehicle speed increases.

As shown in FIG. 5, the current command value limiter 35 receives the motor angular velocity ω _m output from the motor angular velocity converter 34 and the steering assist current command value I _ref calculated by the steering assist current command value calculator 31. And a command value for generating a d-axis command value and a q-axis command value on dq coordinates for limiting the upper limit value of the steering assist current command value I _ref based on the motor angular velocity ω _m and making the motor torque constant. A limiting unit 41 is included.
The current command value limiting unit 35 includes a d-axis current command value DC component calculation unit 42 that calculates a d-axis current command value DC component I _dDC based on the limited command value output from the command value limiting unit 41. Similarly, a q-axis current command value DC component calculation unit 43 that calculates a q-axis current command value DC component I _qDC based on the limited command value output from the command value limiting unit 41 is provided.

Further, the current command value limiting unit 35 is configured to output the d-axis current command value DC component I _dDC calculated by the d-axis current command value DC component calculation unit 42 and the q-axis current command value calculated by the q-axis current command value DC component calculation unit 43. A d-axis current command value 6n-order component amplitude calculation unit 44 for calculating the amplitudes i _{d6 to} i _d6n of the d-axis current command value 6n-order component based on the value DC component I _qDC , and similarly the d-axis current command value DC component I a q-axis current command value 6n-order component amplitude calculation unit 45 that calculates the amplitudes i _{q6 to} i _q6n of the q-axis current command value 6n-order component based on the _dDC and the q-axis current command value DC component I _qDC .

Then, the d-axis command value DC component I _dDC and the d-axis command value 6n-order components i _{d6 to} i _d6n output from the d-axis current command value DC-component calculation unit 42 and the d-axis current command value 6n-order component amplitude calculation unit 44. Is output as a d-axis current command value I _{d in the} form represented by the following equation (1).
Also, the q-axis command value DC component I _qDC and the q-axis command value 6n-order component i _{q6 to} i _q6n output from the q-axis current command value DC-component calculation unit 43 and the q-axis current command value 6n-order component amplitude calculation unit 45. Is output as the q-axis current command value I _{q in the} form represented by the following equation (2).
I _d = [I _dDC i _d6 ...... i _d6n ] (1)
I _q = [I _qDC i _q6 ... I _q6n ] (2)
Here, in the current command value limiter 35, the d-axis command value DC component I _dDC , the d-axis command value 6n-order components i _{d6 to} i _d6n, and the q-axis command value DC component I _qDC , q-axis command that make the motor torque constant The calculation of the value 6n-order components i _{q6 to} i _q6n is performed as follows.

That is, a constant torque equation from the motor energy balance equation can be expressed by the following equation (3).
(2/3) K _t I _ref ω _m = i _q e _q + i _d e _d (3)
Where K _t is a torque constant, I _ref is a current command value from the torque system, ω _m is a motor rotation speed (mechanical angle), i _q is a q-axis current, _id is a d-axis current, and e _q is a q-axis motor. The induced voltage, _ed, is a d-axis motor induced voltage.

Here, the motor induced voltage _{e q} and _{e d} is the speed sensitive, when the induced voltage when 1 rad / sec of the dq axis _{E d} and _{E q,}
e _q = E _q ω _m (4)
e _d = E _d ω _m (5)
Therefore, if you substitute these into equation (3) and rewrite them,
i _q = {(2/3) K _t I _{ref −E d} i _d } / E _q (6)
It becomes.

Induced voltage e _a motor, for example a phase if it contains higher components, represented by the following equation (7).
e _a = E _a ω _m = E ₁ ω _m sin (ω _e t) + E ₃ ω _m sin (3ω _e t) + E ₅ ω _m sin (5ω _e t)
+ E ₇ ω _m sin (7ω _e t) ＋ ……………… （7）
Where E _a is the induced voltage of the a phase at 1 rad / sec, E ₁ , E ₃ , E ₅ , E ₇ ...... is the induction of 1, 3, 5, 7, ... components at 1 rad / sec The voltage coefficient, ω _e, is the motor electrical angular velocity, and the b phase and c phase are omitted because they are only 120 ° out of phase.

When the phase induced voltage e _{a in} the above equation (7) is dq converted, the induced voltages E _d and E _q are omitted for the 9th order or more in order to simplify the explanation.
E _d = (E ₅ + E ₇ ) sin6ω _e t (8)
E _q = E ₁ − (E ₅ + E ₇ ) cos 6ω _e t (9)
It becomes.

As described above, the current command value I _ref is calculated by the steering assist current command value calculation unit, and the DC component I _dDC of the d-axis current is calculated based on the current command value I _ref ′ after the current limit, as shown in FIG. Calculated with reference to the map.
And when considering the AC d-axis, the d-axis current i _d is omitted if the 12th order or more is omitted in order to simplify the explanation.
i _d = I _dDC + (i _dc6 cos6ω _e t−i _ds6 sin6ω _e t) (10)
It expresses. Here, i _dc and i _ds are the cosine and sine component sixth-order d-axis currents.

Then, substituting the equations (8) to (10) into the equation (6), Taylor expansion of 1 / E _q , and omitting the 12th order or higher,
i _q = I _qDC + (i _qc6 cos6ω _e t−i _qs6 sin6ω _e t) (11)
Is obtained.
However, I _qDC = (2/3) K _t I _ref (12)
i _qc6 = (E ₅ −E ₇ ) I _qDC / E ₁ (13)
i _qs6 = (E ₅ + E ₇ ) _{Id DC} / E ₁ (14)
Since the 6th-order d-axis depends on the 6th-order q-axis,
i _dc6 ∝i _qc6 ………… (15)
i _ds6 ∝i _qs6 ………… (16)
It becomes.

Accordingly, the d-axis current command value DC component calculation unit 42 calculates the d-axis current command value DC component I _dDC with reference to the DC component calculation map of FIG. 6 based on the limited current command value I _ref ′, and the q-axis The current command value DC component calculation unit 43 calculates the q-axis current command value DC component I _qDC by performing the calculation of the equation (12), and the d-axis current command value 6n-order component amplitude calculation unit 44 and the q-axis current command. The value 6n-order component amplitude calculation unit 45 calculates the 6th-order component amplitudes i _d6 and i _q6 based on the equations (13) to (16), and the 12th-order or higher 6n-order component amplitudes i _{d12 to} i _d6n and i _{q12 to} i _q6n are calculated.

Further, as shown in FIG. 7, the advance angle and amplitude compensation calculation unit 36 includes a fundamental wave component advance amount calculation unit 51 that performs an advance calculation of the fundamental wave component based on the electrical angle θ _e and the motor angular velocity ω _m. I have.
Further, the advance angle and amplitude compensation calculation unit 36, based on the d-axis current command value Id and the motor angular speed omega _m, d-axis current command NeSusumu for calculating the advance amount phi _d6 to [phi] _D6N of d-axis current command value Angular amount calculation unit 52, 6n-order d-axis current command value amplitude compensation gain calculation unit 53, q-axis current command for calculating amplitude compensation gains G _{d6 to} G _d6n of amplitudes i _{d6 to} i _d6n of 6n-order d-axis current command values calculating the amplitude compensation gain _G q6 _{~G Q6n} amplitude _i q6 _{through i Q6n} of advance amount phi _q6 to [phi] q-axis current command value _Q6n calculates the advance angle calculation section 54 and 6n following q-axis current command value value Iq 6nth order q-axis current command value amplitude compensation gain calculating section 55 is provided.

Furthermore, the advance and amplitude compensation calculation unit 36 includes a DC current command value amplitude compensation gain calculation unit 56 that calculates a DC current command value amplitude compensation gain Gdc based on the motor angular velocity ω _m .
Whether the d-axis current command value advance amount calculation unit 52 is in a fundamental wave advance angle control state in which only the fundamental wave component that continues “0” is advanced based on the d-axis current command value Id. It is determined whether or not it is a full advance control state in which the fundamental wave component in which the d-axis current command value Id changes includes the 6n-order harmonic component. When the result of this determination is a fundamental wave advance angle control state in which the d-axis current command value Id continues “0”, taking the advance amount of the sixth-order d-axis current command value as an example, FIG. as shown in the motor when a predetermined positive advance amount phi _d1 becomes regardless of the value of the motor angular speed omega _m when the motor angular speed omega _m is a positive value including "0", the motor angular speed omega _m is a negative value Regardless of the value of the angular velocity ω _m, an advance amount calculation map that becomes a negative predetermined advance amount −φ _d1 is used, and the advance amount Φ _d6 is referred to by referring to the advance amount calculation map based on the motor angular velocity ω _m. Similarly, the advance amount of the 6n-order d-axis current φ _d12 , φ _d18 ... Φ _d6n is calculated with reference to an advance amount calculation map (not shown).

Further, when the determination result is a full advance control state in which the d-axis current command value Id changes, taking the advance amount of the sixth d-axis current command value as an example, as shown in FIG. In addition, when the motor angular velocity ω _m increases from “0” to a positive value, when the motor angular velocity ω _m is “0”, the positive predetermined advance amount φ _{d1 is obtained} . From this, the motor angular velocity ω _m is predetermined. while increases to a value ωa1 decreases relatively slowly, and then while the motor angular speed omega _m is increased to a predetermined value ωa2 decreases relatively rapidly, with the motor angular speed omega _m is a predetermined value ωa2 above, relatively slowly When the decreasing characteristic curve L1 is set and the motor angular velocity ω _m increases from “0” to a negative value, the origin (advance amount φ _d6 = 0, motor angular velocity ω _m = 0) with respect to the characteristic curve L1 Point-symmetric characteristic curve L2 centered on Using the angle amount calculation map, with reference to the advancing amount calculation map on the basis of the motor angular speed omega _m to calculate the advancing amount phi _d6, advance amount Likewise 6n following d-axis current command value phi _With respect to _d12 , _φd18 ... _φd6n , the advance amount is calculated with reference to an advance amount calculation map (not shown).

Also, the 6n-th order d-axis current command value amplitude compensation gain calculation unit 53 also changes the d-axis current command value Id based on the d-axis current command value Id, whether this is a fundamental wave advance angle control state in which “0” continues. It is determined whether or not it is a full advance angle control state. When this determination result is a fundamental wave advance angle control state in which the d-axis current command value Id continues “0”, the amplitude compensation gain Gd ₆ of the sixth-order d-axis current command value is taken as an example in FIG. As shown in a), the amplitude compensation gain Gd ₆ is maintained at “0” regardless of the value of the motor angular velocity ω _m , and the sixth current command value amplitude compensation gain calculation map is used to obtain the motor angular velocity ω _m based on the map. The sixth current command value amplitude compensation gain calculation map is referred to to calculate the sixth current command value amplitude compensation gain Gd ₆ = 0. Similarly, although not shown, refer to the 6nth current command value amplitude compensation gain calculation map 6n-order current command value amplitude compensation gains Gd ₁₂ , Gd ₁₈ ... Gd _6n are calculated.

Further, when the determination result is the fundamental wave advance angle control state in which the d-axis current command value Id changes, the amplitude gain Gd ₆ of the sixth-order d-axis current command value is taken as an example in FIG. 9B. As shown, when the motor angular velocity ω _m increases from “0” to a positive value, the amplitude compensation gain Gd becomes “0” while the motor angular velocity ω _m is between “0” and a positive predetermined value ωg1. , amplitude compensation gain increases relatively sharply with an increase in the motor angular speed omega _m between until the motor angular speed omega _m reaches the predetermined value Omegaji2 exceeds a predetermined value Omegaji1, the motor angular speed exceeds a predetermined value Omegaji2 omega _When a characteristic line L3 on a polygonal line in which the amplitude compensation gain increases relatively slowly as _m increases and the motor angular velocity ω _m increases from “0” to a negative value, the characteristic curve L3 Origin (advance amount φ _d6 = 0, motor angular velocity ω _m = 0), the amplitude compensation gain Gd ₆ is calculated with reference to the sixth-order current command value amplitude compensation gain calculation map in which the point-symmetric characteristic curve L4 is set. 6n-order current command value amplitude compensation gains Gd ₁₂ , Gd ₁₈ ... Gd _6n are calculated with reference to the value amplitude compensation gain calculation map.

Further, the q-axis current command value advance amount calculation unit 54 is based on the d-axis current command value Id and is in a fundamental wave advance angle control state in which this continues “0” or the d-axis current command value Id changes. It is determined whether the advance angle control state is set. When this determination result is a fundamental wave advance angle control state in which the d-axis current command value Id continues to be “0”, the advance amount φ _q6 of the sixth-order q-axis current command value is taken as an example in FIG. As shown in a), the advance amount Φ _q6 is maintained at “0” regardless of the value of the motor angular velocity ω _m , and the d-axis current command value advance amount calculation map is used to determine the advance amount Φ _m6 based on the motor angular velocity ω _m . The d-axis current command value advance amount Φ _q6 is calculated with reference to the advance amount calculation map, and similarly, the advance amounts Φ _q12 , Φ _q18 ...... Φ _q6n of the 6n-th order d-axis current are also not shown. The advance amount is calculated with reference to the angular amount calculation map.

In addition, when the determination result is the full advance control state in which the d-axis current command value Id changes, taking the advance amount φ _q6 of the sixth-order q-axis current command value as an example, FIG. As shown, when the motor angular speed ω _m increases from “0” to a positive value, the advance amount φ _q6 is “0” while the motor angular speed ω _m is between “0” and a positive predetermined value ωa3. next, parabolic characteristic curve L5 motor angular speed omega _m is the q JikuSusumukakuryou phi _q6 is increased in the negative direction with an increase in the motor angular speed omega _m exceeds a predetermined value ωa3 is set, the motor angular speed omega _m is When the value increases from “0” to a negative value, a point-symmetric characteristic curve L6 centered on the origin ( _{advance amount} φ _q6 = 0, motor angular velocity ω _m = 0) is set with respect to the characteristic curve L5. It uses the advance amount calculation map, referring to the map for advancing amount calculated based on the motor angular speed omega _m Calculating a q JikuSusumukakuryou phi _q6 Te similarly calculated advance angle φ _q12, φ _{q18 ......} φ advance amount with reference to the advancing amount computation map, not shown also _q6n of 6n following d-axis current To do.

Furthermore, in the 6nth-order q-axis current command value amplitude compensation gain calculation unit 55, based on the d-axis current command value Id, whether this is the fundamental wave advance angle control state in which “0” is continued or the d-axis current command value Id is It is determined whether or not it is a changing total advance angle control state. When this determination result is a fundamental wave advance angle control state in which the d-axis current command value Id continues “0”, the amplitude compensation gain of the sixth-order q-axis current command value is taken as an example in FIG. As shown in FIG. 4, when the value of the motor angular velocity ω _m is “0”, the amplitude compensation gain G _q6 becomes a predetermined positive value _{Ga2, and} when the value of the motor angular velocity ω _m increases in the positive direction, the increase Accordingly, a parabolic characteristic curve L7 in which the amplitude compensation gain _Gq6 increases nonlinearly is set, and when the motor angular velocity ω _m increases in the negative direction from “0”, the amplitude compensation gain axis with respect to the characteristic curve L7 6th-order q-axis current command value amplitude compensation gain based on the motor angular velocity ω _m using a map for calculating the 6th-order q-axis current command value amplitude compensation gain in which a line-symmetric characteristic curve L8 is set. calculating a G _q6, although not shown, in a similar way 6n following current command value amplitude compensation gain _Gq 12 with reference to the n-th current command value amplitude compensation gain calculation map _to calculate the _Gq 18 ...... Gq _6n.

Further, when the determination result is the full advance control state in which the d-axis current command value Id changes, the amplitude compensation gain of the sixth-order q-axis current command value is taken as an example as shown in FIG. to, using Figure 11 (a) the same parabolic curve L9 and L10 are set 6n following q-axis current command value amplitude compensation gain calculation map, the amplitude compensation gain based on the motor angular speed omega _m The 6n-order q-axis current command value amplitude compensation gain G _q6 is calculated with reference to the calculation map. Similarly, although not shown, the 6n-order current command value amplitude compensation gain calculation map is referred to with reference to the 6n-order current command value amplitude compensation gain calculation map. Gains Gq ₁₂ , Gq ₁₈ ... Gq _6n are calculated.

Furthermore, as shown in FIG. 12, the DC current command value amplitude compensation gain calculation unit 56, when the motor angular velocity ω _m is “0” regardless of the fundamental wave advance angle control state and the full advance angle control state, The amplitude compensation gain GD having a predetermined value GD1, and when the motor angular velocity ω _m increases in the positive or negative direction, the amplitude compensation gain GD increases in a non-linear manner, and the parabolic characteristic curves L11 and L12 are set. It has a gain calculation map to calculate the DC command value amplitude compensation gain GD based on the motor angular speed omega _m by referring to a map for amplitude compensation gain calculation.

In the control maps shown in FIGS. 8 to 12, each characteristic line is set so as to compensate for these attenuation components in consideration of attenuation of each harmonic component in the current control system, that is, the current control unit 40. .
Then, the electrical angle θ _e ′ input from the electrical angle conversion unit 32 and advanced by the fundamental wave component advance amount calculation unit 51 is multiplied by 6n by the multiplier 57 and then the d-axis advance calculation unit 58 and the q-axis The fundamental wave component advanced electrical angle 6θ _e ′ supplied to the advance angle calculation unit 59 and multiplied by 6n by the d axis advance angle calculation unit 58 and the d axis advance angle calculated by the d axis current command value advance amount calculation unit 52 performs advance angle computation based on the amount _{φ d6 ~φ d6n, sin (6θ} e '+ φ d6), sin (12θ e' + φ d12), ......, calculates the _{sin (6nθ e '+ φ d6n} ), these Are output individually.

Similarly, q JikuSusumu angle q JikuSusumukakuryou phi _q6 to [phi] _Q6n calculated by the arithmetic unit 58 6n multiplied by the fundamental wave component advance angle after electrical angle 6θ _{e 'in} the q-axis current command value advance amount calculating section 54 performs advance angle computation based on the _{bets, sin (6θ e '+ φ} q6), sin (12θ e' + φ q12), ......, calculates the _{sin (6nθ e '+ φ q6n} ).
On the other hand, the d-axis current command value 6n-order components i _{d6 to} i _d6n output from the 6n-order d-axis current command value amplitude compensation gain calculation unit 53 are supplied to the multipliers MULd1 _{6 to} MULd1 _6n , and these multipliers MULd1 ₆ to in MULd1 _6n, the d-axis current command value 6n-order components _i d6 _{through i D6N} to 6n following d-axis current command value amplitude compensation gain _Gd d6 to GD of 6n following d-axis current command value output from the amplitude compensation gain calculation unit 53 Multiply by _d6n .

The multiplication outputs i _d6 , Gd _{d6 to} i _d6n, and Gd _d6n output from the multipliers MULd1 _{6 to} MULd1 _6n are the multipliers MULd2 _{6 to} MULd2 _6n and the advance output sin ( _{6 [} theta] _e '+ [phi] _d6 ) to sin (6n [theta] _e ' + [phi] _d6n ). On the other hand, the d-axis current command DC component I _dDC output from the current command value limiting unit 35 is supplied to the multiplier MULd3, and the multiplier MULd3 uses the d-axis current command value DC component I _dDC to compensate the DC current command value amplitude. amplitude compensation gain _{G DC} calculated by the gain calculation unit 56 is multiplied, the multiplied value _I dDC · _{G DC} is added to the multiplication output of the multiplier MULd2 ₆ ~MULd2 _6n are supplied to the adder ADDd by the following (17 The post-compensation d-axis current command value i _d ′ represented by the formula (1) is calculated.
i _d ′ = I _dDC · G _DC + i _d6 · G _d6 sin (6θ _e ′ + φ _d6 )
+ I _d12 · G _d12 sin (12θ _e ′ + φ _d12 ) + ……
+ I _d6n · G _d6n sin (6nθ _e ′ + φ _d6n ) (17)

Similarly, the compensated q-axis current command value i _q ′ is also multiplied and added by the multipliers MULq1 _{6 to} MULq1 _6n , MULq2 _{6 to} MULq2 _6n , MULq3 and the adder ADDq. The compensated q-axis current command value i _q ′ shown is calculated.
i _q ′ = I _qDC · G _DC + i _q6 · G _q6 sin (6θ _e ′ + φ _q6 )
+ I _q12 · G _q12 sin (12θ _e ′ + φ _q12 ) + ……
+ I _q6n · G _q6n sin (6nθ _e '+ φ _d6n ) (18)

Then, the post-advance electrical angle θ _e ′, the post-compensation d-axis current command value i _d ′, and the post-compensation q-axis current command value i _q ′ output from each harmonic component advance angle and amplitude compensation calculation unit 36 are obtained. It is supplied to the two-phase / three-phase conversion unit 37 and converted into current command values Iu ^* , Iv ^*, and Iw ^* of the U phase, V phase, and W phase of the brushless motor 12.
The current control unit 40 detects a motor phase current detection value Iud that flows in each phase coil Lu, Lv, Lw detected by the current detection circuit 22 from the current command values Iu ^* , Iv ^* , Iw ^* supplied from the target current setting unit 30. , Ivd, Iwd are subtracted to obtain respective phase current errors ΔIu, ΔIv, ΔIw and subtractors 71u, 71v, 71w are obtained, and the obtained phase current errors ΔIu, ΔIv, ΔIw are subjected to proportional-integral control and phase voltage is controlled. And a PI control unit 72 that calculates command values Vu, Vv, and Vw.

Then, phase voltage command values Vu, Vv, Vw output from the PI control unit 72 are supplied to the FET gate drive circuit 25.
As shown in FIG. 2, the motor drive circuit 24 includes switching elements Qua, Qub, Qva, Qvb and Qwa, Qwb composed of N-channel MOSFETs connected in series corresponding to the phase coils Lu, Lv and Lw. Are connected in parallel, and the connection points of the switching elements Qua and Qub, the connection points of Qva and Qvb, and the connection points of Qwa and Qwb are opposite to the neutral point Pn of the phase coils Lu, Lv and Lw, respectively. Connected to the side.
A PWM (pulse width modulation) signal output from the FET gate drive circuit 25 is supplied to the gates of the switching elements Qua, Qub, Qva, Qvb and Qwa, Qwb that constitute the motor drive circuit 24.

Next, the operation of the above embodiment will be described.
Now, when the steering wheel 1 is steered, the steering torque T at that time is detected by the steering torque sensor 3, and the vehicle speed detection value Vs is detected by the vehicle speed sensor 21. Then, the detected steering torque T and the detected vehicle speed Vs are input to the steering assist current command value calculator 31 in the target current setting unit 30 of the control calculator 23, so that the steering assist current command value calculator 31 Then, the steering assist current command value I _ref is calculated with reference to the steering assist current command value calculation map of FIG.

The calculated steering assist current command value I _ref is limited by the current command value limiter 35 based on the motor angular velocity ω _m and the d-axis current command expressed by the above-described equation (1). D-axis current command value I _d composed of value direct current component I _dDC and d-axis current command value 6n-order component amplitudes i _{d6 to} i _d6n , and q-axis current command value direct current represented by equation (2) above and output the components _{I QDC} and the q-axis electric current command value 6n-order components of the amplitude _i q6 _{through i Q6n} q-axis electric current command value _{I q} and each order harmonic component advance angle and amplitude compensation operation unit 36 composed of.

In each of the higher harmonic component advance angle / amplitude compensation calculation unit 36, the fundamental wave component advance amount calculation unit 51 calculates a compensated fundamental wave component advance amount θ _e ′ and calculates a d-axis current command value advance amount. Unit 52, 6n-order d-axis current command value amplitude compensation gain calculation unit 53, q-axis current command value advance amount calculation unit 54, 6n-order q-axis current command value amplitude compensation gain calculation unit 55, and DC current command value amplitude compensation gain refers to the control map based on each motor angular speed omega _m in the arithmetic unit 56 d JikuSusumukakuryou _{φ d6} ~φ d6n, d-axis amplitude compensation gain _G d6 _{~G d6n,} q JikuSusumukakuryou phi _q6 to [phi] _Q6n Q-axis amplitude compensation gains G _{q6 to} G _q6n are calculated, and based on these, the compensated d-axis current command value i _d ′ and the compensated q-axis current represented by the above-described equations (17) and (18) are calculated. The command value i _q ′ is calculated and the calculated fundamental wave component advance amount θ after compensation _e ′, the compensated d-axis current command value i _d ′, and the compensated q-axis current command value i _q ′ are supplied to the two-phase / three-phase converter 37, and the three-phase current command values Iu ^* , Iv ^* and Iw ^* is calculated.

Then, the calculated three-phase current command values Iu ^* , Iv ^* and Iw ^* are input to the current control unit 40. In this current control unit 40, PI control is performed on deviations ΔIu, ΔIv, and ΔIw between the three-phase current command values Iu ^* , Iv ^*, and Iw ^* and the phase current detection values Iud, Ivd, and Iwd detected by the motor current detection circuit 22. The voltage command values Vu, Vv, and Vw are calculated by PI control in the unit 72 and supplied to the FET gate drive circuit 25, whereby a pulse width modulation signal having a duty ratio corresponding to the voltage command values Vu, Vv, and Vw. Is supplied to the motor drive circuit 24, and a motor drive current is supplied from the motor drive circuit 24 to the brushless motor 12.
As a result, a steering assist force corresponding to the steering torque T is generated by the brushless motor 12, and this steering assist force is transmitted to the steering shaft 2 via the reduction gear mechanism 11, so that the steering wheel 1 can be operated with a light steering force. Can be steered.

At this time, each harmonic component advance angle / amplitude compensation calculation unit 36 refers to the control map based on the motor angular velocity ω _m , and d-axis advance _amount φ _{d6 to} φ _d6n , d-axis amplitude compensation gain G _d6. ~G _D6N, q JikuSusumukakuryou _{φ q6} ~φ q6n, calculates the q-axis amplitude compensation gain _G q6 _{~G q6n,} described above on the basis of these (17) and (18) after the compensation represented by the formula The d-axis current command value i _d ′ and the compensated q-axis current command value i _q ′ are calculated. During fundamental wave advance control where the d-axis current command value I _d is “0”, the d-axis advance angle φ _d6 to [phi] _D6N when the motor angular speed omega _m is a positive value including zero maintaining a predetermined value phi _d1 regardless of its value, when the motor angular speed omega _m is a negative value d-axis regardless of its value The advance amounts φ _{d6 to} φ _d6 n of the _second angle are maintained at a predetermined value −φ _d1 . However, since the 6n-th order d-axis current command value amplitude compensation gains G _{d6 to} G _d6n are “0”, the d-axis current command value 6n-order component is “0”, and the d-axis current DC component I _dDC is also “0”. Therefore, the d-axis current command value i _d ′ output from the adder ADDd also maintains zero.

For relative q-axis current command value _{i q} 'including but is set to q JikuSusumukakuryou phi _q6 to [phi] _Q6n is "0", the q-axis 6n following current command value amplitude compensation gain is at least q-axis As shown in the control map of FIG. 11, the _sixth current command value amplitude compensation gain Gq ₆ becomes a predetermined positive value Ga2 when the motor angular velocity ω _m is “0”, and the absolute value of the motor angular velocity ω _m increases from this. As a result, the q-axis sixth-order current command value amplitude compensation gain increases relatively steeply in the positive direction, and the other q-axis 6n-order current command value amplitude compensation gains have the same tendency. A compensated d-axis current command value i _q ′ in which the 6n-order component of the value is compensated for increase can be obtained.

Further, during full advance control where the d current command value I _d is not “0”, the d-axis _advance amount φ _{d6 to} φ _d6n decreases as the motor angular velocity ω _m increases, but the d-axis 6n-order component. By increasing the amplitude compensation gains G _{d6 to} G _d6n , a compensated q-axis current compensation value i _q ′ in which the 6n-order harmonic component of the d-axis current command value is compensated for increase can be obtained.
Further, with respect to the DC components I _dDC and I _qDC of the d-axis current command value and the q-axis current command value, similarly, the compensation gain GD increases as the motor angular velocity ω _m increases as in the q-axis current command value 6n-order component. By increasing, the DC components I _dDC and I _qDC are increased and compensated, and the compensated d-axis current command value i _d ′ and the compensated q-axis current command value i _q ′ are increased.

For this reason, while maintaining a constant torque state, the attenuation amount of each harmonic component due to the characteristics of the current control system can be compensated, and an ideal current as designed can be supplied to the brushless motor 12. The high-order harmonics in the case of high-speed rotation control can be reliably prevented from being attenuated, and the output reduction and torque fluctuation of the brushless motor 12 can be reliably prevented.
In addition, since the higher-order harmonic components are compensated on the dq coordinates, it is not necessary to separate the three-phase current command values from each harmonic component using a filter or the like. Therefore, it is possible to reduce the calculation load of the arithmetic processing unit, and it is possible to compensate for higher-order harmonic components without applying an expensive arithmetic processing unit. Moreover, in the dq coordinate system, when the 6n ± 1st order (n = 1, 2, 3,...) Harmonics of the three-phase current are coordinate-transformed, the 6nth order (n = 1, 2, 3,. Therefore, the equivalent advance amount and amplitude compensation gain of each order component including the fundamental wave can be easily set in advance by calculation or experiment.

  Furthermore, the d-axis command value advance amount calculation unit 52, the 6n-order d-axis current command value amplitude compensation gain calculation unit 53, the q-axis current command value advance amount calculation unit 54, and the 6n-order q-axis current command value amplitude compensation gain calculation 55 and DC current command value amplitude compensation gain calculation unit 56 use control maps that take into account the attenuation of harmonic components in the current control system, and refer to the control map based on the motor angular velocity to advance the amount of advancement. Since the amplitude compensation gain is calculated, the advance amount and the amplitude compensation gain can be accurately and easily calculated.

  Further, as described above, the brushless motor 12 can be driven while reliably reducing the output and torque fluctuation, so that the steering assist force generated by the brushless motor 12 is supplied to the steering shaft 2 via the reduction gear mechanism 11. Thus, it is possible to reliably prevent torque fluctuations from occurring in the steering wheel and to give the driver a good steering feeling.

In the first embodiment, each of the higher harmonic component advance angle and amplitude compensation calculation unit 36 uses the _advance angle amounts φ _{d6 to} φ _{d6 n} and φ _{q6 to} φ q6 _n and the 6 n order component compensation gain G _{d6 to} G. _{Although the case where d6n} and _{Gq6 to Gq6n} and the DC component compensation gain GD are calculated has been described, the present invention is not limited to this, and either one of the advance amount and the 6nth-order component compensation gain is omitted. May be.

In the first embodiment, the compensated d-axis current command value i _d ′ and the compensated q-axis current command value i _q ′ are converted into the three-phase target currents Iu ^* and Iv by the two-phase / three-phase conversion unit 37. ^* and has been described the case of supplying to the current controller 40 after converting to Iw ^*, is not limited to this, it omits the 2-phase / 3-phase conversion unit 37, a current detection circuit in place of this The motor currents Idu, Idv, and Idw detected in 22 are supplied to the three-phase / two-phase converter to convert them into d-axis detection current and q-axis detection current, and the converted d-axis detection current and q-axis detection current After calculating the deviation between the compensated d-axis current command value i _d ′ calculated by the current setting unit 30 and the q-axis current command value i _q ′, the deviation is converted into two-phase / 3-phase to calculate the phase control voltage. You may do it.
Furthermore, although the case where the electric motor is a three-phase motor has been described in the first embodiment, the present invention is not limited to this, and the present invention is also applied to a multiphase motor having a number of phases exceeding three. can do.

Next, a second embodiment of the present invention will be described with reference to FIG.
In the second embodiment, the 6n ± 1st harmonic component is extracted from the d-axis command value and the q-axis command value, and the extracted 6n ± 1st harmonic component is individually amplitude compensated and then added. The first phase target current to the m-th phase target current of the m-phase brushless motor are calculated.

That is, in the second embodiment, the target current setting unit 30 is configured as shown in FIG. Similar to the first embodiment described above, the target current setting unit 30 inputs the steering assist current command value I _ref output from the steering assist current command value calculation unit 31 to the current command value limit 32, and this In the current command value limiting unit 82, the d-axis current command value _Id and the q-axis current in the same vector format as the above-described equations (1) and (2) expressed by the following equations (19) and (20): The command value I _q is calculated, and the calculated d-axis current command value I _d and q-axis current command value I _q are input to each order component extractor 83, and each order component extractor 83 determines the fundamental wave component and Dq-axis current command values I _d1 and I _q1 equivalent to the first-order phase current, dq-axis current command values I _d5 and I _q5 equivalent to the fifth-order harmonic phase current, dq-axis equivalent to the seventh-order harmonic phase current Current command values I _d7 , I _q7 ,..., Dq axis power equivalent to 6n ± 1st harmonic phase current The flow command values I _{d6n ± 1} and I _{q6n ± 1} are separated and extracted.
I _d = [I _d1 I _d5 I _d7 ...... I _{d (6n ± 1)} ] ............ (19)
I _q = [I _q1 I _q5 I _q7 ... I _{q (6n ± 1)} ] (20)
Here, I _{d1 to} I _{d (6n ± 1)} are primary phase current equivalent d-axis current to 6n ± primary phase current equivalent d-axis current, and I _{q1 to} I _{q (6n ± 1)} are equivalent to primary phase current. q-axis current to 6n ± primary phase current equivalent q-axis current.

Then, the separated and extracted dq-axis current command values I _d1 , I _q1 ; I _d5 , I _q5 ; I _d7 , I _q7 ; ...; I _{d (6n ± 1)} , I _{q (6n ± 1)} are individually obtained. Primary 2-phase / m-phase converter TR _{1 for} converting dq 2-phase signal into m-phase signal of brushless motor 12, 5-order 2-phase / m-phase converter TR ₂ ,..., 6n ± primary 2-phase / m-phase Supply to the converter TR _p . The converters TR _{1 to} TR _p receive the electrical angle θ _e and the motor angular velocity ω _m and calculate the respective advance amounts φ _{1 to} φ _{6n ± 1} to calculate the respective secondary component electrical angles ω _e t to. output from the (6n ± 1) ω _e advanced after each next electric angle obtained by adding to _{t (ω e t + φ 1} ) ~ ((6n ± 1) ω e t + φ 6n ± 1) advancing amount computation unit 84 for calculating the that advances the next electric angle dorsal horn _{(ω e t + φ 1)} ~ ((6n ± 1) ω e t + φ 6n ± 1) are inputted, these post the advance each next electric angle _{(ω e t + φ 1)} ~ ((6n _{± 1) ω e t + φ} 6n ± 1) to perform the two-phase / m-phase conversion on the basis of the following (21) each phase primary phase currents _{i 1} ~ phase 6n ± 1-order phase current _{i 6n ±} in the expression ₁ is calculated.

The calculated primary phase currents i ₁ to 6n ± primary phase currents i _{6n ± 1} are individually supplied to the primary component amplitude compensators AC _{1 to} 6n ± primary component amplitude compensators AC _p. Then, in these primary component amplitude compensators AC _{1 to} 6n ± primary component amplitude compensators AC _p , as shown in the following equation (22), an amplitude gain for compensating for the amplitude attenuation of the harmonic component generated in the current control system Dividing g _{1 to} g _{6n ± 1} to calculate each phase primary phase current i ₁ ′ to each phase 6n ± primary phase current i _{6n ± 1} ′.

Then, each phase current is individually extracted from the calculated post-compensation primary phase current I ₁ ′ to each phase 6n ± primary phase current I _{6n ± 1} to obtain first phase component adders ADD ₁ to m-th. By inputting to the phase component adder ADD _m , the first phase target current i ₁ ^* to the m-th phase target current i _m ^* are calculated and output to the voltage control unit 40.
Here, as shown in FIG. 14, the advance amount calculation unit 84 takes the motor angular velocity ω _m on the horizontal axis, the advance amount on the vertical axis, and sets each of the advance amounts φ _{1 to} φ _{6n ± 1} . It has a control map in which a characteristic straight line as a parameter is set. The control map, taking into account the attenuation of the harmonic components in the current control system, 1 TsugiSusumukakuryou for phi ₁ is represented by a characteristic line L ₁ of the relatively gentle slope, the 1 TsugiSusumukakuryou phi Characteristic straight lines L ₅ , L ₇ ... Are set such that the inclination increases as the degree of advance amount increases from ₁ . Then, referring to the control map of FIG. 14 based on the motor angular velocity ω _m , the primary advance angle φ ₁ , the fifth advance angle φ ₅ , the seventh advance angle φ ₇ ,..., 6n ± primary to calculate the advance amount φ _{6n ± 1,} calculated 1 TsugiSusumukakuryou φ ₁ ~6n ± ₁ TsugiSusumukakuryou φ _{6n ± 1} of each following electrical angle ω _e t~ (6n ± 1) to ω _e t By adding, each compensated primary electrical angle θ _e1 ′ to θ _{e6n ± 1} ′ is calculated, and the calculated compensated primary electrical angles θ _e1 ′ to θ _{e6n ± 1} ′ are converted into a primary two-phase / m-phase conversion. vessel _TR 1 _~6n ± supplying the primary two-phase / m converters TR _P.

Also, the first-order component amplitude compensator _AC 1 ~6n ± 1-order component amplitude compensator AC _p, a control map showing the relation between the motor angular velocity omega _m of FIG. 15 on the basis of the motor angular speed omega _m and the compensation gain The compensation gains g _{1 to} g _{6n ± 1} of the primary to _{6n ± 1st} order components are calculated with reference to the reference. As shown in FIG. 15, this control map is set by a characteristic curve on a parabola with the motor angular velocity ω _m on the horizontal axis, the compensation gain value on the vertical axis, and the order as a parameter. Is set so that the curvature of the characteristic curve decreases as the order increases from the first order.

Next, the operation of the second embodiment will be described.
The steering assist current command value I _ref corresponding to the steering torque T calculated by the steering assist current command value calculation unit 31 is converted into a d-axis expressed by the current command value limiting unit 82 and expressed by the equations (19) and (20). The current command value I _d and the q-axis current command value I _q are calculated, and the calculated d-axis current command value I _d and the q-axis current command value I _q are respectively calculated by the component extractor 83 as the primary phase current equivalent dq current I _d1. , I _{q1 to} 6n ± 1 primary phase current equivalent dq axis currents I _{d (6n ± 1)} , I _{q (6n ± 1)} are separated and extracted, and these extracted dq axis currents are primary 2 phase / m phase Phase conversion based on post- _advanced electrical angles θ _e1 ′ to θ _{e6n ± 1} ′ calculated by the advance angle calculation unit 84 in the converters TR _{1 to} 6n ± primary 2-phase / m-phase converter TR _p To calculate each phase primary phase current to each phase 6n ± primary phase current, and calculate each phase primary phase current to each phase 6n ± primary phase current. Amplitude compensation is individually performed by the amplitude compensation units AC _{1 to} AC _p .

For this reason, generally, when the current command value for the A phase (first phase) of the brushless motor 12, for example, is i ₁ ^* , this A phase current command value i ₁ ^* is expressed by the following equation (23). be able to.
^{_{i 1 * = I 1 sinω e}} t + I 5 sin5ω e t + ...... + I 6n ± 1 sin (6n ± 1) ω e t
……………… (23)
Where I ₁ , I ₅ ,... I _{6n ± 1} are the amplitudes of the currents of the respective components, ω _e is the motor electrical angular velocity, t is the time, and n is a natural number.

This equation (23) is a current waveform obtained from a constant torque equation, that is, an ideal current waveform, but the actual current actually supplied to the brushless motor 12 depends on the characteristics of the current control system (24 ) As shown in the equation, it attenuates.
i ₁ = g ₁ I ₁ sin (ω _e t−φ ₁ ) + g ₅ I ₅ sin (5ω _e t−φ ₅ )
+ …… + g _{6n ± 1} I _{6n ± 1} sin {(6n ± 1) ω _e t−φ _{6n ± 1} } …… (24)
However, g _{1 to} g _{6n ± 1} are amplitude reduction rates of the respective order components, and φ _{1 to} φ _{6n ± 1} are phase delays of the respective order components.
From this equation (24), it can be seen that the actual current i ₁ that actually flows through the brushless motor 12 is not an ideal waveform.

However, in the present embodiment, the advance amount calculation unit 84 calculates the advance amount that eliminates the phase lag due to the characteristics of the current control system, and each of the component amplitude compensators AC _{1 to} AC _P uses the amplitude reduction rate g. _Since the amplitude compensation is performed by dividing ₁ to g _{6n ± 1} , the A-phase target current I ₁ ^* output from the adder ADD ₁ is expressed by the following equation (25). The actual current supplied to the brushless motor 12 by compensating for the phase lag and amplitude reduction at, can be made the ideal current as expressed by the above-described equation (23).
i ₁ = (1 / g ₁ ) I ₁ sin (ω _e t + φ ₁ ) + (1 / g ₅ ) I ₅ sin (5ω _e t + φ ₅ )
_{+ ...... + (1 / g 6n} ± 1) I 6n ± 1 sin {(6n ± 1) ω e t + φ 6n ± 1} ...... (25)

  Therefore, while maintaining a constant torque state, it is possible to compensate for the attenuation of each harmonic component due to the characteristics of the current control system, and to supply the ideal current as designed to the brushless motor 12 and output the brushless motor. It is possible to control the driving in the optimum state to reduce the torque and to generate the torque ripple. In addition, since the advance angle amount and the amplitude compensation gain are calculated with reference to the control map based on the motor angular velocity, the advance angle amount and the amplitude compensation gain can be accurately and easily calculated.

Therefore, by applying the brushless motor 12 as a drive motor of an electric power steering device that generates a steering assist force with respect to the steering system, it is possible to improve responsiveness at high speed steering and suppress torque ripple. The steering feeling can be improved.
In the second embodiment, the case where both the advance angle amount and the amplitude are compensated has been described. However, the present invention is not limited to this, and only one of the advance angle amount and the amplitude is compensated. It may be.

  In the first and second embodiments, the case where the present invention is applied to an electric power steering apparatus has been described. However, the present invention is not limited to this and is applied to an electric tilt apparatus, an electric telescopic apparatus, and the like. The present invention can be applied to an in-vehicle electric motor control device and an electric motor control device that is applied to other devices including a general electric motor.

It is a whole lineblock diagram showing one embodiment of the present invention. It is a block diagram which shows an example of a steering assistance control apparatus. It is a block diagram which shows the specific structure of the control arithmetic unit 23 of FIG. It is a characteristic diagram which shows a steering auxiliary current command value calculation map. It is a block diagram which shows the specific structure of the electric current command value limiting part of FIG. FIG. 7 is a characteristic diagram showing a d-axis current command value DC component calculation map representing a relationship between a post-limit current command value I _ref ′ and a d-axis current command value DC component i _dDC . FIG. 4 is a block diagram illustrating a specific configuration of each time harmonic component advance angle and amplitude compensation calculation unit of FIG. 3. It is a characteristic diagram which shows the control map which calculates d-axis current command value advance amount. It is a characteristic diagram which shows the control map which calculates 6th d-axis current command value amplitude compensation gain. It is a characteristic diagram which shows the control map which calculates q-axis current command value advance amount. It is a characteristic diagram which shows the control map which calculates 6nth order q-axis current command value amplitude compensation gain. It is a characteristic diagram which shows the control map which calculates a direct current command value amplitude compensation gain. It is a block diagram which shows the specific structure of the target electric current setting part which shows the 2nd Embodiment of this invention. It is a characteristic diagram which shows the control map which calculates the amount of advance of each next-order component in 2nd Embodiment. It is a control map which calculates the amplitude compensation gain in a 2nd embodiment.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 ... Steering wheel, 2 ... Steering shaft, 3 ... Torque sensor, 10 ... Steering assistance mechanism, 11 ... Reduction gear, 12 ... Three-phase brushless motor, 20 ... Steering assistance control device, 21 ... Vehicle speed sensor, 24 ... Motor drive circuit 25 ... FET gate drive circuit, 30 ... target current setting unit, 31 ... steering auxiliary current command value calculation unit, 32 ... electrical angle conversion unit, 33 ... differential circuit, 34 ... motor angular velocity conversion unit, 35 ... current command value limit , 36... Each harmonic component advance and amplitude compensation calculation unit, 37... 2 phase / 3 phase conversion unit, 40... Current control unit, 41 command value limiting unit, 42... D-axis current command value DC component calculation unit , 43... Q-axis current command value DC component calculation unit, 44... D-axis current command value 6n-order component amplitude calculation unit, 45... Q-axis current command value 6n-order component amplitude calculation unit, 51. Part, 52 ... d-axis Flow command value advance angle calculation unit, 53... 6n-order d-axis current command value amplitude compensation gain calculation unit, 54... Q-axis current command value advance amount calculation unit, 55. parts, 56 ... DC current command value amplitude compensation gain calculation unit, 58, 59 ... advance angle calculation _{section, MULd1 6 ~MULd1 6n, MULd2 6} ~MULd 6n, MULq1 6 ~MULq1 6n, MULq2 6 ~MULq2 6n, MULd3, MULq3 ... multiplier, ADDD, addq ... adder, 82 ... current command value limiting section, 83 ... each component _extractor, TR 1 to Tr _p ... 2-phase / m-phase converter, _AC 1 to Ac p _... amplitude compensator, ADD _{1 to} ADD _m ... adder

Claims (5)

An electric motor control device comprising motor control means having a current control system for controlling an electric motor based on a current command value and a motor current,
Motor angular velocity detecting means for detecting a motor angular velocity of the electric motor, wherein the motor control means considers the characteristics of the current control system based on the motor angular speed detected by the motor angular velocity detecting means, and makes the torque constant An electric motor control device configured to calculate a command value. An electric motor control device comprising motor control means having a current control system for controlling a multiphase electric motor having an induced voltage including high-order harmonics based on a current command value and a motor current,
The motor control means calculates the advance amount and amplitude compensation value of the current command value of each harmonic component so that the torque is constant in consideration of the characteristics of the current control system, and the calculated advance amount and amplitude An electric motor control device comprising current command value compensation means for compensating a current command value of the multiphase electric motor based on a compensation value. An electric motor control device comprising motor control means having a current control system for controlling a multiphase electric motor having an induced voltage including high-order harmonics based on a current command value and a motor current,
The motor angular velocity detecting means for detecting the motor angular velocity of the electric motor is provided, and the motor control means has a motor angular velocity detected by the motor angular velocity detecting means on a dq coordinate rotating at a frequency corresponding to the motor angular velocity. Based on the characteristics of the current control system, the advance amount and amplitude compensation value of the current command value in each harmonic component of the dq coordinate of the multiphase motor are calculated so that the torque is constant. An electric motor control device comprising: current command value compensation means for compensating the current command value of the multiphase electric motor based on the advance amount and amplitude compensation value.   The current command value compensation means has a control map representing the relationship between the motor angular velocity, the advance amount, and the amplitude compensation gain that suppresses the attenuation of the harmonic component in the current control system, and based on the motor angular velocity The electric motor control device according to claim 2 or 3, wherein an advance amount and an amplitude compensation gain are calculated with reference to a control map.   5. An electric power steering apparatus, wherein an electric motor that generates a steering assist force for a steering mechanism based on a steering torque is controlled by the electric motor control apparatus according to claim 1.

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