JP2007159177A - Switching power supply - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a switching power supply which can prevent decline in efficiency resulting from a recovery current. <P>SOLUTION: The switching power supply comprises voltage converting sections 1A and 1B connected in parallel, a smoothing capacitor 3 connected to the post-stage of these voltage converting sections 1A and 1B, and a control section 4. The voltage converting section 1A consists of transformation inductors 11A wound around a common core 20 to have the same polarity, a diode 12A and a rectifier type switching element 13A, and the voltage converting section 1B consists of transformation inductors 11B wound around the common core 20 to have the same polarity, a diode 12B and a rectifier type switching element 13B. The control section 4 operates the switching elements 13A and 13B sequentially with mutually different phases. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

  The present invention relates to a switching power supply device that transforms an input DC voltage by switching and outputs an output DC voltage, and more particularly to a switching power supply device that is suitable for a device having a high switching frequency.

  Conventionally, various types of switching power supply devices have been proposed and put into practical use. As one of them, in Patent Document 1, as shown in FIG. 39, one voltage converter 101 including an inductor 111, a diode 112 and a switching element 113 wound around a core 114, and a smoothing capacitor 103 are provided. A device including a step-up chopper circuit and a control unit 104 is disclosed. This switching power supply device functions as a DC-DC converter that converts the DC input voltage Vin supplied from the DC power supply P into a higher DC output voltage Vout and supplies it to the load L.

Japanese Patent Publication No. 63-43766

In the switching power supply device incorporating such a chopper circuit, as shown in FIG. 40, when the control signal S ₁₀₁ is input from the control unit 104 to the switching element 113 and the switching element 113 is turned on, a current flows. The voltage on the anode side of the diode 112 becomes substantially the same as that on the negative electrode side of the DC power supply P, and is smaller than the voltage on the cathode side of the diode 112 (output DC voltage Vout). As a result, a large reverse bias is applied to the diode 112 through which current flows, and the diode 112 shifts from on to off.

  At this time, since there is a period (recovery period) in which current can flow in the reverse direction at the moment when the diode 112 shifts from on to off, current (recovery current) flows through the diode 112 during the recovery period. This recovery current flows through the switching element 113 directly connected to the diode 112 to generate a ripple (see within a one-dot chain line frame in FIG. 40). As a result, a switching loss (specifically, a loss at turn-on) occurs in the switching element 113, so that the efficiency of the switching power supply device decreases.

  Such a recovery current is not limited to a step-up chopper circuit, but is common to any switching power supply that has a circuit configuration in which a switching element and a diode are directly connected, such as a step-down type or a step-up / step-down type. It was difficult to prevent the efficiency from being reduced due to the recovery current.

  The present invention has been made in view of such problems, and an object thereof is to provide a switching power supply device capable of preventing a reduction in efficiency due to a recovery current.

  The first switching power supply device of the present invention includes a plurality of transformer chopper circuits and control means. Each transformer chopper circuit includes a transformer inductor, a rectifier element, and a switching element, and is connected in parallel to each other. The control means sequentially operates the switching elements at different phases. One end of each of the transformer inductor, the rectifier element, and the switching element is commonly connected at a common connection point. Between a plurality of transformer chopper circuits, transformer inductors are magnetically coupled so as to have the same polarity. Here, “phases different from each other” means giving a phase difference that does not overlap the ON periods.

  In the first switching power supply device of the present invention, the transformer inductors are magnetically coupled so that the transformer inductors have the same polarity between the plurality of transformer chopper circuits, so that each transformer inductor constitutes a transformer. This transformer is equivalent to a circuit in which one end of the mutual inductor is connected in common to one end of each leakage inductor of each inductor. This allows mutual inductors to be shared by each transformer chopper circuit, while leakage inductors are included in each transformer chopper circuit, so that if the transformer inductor, rectifier element and switching element are properly connected, then from off It is also possible to apply a voltage (reverse bias) that does not pass a current to the rectifying element connected to the one switching element that is turned on.

  Specifically, by connecting each one end of the transformer inductor, the rectifying element, and the switching element at a common connection point (for example, by performing the following first to third connection modes), It is also possible to apply a voltage (reverse bias) that does not allow a current to flow to the rectifying element connected to one switching element that is switched from OFF to ON.

  As a first connection mode, the transformer chopper circuit includes a pair of input terminals and a pair of output terminals, one end of the transformer inductor is connected to one input terminal, and the other end is connected to a rectifier element at a common connection point. A step-up type transformer chopper circuit is commonly connected to each one end of the switching element, the other end of the rectifying element is connected to one output terminal, and the other end of the switching element is connected to the other input terminal and the other output terminal. It is done.

  As a second connection mode, the transformer chopper circuit includes a pair of input terminals and a pair of output terminals, one end of the switching element is connected to one input terminal, and the other end is connected to a rectifier element at a common connection point. A step-down transformer chopper circuit in which one end of the transformer inductor is connected in common, the other end of the transformer inductor is connected to one output terminal, and the other end of the rectifier element is connected to the other input terminal and the other output terminal. It is done.

  As a third connection mode, when the transformer chopper circuit includes a pair of input terminals and a pair of output terminals, one end of the switching element is connected to one input terminal, and the other end is connected to the rectifier element at the common connection point. A step-up / step-down transformer chopper circuit is commonly connected to each one end of the transformer inductor, the other end of the rectifier element is connected to one output terminal, and the other end of the transformer inductor is connected to the other input terminal and the other output terminal. It is done.

  The second switching power supply device of the present invention includes a plurality of transformer chopper circuits, control means, and a common inductor. Each transformer chopper circuit includes an inductor, a rectifier element, and a switching element, and is connected in parallel to each other. The control means sequentially operates the switching elements at different phases. The common inductor is provided in common to the plurality of transformer chopper circuits, and is commonly connected to each end of the inductor of each transformer chopper circuit.

  In the second switching power supply device of the present invention, since the common inductor is shared by each transformer chopper circuit, and the inductor is included in each transformer chopper circuit, the inductor, common inductor, rectifier element, and switching element are provided. When properly connected, a voltage (reverse bias) that does not allow a current to flow to the rectifier connected to the next switching element that switches from off to on may be applied to the rectifier. Is possible.

  For example, the inductor of one transformer chopper circuit allows a current to flow through the rectifier element of the transformer chopper circuit when the switching element of the transformer chopper circuit shifts from on to off, and the switching element of another transformer chopper circuit is off. After switching from ON to ON, the common inductor acts to keep current flowing through the rectifying element of one transformer chopper circuit, and the common inductor accumulates energy by the current flowing through each switching element during the ON period. It is also possible to connect the inductor, the common inductor, the rectifier element and the switching element so as to act so as to release the stored energy during the period when all the switching elements are off.

  In that case, the inductor of one transformer chopper circuit has a voltage (forward bias) that immediately turns on the rectifier element of the transformer chopper circuit when the switching element of the transformer chopper circuit shifts from on to off. ) Is applied to the rectifying element. Next, the inductor of one transformer chopper circuit has a voltage (forward bias) that does not immediately turn off the rectifier element of one transformer chopper circuit when the switching element of another transformer chopper circuit shifts from OFF to ON. It acts to be applied to the rectifying element. As a result, a voltage (reverse bias) is applied to the rectifying element so that no current flows through the rectifying element of the other transformer chopper circuit that next shifts from OFF to ON.

  According to the first switching power supply device of the present invention, the transformer inductors are magnetically coupled to each other so that the transformer inductors have the same polarity between the plurality of transformer chopper circuits. When the elements are appropriately connected, it is possible to prevent the recovery current from being generated in each rectifying element when any switching element shifts from OFF to ON. Thereby, since the switching loss in each switching element is reduced, it is possible to prevent a decrease in efficiency due to the recovery current.

  According to the second switching power supply device of the present invention, since the common inductor is shared by each transformer chopper circuit and the inductor is included in each transformer chopper circuit, the inductor, the common inductor, the rectifier element, and When the switching elements are appropriately connected, it is possible to prevent the recovery current from being generated in each rectifying element when any switching element shifts from OFF to ON. Thereby, since the switching loss in each switching element is reduced, it is possible to prevent a decrease in efficiency due to the recovery current.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

[First Embodiment]
FIG. 1 shows a circuit configuration of a switching power supply apparatus according to the first embodiment of the present invention. This switching power supply device functions as a DC-DC converter that converts the DC input voltage Vin supplied from the DC power supply P into a higher DC output voltage Vout and supplies it to the load L.

  This switching power supply device includes voltage conversion units 1A and 1B, a smoothing capacitor 3, and a control unit 4 connected in parallel to each other. Here, the circuit composed of the voltage converter 1A and the smoothing capacitor 3 constitutes one boost chopper circuit, and the circuit composed of the voltage converter 1B and the smoothing capacitor 3 constitutes another boost chopper circuit. That is, this switching power supply device is configured by connecting boost type chopper circuits in parallel, and has a configuration in which the smoothing capacitor 3 is shared by both chopper circuits.

  The voltage converter 1A includes a transformer inductor 11A wound around the core 20, a diode 12A, and a rectifying switching element 13A. The voltage converter 1B includes a transformer inductor 11B wound around the core 20, It has a diode 12B and a rectifying switching element 13B. The transformer inductors 11A and 11B are wound around the core 20 so as to have the same polarity, and constitute a transformer.

  The step-up chopper circuit described above corresponds to the “transformer chopper circuit” of the present invention, and the control unit 4 corresponds to “control means” of the present invention. Further, the transformer inductors 11A and 11B correspond to “transformer inductors” of the present invention, the diodes 12A and 12B correspond to “rectifier elements” of the present invention, and the rectifier type switching elements 13A and 13B correspond to “switching elements” of the present invention. It corresponds to.

  In the voltage converter 1A, one end of the transformer inductor 11A, the anode of the diode 12A, and the collector of the rectifying switching element 13A are connected to each other at the connection point J1, and in the voltage converter 1B, one end of the transformer inductor 11B, the anode of the diode 12B, and The collectors of the rectifying switching element 13B are connected to each other at the connection point J2. The other end of each of the transformer inductors 11A and 11B is connected to an input line Lin extending from the input terminal T1, and the other end of each of the diodes 12A and 12B is connected to an output line Lout extending from the output terminal T3. The emitters of the type switching elements 13A and 13B are connected to a common line Lc. This common line is for electrically connecting the Lc input terminal T2 and the output terminal T4. Here, the input terminal T1 is a terminal to which the positive electrode of the DC power supply P is connected, the input terminal T2 is a terminal to which the negative electrode of the DC power supply P is connected, and the output terminal T3 is a terminal to which the high voltage side of the load L is connected. The output terminal T4 is a terminal to which the low voltage side of the load L is connected.

  The core 20 is preferably made of a material that is less likely to be magnetically saturated by the current flowing through the transformer inductors 11A and 11B. For example, the core 20 is made of ferrite and amorphous that can handle a low current, or a silicon steel plate that can handle a large current. Is done. The core 20 made of ferrite, amorphous, silicon steel plate or the like can be generally reduced in size although the frequency of the current flowing through the transformer inductors 11A and 11B is increased by individual magnetic materials.

  The diodes 12A and 12B are preferably high-speed diodes having a very short period (recovery period) in which a current (recovery current) can flow when the diodes are switched from on to off, but may not be high-speed diodes. The recovery current is a phenomenon that appears prominently when using a control method that shifts the diode from on to off by applying a large reverse bias when the current is flowing through the diode. This is because the power supply apparatus does not take such a control method as will be described later, and a recovery current hardly occurs. Note that elements similar to the rectifying switching elements 13A and 13B can be used instead of the diodes 12A and 12B.

  The rectifying switching elements 13A and 13B are power elements having a rating larger than the magnitude of the current flowing through the rectifying switching elements 13A and 13B. For example, a MOS-FET (Metal Oxide Semiconductor-Field Effect Transistor) capable of handling high frequency and small power Insulated Gate Bipolar Transistor (IGBT) that can handle low frequency (˜20 kHz) and high power (several tens of kW). For the purpose of protecting the rectifying switching elements 13A and 13B, a diode (not shown) is connected to the rectifying switching elements 13A and 13B so that the rectifying direction is opposite to the rectifying direction of the rectifying switching elements 13A and 13B. May be connected in parallel.

The transformer inductors 11A and 11B constitute a transformer as described above, and this transformer is represented by an equivalent circuit as shown in FIG. The equivalent circuit consists of a transformer inductor 11A, the mutual inductance Lm of 11B, and the leakage inductor L _11A of the transformer inductor 11A, the leakage inductance L _11B of the transformer inductor 11B. One end of each of the mutual inductor Lm, the leakage inductor L _11A and the leakage inductor L _11B is connected in common, the other end of the mutual inductor Lm is connected to the input line Lin, and the other end of the leakage inductor L _11A is connected to the connection point J1. The other end of the leakage inductor L _11B is connected to the connection point J2.

  The mutual inductor Lm mainly temporarily supplies the energy supplied from the DC power source P during the period when the rectifying switching element 13A or 13B is on (on period, period a, b, d, e described later). The accumulated energy is temporarily released to the load L during a period in which the rectifying switching element 13A or 13B is off (off period, periods c and f described later). Further, since the electromotive force is generated in the mutual inductor Lm so as to maintain the magnitude of the current flowing in the mutual inductor Lm during the period in which the mutual inductor Lm releases energy to the load L (off period), the rectifying switching element During the OFF period of 13A, a voltage obtained by subtracting the ON voltage of the rectifying switching element 13A from the voltage obtained by adding the electromotive force of the mutual inductor Lm to the DC input voltage Vin of the DC power supply P is applied to the load L, and the rectifying switching element 13B During the OFF period, a voltage obtained by subtracting the ON voltage of the rectifying switching element 13B from the voltage obtained by adding the electromotive force of the mutual inductor Lm to the DC input voltage Vin of the DC power supply P is applied to the load L. Thus, the voltage conversion units 1A and 1B have a function of boosting the DC input voltage Vin of the DC power supply P.

Leakage inductors L _11A and L _11B mainly function to keep the current flowing through themselves when the rectifying switching element 13A or 13B shifts from on to off. Specifically, when the rectifying switching element 13A or 13B shifts from on to off, the leakage inductor L _11A passes a current flowing through the rectifying switching element 13A through the diode 12A, while the leakage inductor L _{11A 11B} is configured to pass a current flowing through the rectifying switching element 13B through the diode 12B.

As a result, the leakage inductor L _11A has a voltage (forward bias) that immediately turns on the diode 12A directly connected to the rectifying switching element 13A when the rectifying switching element 13A shifts from on to off. It acts to be applied to 12A. Also, the leakage inductor L _11A is such that the diode 12A directly connected to the rectifying switching element 13A is not immediately turned off when another rectifying switching element 13B different from the rectifying switching element 13A shifts from OFF to ON. Such a voltage (forward bias) is applied to the diode 12A. As a result, a voltage (reverse bias) is applied to the diode 12B so that no current flows through the diode 12B directly connected to the other rectifying switching element 13B that is next switched from OFF to ON. On the other hand, the leakage inductor L _11B has such a voltage (forward bias) that the diode 12B directly connected to the rectifying switching element 13A is immediately turned on when the rectifying switching element 13B shifts from on to off. It acts to be applied to. The leakage inductor L _11B also prevents the diode 12B directly connected to the rectifying switching element 13B from being immediately turned off when another rectifying switching element 13A different from the rectifying switching element 13B shifts from off to on. Such a voltage (forward bias) is applied to the diode 12B. As a result, a voltage (reverse bias) is applied to the diode 12A so that no current flows through the diode 12A directly connected to the other rectifying switching element 13A that is subsequently switched from OFF to ON. As a result, no recovery current is generated in any of the diodes 12A and 12B.

  The smoothing capacitor 3 is for smoothing the output voltage from the voltage conversion units 1A and 1B, and is connected to the output line Lout and the common line Lc after the voltage conversion units 1A and 1B. Yes.

The control unit 4 is for operating the rectifying switching elements 13A and 13B sequentially in different phases. Specifically, the period when the rectifying switching element 13A is on and the rectifying switching element 13B is off (the first on period, the period of states a and b described later), and the rectifying switching element 13A is off. The period during which the rectifying switching elements 13B are on (second on period, periods d and e described later) and the period during which both the rectifying switching elements 13A and 13B are off (off period, states described later) The control signals S _A and S _B whose duty ratio D is controlled are output to the rectifying switching elements 13A and 13B so as to be alternately repeated through the periods c) and f). That is, the control unit 4 does not use a control method in which the rectifying switching elements 13A and 13B are simultaneously turned on or the on periods overlap.

Here, the duty ratio D of the control signals S _A and S _B output to the rectifying switching elements 13A and 13B is set to be smaller than 50% in order to sequentially operate the rectifying switching elements 13A and 13B at different phases. Therefore, the control signals S _A and S _B are set to a duty ratio D in a range larger than 0% and smaller than 50%. When the duty ratio D is increased, the step-up ratio (Vout / Vin) increases, and the magnitude of the output DC voltage Vout output to the load L can be increased. Conversely, when the duty ratio D is reduced, the step-up ratio (Vout / Vin) is reduced, and the magnitude of the output DC voltage Vout output to the load L can be reduced.

For example, the input DC voltage Vin is 200 V, the inductance of the mutual inductor Lm is 285 μH, the inductance of the leakage inductors L _11A and L _11B is 15 μH, the frequency of the control signals S _A and S _B is 20 kHz, and the current flowing through the load L is 10 _A. FIG. 3 shows the relationship between the duty ratio D and the input / output voltage ratio (Vout / Vin). As can be seen from FIG. 3, when the duty ratio D is about 0.27, the input / output voltage ratio (Vout / Vin) is 2.

  Next, the operation of the switching power supply device configured as described above will be described. FIG. 4 is a timing chart of main signals in the switching power supply apparatus, and FIGS. 5 to 10 schematically show signal flows in each of the states a to f in FIG. Hereinafter, for convenience of description, description will be made in order from state b after a while since the rectifying switching element 13A is turned on.

In the state b, the control unit 4 outputs a control signal S _A to the rectification switching element 13A, and stops outputting the control signal S _B. Therefore, as shown in FIG. 5, the current I _13A flows through the rectifying switching element 13A, and the rectifying switching element 13B is off. At this time, the same voltage as that on the negative electrode side of the DC power source P is applied to the connection point J1, and the voltage obtained by dividing the DC input voltage Vin of the DC power source P by the impedance of the mutual inductor Lm and the leakage inductor L _11A is passed through the leakage inductor L _11B . To the connection point J2. The voltage on the anode side of the diode 12A is the same voltage as that of the connection point J1, and is lower than the output DC voltage Vout on the cathode side of the diode 12A. Further, the voltage on the anode side of the diode 12B is the same voltage as that of the connection point J2, and is lower than the output DC voltage Vout on the cathode side of the diode 12B. Therefore, reverse bias is applied to the diodes 12A and 12B during this period. Accordingly, the current I _m flowing through the mutual inductance Lm flows to the DC power source P through the leakage inductor L _11A and rectification switching element 13A. At this time, it increases the current I _m flowing through the mutual inductance Lm is gradually energy is gradually accumulated in the mutual inductance Lm.

Next, the control unit 4 stops the output of the control signal S _A and turns off both the rectifying switching elements 13A and 13B. As a result, the state b is shifted to the state c. At this time, as shown in FIG. 6, the leakage inductor L _11A acts so as to turn on the diodes 12A, current I _12A flows to the diode 12A, releasing the energy stored in the mutual inductance Lm is the load L Is done. As a result, the currents I _m , I _11A and I _12A gradually decrease. Further, since the output DC voltage Vout is applied to the cathode of the diode 12B and the reverse bias is applied to the diode 12B, the diode 12B maintains the OFF state.

Next, the control unit 4 turns on the rectifying switching element 13B outputs a control signal S _B. As a result, the state c is shifted to the state d. At this time, as shown in FIG. 7, the voltage at the connection point J2 side of the leakage inductance L _11B becomes substantially equal to the voltage of the negative side of the DC power source P, it becomes lower than the other end of the leakage inductance L _11B, The currents I _11B and I _13B begin to flow through the leakage inductor L _11B and the rectifying switching element 13B, and the currents I _m , I _11A and I _12A continue to decrease.

By the way, the diode 12A in which current flows immediately before the rectifying switching element 13B is turned on does not immediately turn off when the rectifying switching element 13B is turned on. This is because the leakage inductor L _11A acts so as to keep current flowing through the diode 12A. In other words, a voltage (forward bias) is applied to the diode 12A so that the diode 12A does not immediately turn off. This is because it works. Thus, since the current I _12A flowing through the diode 12A gradually decreases after the rectifying switching element 13B is turned on, there is no room for generating a recovery current in the diode 12A.

  On the other hand, when the rectifying switching element 13B is turned on, the diode 12B directly connected to the rectifying switching element 13B is already turned off, and no current flows. This is because, as a result of applying a voltage (forward bias) that does not immediately turn off the diode 12A to the diode 12A, a voltage (reverse bias) that does not flow current to the diode 12B is applied to the diode 12B. It is. Thereby, there is no room for generating a recovery current in the diode 12B even if the rectifying switching element 13B shifts from OFF to ON.

  Thus, there is no room for generating a recovery current in either of the diodes 12A and 12B.

Thereafter, as shown in FIG. 8, the leakage inductor L _11A and the currents I _11A and I _{12A of} the diode 12A become zero, thereby shifting from the state d to the state e. At this time, and the current I _11B of the leakage inductor L _11B, along with the current I _13B of rectification switching element 13B is gradually increased, the current also I _m of the mutual inductor Lm increases, energy is accumulated in the mutual inductance Lm To go.

Next, the control unit 4 turns off both rectification switching element 13A, 13B and stops the output of the control signal S _B. As a result, the state e changes to the state f. At this time, as shown in FIG. 9, since the leakage inductor L _11B acts to turn on the diode 12B, the current I _12B flows through the diode 12B, and the energy accumulated in the mutual inductor Lm is released to the load L. Is done. As a result, the currents I _m , I _11B and I _12B are gradually decreased. In addition, since the output DC voltage Vout is applied to the cathode of the diode 12A and the reverse bias is applied to the diode 12A, the diode 12A maintains the OFF state.

Next, the control unit 4 shifts from a state f the state a by turning on the rectification switching element 13A outputs a control signal S _A. At this time, as shown in FIG. 10, the voltage at the connection point J1 side of the leakage inductance L _11A is substantially equal to the voltage of the negative side of the DC power source P, it becomes lower than the other end of the leakage inductance L _11A, The currents I _11A and I _13A begin to flow through the leakage inductor L _11A and the rectifying switching element 13A, and the currents I _m , I _11B and I _12B continue to decrease.

By the way, the diode 12B in which current flows immediately before the rectifying switching element 13A is turned on does not immediately turn off when the rectifying switching element 13A is turned on. This is because the leakage inductor L _11B acts to keep current flowing through the diode 12B. In other words, a voltage (forward bias) is applied to the diode 12B so that the diode 12B does not turn off immediately. This is because it works. Thus, since the current I _12B flowing through the diode 12B gradually decreases after the rectifying switching element 13A is turned on, there is no room for generating a recovery current in the diode 12B.

  On the other hand, when the rectifying switching element 13A is turned on, the diode 12A directly connected to the rectifying switching element 13A is already off and no current flows. This is because, as a result of applying a voltage (forward bias) to the diode 12B that does not immediately turn off the diode 12B, a voltage (reverse bias) that does not flow current to the diode 12A is applied to the diode 12A. It is. As a result, there is no room for a recovery current to be generated in the diode 12A even when the rectifying switching element 13A shifts from OFF to ON.

  Thus, there is no room for generating a recovery current in either of the diodes 12A and 12B.

  In this way, the switching power supply device transforms (boosts) the DC input voltage Vin supplied from the DC power supply P to the DC output voltage Vout, and supplies the transformed DC output voltage Vout to the load L.

  Next, the effect of the switching power supply device of the present embodiment will be described in comparison with the conventional example.

  As shown in FIG. 39, the switching power supply according to the conventional example has one voltage booster composed of a voltage conversion unit 101 including an inductor 111, a diode 112 and a switching element 113 wound around a core 114, and a smoothing capacitor 103. Transformer inductors 11A and 11B wound around a common core 20 so as to have the same polarity as that of the step-up chopper circuit and the control unit 104. This is different from the switching power supply device of the present embodiment in that it does not include a transformer.

In the conventional example, as shown in FIG. 40, when the control signal S ₁₀₁ is input from the control unit 104 to the switching element 113 and the switching element 113 is turned on, the voltage on the anode side of the diode 112 through which current flows is DC. The voltage is substantially the same as that on the negative side of the power supply P, and is smaller than the voltage on the cathode side of the diode 112 (output DC voltage Vout). As a result, a large reverse bias is applied to the diode 112 through which current flows, and the diode 112 shifts from on to off. At this time, since there is a period (recovery period) in which current can flow in the reverse direction at the moment when the diode 112 shifts from on to off, current (recovery current) flows through the diode 112 during the recovery period. This recovery current flows through the switching element 113 directly connected to the diode 112 to generate a ripple (see within a one-dot chain line frame in FIG. 40). As a result, a switching loss (specifically, a loss at turn-on) occurs in the switching element 113, so that the efficiency of the switching power supply device according to the conventional example is lowered.

In addition, since the current I ₁₁₁ flowing through the inductor 111 has a low frequency according to the on / off state of the control signal S ₁₀₁ , it is difficult to reduce the size of the core 114, thereby preventing the switching power supply from being reduced in size and weight. Become.

On the other hand, in the present embodiment, as described above, the diode 12B or 12A in which a current flows immediately before the rectifying switching element 13A or 13B is turned on is connected to the rectifying switching element by the action of the leakage inductor L _11B or L _11A. When the element 13A or 13B is turned on, the element 13A or 13B is not turned off immediately, and the room for generating a recovery current in the diode 12B or 12A is eliminated. Further, when the rectifying switching element 13A or 13B is turned on, the diode 12B or 12A directly connected to the rectifying switching element 13A or 13B is turned off in advance, and there is room for a recovery current to be generated in the diode 12B or 12A. Is eliminated.

  As described above, in the switching power supply according to the present embodiment, any one of the switching elements 13A and 13B is provided by including a transformer including the transformer inductors 11A and 11B wound around the common core 20 so as to have the same polarity. Even when the diode is turned from OFF to ON, generation of a recovery current in each of the diodes 12A and 12B is prevented. Thereby, since the switching loss in each switching element 13A, 13B reduces, the fall of the efficiency resulting from a recovery current can be prevented.

In addition, a transformer including transformer inductors 11A and 11B wound around a common core 20 so as to have the same polarity is provided, and the control unit 4 sequentially operates the rectifying switching elements 13A and 13B in different phases. by, the current I _m flowing through the mutual inductance Lm, as shown in FIG. 4, the control signal S _a, on the S _B, twice the frequency of the frequency corresponding to the off, thereby, the size of the core 20 Can be reduced. As a result, the switching power supply device can be reduced in size and weight.

By the way, it is possible to reduce the size of the core 20 by increasing the driving frequency of the rectifying switching elements 13A and 13B. In this case, however, the switching loss in the rectifying switching elements 13A and 13B is increased by increasing the driving frequency. Increases, and heat generation also increases. In order to dissipate the heat, a large heat sink is required. However, if such a large heat sink is attached, the switching power supply cannot be downsized. On the other hand, in this embodiment, the rectification switching element 13A, without increasing the driving frequency of 13B, it is possible to increase the frequency of the current I _m flowing through the mutual inductance Lm. Further, rectification switching element 13A, even the upper limit of the driving frequency of 13B is a case low, the frequency of the current I _m flowing through the mutual inductance Lm can be larger than the upper limit. Therefore, the switching loss in each of the rectifying switching elements 13A and 13B does not increase significantly as in the previous example. As a result, since it is not necessary to attach a large heat sink, the switching power supply device can be easily reduced in size and weight.

  As described above, the switching power supply according to the present embodiment can prevent the efficiency from being reduced due to the recovery current and can be reduced in size and weight.

[Modification]
Next, a modification of the above embodiment will be described.

  In the above embodiment, the case where the present invention is applied to a switching power supply device in which two boost chopper circuits are connected in parallel has been described, but even when three or more boost chopper circuits are connected in parallel, Applicable. For example, as shown in FIG. 11, when four step-up chopper circuits are connected in parallel, the switching power supply unit includes a voltage converter 1C including a transformer inductor 11C, a diode 12C, and a rectifier switching element 13C, A voltage converter 1D composed of an inductor 11D, a diode 12D and a rectifying switching element 13D is connected in parallel to the voltage converters 1A and 1B, and the transformer inductors 11A, 11B, 11C and 11D have the same polarity. It is configured by winding around a common core 20. Even in such a switching power supply device, even when any of the switching elements 13A to 13D is switched from OFF to ON, generation of a recovery current in each of the diodes 12A, 12B, 12C, and 12D is prevented. . Thereby, since the switching loss in each switching element 13A, 13B, 13C, 13D reduces, the fall of the efficiency resulting from a recovery current can be prevented.

In addition, a transformer including transformer inductors 11A, 11B, 11C, and 11D wound around a common core 20 so as to have the same polarity is provided, and each rectifying switching element 13A, 13B, 13C, and 13D is controlled by the control unit 4. Are sequentially operated in different phases, the current flowing through the transformer inductors 11A, 11B, 11C, and 11D has a frequency that is four times the frequency corresponding to the on / off state of the control signal, as shown in FIG. . As described above, the number n of boosting chopper circuits (n is an integer of 2 or more) is increased, and each rectifying switching element is sequentially turned on and off by the control unit 4, thereby increasing the number of boosting chopper circuits. In comparison, the frequency can be increased by n times. As a result, the switching power supply device can be reduced in size and weight as long as the size of the core can be reduced more than the increase in the occupied volume due to the increase in the step-up type chopper circuit. Also, as in the above embodiment, the rectification switching element 13A, 13B, @ 13 C, even without increasing the drive frequency of 13D, it is possible to increase the frequency of the current I _m flowing through the mutual inductance Lm, each The switching loss in the rectifying switching elements 13A, 13B, 13C, and 13D does not increase significantly as in the previous example. As a result, since it is not necessary to attach a large heat sink, the switching power supply device can be easily reduced in size and weight.

  In the above embodiment, it is assumed that the input is performed from the input terminals T1 and T2 and the output is performed from the output terminals T3 and T4. However, as shown in FIG. When the elements 14A and 14B are connected in parallel with reverse polarity, and the diodes 15A and 15B are connected in parallel with reverse polarity to the rectifying switching elements 13A and 13B, they are input from the output terminals T3 and T4, and input terminal T1 , T2 can also be output. However, when input is made from the output terminals T3 and T4, the switching power supply device is a step-down type.

  In the above embodiment, the case where the present invention is applied to a switching power supply device incorporating a step-up chopper circuit has been described. However, the present invention is as shown in FIGS. 14 and 15 (equivalent circuit of FIG. 14). This is applicable even when a step-down chopper circuit is incorporated. In the switching power supply device at this time, one end of the transformer inductors 11A and 11B is connected to the output line Lout, the anodes of the diodes 12A and 12B are connected to the common line Lc, and the collectors of the rectifying switching elements 13A and 13B are connected to the input line Lin. It is different from the above embodiment in that it is connected to. In this switching power supply device, the waveform is the same as that in FIG. Therefore, as in the case of the above-described embodiment, it is possible to prevent a reduction in efficiency due to the recovery current and to reduce the size and weight.

  Further, the present invention is applicable even when a step-up / step-down chopper circuit is incorporated as shown in FIGS. 16 and 17 (equivalent circuit of FIG. 16). In this switching power supply device, one end of the transformer inductors 11A and 11B is connected to the common line Lc, the anodes of the diodes 12A and 12B are connected to the output line Lout, and the collectors of the rectifying switching elements 13A and 13B are connected to the input line Lin. It is different from the above embodiment in that it is connected to. In this switching power supply device, the waveform is the same as that in FIG. Therefore, as in the case of the above-described embodiment, it is possible to prevent a reduction in efficiency due to the recovery current and to reduce the size and weight.

[Second Embodiment]
FIG. 18 shows a circuit configuration of a switching power supply according to the second embodiment of the present invention. This switching power supply device has a circuit configuration equivalent to the equivalent circuit of this transformer (see FIG. 2) instead of the transformer composed of the transformer inductors 11A and 11B wound around the common core 20 so as to have the same polarity. It is provided. Specifically, instead of the leakage inductor L _11A, an inductor L1 having an inductance equivalent to the leakage inductor L _11A provided, instead of the leakage inductor L _11B, provided the inductor L2 having a leakage inductor L _11B equivalent inductance Instead of the mutual inductor Lm, a common inductor L3 having an inductance equivalent to the mutual inductor Lm is provided. As described above, in the switching power supply device according to the present embodiment, since the circuit configuration equivalent to the equivalent circuit of the transformer is provided, as in the above embodiment, a reduction in efficiency due to the recovery current is prevented, and It can be reduced in size and weight. Note that the circuit configuration equivalent to the equivalent circuit of the transformer is provided in the same manner as in the present embodiment even in the step-down chopper circuit of FIG. 14 and the switching power supply device incorporating the step-up chopper circuit of FIG. Of course, this is possible (FIGS. 19 and 20).

  Here, when the common inductor L3 is removed and the inductances of the inductors L1 and L2 are made equal to that of the mutual inductor in the circuits of FIGS. 18, 19 and 20, the same as the circuit of FIG. Since it operates, a recovery current is generated and efficiency is lowered. 18, 19, and 20, the common inductor L <b> 3 is removed, and the inductances of the inductors L <b> 1 and L <b> 2 are equal to the mutual inductor, or the circuits of FIGS. 18, 19, and 20. When the common inductor L3 is removed and the inductances of the inductors L1 and L2 are made equal to the leakage inductor, a current having a frequency equal to the drive frequency of the rectifying switching elements 13A and 13B flows through the inductors L1 and L2. Since it is on the wiring, to reduce the size of the core 20, the drive frequency of the rectifying switching elements 13A and 13B must be increased. Therefore, when the upper limit of the driving frequency of the rectifying switching elements 13A and 13B is low, the core 20 cannot be downsized. On the other hand, since the circuits of FIGS. 18, 19 and 20 have the same operations and effects as those of the above-described embodiment, it can be said that the circuits have unique operations and effects different from these circuits.

[Third Embodiment]
FIG. 21 shows a circuit configuration of a switching power supply apparatus according to the third embodiment of the present invention. In this switching power supply device, an auxiliary inductor 30A is provided in the switching power supply device of FIG.

  The auxiliary inductor 30A has an inductance value smaller than that of the mutual inductor Lm, and is arranged between the connection point J1 of the voltage conversion unit 1A and the connection point J2 of the voltage conversion unit 1B.

  The auxiliary inductor 30A mainly functions to keep the current flowing through itself when the rectifying switching element 13A or 13B shifts from on to off. Specifically, when the rectifying switching element 13A or 13B shifts from on to off, the auxiliary inductor 30A causes the current flowing through the rectifying switching element 13A to flow through the diode 12A or through the diode 12A. The current flowing through the rectifying switching element 13B is caused to flow through the diode 12B.

  As a result, the auxiliary inductor 30A has a voltage (for example, such that when one rectifying switching element 13A shifts from on to off, the diode 12A directly connected to the one rectifying switching element 13A immediately turns on ( Forward bias) is applied to the diode 12A. The auxiliary inductor 30A includes a diode 12A directly connected to one rectifying switching element 13A when another rectifying switching element 13B different from the one rectifying switching element 13A shifts from OFF to ON. A voltage (forward bias) that does not immediately turn off is applied to the diode 12A. As a result, a voltage (reverse bias) is applied to the diode 12B so that no current flows through the diode 12B directly connected to the other rectifying switching element 13B that is next switched from OFF to ON. As a result, no recovery current is generated in any of the diodes 12A and 12B.

  Next, the operation of the switching power supply device configured as described above will be described. FIG. 22 is a timing chart of main signals in the switching power supply apparatus, and FIGS. 23 to 30 schematically show the flow of signals in each of the states g to p in FIG. Hereinafter, for convenience of explanation, description will be made in order from a state i after a while since the rectifying switching element 13A is turned on.

In the state i, the control unit 4 outputs a control signal S _A to the rectification switching element 13A, and stops outputting the control signal S _B. Therefore, as shown in FIG. 23, the current I _13A flows through the rectifying switching element 13A, and the rectifying switching element 13B is off. At this time, the same voltage as the negative side of the DC power supply P is applied to the connection point J1 side of the auxiliary inductor 30A, and the voltage obtained by dividing the DC input voltage Vin of the DC power supply P by the impedance of the transformer inductor 11B and the auxiliary inductor 30A is connected. Take the J2 side. The voltage on the anode side of the diode 12A is the same voltage as that of the connection point J1, and is lower than the output DC voltage Vout on the cathode side of the diode 12A. Further, the voltage on the anode side of the diode 12B is the same voltage as that of the connection point J2, and is lower than the output DC voltage Vout on the cathode side of the diode 12B. Therefore, reverse bias is applied to the diodes 12A and 12B during this period. As a result, the current I _11A flowing through the transformer inductor 11A flows to the DC power source P via the rectifying switching element 13A, and the current I _11B flowing through the transformer inductor 11B passes through the transformer inductor 11B, the auxiliary inductor 30A, and the rectifying switching element 13A. And flows to the DC power source P. At this time, currents I _11A and I _11B flowing through the transformer inductors 11A and 11B gradually increase, and energy is gradually stored in the transformer inductors 11A and 11B.

Next, the control unit 4 stops the output of the control signal S _A and turns off both the rectifying switching elements 13A and 13B. As a result, the state i is shifted to the state j. At this time, as shown in FIG. 24, the auxiliary inductor 30A flows the current I _30A from the connection point J2 side to the connection point J1 side, and acts to turn on the diode 12A. Therefore, the current I _12A flows to the diode _12A. Flows, and the energy stored in the transformer inductors 11A and 11B is released to the load L. As a result, the currents I _11A , I _11B and I _30A gradually decrease. Further, since the output DC voltage Vout is applied to the cathode of the diode 12B and the reverse bias is applied to the diode 12B, the diode 12B maintains the OFF state.

Next, the control unit 4 turns on the rectifying switching element 13B outputs a control signal S _B. As a result, the state j is shifted to the state k. At this time, as shown in FIG. 25, the voltage on the connection point J2 side of the auxiliary inductor 30A is substantially the same as the voltage on the negative electrode side of the DC power supply P, and is lower than the connection point J1 side. Current I _13B starts to flow, current I _30A flowing from the connecting point J2 side of the auxiliary inductor 30A to the connecting point J1 side starts to decrease, the current I _11A of the transformer inductor 11A continues to decrease, and the current I of the transformer inductor 11B decreases. _11B begins to increase.

By the way, the diode 12A in which current flows immediately before the rectifying switching element 13B is turned on does not immediately turn off when the rectifying switching element 13B is turned on. This is because the auxiliary inductor 30A acts so as to keep current flowing through the diode 12A. In other words, a voltage (forward bias) is applied to the diode 12A so that the diode 12A is not immediately turned off. It is to do. Thus, since the current I _12A flowing through the diode 12A gradually decreases after the rectifying switching element 13B is turned on, there is no room for generating a recovery current in the diode 12A.

  On the other hand, when the rectifying switching element 13B is turned on, the diode 12B directly connected to the rectifying switching element 13B is already turned off, and no current flows. This is because, as a result of applying a voltage (forward bias) that does not immediately turn off the diode 12A to the diode 12A, a voltage (reverse bias) that does not flow current to the diode 12B is applied to the diode 12B. It is. Thereby, there is no room for generating a recovery current in the diode 12B even if the rectifying switching element 13B shifts from OFF to ON.

  Thus, there is no room for generating a recovery current in either of the diodes 12A and 12B.

Thereafter, as shown in FIG. 26, the current I _30A flowing from the connection point J2 side of the auxiliary inductor 30A to the connection point J1 side becomes zero, the direction of the current I _30A of the auxiliary inductor 30A is reversed, and from the connection point J1 side. By starting to flow toward the connection point J2, the state transitions from state k to state m. At this time, and the current I _12A of diode 12A with the current I _30A which flows from the connection point J1 side of the auxiliary inductor 30A to the connection point J2 side increases, current I _11A of the transformer inductor 11A is reduced, the transformer inductor 11B The current I _11B and the current I _13B of the rectifying switching element 13B increase.

Thereafter, as shown in FIG. 27, when the current I _{12A of the} diode 12A becomes zero, the state m is shifted to the state n. At this time, the auxiliary and the current I _30A of inductor 30A, along with the current I _11B of the transformer inductor 11B is gradually increased, the transformer inductor 11A, the current I _11A flowing in 11B, I _11B increases, transformers inductors 11A, 11B The energy is accumulated in.

Next, the control unit 4 turns off both rectification switching element 13A, 13B and stops the output of the control signal S _B. As a result, the state n is shifted to the state p. At this time, as shown in FIG. 28, the auxiliary inductor 30A flows the current I _30A from the connection point J1 side to the connection point J2 side, and acts to turn on the diode 12B, so that the current I _12B flows to the diode _12B. Flows, and the energy stored in the transformer inductors 11A and 11B is released to the load L. As a result, the currents I _11A , I _11B and I _30A gradually decrease. In addition, since the output DC voltage Vout is applied to the cathode of the diode 12A and the reverse bias is applied to the diode 12A, the diode 12A maintains the OFF state.

Next, the control unit 4 shifts from the state p to the state g by turning on the rectification switching element 13A outputs a control signal S _A. At this time, as shown in FIG. 29, the voltage on the connection point J1 side of the auxiliary inductor 30A is substantially the same as the voltage on the negative electrode side of the DC power supply P and is lower than the connection point J2 side. Current I _13A starts to flow, current I _30A flowing from the connection point J1 side to the connection point J2 side of the auxiliary inductor 30A starts to decrease, the current I _11B of the transformer inductor 11B continues to decrease, and the current I of the transformer inductor 11A decreases. _11A begins to increase.

By the way, the diode 12B in which current flows immediately before the rectifying switching element 13A is turned on does not immediately turn off when the rectifying switching element 13A is turned on. This is because the auxiliary inductor 30A acts so as to keep a current flowing through the diode 12B. In other words, a voltage (forward bias) that does not immediately turn off the diode 12B is applied to the diode 12B. It is to do. Thus, since the current I _12B flowing through the diode 12B gradually decreases after the rectifying switching element 13A is turned on, there is no room for generating a recovery current in the diode 12B.

  On the other hand, when the rectifying switching element 13A is turned on, the diode 12A directly connected to the rectifying switching element 13A is already off and no current flows. This is because, as a result of applying a voltage (forward bias) to the diode 12B that does not immediately turn off the diode 12B, a voltage (reverse bias) that does not flow current to the diode 12A is applied to the diode 12A. It is. As a result, there is no room for a recovery current to be generated in the diode 12A even when the rectifying switching element 13A shifts from OFF to ON.

  Thus, there is no room for generating a recovery current in either of the diodes 12A and 12B.

Thereafter, as shown in FIG. 30, the current I _30A flowing from the connection point J1 side to the connection point J2 side of the auxiliary inductor 30A becomes zero, the direction of the current I _30A flowing through the auxiliary inductor 30A is reversed, and the connection point J2 side Starts to flow to the connection point J1 side, thereby shifting from the state g to the state h. At this time, and the current I _12B of diode 12B with the current I _30A which flows from the connection point J2 side of the auxiliary inductor 30A to the connection point J1 side increases, current I _11B of the transformer inductor 11B is reduced, the transformer inductor 11A The current I _11A and the current I _13A of the rectifying switching element 13A increase. Thereafter, when the current I _12B flowing through the diode 12B becomes zero, the state i is entered.

  In this way, the switching power supply device transforms (boosts) the DC input voltage Vin supplied from the DC power supply P to the DC output voltage Vout, and supplies the transformed DC output voltage Vout to the load L.

  Next, effects of the present switching power supply device will be described. In the present embodiment, as described above, the diode 12B or 12A in which the current flows immediately before the rectifying switching element 13A or 13B is turned on is turned on by the action of the auxiliary inductor 30A. Therefore, the diode 12B or 12A eliminates a room for generating a recovery current. Further, when the rectifying switching element 13A or 13B is turned on, the diode 12B or 12A directly connected to the rectifying switching element 13A or 13B is turned off in advance, and there is room for a recovery current to be generated in the diode 12B or 12A. Is eliminated.

  As described above, the switching power supply according to the present embodiment includes not only the transformer composed of the transformer inductors 11A and 11B wound around the common core 20 so as to have the same polarity but also the auxiliary inductor 30A. This has the same effect as the first embodiment described above.

[Modification]
Next, a modification of the second embodiment will be described.

  In the second embodiment, the case where the present invention is applied to a switching power supply device in which two boost chopper circuits are connected in parallel has been described. However, this is the case where three or more boost chopper circuits are connected in parallel. Is applicable. For example, as shown in FIG. 31, when the voltage converters 1A, 1B, 1C, 1D in the switching power supply device of FIG. 11 are connected to each other by the auxiliary inductors 30A, 30B, 30C, 30D, the switching elements 13A to 13D Even when any of these switches from OFF to ON, generation of a recovery current in each of the diodes 12A, 12B, 12C, and 12D is prevented. Thereby, since the switching loss in each switching element 13A, 13B, 13C, 13D reduces, the fall of the efficiency resulting from a recovery current can be prevented. Further, as with the switching power supply device of FIG. 11, a reduction in size and weight can be easily realized.

  As shown in FIG. 32, even when the auxiliary inductors 30A, 30B, 30C, and 30D in FIG. 31 are connected in a star shape, they have the same operations and effects as the circuit in FIG.

  In the above embodiment, it is assumed that the input is performed from the input terminals T1 and T2 and the output is performed from the output terminals T3 and T4. However, as illustrated in FIG. 33, each of the switching power supply devices illustrated in FIG. When the voltage converters 1A and 1B are connected to each other by the auxiliary inductor 30A, it is possible to input from the output terminals T3 and T4 side and to output from the input terminals T1 and T2 side. However, when input is made from the output terminals T3 and T4, the switching power supply device is a step-down type.

  Further, in the above embodiment, the case where the present invention is applied to a switching power supply device incorporating a boost chopper circuit has been described, but the present invention may include a step-down chopper circuit as shown in FIG. As shown in FIG. 35, the present invention can be applied even when a buck-boost chopper circuit is incorporated. These switching power supplies are configured by connecting the voltage converters 1A and 1B in FIGS. 14 and 16 to each other by an auxiliary inductor 30A. Therefore, similarly to the switching power supply devices of FIGS. 14 and 16, it is possible to prevent the efficiency from being reduced due to the recovery current and to reduce the size and weight.

  In the above embodiment, the transformer is formed of the transformer inductors 11A and 11B wound around the common core 20 so as to have the same polarity. However, instead of this transformer, the equivalent of this transformer is provided. You may make it provide the circuit structure equivalent to a circuit. Specifically, as shown in FIG. 36, the voltage converters 1A and 1B in the switching power supply device of FIG. 18 may be connected to each other by an auxiliary inductor 30A. It is of course possible to provide a circuit configuration equivalent to the equivalent circuit of the transformer in the step-down chopper circuit of FIG. 34 and the switching power supply device incorporating the step-up chopper circuit of FIG. 37, FIG. 38). The circuits shown in FIGS. 36, 37, and 38 have the same operations and effects as the circuits shown in FIGS. 18, 19, and 20.

  While the present invention has been described with reference to the embodiments, the present invention is not limited to these and various modifications can be made.

1 is a circuit diagram of a switching power supply device according to a first embodiment of the present invention. It is an equivalent circuit diagram of a switching power supply device. It is a relationship figure for demonstrating the relationship between a duty ratio and an input DC voltage ratio. It is a wave form diagram for demonstrating operation | movement of a switching power supply device. FIG. 6 is a circuit diagram for explaining an operation in a state b. It is a circuit diagram for demonstrating the operation | movement in the state c. It is a circuit diagram for demonstrating the operation | movement in the state d. FIG. 6 is a circuit diagram for explaining an operation in a state e. FIG. 6 is a circuit diagram for explaining an operation in a state f. It is a circuit diagram for demonstrating the operation | movement in the state a. It is a circuit diagram of the switching power supply device concerning one modification. It is a wave form diagram for demonstrating operation | movement of the switching power supply device of FIG. It is a circuit diagram of the switching power supply concerning other modifications. It is a circuit diagram of the switching power supply concerning other modifications. FIG. 15 is an equivalent circuit diagram of the switching power supply device of FIG. 14. It is a circuit diagram of the switching power supply concerning other modifications. FIG. 17 is an equivalent circuit diagram of the switching power supply device of FIG. 16. It is a circuit diagram of the switching power supply device concerning the 2nd Embodiment of this invention. . It is a circuit diagram of the switching power supply device concerning one modification. It is a circuit diagram of the switching power supply concerning other modifications. It is a circuit diagram of the switching power supply device which concerns on the 3rd Embodiment of this invention. It is a wave form diagram for demonstrating operation | movement of a switching power supply device. 6 is a circuit diagram for explaining an operation in a state i. FIG. 6 is a circuit diagram for explaining an operation in a state j. FIG. It is a circuit diagram for demonstrating the operation | movement in the state k. 6 is a circuit diagram for explaining an operation in a state m. FIG. 6 is a circuit diagram for explaining an operation in a state n. FIG. 6 is a circuit diagram for explaining an operation in a state p. FIG. FIG. 6 is a circuit diagram for explaining an operation in a state g. It is a circuit diagram for demonstrating the operation | movement in the state h. It is a circuit diagram of the switching power supply device concerning one modification. It is a circuit diagram of the switching power supply concerning other modifications. It is a circuit diagram of the switching power supply concerning other modifications. It is a circuit diagram of the switching power supply concerning other modifications. It is a circuit diagram of the switching power supply concerning other modifications. It is a circuit diagram of the switching power supply concerning other modifications. It is a circuit diagram of the switching power supply concerning other modifications. It is a circuit diagram of the switching power supply concerning other modifications. It is a circuit diagram of the conventional switching power supply device. It is a wave form diagram for demonstrating operation | movement of the switching power supply device of FIG.

Explanation of symbols

1A, 1B, 1C, 1D ... Voltage conversion unit, 3, C ... Smoothing capacitor, 4 ... Control unit, 11A, 11B, 11C, 11D ... Transformer inductor, 12A, 12B, 12C, 12D ... Diode, 13A, 13B, 13C , 13D: Rectification switching element, 20: Core, 30A, 30B, 30C, 30D ... Auxiliary inductor, J1, J2 ... Connection point, L ... Load, L1, L2 ... First inductor, L3 ... Second inductor, _L11A , L _11B , L _11C , L _11D ... Leakage inductor, Lc ... Common line, Lin ... Input line, L _m ... Mutual inductor, Lout ... Output line, P ... DC power supply, T1, T2 ... Input terminal, T3, T4 ... Output terminal, Vin: input DC voltage, Vout: output DC voltage.

Claims (6)

A plurality of transformer chopper circuits each including a transformer inductor, a rectifier element and a switching element and connected in parallel to each other;
Control means for sequentially operating the respective switching elements in mutually different phases;
With
Each one end of the transformer inductor, the rectifier element and the switching element is commonly connected at a common connection point,
The switching power supply device, wherein the plurality of transformer chopper circuits are magnetically coupled so that transformer inductors have the same polarity. The transformer chopper circuit includes a pair of input terminals and a pair of output terminals,
One end of the transformer inductor is connected to one input terminal, and the other end is commonly connected to each end of the rectifier element and the switching element at a common connection point, and the other end of the rectifier element is connected to one output terminal. The switching power supply according to claim 1, wherein the other end of the switching element is connected to the other input terminal and the other output terminal. The transformer chopper circuit includes a pair of input terminals and a pair of output terminals,
One end of the switching element is connected to one input terminal and the other end is commonly connected to each end of the rectifier element and the transformer inductor at a common connection point, and the other end of the transformer inductor is connected to one output terminal. The switching power supply according to claim 1, wherein the other end of the rectifying element is connected to the other input terminal and the other output terminal. The transformer chopper circuit includes a pair of input terminals and a pair of output terminals,
One end of the switching element is connected to one input terminal and the other end is commonly connected to each end of the rectifying element and the transformer inductor at a common connection point, and the other end of the rectifying element is connected to one output terminal. The switching power supply according to claim 1, wherein the other end of the transformer inductor is connected to the other input terminal and the other output terminal. A plurality of transformer chopper circuits each including an inductor, a rectifier element and a switching element and connected in parallel to each other;
Control means for sequentially operating the respective switching elements in mutually different phases;
A switching power supply comprising: a common inductor provided in common to a plurality of transformer chopper circuits and commonly connected to one end of each of the inductors. An inductor of one transformer chopper circuit causes a current to flow through the rectifier element of the transformer chopper circuit when the switching element of the transformer chopper circuit shifts from on to off, and the switching element of the other transformer chopper circuit switches from off to on. Even after the transition to, the current continues to flow through the rectifying element of the one transformer chopper circuit,
The common inductor stores energy by a current flowing through the switching element while each switching element is on, and discharges the stored energy when all switching elements are off. 5. The switching power supply device according to 5.

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