JP2007068210A - Transmitting/receiving device, transmission method, and cell search method - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To improve channel estimation precision without an interference from DPCH. <P>SOLUTION: A transmission method, for spreading a transmission signal sequence over a plurality of subcarriers in a frequency direction and transmitting the subcarriers, is provided. The method comprises the steps of spreading the transmission signal sequence constituting a common pilot channel and separate channels over a two-dimensional (time and frequency) region using respective channelization codes, masking the spread signal with a scramble code for discrimination of stations, and transmitting the masked transmission signal sequence. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

  The present invention relates to a transmission / reception apparatus, a transmission method, and a cell search method. The present invention relates to a transmission / reception apparatus that masks a multiplexed signal with a scrambling code of a base station and transmits / receives the masked signal, a transmission method thereof, and a cell search method.

  As a next-generation mobile communication system, a multicarrier modulation system has attracted attention. By using the multi-carrier modulation scheme, not only can broadband high-speed data transmission be realized, but also the influence of frequency selective fading can be reduced by narrowing each subcarrier. In addition, by using orthogonal frequency division multiplexing (OFDM), frequency utilization efficiency can be increased, and by providing a guard interval for each OFDM symbol, the influence of intersymbol interference can be eliminated. it can.
  In recent years, research on multi-carrier CDMA (MC-CDMA) has been actively conducted, and application to next-generation broadband mobile communication is being studied. In MC-CDMA, transmission data is divided into a plurality of subcarriers by performing serial / parallel conversion of transmission data and frequency domain orthogonal code spreading.
  In addition, orthogonal frequency / code division multiple access (OFDM / CDMA) systems combining OFDM and CDMA are also being studied. This is a system that improves frequency utilization efficiency by performing IFFT processing on signals divided into subcarriers by MC-CDMA and orthogonal frequency multiplexing.
  ・ Principle of multi-carrier CDMA system
  As shown in FIG. 22, the principle of the multi-carrier CDMA system is to create N copy data from transmission data D of one symbol with a period Ts, and configure each code constituting a spreading code (orthogonal code) that is a channelization code. C₁~ C_NMultiplier 1 for each copy data individually₁~ 1_NMultiply by and each multiplication result D ・ C₁~ D ・ C_NIs the frequency f shown in FIG.₁~ F_NMulticarrier transmission with N subcarriers. The above is a case where 1-symbol transmission data is transmitted by multicarrier. Actually, as will be described later, the transmission data is converted into M-symbol parallel data, and the processing shown in FIG. 22 is performed on each of the M symbols. , M × N all multiplication results are expressed as frequency f₁~ F_NMMulticarrier transmission using M × N subcarriers. The total number MN of subcarriers is (spreading factor N) × (number of parallel sequences M). Also, an orthogonal frequency / code division multiple access scheme (OFDM / CDMA) can be realized by using subcarriers having the frequency arrangement shown in FIG. 23 (b).
  MC-CDMA transmitter side (base station) configuration
  FIG. 24 is a block diagram of the MC-CDMA transmission side (base station). The data modulating unit 10 for the first user (first channel) modulates the transmission data of the first user and converts it into a complex baseband signal (symbol) composed of an in-phase component and a quadrature component. The user data has 32 × M symbols per frame as shown in FIG. As shown in FIG. 25, the serial / parallel conversion unit 11 converts the input data into parallel data of M symbols, and the branching unit 12 N-branches each of the M symbols and inputs them to the first spreading unit 13. The first spreading unit 13 includes M multiplication units 14₁~14_MAnd each multiplication unit 13₁~13_MAre orthogonal codes C of the first user, respectively.₁, C₂,. . C_NAre individually multiplied by the branch symbol and output. The orthogonal code is a Walsh code that is different for each user and pilot. As a result, N × M subcarriers f₁~ F_MNSubcarrier signal S for multicarrier transmission in₁~ S_MNIs input from the first spreading unit 13 to the code multiplexing unit 14. That is, the first spreading unit 13 receives the orthogonal code C of the first user.₁, C₂,. . . C_NIs multiplied in the frequency direction by multiplying the symbol for each parallel series and input to the code multiplexing unit. Note that MN subcarrier signals S₁~ S_MNConstitutes an OFDM symbol.
  In the code multiplexing unit 14, the subcarrier signal S corresponding to the 2nd to nth users created in the same way₁~ S_MNSubcarrier signal P according to pilot₁~ P_MNEnter. Since the frame structure of the signal transmitted from the base station is as shown in FIG. 26, the first to nth user subcarrier signals and the pilot subcarrier signals are added and output for each subcarrier. That is, the frame code multiplexing unit 14 adds an adder AD for each subcarrier.₁~ AD_MNAnd an adder AD according to the j-th subcarrier_j27, the jth subcarrier signal S of the 1st to nth users as shown in FIG._jAnd pilot j-th subcarrier signal P_jAre added and output.
  In the frame configuration diagram of FIG. 26, MN subcarriers f in the frequency direction (lateral direction)₁~ F_MNAre arranged, Walsh codes of 1st to nth users (dedicated channel DPCH) and pilot (common pilot channel CPICH) are arranged in the code direction (vertical direction), and 1 to Ns (for example, 32 in the time direction (depth direction)) ) OFDM symbol times are arranged, and the data of the synchronization channel (SCH) is divided into a plurality of discrete (N_SCHMultiplexed) subcarriers.
  Multiplying unit MP of second spreading unit 15₁~ MP_MNIs a code multiplexed signal S ′ corresponding to each subcarrier input from the code multiplexer 14.₁~ S '_MNCell specific scrambling code (CSSC) G₁~ G_MNMultiply and output. The synchronization channel (SCH) data multiplexing unit 16 performs synchronization channel data D₁~ D_NSCHDiscrete multiple (N_SCHMultiple) subcarriers and input to IFFT 17. FIG. 28 is a block diagram of the SCH data generation unit. The SCH data pattern generator 1 sequentially generates SCH data patterns at symbol periods, the multiplier 2 multiplies the SCH data pattern by a common scramble code, and the branching unit 3 branches one input symbol to N_SCHPieces of SCH data D₁~ D_SCHIs input to the SCH data multiplexing unit 16.
  An IFFT (Inverse Fast Fourier Transform) unit 17 performs IFFT (Inverse Fourier Transform) processing on the subcarrier signals input in parallel to convert them into OFDM signals (real part signal, imaginary part signal) on the time axis. The guard interval insertion unit 18 inserts a guard interval into the OFDM signal, the quadrature modulation unit 19 performs quadrature modulation on the OFDM signal into which the guard interval is inserted, and the radio transmission unit 20 up-converts to a radio frequency and amplifies the frequency. And transmit from the antenna.
  FIG. 29 is an explanatory diagram of guard interval insertion. The guard interval insertion is to copy the tail part to the head part of IFFT output signal corresponding to M × N subcarrier samples (= 1 effective OFDM symbol) as one unit. By inserting the guard interval GI, it becomes possible to eliminate the influence of intersymbol interference due to multipath.
  MC-CDMA receiver configuration
  FIG. 30 is a block diagram of the receiving side of an MC-CDMA mobile station. The radio reception unit 21 performs frequency conversion processing on the received multicarrier signal, and the orthogonal demodulation unit 22 performs orthogonal demodulation processing on the received signal. The timing synchronization / guard interval removing unit 23 obtains timing synchronization of the received signal, and then removes the guard interval GI from the received signal and inputs the guard interval GI to an FFT (Fast Fourier Transform) unit 24. The FFT unit 24 performs FFT calculation processing at the FFT window timing and converts the time domain signal to N × M subcarrier signals (subcarrier samples) SP.₁~ SP_MNThe channel estimation unit 25a performs channel estimation for each subcarrier using a pilot multiplexed on each subcarrier, and the channel compensation unit 25b performs channel estimation value CC for each subcarrier.₁~ CC_MNIs multiplied by the FFT output to compensate for fading.
  The channel estimation unit 25a multiplies the subcarrier component of the pilot symbol output from the FFT unit 24 by the cell identification scrambling code CSSC, adds the multiplication result for each subcarrier, and calculates the channel estimated value CC of each subcarrier by the average value.₁~ CC_MNIs calculated. That is, the channel estimation unit 25a estimates the phase effect exp (jφ) due to fading of each subcarrier using the pilot signal, and the channel compensation unit 25b multiplies the subcarrier signal component of the transmission symbol by exp (−jφ). To compensate for fading.
  The first despreading unit 26 applies the cell identification scramble code G to each of the M × N subcarrier signal components subjected to fading compensation.₁~ G_MNMultiply and output. That is, the fading-compensated signal is despread by the cell identification scramble code, and a signal transmitted by the base station to which the mobile station belongs is extracted from the code-multiplexed signal. The second despreading unit 27 includes M multiplication units 27.₁~ 27_MThe multiplication unit 27₁Is the orthogonal code (Walsh code) assigned to the user (mobile station)
C₁, C₂,. . . C_NN subcarrier signals are individually multiplied and output, and the other multiplier 27₂~ 27_MPerforms the same arithmetic processing. As a result, the signal transmitted from the base station to which the mobile station belongs is despread by the spreading code assigned to the user, and the signal addressed to the user is extracted from the code-multiplexed signal by this despreading. Synthesis unit 28₁~ 28_MIs the multiplication unit 27₁~ 27_MThe N multiplication results output from are added to create parallel data consisting of M symbols, the parallel-serial conversion unit 29 converts the parallel data into serial data, and the data demodulation unit 30 demodulates the transmission data .
  ・ Scramble code array for cell identification
  FIG. 31 (a) shows a scramble code G for cell identification.₁~ G_MNFIG. 5 is an explanatory diagram of an arrangement of (MN = 512), and is shifted by 8 codes in the subcarrier direction for each OFDM symbol. The reason for shifting the code is as follows. If the codes are not shifted, if the m-th subcarrier code of the cell identification scrambling code CSSC of the two stations is the same, the channel estimation value of the m-th subcarrier of the two cells cannot be distinguished, and one channel estimation value Is mistaken for the other channel estimation value. For this reason, as shown in FIG.
The scramble code for cell identification is shifted in the subcarrier direction for each OFDM symbol.
  ・ Necessity of cell search and AFC
  As described above, in the OFDM-CDMA system, cells can be separated using a scramble code. Therefore, the same frequency can be used in all cells, and the frequency utilization efficiency can be improved. However, the scramble code CSSC of the cell (base station) must be identified at the time of reception (cell search).
  In general, since the number of users in each base station is different, the transmission power in each base station is different. Cell search is a technology that detects the base station with the smallest amount of radio wave attenuation (path loss) between the base station and mobile station, in other words, the base station with the largest received power per user. Since the transmission power is different, the base station with the maximum received power is not necessarily the base station with the minimum path loss even if the total received power from each base station is simply detected. Therefore, an effective frame configuration for detecting a base station with the smallest path loss in a mobile station and a cell search method for detecting the base station are required.
  Also, in the OFDM scheme or OFDM-CDMA scheme, a transmission signal is divided into a plurality of subcarriers and transmitted. Since the symbol length of each subcarrier signal can be increased by dividing the subcarrier into a plurality of subcarriers, it is possible to reduce the influence of the delayed wave. However, when the symbol length is increased, there is a disadvantage that it is more easily affected when a carrier frequency offset is present, and an AFC technique and / or a carrier frequency offset compensation technique is required.
  In addition, when a signal is transmitted from a mobile station, high accuracy is required for the carrier frequency of the transmission signal. Normally, this transmission carrier frequency is generated by dividing the frequency-controlled oscillator (VCO) controlled by the AFC at the time of reception at the mobile station, so high accuracy is required for the AFC at the time of reception. .
  ・ Conventional AFC technology
  In communication employing the OFDM scheme, the frequency of the reference clock signal on the receiving side (mobile station) must match the frequency of the reference clock signal on the transmitting side (base station). However, there is usually a frequency deviation Δf between the two. This frequency deviation Δf interferes with adjacent carriers and becomes a factor that impairs orthogonality. For this reason, it is necessary to suppress interference by performing AFC control immediately after power-on of the receiving apparatus to reduce the frequency deviation (frequency offset).
  FIG. 32 is a main part configuration diagram of a conventional receiving apparatus including an AFC (Automatic Frequency Control) unit that matches the oscillation frequency of the local oscillator with the frequency on the transmission side. The high frequency amplifier of the radio reception unit 21 amplifies the received radio signal, and the frequency conversion / orthogonal demodulation unit 22 uses the clock signal input from the local oscillator 31 to perform frequency conversion processing and orthogonal demodulation processing on the received signal. The AD converter 32 AD-converts the quadrature demodulated signal (I, Q complex signal), and the OFDM symbol extraction unit (timing synchronization / guard interval removal unit) 23 extracts one OFDM symbol from which the guard interval GI has been removed, and sends it to the FFT unit 24. input. The FFT unit 24 performs FFT calculation processing at the FFT window timing to convert a time domain signal into a frequency domain signal. The AFC unit 33 detects the phase error Δθ corresponding to the frequency deviation Δf using the reception data which is a complex signal input from the AD converter, and inputs the AFC control signal corresponding to the phase error Δθ to the local oscillator 31. Control the oscillation frequency to match the oscillation frequency on the transmission side. That is, the AFC unit 33 calculates a correlation value between the time profile in the guard interval added to the OFDM symbol and the time profile of the OFDM symbol portion copied to the guard interval, and calculates the phase of the correlation value (complex number) as a transmission device and The frequency deviation Δf between the receiving devices is obtained, and the oscillation frequency is controlled based on the phase to match the oscillation frequency on the transmission side.
  FIG. 33 is a configuration diagram of the AFC unit 33, and FIG. 34 is an operation explanatory diagram of the AFC unit.
  As shown in FIG. 34 (a), the guard interval GI has a number N of samples at the beginning of an OFDM effective symbol having Nc samples._GSince the last part is copied, the guard interval GI is calculated as shown in FIG. 34 (b) by calculating the correlation between the received signal before 1 OFDM effective symbol (before Nc samples) and the current received signal. The correlation value is maximum in the part. Since the maximum correlation value is a value having a phase depending on the frequency deviation, the phase, that is, the frequency deviation can be detected by detecting the maximum correlation value.
  In FIG. 33, the delay unit 33a delays the received signal by one OFDM effective symbol (number of samples Nc = 512), and the multiplier 33b receives the received signal P before one OFDM effective symbol.₂Complex conjugate P₂ ^*And current received signal P₁And the multiplication result is output. The shift register 33c has N guard intervals._GThe latest N with sample (= 100 samples) length_GStores (= 100) multiplication results, and the adder 33d is N_GN multiplied by the multiplication results_GOutput correlation value of sample width. The correlation value storage unit 33e is shifted by one sample output from the adder 33d (N_G+ Nc) (= 612) correlation values are stored, and the adder 33f accumulates the correlation values over 32 symbols and a plurality of frames in the frame in order to improve the S / N ratio, and stores them in the correlation value storage unit 33e. Remember.
  Since the received signal before one OFDM effective symbol and the current received signal in the guard interval period are ideally the same, as the number of multiplication results of the guard interval period stored in the shift register 33c increases, FIG. 34 (b) As the correlation value increases gradually, N in the guard interval period_GWhen all the multiplication results are stored in the shift register 33c, the correlation value becomes maximum, and thereafter, the number of multiplication results in the guard interval period stored in the shift register 33c decreases, and the correlation value gradually decreases.
  Also, assuming that there is no noise when the frequency offset Δf = 0, as shown in FIG.₁And P₂Are the same vector and the output P of the multiplier 33b₁・ P₂ ^*The lie part of is 0. However, assuming that there is no noise when the frequency deviation Δf = a, as shown in FIG.₁And P₂Are not the same vector and P₁And P₂In the meantime, a phase rotation θ corresponding to the frequency deviation Δf occurs. As a result, the output P of the multiplier 33b₁・ P₂ ^*Is rotated by θ compared to the case of Δf = 0, P₁・ P₂ ^*The lie part of has a non-zero value.
  From the above, the correlation value output from the adder 33d is N in the guard interval period._GWhen all the multiplication results are stored in the shift register 33c, the maximum value is obtained, and the maximum value is a complex number having a phase difference θ corresponding to the frequency offset Δf.
  The peak detection unit 33g is stored in the correlation value storage unit 33e (N_G+ Nc) Among the correlation values, the peak correlation value Cmax with the maximum correlation power is detected, and the phase detection unit 33h uses the real part Re [Cmax] and the imaginary part Im [Cmax] of the correlation value (complex number). The following formula
  θ = tan^-1{Im [Cmax] / Re [Cmax]} (1)
To calculate the phase θ. Since the phase θ is caused by the frequency deviation Δf, the control signal of the local oscillator 31 (FIG. 32) is fed back based on the phase θ. The variable damping coefficient α (0 <α <1) is controlled so as not to follow the instantaneous response by multiplying the phase θ by the multiplier 33i, and is integrated and smoothed by the integrating unit 33j to the local oscillator 31. Feedback is performed, and the frequency of the clock signal output from the local oscillator 33 is controlled. A feature of this method is that carrier frequency synchronization can be performed before FFT.
  By the way, in addition to the code multiplexing coding method described in FIG. 26, the pilot multiplexing method includes a time division multiplexing method in which pilots are time division multiplexed in transmission data. When this time division multiplexing method is employed, AFC control can be performed so that the carrier frequency error is detected from the phase rotation amount of the received pilot signal after the FFT and the oscillation frequency of the local oscillator is controlled. FIG. 36 is a time-division multiplexing frame configuration diagram, in which pilot P is time-multiplexed ahead of one frame of transmission data. The pilot P can also be distributed within the frame. For example, if the pilot per frame is 4 × M symbols and the transmission data is 28 × M symbols, the serial / parallel converter 12 (FIG. 24) outputs pilot M symbols up to the first four times as parallel data, and thereafter Then, M symbols of 28 times transmission data are output as parallel data. As a result, the pilot can be time-multiplexed on all subcarriers and transmitted four times in one frame period, and AFC control and channel estimation / channel compensation for each subcarrier can be performed on the receiving side using the pilot.
  FIG. 37 shows a configuration example of AFC control in the time division multiplexing system, in which two pilot signals of the same symbol are inserted over all subcarriers. On the receiving side, after the FFT calculation in the FFT calculation unit 24, for each subcarrier, one of the two pilot signals is multiplied by the other complex conjugate. That is, for each subcarrier, the delay unit DLY delays the first pilot signal by one OFDM symbol, the CNJ calculates the complex conjugate of the delayed pilot signal, and the multiplication unit MPL is the first pilot among the continuous pilot signals. Multiply the complex conjugate of the signal by the second pilot signal and output. If a phase difference θ corresponding to the frequency deviation Δf exists between the first and second pilot signals, the multiplication result includes an imaginary part for the same reason as described in FIG. 35B, and the multiplication result is Δf Compared with the case of = 0, it rotates θ. Even if the pilot symbols in the respective subcarriers are different, the same symbol is transmitted in each subcarrier, so that the calculation result is ideally the same signal. Therefore, the average value calculation unit MEN calculates the average value for each real part and imaginary part of the multiplication result of all subcarriers, and the phase detection unit PDT uses the real part Re [Cmax] and the imaginary part Im [Cmax]. The phase θ is calculated by the equation (1). Since the phase θ is caused by the frequency deviation Δf, the oscillation frequency of the local oscillator 31 (FIG. 32) is controlled based on the phase θ.
  ・ Conventional cell search technology
  The cell search for the frame configuration shown in FIG. 26 consists of three steps as shown in FIG. 38. The FFT window timing is detected in the first step (step S1), and the SCH frame timing is detected in the second step ( In step S2), CSSC (scramble code for cell identification) is identified in the third stage (step S3).
  In the first stage, the guard interval correlation (see FIG. 39) is calculated with the same configuration as the AFC unit in FIG. 33 to detect two correlation peak values equal to or higher than the set value TH1, and each of them is detected as an FFT window timing candidate 1, 2.
  In the second stage, optimal frame timing is detected from 32 different frame phases using SCH for each FFT window timing candidate. Specifically, the following processing is performed on each of the two FFT window timing candidates detected in the first stage. That is, first, FFT is performed on the received wave at the first timing of the FFT window timing candidate, and the signal after FFT is stored in the memory. As shown in the SCH frame format in FIG. 40, SCH data is multiplexed with user data and pilot data in a plurality of discrete subcarriers. Therefore, for the subcarrier to which each SCH is assigned after FFT, the time correlation between the post-FFT signal stored in the memory and the sequence signal obtained by sequentially shifting the phase of the known SCH signal sequence is one frame. Calculate minutes (= 32 OFDM symbols). Then, the average value of the correlation values for the eight subcarriers is obtained by power average, and the average power for a plurality of frames is calculated as necessary. The correlation value is calculated for each pattern in which the phase of the known SCH signal sequence is shifted by 0, 1, 2,..., 31 OFDM symbols, and the correlation power values for the 32 phase frame timing candidates are stored in the memory. Thereafter, the largest frame timing is selected from the correlation power values of the 32 phases, and it is set as a frame timing candidate for one FFT window timing candidate. The same frame timing candidate determination process is performed for each FFT window timing candidate.
  In the third stage, CSSC number detection and FFT window timing and frame timing are determined for the candidates detected up to the second stage. That is, by multiplying each received subcarrier by multiplying a number of known CSSC candidates and each complex conjugate of CPICH and integrating in the time direction, a correlation value is obtained for each subcarrier, and a subcarrier block consisting of several subcarriers. The correlation value is averaged, and the metric for the CSSC candidate is obtained by adding power to the correlation average value over all subcarriers, and the CSSC candidate having the maximum metric is set as the CSSC of the base station to be connected.
  In AFC control that extracts the phase error due to the carrier frequency offset by the guard interval correlation described in FIGS. 32 to 35, the carrier frequency offset is used to correlate signals that are separated by Nc sample length (1 OFD symbol). When becomes smaller, the phase error is buried in noise and the phase error cannot be detected accurately. In particular, in AFC control, the time constant is set to be long for stable operation. Therefore, if there is a residual carrier frequency offset, even if the carrier phase rotation is small in one OFDM symbol period, the beginning of one frame consisting of several tens of OFDM symbols. There is a problem that the carrier phase is greatly rotated between the first symbol and the last symbol.
  Further, when the carrier frequency of the transmission signal on the mobile station side is required to be higher than the accuracy of the carrier frequency obtained by guard interval correlation and AFC control, there arises a problem that carrier frequency synchronization is insufficient.
  In addition, as shown in FIG. 37, the AFC method in which a known signal is inserted in a transmission signal in advance and the carrier frequency error is detected from the phase rotation amount of the received known signal after FFT, the channel model is multipath fading. Limited to noise. For this reason, in an environment where co-channel interference is also received from other cells as in the OFDM-CDMA system, there is a problem that the detection accuracy of the phase error due to the carrier frequency offset deteriorates because the known signal is also affected by the co-channel interference. In addition, in the method shown in FIG. 37, since the phase error between 1 OFDM symbols is estimated, it is difficult to estimate the phase error with higher accuracy, and even if the interval between pilot symbols is widened, phase error detection is possible. The accuracy is fixed at the pilot symbol interval, and the flexibility is poor.
  Also, in the conventional cell search, the cell search algorithm is usually started after the local oscillator (Voltage Controlled Oscillator: VCO) is first pulled into the carrier frequency by AFC, but in order to detect various timings in the cell search A voltage addition operation is used for the above. In particular, in a conventional cell search, a correlation operation between a sequence signal obtained by sequentially shifting the phase of a known SCH signal sequence and a received SCH signal is performed over one frame length. For this reason, if carrier frequency synchronization by AFC is insufficient, the received signal rotates due to residual frequency error, and the effect of voltage addition in the correlation calculation is reduced, so that it is expected that sufficient frame timing detection characteristics cannot be obtained. Is done.
  Further, in the code multiplexing frame configuration of FIG. 26, a SCH (Synchronous) is further added to a signal in which DPCH (Dedicated Physical Channel) and CPICH (Common Pilot Channel) are multiplexed.
Channel). However, since SCH is not orthogonal to DPCH and CPICH, when SCH frame timing is detected by correlation calculation, it receives interference from DPCH and CPICH. In order to reduce this influence, it is necessary to add power to the interphase value over several frames, and it takes time to detect the SCH frame timing. Also, when combining SCHs multiplexed on multiple subcarriers, the phase of the channel response at each SCH is different, so voltage addition cannot be performed, and it is necessary to combine by power addition. There is a problem that it cannot be obtained sufficiently. FIG. 41 is an explanatory diagram of a voltage addition region and a power addition region in the SCH frame timing detection process. Area where the phase difference of the correlation value is small, that is, m × n areas specified by n OFDM symbol intervals (for example, n = 4) in the time axis direction and m subcarrier intervals (for example, m = 8) in the subcarrier direction The correlation value in is a voltage addition region where an average value can be obtained by voltage addition. However, since the correlation value in the voltage addition region and the correlation value outside the voltage addition region have a large phase difference, the voltage cannot be added and averaged, and the power is added to obtain the average value. That is, the area outside the voltage addition area becomes the power addition area.
  Further, in the conventional example, in parallel with the CSSC identification processing, the complex conjugate of the CSSC candidate and the CPICH are multiplied with each received subcarrier signal and integrated in the time direction. When CSSC is correct, the obtained integral value is used as a channel estimation value for each subcarrier. However, when the number of users is large, there is a problem that interference from the DPCH increases and the error of the channel estimation value increases. In addition, there is a problem that it is necessary to increase the number of integration symbols in the time direction in order to improve the accuracy of channel estimation. This occurs because the diffusion area SPA and the channel response estimation area CEA do not match as shown in FIG. Further, when the received subcarrier signal is rotated due to the residual frequency offset, the channel estimation accuracy is deteriorated, so that the CSSC identification probability is expected to deteriorate.

  Accordingly, an object of the present invention is to enable accurate channel estimation and CSSC identification without receiving interference from DPCH.

The present invention is a transmission method for transmitting a transmission signal sequence by spreading the transmission signal sequence to a plurality of subcarriers in the frequency direction. The transmission signal sequences of the common pilot channel and the dedicated channel constituting the transmission signal sequence are respectively channelized codes. And spreading to a two-dimensional region of time and frequency, masking the spread signal with a scramble code for base station identification, and transmitting a masked transmission signal sequence.
The present invention spreads and multiplexes transmission signal sequences of common pilot channels and dedicated channels in a two-dimensional area of time and frequency using channelization codes on the transmission side, and masks the multiplexed signals with a scramble code of a base station. And a cell search method for a receiving apparatus for receiving the masked signal, wherein each subcarrier signal extracted from the received signal is multiplied by a complex conjugate of a common pilot and a scramble code candidate, A step of obtaining a channel response by adding the multiplication results in the two-dimensional diffusion region, a voltage addition of the channel responses of a plurality of the two-dimensional diffusion regions belonging to the enlarged two-dimensional region, and averaging the power of the averaged channel response Calculating power, adding power to the channel response of each enlarged two-dimensional region, Force addition value has a determining that the scramble code of the base station a scrambling code candidate having the maximum.
The present invention is a transmission apparatus for transmitting a transmission signal sequence by spreading it over a plurality of subcarriers in the frequency direction, and using a channelization code for each of the transmission signal sequences of the common pilot channel and the dedicated channel constituting the transmission signal sequence. A spreading unit that spreads in a two-dimensional region of time and frequency, a masking unit that masks the spread signal with a scramble code for each base station, and a transmission unit that transmits the masked transmission signal sequence.
In the present invention, transmission signal sequences of a common pilot channel and a dedicated channel are spread and multiplexed in a two-dimensional region of time and frequency using channelization codes on the transmission side, and the multiplexed signals are masked with a scramble code of a base station. A reception device that receives the masked signal, multiplies each subcarrier signal extracted from the received signal by the complex conjugate of each of the common pilot and the scramble code candidate, and adds the multiplication result in the two-dimensional spreading region A channel response calculating unit for calculating a channel response, a two-dimensional voltage adding unit for averaging the channel responses of the plurality of two-dimensional diffusion regions belonging to the enlarged two-dimensional region by voltage addition, an average of the two-dimensional diffusion regions 2D power to calculate the power of the normalized channel response and add power to the channel response of each expanded 2D region Adding unit, power sum is equipped with a cell determining unit determines that the scrambling code of the base station a scrambling code candidate having the maximum.

According to the present invention, the signal sequence of DPCH and CPICH is spread in a two-dimensional region of frequency and time, and the spread signal in this two-dimensional region is multiplied by each complex conjugate of pilot and cell identification scramble code CSSC and added. Thus, the correlation calculation is performed, and the channel response of the two-dimensional region is estimated based on the correlation value, so that the diffusion region and the channel estimation region can be made the same. For this reason, there is no interference from DPCH, channel estimation accuracy can be improved, and accurate channel compensation becomes possible.
In addition, according to the present invention, the correlation voltage of the plurality of two-dimensional regions is added and averaged, and the power of the plurality of average correlation voltages is added to the CSSC of the cell to be connected to the other cells. The difference with the power value for CSSC can be increased, and CSSC identification can be performed accurately. In addition, since the cell detection accuracy can be improved, it greatly contributes to the performance improvement of the terminal device. Further, the carrier frequency synchronization of the present invention contributes to the performance improvement of the terminal device because the operation is continued during the data demodulation period after the cell search is completed.

で表される。キャリア周波数オフセットが存在する場合にはm番目のサブキャリア信号は
、

となる。ここで、h_mはm番目のサブキャリアにおけるチャネル応答、Δf_cはキャリア周波
数オフセット、T_SBはシンボルブロック時間である。この受信信号系列に対して差動復号
を行うと、

となる。
受信信号系列の位相φが未知であるため、差動符号化前の信号系列b(k)の位相を順次シフトさせた系列b(k+Ψ)との相関演算を行うと相関器出力は、

で表される。相関器で与えるb(k)の位相Ψが受信信号系列の位相φと一致していないとb(k+φ)・b(k+Ψ)は1あるいは−1となりN_Sシンボルに渡って加算してもN_S 倍とはならない
。一方、b(k)の位相Ψが受信信号系列の位相と一致したときb(k+φ)・b(k+Ψ)は常に1と
なり、N_S シンボルすべてに渡って加算されるため｜R_m(Ψ)｜²は最大となる。従って、R_m(Ψ)の電力が最大となる位相をフレームタイミングとして検出することができる。
また、キャリア周波数オフセットΔfcによりサブキャリア信号ｒ_m(k)は回転しているが差動復号を行うことにより差動復号毎に同じ位相シフト量に変換される。従って、相関器で加算しても電力の損失がないためフレームタイミングを正確に検出可能となる。
更に、２セル(2基地局)からの電波を強く受けている場合には、

となり、同様に差動復号を行うと、

となる。受信信号系列の位相φあるいは

が未知であるため、差動符号化前の信号系列b(k)の位相を順次シフトさせた系列b(k+Ψ)
との相関演算を行うと相関器出力は、

で表される。b(k)の位相Ψが受信信号系列の位相φと一致したとき

となり、b(k)の位相Ψが受信信号系列の位相

と一致したとき

となる。ここで、ξ ₁とξ₂は相関で抑圧しきれない信号成分である。従って、相関器出
力の電力が最大となる位相をSCHフレームタイミングするとチャネル応答の大きい方、す
なわちパスロスが最小の基地局からの送信信号のフレームタイミングを検出することが可能となる。
更に複数のサブキャリアにSCHが割り当てられている場合には、差動復号器出力を合成
すると相関器出力は、

となるため、短区間フェージングの影響でパスロスが最小でないセルからのフレームタイミングを検出してしまう確率を減少させることが可能となる。
図4は、本発明の受信装置におけるSCHの合成及びフレームタイミング検出構成図である。差動復号部72₁〜72_NSCHは、FFT演算部71より出力する多数のサブキャリアのうちSCHが
割り当てられているサブキャリアF₁〜F_NSCHの信号を差動復号し、加算部73は各差動復号
信号を加算する。ついで、相関器74は、加算器73から出力するSCH合成信号系列と既知のSCH信号系列との相関演算を、後者の位相を位相シフト部75で順次シフトさせながら行い、最大値検出部76は相関器出力が最大となる位相を、フレームタイミングとして検出する。尚、相関演算部74は(8),(10)式の相関演算を行うもこである。又、各差動復号部72₁〜72_NSCHは1シンボル遅延部SDL、複素共役発生部CNJ、乗算部MPLで構成されている。
（C）CMAによるフレームタイミングの検出
本発明ではSCHを複数の専用サブキャリアに割り当てており、また、SCHは全セル共通である。このため、基地局間でのタイミングが非同期であると、タイミングとチャネル応答が異なった複数のSCH信号が重畳して受信される。これは、１つの信号系列を送信して直
接波と遅延波が複数のアンテナで受信されるモデルと同じである。他のパスを抑圧しながら最大電力を持つパスを抽出するアルゴリズムとしてCMA(Constant Modulus Algorithm)
が知られている。そこで本発明では複数のSCH用サブキャリアでの受信信号に複素数の重
みづけを行って合成し、合成信号電力と希望電力の誤差が最小となるように重み係数を更新する。
図５は本発明におけるConstant Modulus Algorithm (CMA)によるSCH合成およびフレー
ムタイミングの検出構成図である。ＣＭＡアルゴリズムでは、最も信号電力が大きいものを選ぶアダプティブアレイのアルゴリズムであり、図5では複数の基地局から送られてく
るSCHのうち最も信号電力が大きいSCHを受信し、他のSCH信号を打ち消すように重み付け
を適応制御により行うものである。
CMA合成部77の乗算部77ａ₁〜77a_NSCHは、FFT演算部71より出力する多数のサブキャリア信号のうちSCHが割り当てられているサブキャリアF₁〜F_NSCHの信号に重み係数w₁(k)〜w_NSCH(k)を乗算し、合成部77bは各乗算器出力信号を合成し、CMA処理部77cは、CMAアルゴリ
ズムに従って合成出力電力と希望出力電力との誤差が小さくなるように重み係数w₁(k)〜w_NSCH(k)を更新する。
すなわち、SCHが割り当てられているm番目のサブキャリアの信号を時刻kにおいてx_m(k)とすると合成部77bの出力信号は

で表され、誤差信号は、

で与えられる。ここで、pとqはアルゴリズムの収束特性を与えるパラメータである。係数ベクトルW=[w₁,w₂, … , w_NSCH]^Tは、

で更新される。ここでμはステップサイズ、＊は複素共役であり、X(k)はSCHが割り当て
られているサブキャリアの受信信号ベクトルである。重み係数が収束すると合成出力にはパスロスが最小のSCH信号系列のみが出力され、他の基地局からの信号は抑圧される。従
って、この合成出力を差動復号すると不要な信号成分が生成されないため後段の相関演算によりSCHフレームタイミング検出をより正確に行うことが可能となる。
重み係数収束後、差動復号部78はCMA合成出力信号を差動復号し、相関器74は、差動復
号部78から出力するＣＭＡ合成信号系列と既知のSCH信号系列との相関演算を、後者の位
相を位相シフトと部75で順次シフトさせながら行い、最大値検出部76は相関器出力が最大となる位相をフレームタイミングとして検出する。なお、差動復号部７８は1シンボル遅
延部SDL、複素共役発生部CNJ、乗算部MPLで構成されている。
（D）高精度位相誤差推定制御
図4、図5では、差動復号を行った後に相関演算を行っているため、検出されたフレームタイミングにおける相関値の位相シフト量はキャリア周波数オフセットの1シンボルブロ
ック間での回転量となる。位相シフト量、例えば、(9a)式における

の位相シフト量Δθ＝2πΔf_cT_SBは次式

により検出できる。ところで、キャリア周波数オフセットが小さいと、雑音や抑圧しきれていない信号成分によりΔθあるいはΔθの極性の検出精度が劣化する。雑音などによりΔθの極性判定の精度が最も劣化しにくいのは、極性判定の境界値である0から最も離れ
たΔθが±π/2のときである。そこで、後段の高精度AFCにおける差動復号のシンボル間
隔数Ｍを

として決定する。このようにすれば、ΔθがM倍されて±π/2になるため、Δθが小さく
てもその極性をより正しく検出でき、キャリア周波数制御を高精度に行うことが可能になる。
以上から、図4、図5の差動復号部において、複数シンボル間隔Mの差動復号を行ってか
ら相関演算を行う。例えば上記のMがM=2であれば、差動復号結果ｄ_m(K)は

となる。フレームタイミングφは既に検出されているので、そのタイミングで相関演算を行うと、

となり、位相シフト量がM倍に拡大される。従って、位相シフト量検出の分解能が雑音な
どにより十分でない場合には、本発明によりに位相シフトの極性判定を大幅に改善することが可能となる。
図6は本発明の高精度位相誤差推定部の構成図であり、前段で差動復号のシンボル間隔
Ｍを決定し、後段で高精度AFC方式に用いられる高精度位相誤差Δθの推定を行う。すな
わち、前段において、相関器74、位相シフト部75、最大値検出部76は、図4、図5で説明したように、SCH信号系列の位相をシフトしながら相関器出力が最大となる位相φをフレー
ムタイミングとして検出する。位相検出部79は前記フレームタイミングにおける相関値の位相シフト量Δθ＝2πΔf_cT_SBを(14)式により演算し、シンボル間隔数計算部80は(15)式によりシンボル間隔数Ｍを決定し、複数シンボル間隔差動復号部81に入力する。
複数シンボル間隔差動復号部81は、CMA合成部77(図5参照)から出力するＣＭＡ合成信号をMシンボル遅延するシンボル遅延部SDL、遅延部SDLの出力信号の複素共役を演算する複
素共役部CNJ、受信SCH信号とMシンボル前のSCH信号の複素共役との乗算を行う乗算部を備え、(16)式の演算を行う。位相シフト部82はSCH信号系列b(k)の位相を既に求まっている
フレームタイミングφ分シフトしてb(k+φ)を出力する。Mシンボル間位相回転量計算部83は、位相シフト部82から入力するMシンボル分のSCH信号系列b(k+φ)，b(k+φ−1)，．．
．b(k+φ−(M−1))を乗算し、乗算結果を出力する。Ｍ＝２であれば、Mシンボル間位相回転量計算部83は、b(k+φ)・b(k+φ−1)を出力する。相関器84は(17)式の演算を行って相
関値Rm(φ)を出力し、位相誤差検出部85は次式

によりM倍された位相誤差（位相シフト量）M・Δθを算出し、図示しないAFC制御部に入
力する。AFC制御部は後述するように位相誤差の極性に基いて局部発振器の発振周波数を
制御し、オフセット周波数が零となるように、すなわち、送信側及び受信側の発振周波数が等しくなるように制御する。
ところで、上記制御おいて、差動復号部のシンボル間隔数Mをセルサーチ期間中固定し
ておく必要はない。AFCが引き込むにつれてキャリア周波数オフセットは小さくなってい
き、位相シフト量Δθも小さくなる。位相シフト量はフレーム毎に得られるので、

の位相M・Δθを(18)式により求め、位相シフト量Δθを次式

により算出し、次式

により次フレームの位相シフト量推定に最適なパラメータＭを決定する。図７はかかる高精度位相誤差推定部の構成図であり、図6と同一部分には同一符号を付している。シンボ
ル間隔計算部86は(19)、(20)式の演算を行って次フレームの差動復号演算におけるシンボル間隔数Mを計算し、複数シンボル間隔差動復号部81に設定する。これにより、複数シン
ボル間隔差動復号部81は設定されたシンボル間隔で差動復号を行う。
（E）残留周波数誤差補正
本発明ではキャリア周波数オフセットによる位相シフト量M・Δθを推定することがで
きるので、推定された位相シフト量を1/M倍して1シンボルブロックあたりの位相回転量Δθを計算し、各受信サブキャリア信号をこの位相回転量で逆回転させて残留周波数誤差を補正することが可能となる。
AFCにおいて、キャリア周波数オフセットを補正しているがAFC制御の時定数は非常に長く、局部発振器（VCO）のキャリア周波数更新量は微量である。そのため、AFCとサブキャリア信号の残留周波数誤差補正とはお互いに妨害しあうことはない。そこで、位相シフト量を用いてAFCと残留キャリア周波数誤差補正の両方を行うことができる。
図8は位相シフト量を用いて残差キャリア周波数オフセットを補正する本発明の構成図
である。図6あるいは図7に示した高精度位相誤差推定部87は(19)式により位相誤差Δθを計算し、位相回転制御部88はexp(−Δθ)を計算し、位相回転部89の各乗算器はFFT演算部71から出力する各サブキャリア信号にexp(−Δθ)を乗算して−Δθだけ逆回転させて残
留周波数誤差を補正する。
（G）二次元拡散
従来は送信データを周波数方向に、すなわち1次元領域に拡散している。しかし、周波
数あるいは時間の1次元領域での拡散ではチャネルの周波数選択性あるいは時変性により
直交性が崩れやすい。そこで、本発明では図9に示すように、個別物理チャネルDPCHと共
通パイロットチャネルCPICHのそれぞれの拡散領域ができるだけチャネル変動の影響を受
けないように、時間と周波数の二次元領域（周波数方向にm、時間方向にｎのm×ｎの領域）において拡散コードを用いて拡散して多重化し、さらにCSSC(Cell Specific Scramble Code)でマスクしている。二次元領域に拡散すると、同じ拡散利得において1次元拡散に比べて周波数領域におけるサブキャリア数を減少することができるので直交性をより保つことができる。
図10、図11は本発明における二次元拡散部を備えた基地局装置の送信側の構成図である。図10では、CPICHとDPCHとを拡散コードを用いて時間と周波数の二次元領域において拡
散して多重化を行い、更に基地局毎のCSSCで乗積（マスキング）する構成になっている。又、図11では、各サブキャリア信号である二次元拡散信号と同期チャネルSCHの信号にIFFT（逆フーリエ変換）処理を施して時間軸上のOFDMシンボル信号に変換し、ガードインタ
ーバルを付加して出力する構成になっている。
図10において、第1ユーザの二次元拡散部50₁におけるシリアル・パラレル変換部55は、該第1ユーザの送信信号をM個の並列データに変換する。拡散部56のM個の乗算部56₁〜56_M
は各並列データに第1ユーザのチャネライゼーションコード（ウォルシュコード）C₁〜C_N
を乗算して拡散し、シリアル・パラレル変換部57₁〜57_Mは、対応する乗算器56₁〜56_Mから入力するN個のシリアルデータをm個づつのn組の並列データ（N=m×ｎ）に変換して順次出力する。これにより、第1ユーザデータは図12に示すようＭ個の二次元領域（周波数方向
にm、時間方向にｎのm×ｎの領域）A₁〜A_Mに拡散される。同様に他ユーザ用の二次元拡散部50₂〜50_K、パイロット用の二次元拡散部50_Pも各ユーザデータ及びパイロットを二次元
領域に拡散する。
加算器58₁₁は、各二次元拡散部50₁〜50_K、50_Pのシリアルパラレル変換部57₁から出力する第1〜第mデータのうち第1データを加算し、加算器58₁₂は同様に、二次元拡散部50₁〜50_K、50_Pのシリアルパラレル変換部57₁から出力する第1〜第mデータのうち第２データを加
算し、同様に加算器58_1mは、二次元拡散部50₁〜50_K、50_Pのシリアルパラレル変換部57₁から出力する第1〜第mデータのうち第ｍデータを加算する。
マスキング部59の各乗算器59₁₁〜59_Mmは各加算器58₁₁〜58_Mmの出力信号にセル識別用スクランブルコードCSSC（G1〜G_Mm）の各コードを乗算して出力する。乗算器59₁₁〜59_1mは
、まず最初の組のｍ個のサブキャリア信号を出力し、次に、第2組のｍ個のサブキャリア
信号を出力し、以後同様にｎ組のサブキャリア信号を出力し、トータルm×n個の二次元拡散信号TDS₁を出力する。同様に図示しない乗算器59₂₁〜59_2mは、まず最初の組のｍ個のサブキャリア信号を出力し、次に、第2組のｍ個のサブキャリア信号を出力し、以後同様に
ｎ組のサブキャリア信号を出力し、トータルm×n個の二次元拡散信号TDS₂を出力する。以下同様に図示しない乗算器59_M1〜59_Mmは、まず最初の組のｍ個のサブキャリア信号を出力し、次に、第2組のｍ個のサブキャリア信号を出力し、以後同様にｎ組のサブキャリア信
号を出力し、トータルm×n個の二次元拡散信号TDS_Mを出力する。これら各二次元拡散信号TDS₁〜TDS_MはＩＦＦＴ演算部53(図11)に入力する。
図11において、差動符号化SCH信号系列発生器51は差動符号化した1フレームNsシンボルのSCH信号系列を発生し、コピー部52は1シンボルのSCH信号よりN_SCH個のコピーシンボルD₁〜D_NSCHを発生し、各コピーシンボルD₁〜D_NSCHをSCH専用のサブキャリアＦ₁〜Ｆ_NSCHの
信号としてIFFT部５３に入力する。また、マスキング部59(図10)より発生した各二次元拡散信号TDS₁〜TDS_Mは上記専用サブキャリアF1〜F_NSCH以外のサブキャリアf₁₁〜f_Mmの信号
としてIFFT部５３に入力する。
IFFT部53は、並列入力するサブキャリア信号にIFFT（逆フーリエ変換）処理を施して時間軸上のOFDMシンボル信号に変換する。ガードインターバル挿入部54は、OFDMシンボル信号にガードインターバルを挿入し、図示しない送信部はガードインターバルが挿入されたOFDMシンボル信号に直交変調を施し、無線周波数にアップコンバージョンすると共に高周波増幅してアンテナより送信する。
（I）チャネル推定及びCSSCの同定
図9に示すように二次元拡散されている場合には、受信側はFFT演算により得られたサブキャリア信号にパイロット（CPICH）とCSSC候補の各複素共役を乗算して積和演算を行い
、すなわち相関演算を行って二次元拡散領域におけるチャネル応答を推定することができる。個の場合、拡散領域とチャネル推定領域が一致しているのでDPCHからの干渉をなくすことができ、チャネル推定精度の向上が可能となる。なお、各二次元拡散領域において1
つのチャネル推定値h(k, n) （kはシンボルインデックス、nサブキャリアインデックス）が得られる。
図13はCSSC候補に対するチャネル応答推定部の構成図であり、71はFFT演算部、89は図8で説明した位相回転部（残留キャリア周波数オフセット補正部）、90₁〜90_Mは二次元相関演算部である。周波数と時間の二次元領域(m×nの二次元領域)にそれぞれ拡散されているCPICHとDPCHの多重データに、パイロット（CPICH）とCSSC候補の各複素共役を乗算して積和演算を行うと、二次元拡散領域におけるチャネル応答推定値が求まる。そこで、二次元相関演算部90₁〜90_Mは上記積和演算を行って二次元拡散領域におけるCSSC候補のチャネル応答推定値h(i,1)〜h(i,M)を演算する。
次に、接続すべきセル(基地局)のCSSCに対して各拡散領域において得られた相関値と他のセルのCSSCに対して得られた相関値との差を大きくするために、相関値を電圧加算して平均化する。本発明では、図9に示す拡散領域を更に拡大した二次元領域BAにおいて平均
化を行う。図14は相関値を電圧加算して平均する構成図であり、図13と同一部分には同一符号を付している。演算部91a〜91ｒは拡大した二次元領域BA内の相関値の電圧平均を演
算して出力する。かかる構成によれば、接続すべきセルのCSSCに対して各拡散領域において得られた平均相関値は、他のCSSCの平均相関値に対して大きくなるのでCSSC同定性能の向上が可能となる。
また、本発明では、二次元領域ＢＡより更に拡大した二次元領域MBAにおいて平均相関
値の電力加算を行うようにする。図１５は平均相関値の電力加算を行う構成図であり、図14と同一部分には同一符号を付している。二次元電力加算部92は、各演算部91a〜91ｒで
演算された二次元領域BA内の二次元電圧平均の電力を計算して加算する。このようにすれば、接続すべきセルのCSSCに対するメトリックと他セルのCSSCに対するメトリックとの差はより大きくなりCSSC同定性能の向上が可能となる。すなわち、各CSSC候補毎に上記の二次元電力加算値を演算し、該電力加算値が最大となるCSSC候補を接続すべきセルのCSSCであると判定する。
なお、接続すべきセルのCSSCとパイロット（CPICH）の各複素共役を二次元相関演算部90₁〜90_Mに入力して得られる相関値は二次元拡散領域におけるチャネル応答推定値となる
から、これらチャネル応答推定値を用いてチャネル補償することができる。
(J)セルサーチ手順
図16は本発明のセルサーチの手順を示す。まず、第１段階では従来のガードインターバル相関によるキャリア周波数同期を行い、粗くキャリア周波数同期を行う（S101）。ただし、システム的にキャリア周波数オフセットがそれほど大きくならない場合には,この第
１段階は必ずしも必要ではない。…S101
第2段階ではガードインターバル相関により相関のピークを検出し、FFTウィンドウタイミング同期を取る。このとき、基地局間が非同期の場合には、得られた相関ピークの前後をマスクして相関電力値が最大となるタイミングを検出し,複数のタイミング候補を保持
する。基地局間でタイミング同期が取れているシステムの場合には、FFTウィンドウタイ
ミングは１つ検出すればよい。…S102
第３段階では、第2段階で得られたFFTウィンドウタイミング候補それぞれにおいて、受信信号をFFTによりサブキャリア信号に変換し、SCHが割り当てられられているサブキャリア信号を差動復号し、相関演算を行う。そして、全FFTタイミング候補を対象にして相関
電力が最大となるSCHフレームタイミングを検出する。…S103
第４段階では得られたSCHフレームタイミングに基づいて、複数シンボル間隔差動復号
を行って高精度に位相誤差を検出し、キャリア周波数同期を取る(S104)。第４段階と平行
して、第５段階では拡散信号部の残留キャリア周波数オフセットを補正しながらCSSCの同定を行う(S105)。
（K）FFTウィンドウタイミングとSCHフレームタイミング検出構成
・第1実施例
図17はFFTウィンドウタイミング検出とSCHフレームタイミング検出に関する実施例である。局部発振器（VCO）61と混合器62で受信信号周波数を中間周波数に変換する。ガード
インターバル相関演算部63は、受信信号に対してガードインターバル相関を行い、同相加算器64は更に同相加算を行って平均化し、ピーク検出部65は2以上のピーク検出を行ってFFTウィンドウタイミングを複数個検出する。以上のFFTウィンドウタイミング候補の検出
制御は従来例と同じである。
ついで、図4で説明した制御によりフレームタイミング、周波数オフセット(位相誤差)
を決定する。すなわち、各候補のFFTウィンドウタイミングにおいて受信信号をFFT演算部71によりサブキャリア信号に変換し、差動復号部72₁〜72_NSCHはSCHが割り当てられているサブキャリア信号に対して差動復号を行い、加算部73は差動復号結果を合成してSCH合成
信号を出力する。相関器74₁，74₂は各FFTウインドウタイミング候補について、SCH合成信号と差動符号化前のSCH信号系列との相関演算を行い、最大値検出部76は、1)相関電力値
が最大となるSCHフレームタイミングφの検出を行うとともに、2)そのSCHフレームタイミングが検出されたときのFFTウィンドウタイミングを検出し、3)更にSCHフレームタイミングが検出された相関値から位相誤差（キャリア周波数オフセット）Δθを検出する。また、最大値検出部76は、高精度キャリア周波数同期制御に使用する複数シンボル間隔差動復号のシンボル間隔数Ｍも決定する。
・第２実施例
図18はFFTウィンドウタイミング検出とSCHフレームタイミング検出に関する第2の実施
例である。図17と同様の制御により、FFTウインドウ候補を決定する。ついで、図５で説
明した制御によりフレームタイミング、周波数オフセット(位相誤差)Δθを決定する。すなわち、各候補のFFTウィンドウタイミングにおいて、FFT演算部7１は受信信号をサブキ
ャリア信号に変換する。CMA合成部77はSCHが割り当てられているサブキャリア信号に対して重み付けを行って合成し、その重み付け係数をCMAアルゴリズムを用いて更新する。重
み付け係数が収束するとCMA合成部77は受信電力が最大となる基地局からのSCHの合成信号のみを出力する。差動復号部78₁，78₂は各FFTタイミング候補のCMA合成信号に対して差動復号を行い、相関器74₁，74₂は差動復号結果と差動符号化前のSCH信号系列との相関演算
を行う。最大値検出部76は、1)相関電力値が最大となるSCHフレームタイミングφの検出
を行うとともに、2)そのSCHフレームタイミングが検出されたときのFFTウィンドウタイミングを検出し、3)更にSCHフレームタイミングが検出された相関値から位相誤差（キャリ
ア周波数オフセット）Δθを検出する。また、最大値検出部76は、高精度キャリア周波数同期制御に使用する複数シンボル間隔差動復号のシンボル間隔数Ｍも決定する。
（L）AFC制御
・第1実施例
図19はAFC制御の第1の実施例構成図である。第１段階では、AFC部６６は従来例と同様
に(図33〜図35参照)、ガードインターバル相関によるピーク検出を行い、ピークタイミングでの相関値の位相誤差を検出し、その位相誤差を小さくするようにスイッチ67を介してVCO構成の局部発振器61の発振周波数を制御する。
ついで、図17の制御でSCHフレームタイミング、FFTウインドウタイミング、差動復号のシンボル間隔Mを決定する。しかる後、FFT演算部71は該FFTウインドウタイミングでFFT演算し、複数シンボル間隔差動復号部72₁〜72_NSCHはSCHが割り当てられているサブキャリア信号に対して複数シンボル間隔差動復号演算、すなわち(16)式の演算を行う。加算部73は各差動復号結果を合成してSCH合成信号を出力する。
一方、位相シフト部82はSCH信号系列b(k)の位相を既に求まっているフレームタイミン
グφ分シフトしてb(k+φ)を出力する。Mシンボル間位相回転量計算部83は、位相シフト部82から入力するMシンボル分のSCH信号系列b(k+φ)，b(k+φ-1)，．．．b(k+φ-(M-1))を
乗算し、乗算結果を出力する。Ｍ＝２であれば、Mシンボル間位相回転量計算部83は、b(k+φ)・b(k+φ-1)を出力する。相関器84は(17)式の演算を行って相関値Rm(φ)を出力し、
位相誤差検出部85は(18)式によりM倍された位相誤差（位相シフト量）M・Δθを算出してランダムウォークフィルタ68に入力する。
ランダムウォークフィルタ68は、位相誤差の極性が正であればカウントアップし、負であればカウントダウンし、カウント値が設定値以上になればカウントを初期値にリセットすると共に、スイッチ67を介して局部発振周波数を所定幅減少し、設定値以下になれば局部発振周波数を所定幅上昇する。このAFC制御によれば、高精度で安定なキャリア周波数
同期を取ることができる。
・第2実施例
図20はAFC制御の第２の実施例構成図で、SCHが割り当てられているサブキャリア信号の合成をCMAアルゴリズムに従って行った場合の構成を示している。後段のキャリア周波数
同期制御は図19と同様な動作となる。
第１段階では、AFC部６６は従来例と同様に(図33〜図35参照)、ガードインターバル相
関によるピーク検出を行い、ピークタイミングでの相関値の位相誤差を検出し、その位相誤差を小さくするようにスイッチ67を介してVCO構成の局部発振器61の発振周波数を制御
する。
ついで、図18の制御でSCHフレームタイミング、FFTウインドウタイミング、差動復号のシンボル間隔Mを決定する。しかる後、FFT演算部71はFFTウインドウタイミングでFFT演算し、CMA合成部77はCMA SCH合成信号を出力する。複数シンボル間隔差動復号部81は、CMA
合成部77から出力するSCH合成信号に対して(16)式の複数シンボル間隔差動復号演算を行
う。
一方、位相シフト部82はSCH信号系列b(k)の位相を既に求まっているフレームタイミン
グφ分シフトしてb(k+φ)を出力する。Mシンボル間位相回転量計算部83は、位相シフト部82から入力するMシンボル分のSCH信号系列b(k+φ)，b(k+φ-1)，．．．b(k+φ-(M-1))を
乗算し、乗算結果を出力する。相関器84は(17)式の演算を行って相関値Rm(φ)を出力し、位相誤差検出部85は(18)式によりM倍された位相誤差（位相シフト量）M・Δθを算出してランダムウォークフィルタ68に入力する。ランダムウォークフィルタ68は、図19で説明したと同様の発振周波数制御を行ってオフセット周波数を零にする。これにより、高精度で安定なキャリア周波数同期を取ることができる。
（M）CSSC同定部の構成
図21はCSSC同定部の実施例構成図であり、図15と同一部分には同一符号を付している。二次元相関演算部90₁〜90_Mは、周波数と時間の二次元領域(m×nの二次元領域)にそれぞれ拡散されているCPICHとDPCHの多重データに、パイロット（CPICH）とCSSC候補の各複素共役を乗算して加算する積和演算を行い、積和演算結果(相関値)を各二次元拡散領域におけるCSSC候補のチャネル応答推定値h(i,1)〜h(i,M)として出力する。演算部91a〜91ｒは拡
大した二次元領域内の相関値の電圧平均を演算し、二次元電力加算部92は、各演算部91a
〜91ｒで演算された二次元電圧平均の電力を計算して加算する。最大値検出部93は各CSSC候補の二次元電力加算値を比較し、該電力加算値が最大となるCSSC候補を接続すべきセルのCSSCであると判定する。以後、検出されたCSSCを用いて受信データを復調する。
以上では、本発明をOFDM-CDMAに適用した場合について説明したが、本発明は、マルチ
キャリア変調方式、直交周波数分割多重(Orthogonal Frequency-Division Multiplexing:
OFDM)方式、あるいは、マルチキャリア符号分割多元接続(Multi-carrier Code-Division
Multiple Access: MC-CDMA)方式等にも適用できるものである。   (A) Frame configuration of the present invention
  FIG. 1 is an explanatory diagram of a frame structure of a signal transmitted from a base station according to the present invention, in which subcarriers are arranged in the frequency direction (horizontal direction) and the first to nth dedicated physical channels (Dedicated) in the code direction (vertical direction) Walsh codes of Physical Channel (DPCH) and Common Pilot Channel (CPICH) are arranged, and OFDM symbols of 1 to Ns (for example, Ns = 32) are arranged in the time direction (depth direction). In addition, the synchronization channel SCH includes a plurality of subcarriers F1 to F dedicated to SCH._NSCHAnd the length of the synchronization channel SCH is the frame length (= 32 OFDM symbol
Le). That is, subcarriers F1 to F_NSCHIs dedicated to the SCH, and other subcarriers are assigned to the spread signals of DPCH and CPICH.
  FIG. 2 shows the dedicated subcarriers F1 to F of the synchronization channel SCH_NSCHFIG. 6 is a diagram illustrating the relationship between DPCH and CPICH subcarriers. The SCH signal sequence generator 51 generates an SCH signal sequence of 1 frame Ns symbols.
The copy unit 52 generates N from the SCH signal of one symbol._SCHCopy symbols D₁~ D_NSCHGenerated
, Each copy symbol D₁~ D_NSCHSubcarrier F₁~ F_NSCHThe signal is input to the IFFT unit 53. Further, a multiplexed signal d of the dedicated physical channel DPCH or the common pilot channel CPICH that is spread in the frequency direction by a channelization code in a spreading unit (not shown) and then masked by a scramble code for station identification.₁~ D_MNAre the above-mentioned dedicated subcarriers F1 to F_NSCHExcept
Subcarrier f₁~ F_MNThe signal is input to the IFFT unit 53. IFFT unit 53 inputs in parallel
The IFFT (Inverse Fourier Transform) process is performed on the subcarrier signal to convert it into an OFDM symbol signal on the time axis. The guard interval insertion unit 54 inserts a guard interval into the OFDM symbol signal, and the transmission unit (not shown) performs orthogonal modulation on the OFDM symbol signal into which the guard interval is inserted, up-converts to a radio frequency, and amplifies the signal at a high frequency. Send.
  As described above, since the synchronization channel SCH is allocated to a dedicated subcarrier, the SCH is DP.
No interference from CH and CPICH. For this reason, the SCH frame timing detection characteristics described later can be improved. Also, when obtaining the same SCH frame timing detection characteristics as the conventional example
Can be detected with a small number of frames, and the frame timing detection time can be shortened.
  (B) SCH signal sequence differential encoding and frame timing detection
  By applying differential encoding processing to the SCH signal sequence,
Even when the rear frequency synchronization is insufficient and the residual carrier frequency error cannot be ignored, the SCH frame timing can be accurately detected.
  FIG. 3 is a block diagram of the differentially encoded SCH signal sequence generator of the present invention. The SCH signal sequence generator 51a generates a sequence having excellent autocorrelation characteristics, the delay unit 51b delays the input signal by one symbol period, and the multiplier 51c multiplies the SCH signal by the output signal one symbol before. Thus, differential encoding is performed on the SCH signal sequence.
  For example, if the SCH signal sequence is a BPSK modulated signal, the signal sequence is b (k) ∈ {1, -1}, k = 1,2, ..., N_s(N_SIs the number of frame symbols per subcarrier), the differentially encoded signal sequence is

It is represented by If there is a carrier frequency offset, the mth subcarrier signal is
,

It becomes. Where h_mIs the channel response in the mth subcarrier, Δf_cIs the carrier frequency
Number offset, T_SBIs the symbol block time. Differential decoding for this received signal sequence
If you do

It becomes.
  Since the phase φ of the received signal sequence is unknown, when performing a correlation operation with the sequence b (k + Ψ) obtained by sequentially shifting the phase of the signal sequence b (k) before differential encoding, the correlator output is

It is represented by If the phase Ψ of b (k) given by the correlator does not match the phase φ of the received signal sequence, b (k + φ) ・ b (k + Ψ) becomes 1 or −1 and N_SN even if added across symbols_S Not double
. On the other hand, when the phase Ψ of b (k) matches the phase of the received signal sequence, b (k + φ) ・ b (k + Ψ) is always 1.
N_S Because it is added over all symbols | R_m(Ψ) |²Is the maximum. Therefore, R_mThe phase where the power of (Ψ) is maximum can be detected as the frame timing.
  Further, the subcarrier signal r is caused by the carrier frequency offset Δfc._mAlthough (k) is rotating, it is converted into the same phase shift amount for each differential decoding by performing differential decoding. Therefore, even if the addition is performed by the correlator, the frame timing can be accurately detected because there is no power loss.
  In addition, if you are strongly receiving radio waves from 2 cells (2 base stations)

Similarly, when differential decoding is performed,

It becomes. Phase φ of the received signal sequence or

Is unknown, so the sequence b (k + Ψ) is obtained by sequentially shifting the phase of the signal sequence b (k) before differential encoding.
When the correlation operation is performed, the correlator output is

It is represented by When the phase Ψ of b (k) matches the phase φ of the received signal sequence

And the phase Ψ of b (k) is the phase of the received signal sequence

When matches

It becomes. Where ξ₁And ξ₂Is a signal component that cannot be suppressed by correlation. Therefore, the correlator output
When the phase where the power of the power is maximum is SCH frame timing, the one with the larger channel response,
In other words, it is possible to detect the frame timing of the transmission signal from the base station with the smallest path loss.
  In addition, when SCH is assigned to multiple subcarriers, the differential decoder output is combined.
Then the correlator output is

Therefore, it is possible to reduce the probability of detecting frame timing from a cell whose path loss is not the minimum due to the influence of short-term fading.
  FIG. 4 is a configuration diagram of SCH combining and frame timing detection in the receiving apparatus of the present invention. Differential decoder 72₁~ 72_NSCHIs the SCH among a number of subcarriers output from the FFT operation unit 71.
Assigned subcarrier F₁~ F_NSCHAre differentially decoded, and the adding unit 73 performs differential decoding.
Add the signals. Next, the correlator 74 performs a correlation operation between the SCH composite signal sequence output from the adder 73 and the known SCH signal sequence while sequentially shifting the latter phase by the phase shift unit 75, and the maximum value detection unit 76 The phase at which the correlator output is maximum is detected as the frame timing. The correlation calculation unit 74 performs the correlation calculation of the equations (8) and (10). Each differential decoder 72₁~ 72_NSCHConsists of a 1-symbol delay unit SDL, a complex conjugate generation unit CNJ, and a multiplication unit MPL.
  (C) Frame timing detection by CMA
  In the present invention, the SCH is assigned to a plurality of dedicated subcarriers, and the SCH is common to all cells. For this reason, if the timing between base stations is asynchronous, a plurality of SCH signals having different timings and channel responses are superimposed and received. This is a direct transmission of one signal sequence.
This is the same model as that in which tangential waves and delayed waves are received by a plurality of antennas. CMA (Constant Modulus Algorithm) as an algorithm to extract the path with the maximum power while suppressing other paths
It has been known. Therefore, in the present invention, a complex number is superimposed on a received signal on a plurality of SCH subcarriers.
The weighting coefficients are updated so that the error between the combined signal power and the desired power is minimized.
  FIG. 5 shows SCH synthesis and frame by Constant Modulus Algorithm (CMA) in the present invention.
FIG. The CMA algorithm is an adaptive array algorithm that selects the signal with the highest signal power. In FIG. 5, the signals are sent from a plurality of base stations.
Receives the SCH with the highest signal power among the SCHs to be received and weights to cancel other SCH signals
Is performed by adaptive control.
  Multiplication unit 77a of CMA composition unit 77₁~ 77a_NSCHIs the subcarrier F to which the SCH is assigned among the many subcarrier signals output from the FFT calculation unit 71.₁~ F_NSCHWeighting factor w₁(k) -w_NSCH(k), the synthesizer 77b synthesizes each multiplier output signal, and the CMA processor 77c
Weighting factor w so that the error between the combined output power and the desired output power decreases according to the₁(k) -w_NSCHUpdate (k).
  That is, the signal of the m-th subcarrier to which the SCH is assigned is x at time k._mAssuming (k), the output signal of the synthesizer 77b is

And the error signal is

Given in. Here, p and q are parameters that give the convergence characteristics of the algorithm. Coefficient vector W = [w₁, w₂,…, W_NSCH]^TIs

It is updated with. Where μ is the step size, * is the complex conjugate, and X (k) is assigned by SCH.
This is the received signal vector of the subcarrier being transmitted. When the weighting factor converges, only the SCH signal sequence with the minimum path loss is output to the combined output, and signals from other base stations are suppressed. Obedience
Thus, if this combined output is differentially decoded, unnecessary signal components are not generated, and therefore, SCH frame timing detection can be performed more accurately by subsequent correlation calculation.
  After the weighting factor convergence, the differential decoding unit 78 differentially decodes the CMA composite output signal, and the correlator 74 performs differential decoding.
The correlation calculation between the CMA composite signal sequence output from the signal unit 78 and the known SCH signal sequence is performed in the latter order.
The phase is shifted while being sequentially shifted by the phase shift unit 75, and the maximum value detection unit 76 detects the phase at which the correlator output is maximum as the frame timing. The differential decoding unit 78 is delayed by one symbol.
An extension SDL, a complex conjugate generator CNJ, and a multiplier MPL are included.
  (D) High-precision phase error estimation control
  In FIGS. 4 and 5, since the correlation calculation is performed after differential decoding, the phase shift amount of the correlation value at the detected frame timing is 1 symbol block of the carrier frequency offset.
The amount of rotation between the hooks. Phase shift amount, for example, in equation (9a)

Phase shift amount Δθ = 2πΔf_cT_SBIs

Can be detected. By the way, when the carrier frequency offset is small, the detection accuracy of the polarity of Δθ or Δθ deteriorates due to noise and signal components that are not suppressed. The accuracy of the polarity judgment of Δθ is most difficult to deteriorate due to noise etc., which is farthest from 0, which is the boundary value for polarity judgment.
This is when Δθ is ± π / 2. Therefore, between the symbols of differential decoding in the subsequent high-precision AFC
Divide M

Determine as. In this way, Δθ is multiplied by M to ± π / 2, so Δθ is small.
However, the polarity can be detected more correctly, and the carrier frequency control can be performed with high accuracy.
  From the above, is the differential decoding unit of FIGS. 4 and 5 performing the differential decoding with a plurality of symbol intervals M?
The correlation calculation is performed. For example, if the above M is M = 2, the differential decoding result d_m(K) is

It becomes. Since the frame timing φ has already been detected, if the correlation calculation is performed at that timing,

Thus, the amount of phase shift is expanded M times. Therefore, the resolution of phase shift amount detection is noisy.
If this is not sufficient, the present invention can greatly improve the polarity determination of the phase shift.
  FIG. 6 is a block diagram of the high-accuracy phase error estimator of the present invention.
M is determined, and the high-accuracy phase error Δθ used in the high-accuracy AFC method is estimated later. sand
That is, in the preceding stage, as described with reference to FIGS. 4 and 5, the correlator 74, the phase shift unit 75, and the maximum value detection unit 76 shift the phase of the SCH signal sequence and maximize the correlator output. φ
This is detected as a system timing. The phase detector 79 is a phase shift amount Δθ = 2πΔf of the correlation value at the frame timing._cT_SBThe symbol interval number calculation unit 80 determines the symbol interval number M by equation (15) and inputs it to the multiple symbol interval differential decoding unit 81.
  The multi-symbol interval differential decoding unit 81 calculates the complex conjugate of the symbol delay unit SDL that delays the CMA synthesis signal output from the CMA synthesis unit 77 (see FIG. 5) by M symbols and the output signal of the delay unit SDL.
A prime conjugate unit CNJ is provided with a multiplication unit that multiplies the received SCH signal and the complex conjugate of the SCH signal before M symbols, and performs the calculation of equation (16). The phase shift unit 82 has already obtained the phase of the SCH signal sequence b (k).
The frame timing φ is shifted and b (k + φ) is output. M symbol phase rotation amount calculation unit 83 receives M symbol SCH signal sequences b (k + φ), b (k + φ−1),. .
. Multiply b (k + φ− (M−1)) and output the multiplication result. If M = 2, the M symbol phase rotation amount calculation unit 83 outputs b (k + φ) · b (k + φ−1). The correlator 84 performs the calculation of equation (17)
The function value Rm (φ) is output, and the phase error detector 85 is

To calculate the phase error (phase shift amount) M · Δθ multiplied by M, and enter the AFC control unit (not shown).
To help. As described later, the AFC control unit sets the oscillation frequency of the local oscillator based on the polarity of the phase error.
Control is performed so that the offset frequency becomes zero, that is, the oscillation frequencies on the transmission side and the reception side are equal.
  By the way, in the above control, the symbol interval number M of the differential decoding unit is fixed during the cell search period.
There is no need to keep it. Carrier frequency offset decreases as AFC pulls
The phase shift amount Δθ is also reduced. Since the phase shift amount is obtained for each frame,

The phase M · Δθ of the phase is obtained from equation (18), and the phase shift amount Δθ is

Calculated by the following formula:

Thus, the optimum parameter M for estimating the phase shift amount of the next frame is determined. FIG. 7 is a block diagram of such a high-accuracy phase error estimator. The same parts as those in FIG. Symbol
The symbol interval calculation unit 86 calculates the symbol interval number M in the differential decoding calculation of the next frame by performing the calculations of the equations (19) and (20), and sets it in the multiple symbol interval differential decoding unit 81. This allows multiple thin
The Bol interval differential decoding unit 81 performs differential decoding at a set symbol interval.
  (E) Residual frequency error correction
  In the present invention, the phase shift amount M · Δθ due to the carrier frequency offset can be estimated.
Therefore, the estimated phase shift amount is multiplied by 1 / M to calculate the phase rotation amount Δθ per symbol block, and each received subcarrier signal is reversely rotated by this phase rotation amount to correct the residual frequency error. It becomes possible.
  In AFC, the carrier frequency offset is corrected, but the time constant of AFC control is very long, and the carrier frequency update amount of the local oscillator (VCO) is very small. Therefore, the AFC and the subcarrier signal residual frequency error correction do not interfere with each other. Therefore, both AFC and residual carrier frequency error correction can be performed using the phase shift amount.
  FIG. 8 is a block diagram of the present invention for correcting the residual carrier frequency offset using the phase shift amount.
It is. 6 or 7 calculates the phase error Δθ by the equation (19), the phase rotation control unit 88 calculates exp (−Δθ), and each multiplication of the phase rotation unit 89 The multiplier multiplies each subcarrier signal output from the FFT operation unit 71 by exp (−Δθ) and rotates it backward by −Δθ.
Correct the residual frequency error.
  (G) Two-dimensional diffusion
  Conventionally, transmission data is spread in the frequency direction, that is, in a one-dimensional area. However, the frequency
In the one-dimensional domain of number or time, the frequency selectivity of the channel or time-varying
The orthogonality tends to collapse. Therefore, in the present invention, as shown in FIG.
Each spreading area of the general pilot channel CPICH is affected by channel fluctuations as much as possible.
In order to avoid spreading, it is spread and multiplexed using a spreading code in a two-dimensional region of time and frequency (m in the frequency direction and n in the time direction), and further in CSSC (Cell Specific Scramble Code) Masking. When spreading in the two-dimensional region, the number of subcarriers in the frequency domain can be reduced as compared with the one-dimensional spreading at the same spreading gain, so that the orthogonality can be further maintained.
  10 and 11 are configuration diagrams on the transmission side of the base station apparatus including the two-dimensional spreading unit according to the present invention. In Fig. 10, CPICH and DPCH are expanded in a two-dimensional region of time and frequency using spreading codes.
Multiplexing is performed in a distributed manner, and product (masking) is performed with CSSC for each base station. Also, in FIG. 11, IFFT (Inverse Fourier Transform) processing is performed on the two-dimensional spread signal as a subcarrier signal and the signal of the synchronization channel SCH to convert it into an OFDM symbol signal on the time axis, thereby
It is configured to output with a global added.
  In FIG. 10, the first user's two-dimensional diffusion unit 50₁The serial / parallel converter 55 converts the transmission signal of the first user into M pieces of parallel data. M multiplication units 56 of the diffusion unit 56₁~ 56_M
Is the first user's channelization code (Walsh code) C for each parallel data₁~ C_N
Multiply and spread, serial to parallel converter 57₁~ 57_MThe corresponding multiplier 56₁~ 56_MThe N serial data input from is converted into n sets of m parallel data (N = m × n) and sequentially output. As a result, the first user data is divided into M two-dimensional regions (frequency direction) as shown in FIG.
M and n in the time direction)₁~ A_MIs diffused. Similarly, the two-dimensional diffusion unit 50 for other users₂~ 50_KPilot two-dimensional diffusion unit 50_P2D each user data and pilot
Spread to the area.
  Adder 58₁₁Each two-dimensional diffusion unit 50₁~ 50_K, 50_PSerial-to-parallel converter 57₁The first data is added from the first to m-th data output from the₁₂Similarly, the two-dimensional diffusion unit 50₁~ 50_K, 50_PSerial-to-parallel converter 57₁2nd data is added to the 1st to mth data output from
As well as adder 58_1mThe two-dimensional diffusion unit 50₁~ 50_K, 50_PSerial-to-parallel converter 57₁The m-th data among the first to m-th data output from is added.
  Each multiplier 59 of the masking unit 59₁₁~ 59_MmIs each adder 58₁₁~ 58_MmCell identification scramble code CSSC (G1-G_Mm) Is multiplied and output. Multiplier 59₁₁~ 59_1mIs
First, output the first set of m subcarrier signals, then the second set of m subcarrier signals
After that, n sets of subcarrier signals are output in the same manner, and a total of m × n two-dimensional spread signals TDS are output.₁Is output. Similarly, a multiplier 59 (not shown)_{twenty one}~ 59_2mOutputs the first set of m subcarrier signals, then outputs the second set of m subcarrier signals, and so on.
Outputs n sets of subcarrier signals, total mxn two-dimensional spread signal TDS₂Is output. Similarly, a multiplier 59 (not shown)_M1~ 59_MmOutputs a first set of m subcarrier signals, then outputs a second set of m subcarrier signals, and thereafter similarly sets of n subcarrier signals.
A total of m × n two-dimensional spread signal TDS_MIs output. Each of these two-dimensional spread signals TDS₁~ TDS_MIs input to the IFFT calculation unit 53 (FIG. 11).
  In FIG. 11, a differentially encoded SCH signal sequence generator 51 generates a differentially encoded 1 frame Ns symbol SCH signal sequence, and a copy unit 52 generates N symbols from a 1 symbol SCH signal._SCHCopy symbols D₁~ D_NSCHFor each copy symbol D₁~ D_NSCHSCH dedicated subcarrier F₁~ F_NSCHof
The signal is input to the IFFT unit 53 as a signal. In addition, each two-dimensional spread signal TDS generated from the masking unit 59 (FIG. 10).₁~ TDS_MIs the above dedicated subcarrier F1-F_NSCHSubcarrier other than f₁₁~ F_MmSignal
Is input to the IFFT section 53.
  IFFT section 53 performs IFFT (Inverse Fourier Transform) processing on the subcarrier signals input in parallel to convert them into OFDM symbol signals on the time axis. The guard interval insertion unit 54 inserts a guard interval into the OFDM symbol signal, and the transmission unit (not shown) performs orthogonal modulation on the OFDM symbol signal into which the guard interval is inserted, up-converts to a radio frequency, and amplifies the signal at a high frequency. Send.
  (I) Channel estimation and CSSC identification
  In the case of two-dimensional spreading as shown in FIG. 9, the receiving side performs a product-sum operation by multiplying the subcarrier signal obtained by the FFT operation by each complex conjugate of the pilot (CPICH) and the CSSC candidate.
That is, the channel response in the two-dimensional diffusion region can be estimated by performing a correlation operation. In this case, since the spreading area and the channel estimation area match, interference from the DPCH can be eliminated, and the channel estimation accuracy can be improved. In each two-dimensional diffusion region, 1
Two channel estimation values h (k, n) (k is a symbol index, n subcarrier index) are obtained.
  FIG. 13 is a block diagram of a channel response estimation unit for CSSC candidates, 71 is an FFT operation unit, 89 is a phase rotation unit (residual carrier frequency offset correction unit) described in FIG.₁~ 90_MIs a two-dimensional correlation calculation unit. Multiply CPICH and DPCH multiplexed data spread in frequency and time two-dimensional regions (m × n two-dimensional region) by multiplying each complex conjugate of pilot (CPICH) and CSSC candidate for product-sum operation Then, an estimated channel response value in the two-dimensional diffusion region is obtained. Therefore, the two-dimensional correlation calculation unit 90₁~ 90_MPerforms the product-sum operation to calculate channel response estimation values h (i, 1) to h (i, M) of CSSC candidates in the two-dimensional diffusion region.
  Next, in order to increase the difference between the correlation value obtained in each spreading area for the CSSC of the cell (base station) to be connected and the correlation value obtained for the CSSC of another cell, Are added to the voltage and averaged. In the present invention, in the two-dimensional area BA further expanding the diffusion area shown in FIG.
Do. FIG. 14 is a block diagram in which the correlation values are averaged by voltage addition. The same parts as those in FIG. 13 are denoted by the same reference numerals. The arithmetic units 91a to 91r calculate the voltage average of the correlation values in the enlarged two-dimensional area BA.
Calculate and output. According to such a configuration, since the average correlation value obtained in each diffusion region with respect to the CSSC of the cell to be connected becomes larger than the average correlation value of other CSSCs, it is possible to improve the CSSC identification performance.
  In the present invention, the average correlation in the two-dimensional area MBA further expanded from the two-dimensional area BA.
Add power of values. FIG. 15 is a block diagram for performing power addition of the average correlation value, and the same parts as those in FIG. The two-dimensional power addition unit 92 is calculated by the calculation units 91a to 91r.
The power of the two-dimensional voltage average in the calculated two-dimensional area BA is calculated and added. In this way, the difference between the metric for the CSSC of the cell to be connected and the metric for the CSSC of another cell becomes larger, and the CSSC identification performance can be improved. That is, the above two-dimensional power addition value is calculated for each CSSC candidate, and it is determined that the CSSC candidate having the maximum power addition value is the CSSC of the cell to be connected.
  Note that the complex conjugates of CSSC and pilot (CPICH) of the cell to be connected are converted into a two-dimensional correlation calculation unit 90.₁~ 90_MCorrelation value obtained by input to the channel becomes the channel response estimate in the two-dimensional diffusion region
Thus, channel compensation can be performed using these channel response estimation values.
  (J) Cell search procedure
  FIG. 16 shows the cell search procedure of the present invention. First, in the first stage, carrier frequency synchronization by conventional guard interval correlation is performed, and carrier frequency synchronization is roughly performed (S101). However, if the carrier frequency offset does not become so large systematically, this
One stage is not always necessary. … S101
  In the second stage, the correlation peak is detected by guard interval correlation, and the FFT window timing is synchronized. At this time, if the base stations are asynchronous, the timing before and after the obtained correlation peak is masked to detect the timing when the correlation power value is maximized, and a plurality of timing candidates are held.
To do. For systems with synchronized timing between base stations, the FFT window
One ming may be detected. … S102
  In the third stage, in each FFT window timing candidate obtained in the second stage, the received signal is converted into a subcarrier signal by FFT, the subcarrier signal to which the SCH is assigned is differentially decoded, and the correlation calculation is performed. Do. Correlation for all FFT timing candidates
The SCH frame timing that maximizes the power is detected. … S103
  In the fourth stage, based on the obtained SCH frame timing, multiple symbol interval differential decoding
To detect the phase error with high accuracy and achieve carrier frequency synchronization (S104). Parallel to stage 4
In the fifth stage, CSSC is identified while correcting the residual carrier frequency offset of the spread signal part (S105).
  (K) FFT window timing and SCH frame timing detection configuration
  ・ First example
  FIG. 17 shows an embodiment related to FFT window timing detection and SCH frame timing detection. A local oscillator (VCO) 61 and a mixer 62 convert the received signal frequency to an intermediate frequency. guard
The interval correlation calculation unit 63 performs guard interval correlation on the received signal, the in-phase adder 64 further performs in-phase addition and averages, and the peak detection unit 65 performs two or more peak detections and performs multiple FFT window timings. Detect. Detection of the above FFT window timing candidates
The control is the same as in the conventional example.
  Next, the frame timing and frequency offset (phase error) are controlled by the control explained in FIG.
To decide. That is, the received signal is converted into a subcarrier signal by the FFT operation unit 71 at each candidate FFT window timing, and the differential decoding unit 72₁~ 72_NSCHPerforms differential decoding on the subcarrier signal to which the SCH is assigned, and the adder 73 combines the differential decoding results to SCH combining
Output a signal. Correlator 74₁, 74₂Performs a correlation operation between the SCH composite signal and the SCH signal sequence before differential encoding for each FFT window timing candidate, and the maximum value detector 76 1) Correlation power value
2) detects the FFT window timing when the SCH frame timing is detected, and 3) further detects a phase error (carrier) from the correlation value at which the SCH frame timing is detected. (Frequency offset) Δθ is detected. Further, the maximum value detecting unit 76 also determines the symbol interval number M of the multi-symbol interval differential decoding used for the high-accuracy carrier frequency synchronization control.
  Second embodiment
  Figure 18 shows the second implementation of FFT window timing detection and SCH frame timing detection.
It is an example. FFT window candidates are determined by the same control as in FIG. Next, the explanation in FIG.
The frame timing and frequency offset (phase error) Δθ are determined by the control described above. That is, at each candidate FFT window timing, the FFT operation unit 71 converts the received signal into a subkey.
Convert to carrier signal. The CMA combiner 77 weights and combines the subcarrier signals to which the SCH is assigned, and updates the weighting coefficient using the CMA algorithm. Heavy
When the weighting coefficient converges, the CMA combining unit 77 outputs only the combined signal of the SCH from the base station with the maximum received power. Differential decoder 78₁, 78₂Performs differential decoding on the CMA composite signal of each FFT timing candidate, and correlator 74₁, 74₂Is the correlation operation between the differential decoding result and the SCH signal sequence before differential encoding
I do. The maximum value detector 76 detects 1) the SCH frame timing φ at which the correlation power value is maximum.
2) The FFT window timing when the SCH frame timing is detected is detected. 3) Further, the phase error (carrier) is calculated from the correlation value where the SCH frame timing is detected.
A) Frequency offset) Δθ is detected. Further, the maximum value detecting unit 76 also determines the symbol interval number M of the multi-symbol interval differential decoding used for the high-accuracy carrier frequency synchronization control.
  (L) AFC control
  ・ First example
  FIG. 19 is a block diagram of a first embodiment of AFC control. In the first stage, the AFC unit 66 is the same as the conventional example.
(See Fig. 33 to Fig. 35), peak detection by guard interval correlation is performed, the phase error of the correlation value at the peak timing is detected, and the local oscillator of the VCO configuration is set via the switch 67 so as to reduce the phase error. Controls the 61 oscillation frequency.
  Next, the SCH frame timing, FFT window timing, and differential decoding symbol interval M are determined under the control of FIG. Thereafter, the FFT operation unit 71 performs an FFT operation at the FFT window timing, and a multiple symbol interval differential decoding unit 72₁~ 72_NSCHPerforms a multi-symbol differential decoding operation on the subcarrier signal to which the SCH is assigned, that is, the operation of equation (16). The adder 73 combines the differential decoding results and outputs a SCH combined signal.
  On the other hand, the phase shift unit 82 is a frame timing that has already obtained the phase of the SCH signal sequence b (k).
Shifts by φ and outputs b (k + φ). M symbol phase rotation amount calculation unit 83 receives M symbol SCH signal sequences b (k + φ), b (k + φ−1),. . . b (k + φ- (M-1))
Multiply and output the multiplication result. If M = 2, the M symbol phase rotation amount calculation unit 83 outputs b (k + φ) · b (k + φ−1). The correlator 84 calculates the equation (17) and outputs a correlation value Rm (φ).
The phase error detection unit 85 calculates the phase error (phase shift amount) M · Δθ multiplied by M by the equation (18) and inputs it to the random walk filter 68.
  The random walk filter 68 counts up when the polarity of the phase error is positive, counts down when the polarity is negative, resets the count to the initial value when the count value exceeds the set value, and locally via the switch 67. The oscillation frequency is decreased by a predetermined width, and if the oscillation frequency is lower than the set value, the local oscillation frequency is increased by a predetermined width. According to this AFC control, highly accurate and stable carrier frequency
Can be synchronized.
  ・ Second embodiment
  FIG. 20 is a configuration diagram of the second embodiment of AFC control, and shows a configuration when the synthesis of subcarrier signals to which SCH is assigned is performed according to the CMA algorithm. Subsequent carrier frequency
The synchronization control is the same operation as in FIG.
  In the first stage, the AFC unit 66 is the same as the conventional example (see FIGS. 33 to 35), and the guard interval phase.
Detects the phase error of the correlation value at the peak timing, and controls the oscillation frequency of the local oscillator 61 in the VCO configuration via the switch 67 so as to reduce the phase error.
To do.
  Next, the SCH frame timing, FFT window timing, and differential decoding symbol interval M are determined under the control of FIG. Thereafter, the FFT operation unit 71 performs an FFT operation at the FFT window timing, and the CMA synthesis unit 77 outputs a CMA SCH synthesis signal. The multiple symbol interval differential decoding unit 81 is a CMA
The multi-symbol interval differential decoding operation of equation (16) is performed on the SCH composite signal output from the combiner 77.
Yeah.
  On the other hand, the phase shift unit 82 is a frame timing that has already obtained the phase of the SCH signal sequence b (k).
Shifts by φ and outputs b (k + φ). M symbol phase rotation amount calculation unit 83 receives M symbol SCH signal sequences b (k + φ), b (k + φ-1),. . . b (k + φ- (M-1))
Multiply and output the multiplication result. The correlator 84 calculates the equation (17) and outputs the correlation value Rm (φ), and the phase error detector 85 calculates the phase error (phase shift amount) M · Δθ multiplied by M according to the equation (18). To the random walk filter 68. The random walk filter 68 performs the same oscillation frequency control as described with reference to FIG. 19 to make the offset frequency zero. Thereby, highly accurate and stable carrier frequency synchronization can be achieved.
  (M) Configuration of CSSC identification unit
  FIG. 21 is a block diagram of an embodiment of the CSSC identification unit. The same parts as those in FIG. 15 are denoted by the same reference numerals. Two-dimensional correlation calculation unit 90₁~ 90_MIs the product of multiplying the CPICH and DPCH multiplexed data spread in the frequency and time two-dimensional regions (m × n two-dimensional regions) by multiplying each complex conjugate of the pilot (CPICH) and CSSC candidates. The sum operation is performed, and the product-sum operation result (correlation value) is output as channel response estimation values h (i, 1) to h (i, M) of CSSC candidates in each two-dimensional diffusion region. The arithmetic units 91a to 91r are expanded.
The voltage average of the correlation values in the large two-dimensional region is calculated, and the two-dimensional power adding unit 92
The power of the two-dimensional voltage average calculated at ~ 91r is calculated and added. The maximum value detection unit 93 compares the two-dimensional power addition values of the CSSC candidates and determines that the CSSC candidate having the maximum power addition value is the CSSC of the cell to be connected. Thereafter, the received data is demodulated using the detected CSSC.
  The case where the present invention is applied to OFDM-CDMA has been described above.
Carrier modulation method, Orthogonal Frequency-Division Multiplexing:
 OFDM) or multi-carrier code-division
  It can also be applied to multiple access (MC-CDMA) systems.

It is frame structure explanatory drawing of the signal transmitted from the base station of this invention. FIG. 6 is a diagram illustrating the relationship between a dedicated subcarrier for a synchronization channel SCH and subcarriers for DPCH and CPICH. FIG. 3 is a block diagram of the differentially encoded SCH signal sequence generator of the present invention. FIG. 4 is a configuration diagram of SCH combining and frame timing detection in the receiving apparatus of the present invention. FIG. 3 is a configuration diagram of SCH synthesis and frame timing detection by the CMA algorithm in the present invention. It is a block diagram of the highly accurate phase error estimation part of this invention. It is a block diagram of the highly accurate phase error estimation part which varies the symbol space | interval M of differential decoding. It is a block diagram of this invention which correct | amends a residual carrier frequency offset using a phase shift amount. It is explanatory drawing at the time of spreading | spreading each of a separate physical channel and a common pilot channel using the spreading code in the two-dimensional area | region of time and frequency. FIG. 3 is a configuration diagram of a first main part on the transmission side in a base station apparatus including a two-dimensional spreading unit according to the present invention. FIG. 6 is a configuration diagram of a second main part on the transmission side in a base station apparatus including a two-dimensional spreading unit according to the present invention. It is explanatory drawing in case N serial data is converted into n sets of m parallel data (N = m × n) and two-dimensional diffusion is realized. It is a block diagram of the channel response estimation part with respect to a CSSC candidate. It is a block diagram which averages by adding a voltage to a correlation value. It is a block diagram which performs the electric power addition of an average correlation value. It is procedure explanatory drawing of the cell search of this invention. It is an Example of this invention regarding a FFT window timing detection and a SCH frame timing detection. It is the 2nd Example of this invention regarding FFT window timing detection and SCH frame timing detection. FIG. 3 is a configuration diagram of a first embodiment of AFC control. It is a 2nd Example block diagram of AFC control. It is an Example block diagram of a CSSC identification part. It is a principle explanatory drawing of a multicarrier CDMA system. It is subcarrier arrangement | positioning explanatory drawing in multicarrier transmission. It is a block diagram of the transmission side (base station) of MC-CDMA. It is user data explanatory drawing. It is the conventional frame structure of the signal transmitted from a base station. It is explanatory drawing of the addition part in a frame code multiplexing part. It is a block diagram of a SCH data generation part. It is guard interval insertion explanatory drawing. It is a receiving side block diagram of the mobile station of MC-CDMA. It is an array explanatory drawing of the scramble code for cell identification. It is a principal part block diagram of the receiver provided with the AFC part which makes the oscillation frequency of a local oscillator correspond with the frequency of a transmission side. It is a block diagram of an AFC part. FIG. 6 is an operation explanatory diagram of an AFC unit. It is phase error explanatory drawing. It is a frame block diagram of a time division multiplexing system. 2 is a configuration example of AFC control in time division multiplexing. It is cell search explanatory drawing. It is explanatory drawing of the peak value of guard interval correlation. SCH frame format. It is explanatory drawing of the voltage addition area | region and electric power addition area | region in a SCH frame timing detection process.

Explanation of symbols

2D diffusion part for 50 ₁ to 50 _K users
Two-dimensional diffusion for 50 _P pilot
58 ₁₁ ~58 _Mm adder
59 Masking part
59 _{11 to} 59 _1m multiplier

Claims (5)

In a transmission method in which a transmission signal sequence is spread and transmitted to a plurality of subcarriers in the frequency direction,
Spreading the transmission signal sequences of the common pilot channel and the dedicated channel constituting the transmission signal sequence into two-dimensional regions of time and frequency using channelization codes, respectively;
Masking the spread signal with a scramble code for base station identification;
Transmitting a masked transmission signal sequence,
A transmission method characterized by the above. In a transmission apparatus for transmitting a transmission signal sequence by spreading it over a plurality of subcarriers in the frequency direction,
A spreading unit that spreads the transmission signal sequences of the common pilot channel and the dedicated channel constituting the transmission signal sequence into a two-dimensional region of time and frequency using channelization codes, respectively
A masking unit that masks the spread signal with a scramble code for each base station,
A transmission unit for transmitting the masked transmission signal sequence;
A transmission device comprising: On the transmission side, the transmission signal sequences of the common pilot channel and the dedicated channel are spread and multiplexed in a two-dimensional area of time and frequency using channelization codes, respectively, and the multiplexed signal is masked with a scramble code of a base station. In the cell search method of the receiving device for receiving the received signal,
Each subcarrier signal extracted from the received signal is multiplied by the complex conjugate of each of the common pilot and the scramble code candidate,
Add the multiplication results in the two-dimensional diffusion region to obtain the channel response,
The channel responses of a plurality of the two-dimensional diffusion regions belonging to the expanded two-dimensional region are averaged by voltage addition, and the power of the averaged channel response is calculated,
Add power to the channel response of each expanded 2D region,
A scramble code candidate that maximizes the power addition value is determined to be a scramble code of the base station.
A cell search method characterized by the above. Each subcarrier signal is multiplied by the complex conjugate of each of the common pilot and the base station scramble code, the result of multiplication in the two-dimensional spreading region is added to estimate the channel response, and the channel response estimation value is used. Channel compensation,
The cell search method according to claim 3, wherein: On the transmission side, the transmission signal sequences of the common pilot channel and the dedicated channel are spread and multiplexed in a two-dimensional area of time and frequency using channelization codes, respectively, and the multiplexed signal is masked with a scramble code of a base station. In the receiving device for receiving the received signal,
A channel response calculator for multiplying each subcarrier signal extracted from the received signal by the complex conjugate of each of the common pilot and the scramble code candidate, and adding a multiplication result in the two-dimensional spreading region to calculate a channel response;
A two-dimensional voltage adding unit for averaging the channel responses of a plurality of the two-dimensional diffusion regions belonging to the enlarged two-dimensional region by voltage addition;
A two-dimensional power adding unit that calculates the power of the averaged channel response of the two-dimensional diffusion region and adds the power of the channel response of each expanded two-dimensional region;
A cell determination unit that determines that a scramble code candidate having the maximum power addition value is a scramble code of a base station;
A receiving apparatus comprising:

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