JP2007043807A - Switching power supply circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a voltage resonance converter equipped with a series resonance circuit on the secondary in which power conversion efficiency is enhanced as a power circuit with a power factor improving function while decreasing the number of circuit components. <P>SOLUTION: A primary switching converter operating by receiving a DC input voltage obtained by rectifying and smoothing a commercial AC power supply through a rectification circuit (including a smoothing capacitor Ci) is constituted as class E resonance type. A power factor improving circuit 10 is arranged such that an operation for inducing a resonance pulse voltage and feeding that voltage back to a smoothing capacitor Ci and an operation for regenerating power of a switching current obtained in the primary winding N1 and feeding it back to the smoothing capacitor Ci are obtained for a choke coil winding N11 for improving power factor from a choke coil winding N10 forming a primary parallel resonance circuit and a switching diode D1 chops a rectification current depending on a switching output thus fed back. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

  The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

JP-A-6-327246 (FIG. 11)

In recent years, the development of switching elements that can withstand relatively high currents and voltages at high frequencies has led to most switching power supply circuits as power supply circuits that rectify commercial power and obtain a desired DC voltage. .
The switching power supply circuit reduces the size of the transformer and other devices by increasing the switching frequency, and is used as a power source for various electronic devices as a high-power DC-DC converter.

By the way, in general, when a commercial power supply is rectified, the current flowing through the smoothing circuit becomes a distorted waveform, which causes a problem that the power factor indicating the utilization efficiency of the power supply is impaired.
Further, there is a need for measures for suppressing harmonics generated by such a distorted current waveform.
Thus, a technique using a so-called active filter has been known as a technique for improving the power factor in the past (see, for example, Patent Document 1 above).

The basic configuration of such an active filter is, for example, as shown in FIG.
In FIG. 13, a bridge rectifier circuit Di is connected to a commercial AC power supply AC. An output capacitor Cout is connected in parallel to the positive / negative line of the bridge rectifier circuit Di. By supplying the rectified output of the bridge rectifier circuit Di to the output capacitor Cout, a DC voltage Vout is obtained as a voltage across the output capacitor Cout. This DC voltage Vout is supplied as an input voltage to a load 110 such as a DC-DC converter in the subsequent stage.

As shown in the figure, the power factor improvement includes an inductor L, a fast recovery type diode D, a resistor Ri, a switching element Q, and a multiplier 111.
The inductor L and the diode D are connected in series and inserted between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the output capacitor Cout.
The resistor Ri is inserted between the negative output terminal (primary side ground) of the bridge rectifier circuit Di and the negative terminal of the output capacitor Cout.
In this case, the switching element Q1 is a MOS-FET, and is inserted between the connection point of the inductor L and the diode D and the primary side ground as shown.

To the multiplier 111, a current detection line LI and a waveform input line Lw are connected as a feedforward circuit, and a voltage detection line LV is connected as a feedback circuit.
The multiplier 111 detects the level of the rectified current that is input from the current detection line LI and flows to the negative output terminal of the bridge rectifier circuit Di.
Further, the rectified voltage waveform at the positive output terminal of the bridge rectifier circuit Di input from the waveform input line Lw is detected. This corresponds to detecting the waveform of the commercial AC power supply AC (AC input voltage) as an absolute value.
Further, the fluctuation difference of the DC input voltage is detected based on the DC voltage Vout of the output capacitor Cout input from the voltage detection line LV.
The multiplier 111 outputs a drive signal for driving the switching element Q.

  A rectified current flowing in the negative output terminal of the bridge rectifier circuit Di is input from the current detection line LI to the multiplier 111. The multiplier 111 detects the rectified current level input from the current detection line LI. Further, the fluctuation difference of the DC input voltage is detected based on the DC voltage Vout of the output capacitor Cout input from the voltage detection line LV. Further, the rectified voltage waveform at the positive output terminal of the bridge rectifier circuit Di input from the waveform input line Lw is detected. This corresponds to detecting the waveform of the commercial AC power supply AC (AC input voltage) as an absolute value.

  The multiplier 111 first multiplies the rectified current level detected from the current detection line LI as described above and the fluctuation difference of the DC input voltage detected from the voltage detection line LV. Then, a current command value having the same waveform as the AC input voltage VAC is generated based on the multiplication result and the waveform of the AC input voltage detected from the waveform input line Lw.

  Further, the multiplier 111 in this case compares the current command value with the actual AC input current level (detected based on the input from the current detection line L1), and performs PWM control on the PWM signal according to this difference. To generate a drive signal based on the PWM signal. The switching element Q is switched by this drive signal. As a result, the AC input current is controlled to have the same waveform as the AC input voltage, and the power factor is improved so that the power factor approaches one. In this case, since the current command value generated by the multiplier is controlled so that the amplitude changes according to the fluctuation difference of the rectified smoothing voltage, the fluctuation of the rectified smoothing voltage is also suppressed. .

  FIG. 14A shows the input voltage Vin and the input current Iin input to the active filter circuit shown in FIG. The voltage Vin corresponds to the voltage waveform as the rectified output of the bridge rectifier circuit Di, and the current Iin corresponds to the current waveform as the rectified output of the bridge rectifier circuit Di. Here, the waveform of the current Iin has the same conduction angle as the rectified output voltage (voltage Vin) of the bridge rectifier circuit Di. This is the waveform of the AC input current flowing from the commercial AC power supply AC to the bridge rectifier circuit Di. Also indicates that the conduction angle is the same as that of the current Iin. That is, a power factor close to 1 is obtained.

FIG. 14B shows a change in energy (power) Pchg input / output to / from the output capacitor Cout. The output capacitor Cout stores energy when the input voltage Vin is high, and releases energy when the input voltage Vin is low to maintain the flow of output power.
FIG. 14C shows the waveform of the charge / discharge current Ichg with respect to the output capacitor Cout. The charge / discharge current Ichg flows corresponding to the energy Pchg accumulation / discharge operation in the output capacitor Cout, as can be seen from the fact that it is in phase with the waveform of the input / output energy Pchg in FIG. Current.

  Unlike the input current Iin, the charge / discharge current Ichg has substantially the same waveform as the second harmonic of the AC line voltage (commercial AC power supply AC). In the AC line voltage, a ripple voltage Vdc is generated in the second harmonic component as shown in FIG. 14D due to the flow of energy with the output capacitor Cout. The ripple voltage Vdc has a phase difference of 90 ° with respect to the charge / discharge current Ichg shown in FIG. The rating of the output capacitor Cout is determined in consideration of processing the second harmonic ripple current and the high frequency ripple current from the boost converter switch that modulates the current.

Further, FIG. 15 shows a configuration example of an active filter having a basic control circuit system based on the circuit configuration of FIG. In addition, about the part made the same as FIG. 13, the same code | symbol is attached | subjected and description is abbreviate | omitted.
A switching pre-regulator 115 is provided between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the output capacitor Cout. In FIG. 13, the switching pre-regulator 115 is a part formed by the switching element Q, the inductor L, the diode D, and the like.

The control circuit system including the multiplier 111 includes a voltage error amplifier 112, a divider 113, and a squarer 114.
In the voltage error amplifier 112, the DC voltage Vout of the output capacitor Cout is divided by the voltage dividing resistor Rvo-Rvd and input to the non-inverting input of the operational amplifier 112a. The reference voltage Vref is input to the inverting input of the operational amplifier 112a. In the operational amplifier 112a, a voltage of a level corresponding to the error of the divided DC voltage Vout with respect to the reference voltage Vref is amplified by an amplification factor determined by the feedback resistor Rvl and the capacitor Cvl, and the divider 113 is used as the error output voltage Vvea. Output to.

  In addition, a so-called feedforward voltage Vff is input to the squarer 114. The feedforward voltage Vff is an output (average input voltage) obtained by averaging the input voltage Vin by the averaging circuit 116 (Rf11, Rf12, Rf13, Cf11, Cf12). The squarer 114 squares the feedforward voltage Vff and outputs it to the divider 113.

In the divider 113, the error output voltage Vvea from the voltage error amplifier 112 is divided by the square value of the average input voltage output from the squarer 114. A signal as a result of the division is output to the multiplier 111.
That is, the voltage loop is composed of a system of a squarer 114, a divider 113, and a multiplier 111. The error output voltage Vvea output from the voltage error amplifier 112 is divided by the square of the average input voltage (Vff) before being multiplied by the rectified input signal Ivac in the multiplier 111. With this circuit, the gain of the voltage loop is kept constant without changing as the square of the average input voltage (Vff). The average input voltage (Vff) has a function of open loop correction that is sent forward in the voltage loop.

The multiplier 111 receives the output obtained by dividing the error output voltage Vvea by the divider 113 and the rectified output (Iac) of the positive output terminal (rectified output line) of the bridge rectifier circuit Di via the resistor Rvac. Here, the rectified output is shown not as a voltage but as a current (Iac). The multiplier 111 multiplies these inputs to generate and output a current programming signal (multiplier output signal) Imo. This corresponds to the current command value described in FIG. The output voltage Vout is controlled by varying the average amplitude of this current programming signal. That is, a PWM signal corresponding to a change in the average amplitude of the current programming signal is generated, and switching drive is performed by a drive signal based on the PWM signal, thereby controlling the level of the output voltage Vout.
Thus, the current programming signal has an average amplitude waveform that controls the input and output voltages. Note that the active filter controls not only the output voltage Vout but also the input current Vin. Since the current loop in the feedforward circuit can be said to be programmed by the rectified line voltage, the input to the subsequent converter (load 110) becomes resistive.

  FIG. 16 shows a configuration example of a power supply circuit in which a current resonance type converter is connected to the subsequent stage of the active filter based on the configuration shown in FIG. The power supply circuit shown in this figure corresponds to a commercial AC power supply input level indicated as AC input voltage VAC = 85V to 264V, and adopts a configuration that can correspond to the condition of load power Po = 300 W to 0 W. Further, the current resonance type converter adopts a configuration of a separately excited half bridge coupling method.

In the power supply circuit shown in FIG. 16, a common mode noise filter including two sets of line filter transformers LFT and three sets of across capacitors CL is provided for the commercial AC power supply AC according to the illustrated connection mode. A bridge rectifier circuit Di is connected.
Also connected to the rectified output line of the bridge rectifier circuit Di is a normal mode noise filter 125 comprising a pair of choke coils LN and two sets of filter capacitors (film capacitors) CN, CN connected as shown in the figure. Is done.

The positive output terminal of the bridge rectifier circuit Di is connected to the positive terminal of the smoothing capacitor Ci via the series connection of the choke coil LN, the inductor Lpc of the power choke coil PCC, and the fast recovery type rectifier diode D20. The smoothing capacitor Ci corresponds to the output capacitor Cout in FIGS. Further, the inductor Lpc and the diode D20 of the power choke coil PCC correspond to the inductor L and the diode D shown in FIG.
In addition, an RC snubber circuit comprising a series connection of a capacitor Csn and a resistor Rsn is connected in parallel to the rectifier diode D20 in this figure.

Switching element Q3 corresponds to switching element Q10 in FIG. That is, in actually mounting the switching element of the active filter, in this case, the switching element Q3 is connected to the connection point between the power choke coil Lpc and the fast recovery type rectifier diode D20, and the primary side ground (via the resistor R3). It is inserted between.
In this case, a MOS-FET is selected as the switching element Q3.

In this case, the power factor / output voltage control IC 120 is an integrated circuit (IC) that controls the operation of the active filter for improving the power factor so that the power factor approaches 1.
In this case, the power factor / output voltage control circuit 20 includes a multiplier, a divider, an error voltage amplifier, a PWM control circuit, and a drive circuit that outputs a drive signal for switching the switching element. . Circuit portions corresponding to the multiplier 111, the voltage error amplifier 112, the divider 113, the squarer 114, and the like shown in FIG. 15 are included in the power factor / output voltage control IC 20.

  In this case, the feedback circuit is formed so that the voltage value obtained by dividing the voltage across the smoothing capacitor Ci (rectified smoothing voltage Ei) by the voltage dividing resistors R5 and R6 is input to the terminal T1 of the power factor / output voltage control IC 20. Is done.

  As the feedforward circuit, a series connection of voltage dividing resistors R1-R2 is provided between the positive output terminal of the bridge rectifier circuit Di and the primary side ground, and the connection point of the voltage dividing resistors R1-R2 is defined as a terminal T5. To connect with. As a result, the rectified output of the bridge rectifier circuit Di is divided and input to the terminal T5. In this way, a feedforward circuit is formed as a line corresponding to the current detection line LI in FIG.

Further, the operating power supply of the power factor / output voltage control IC 20 is supplied to the terminal T4. In this terminal T4, the alternating voltage excited in the winding N5 transformer-coupled with the inductor Lpc in the power choke coil PCC is converted into a low-voltage DC voltage by a half-wave rectifier circuit comprising a diode D11 and a capacitor C11 as shown. Supplied.
The terminal T4 is connected to the positive output terminal of the bridge rectifier circuit Di via the starting resistor Rs. When the commercial AC power supply AC is turned on and the power supply circuit is activated, the rectified output obtained at the positive output terminal of the bridge rectifier circuit Di is supplied to the terminal T4 via the activation resistor Rs. The power factor / output voltage control IC 20 starts the operation using the rectified output supplied in this way as a starting power source.

A drive signal (gate voltage) for driving the switching element is output from the terminal T3 to the gate of the switching element Q3.
The switching element Q3 performs a switching operation according to the applied drive signal.

  Further, a resistor R3 is inserted between the source of the switching element Q3 and the primary side ground, and the connection point between the source of the switching element Q3 and the resistor R3 is connected to the terminal T2. The resistor R3 in this case is provided for detecting an overcurrent flowing through the switching element Q3 as a voltage. When a voltage at a level corresponding to the overcurrent is detected at the terminal T2, the power factor / output voltage control IC 20 performs a protection operation such as stopping the switching operation of the switching element Q3, for example. .

  The switching drive of the switching element Q3 is a drive based on PWM control so that the conduction angle of the rectified output current is substantially the same as the rectified output voltage waveform as described with reference to FIGS. Done by signal. The conduction angle of the rectified output current is substantially the same as the rectified output voltage waveform. That is, the conduction angle of the AC input current flowing from the commercial AC power supply AC is substantially the same as the waveform of the AC input voltage VAC. As a result, the power factor is controlled to be approximately 1. That is, power factor improvement is achieved.

Here, FIG. 17 and FIG. 18 show the power factor improvement operation of the active filter having the above-described configuration in practice.
First, FIG. 17 shows the switching operation of the switching element Q3 according to the load variation and the current I1 flowing through the inductor Lpc of the power choke coil PCC. FIG. 17A shows the operation at a light load, FIG. 17B shows the operation at an intermediate load, and FIG. 17C shows the operation at a heavy load.
As can be seen by comparing FIGS. 17A, 17B, and 17C, the switching element Q3 has a switching cycle that is constant, and the ON period becomes longer as the load tends to be heavy. It is operating. That is, switching driving by PWM control is performed.
In response to such a switching operation, the current I1 flows in a discontinuous mode in which a non-conducting period is formed at light load. Further, under the condition of an intermediate load, it flows in a critical mode. And in the load condition range made into heavy load, it is made to flow in a continuous mode. Such a transition of the operation mode of the current I1 is obtained by selecting the inductor Lpc. For example, in actuality, Lpc = 140 μH is selected in order to correspond to the input of a 100V commercial AC power supply. In this way, by adjusting the current I1 flowing into the smoothing capacitor Ci through the inductor Lpc according to the load condition, the rectified smoothing voltage with respect to the level fluctuation and load fluctuation of the commercial AC power supply AC (AC input voltage VAC) Ei can be stabilized. In this case, the AC input voltage VAC is set to a constant voltage of 380 V with respect to the range of 85 V to 264 V. The rectified and smoothed voltage Ei is a voltage across the smoothing capacitor Ci and corresponds to Vout in FIG. 15, and is a DC input voltage for the current resonance converter at the subsequent stage.

FIG. 18 shows the waveforms of the AC input current IAC and the rectified and smoothed voltage Ei by comparison with the AC input voltage VAC. In this figure, experimental results when the AC input voltage VAC = 100 V are shown.
As shown in this figure, the AC input current IAC corresponding to the input of the AC input voltage VAC = 100 V has a waveform that substantially matches the conduction period of the AC input voltage VAC. That is, the power factor is improved.
Further, it is shown that the rectified and smoothed voltage Ei is stabilized at an average value of 380 V along with the improvement of the power factor. Further, as shown in the figure, the actual waveform of the rectified and smoothed voltage Ei has a ripple fluctuation of 10 Vp-p with respect to 380 V.

  The current resonance type converter at the latter stage of the active filter performs the switching operation for power conversion by inputting the rectified and smoothed voltage Ei as a DC input voltage. As shown in the figure, two-stone switching elements Q1, Q2 Comprising. In this case, a half-bridge connection is made such that the switching element Q1 is on the high side and the switching element Q2 is on the low side, and the switching element Q1 is connected in parallel with the rectified and smoothed voltage Ei (DC input voltage). That is, a current resonance type converter by a half bridge coupling method is formed.

The current resonance type converter in this case is a separately excited type, and correspondingly, MOS-FETs are used for the switching elements Q1 and Q2. Clamp diodes DD1 and DD2 are connected in parallel to these switching elements Q1 and Q2, respectively, thereby forming a switching circuit. These clamp diodes DD1 and DD2 form a path through which a reverse current flows when the switching elements Q1 and Q2 are turned off.
The switching elements Q1 and Q2 are driven to be switched at a required switching frequency by the oscillation / drive circuit 2 at an on / off timing. The oscillation / drive circuit 2 operates so as to variably control the switching frequency based on the control according to the level of the secondary side DC output voltage Eo executed by the control circuit 1 shown in the figure. The DC output voltage Eo is stabilized.

The insulating converter transformer PIT is provided to transmit the switching outputs of the switching elements Q1, Q2 from the primary side to the secondary side.
One end of the primary winding N1 of the insulating converter transformer PIT is connected to the connection point (switching output point) of the switching elements Q1 and Q2, and the other end is connected to the primary side via the series resonance capacitor C1. Connected to ground. Here, the series resonant capacitor C1 forms a series resonant circuit by its own capacitance and the leakage inductance (L1) of the primary winding N1. This series resonance circuit causes a resonance operation when the switching outputs of the switching elements Q1 and Q2 are supplied. By this, the operation of the switching circuit composed of the switching elements Q1 and Q2 is made a current resonance type.

A secondary winding N2 is wound on the secondary side of the insulating converter transformer PIT.
In this case, the secondary winding N2 is provided with a center tap as shown in the figure, the center tap is connected to the secondary side ground, and then the rectifier diodes Do1, Do2 and the smoothing as shown in the figure. A double-wave rectifier circuit is formed by providing the capacitor Co. Thereby, the secondary side DC output voltage Eo is obtained as the voltage across the smoothing capacitor Co. The secondary side DC output voltage Eo is supplied to a load side (not shown) and is also branched and input as a detection voltage for the control circuit 1. The control circuit 1 supplies a control signal corresponding to the level of the input secondary side DC output voltage Eo to the oscillation / drive circuit 2. The oscillation / drive circuit 2 drives the switching elements Q1, Q2 so as to vary the switching frequency so that the secondary side DC output voltage Eo is stabilized in accordance with the control signal. That is, stabilization by the switching frequency control method is performed.

  FIG. 19 shows characteristics of AC → DC power conversion efficiency (overall efficiency), power factor, and rectified smoothing voltage Ei with respect to load fluctuation. In this figure, the characteristic with respect to the fluctuation | variation of load electric power Po = 300W-0W at the time of alternating current input voltage VAC = 100V (AC100V type | system | group) is shown. FIG. 20 shows characteristics of AC → DC power conversion efficiency (overall efficiency), power factor, and rectified smoothing voltage Ei with respect to fluctuations in the AC input voltage VAC. In this figure, the characteristic with respect to the fluctuation | variation of alternating current input voltage VAC = 85V-264V under load conditions Po = 300W and fixed load conditions is shown.

First, the AC → DC power conversion efficiency (ηAC → DC) decreases as the load power Po becomes a heavy load condition as shown in FIG. Further, with respect to the fluctuation of the AC input voltage VAC, under the same load condition, as shown in FIG. 20, the level of the AC input voltage VAC tends to increase as the level increases. ing.
Actually, under the load condition of load power Po = 300W, when AC input voltage VAC = 100V, ηAC → DC = about 83.0%, and when AC input voltage VAC = 230V, ηAC → DC = about 89.0%. The result is obtained.

Further, as shown in FIG. 19, the power factor PF has a characteristic that is substantially constant with respect to fluctuations in the load power Po. Further, the fluctuation characteristic of the power factor PF with respect to the fluctuation of the AC input voltage VAC is a characteristic that can be regarded as almost constant as shown in FIG. 20, although it tends to decrease as the AC input voltage VAC increases. I understand that.
Actually, under the load condition of load power Po = 300 W, power factor PF = 0.96 is obtained when AC input voltage VAC = 100 V, and power factor PF = 0.94 is obtained when AC input voltage VAC = 230 V.

  Further, as shown in FIGS. 19 and 20, the rectified and smoothed voltage Ei has a result that is constant with respect to fluctuations in the load power Po and the AC input voltage VAC.

  As can be seen from the above description, the power supply circuit shown in FIG. 16 is configured by mounting the conventionally known active filter shown in FIGS. 13 and 15. By adopting such a configuration, the power factor is improved.

However, the power supply circuit having the configuration shown in FIG. 16 has the following problems.
First, as shown in the figure, the power conversion efficiency in the power supply circuit shown in FIG. 16 is the AC → DC power conversion efficiency corresponding to the active filter in the previous stage, and the DC → DC power conversion efficiency of the current resonance converter in the subsequent stage. It will be a synthesis of
That is, the total power conversion efficiency of the circuit shown in FIG. 16 is a value obtained by multiplying the values of these power conversion efficiencies and tends to decrease accordingly.
According to the experiment, the AC → DC power conversion efficiency in the part corresponding to the active filter in the circuit of FIG. 16 is about ηAC → DC = 88.5% when the AC input voltage VAC = 100V, and the AC input voltage VAC = 230V. Under the conditions, ηAC → DC = 95%. The DC → DC power conversion efficiency on the current resonance type converter side is about ηDC → DC = 94% when the load power Po = 300 W and the rectified smoothing voltage Ei = 380V.
Accordingly, the overall AC → DC power conversion efficiency in the circuit of FIG. 16 decreases to about ηAC → DC = 83.0% when the AC input voltage VAC = 100 V, as described above, and the AC input Even when the voltage VAC = 230 V, the voltage decreases to ηAC → DC = 89.0%.

In addition, since the active filter circuit is a hard switching operation, the level of noise generation is very high, and thus a relatively severe noise suppression measure is required.
For this reason, in the circuit shown in FIG. 16, a noise filter is formed by two sets of line filter transformers and three sets of across capacitors for the line of the commercial AC power supply AC. That is, two or more stages of filters are required.
Further, a normal mode noise filter including one set of choke coils LN and two sets of filter capacitors CN is provided for the rectified output line. Further, an RC snubber circuit is provided for the fast recovery diode D20 for rectification.
In this way, noise countermeasures due to an extremely large number of parts are necessary for an actual circuit, resulting in an increase in cost and an increase in the mounting area of the power supply circuit board.

  Furthermore, the switching frequency of the switching element Q3 operated by the power factor / output voltage control IC 120 as a general-purpose IC is fixed at 60 kHz, whereas the switching frequency of the subsequent current resonance type converter is in the range of 80 kHz to 200 kHz. Variable. Since the switching timings of the two are independently performed in this way, the primary side ground potential interferes and becomes unstable due to the switching operation of the two, and abnormal oscillation is likely to occur, for example. As a result, problems such as difficulty in circuit design and deterioration of reliability are also caused.

In view of the above problems, the present invention is configured as a switching power supply circuit as follows.
In other words, it is formed with at least a rectifying element and a smoothing capacitor, and a rectifying / smoothing circuit that generates a rectifying / smoothing voltage as a voltage across the smoothing capacitor by inputting a commercial AC power supply and rectifying / smoothing, and inputting the rectifying / smoothing voltage A switching element that performs switching, and switching drive means that performs switching driving of the switching element.
Further, the first inductor inserted in series with respect to the path through which the rectified and smoothed voltage is input to the switching element is connected in parallel with the switching element, and at least the inductance of the first inductor and its own The primary side parallel resonant capacitor which forms a primary side parallel resonant circuit by capacitance is provided.
Further, by connecting the second inductor and the second inductor in series, a primary side series resonant circuit is formed by at least the inductance of the second inductor and its own capacitance, and the second inductor A primary side series resonance capacitor is provided so that a series connection circuit of the inductor and itself is connected in parallel with the switching element.
In addition, the second inductor is wound as a primary winding, and a secondary winding in which an alternating voltage is induced by the switching output obtained by the primary winding is wound, and is regarded as loosely coupled. A converter transformer configured to obtain a required coupling coefficient between the primary side and the secondary side is provided.
In addition, an operation in which the alternating voltage induced in the power factor improving inductor from the first inductor is fed back to the smoothing capacitor forming the rectifying and smoothing circuit, and a current obtained in the primary winding in accordance with the switching operation of the switching element is obtained. The operation is regenerated as electric power and fed back to the smoothing capacitor via the first inductor, and the rectified current obtained by the rectifying operation in the rectifying and smoothing circuit is changed according to the switching output fed back by these operations. The power factor improving means configured to be intermittent by the power factor improving switching element is provided.

  In the present invention, the “coupling coefficient” indicates the degree of electromagnetic coupling. A numerical value of 1 indicates the highest degree of coupling, and a numerical value of 0 indicates the lowest degree of coupling (coupling) Not).

The power supply circuit having the above configuration forms a circuit configuration as a class E switching converter as a basic configuration of the primary side switching converter. The class E switching converter is a type of soft switching converter called a composite resonance type including a parallel resonance circuit (primary side parallel resonance circuit) and a series resonance circuit (primary side series resonance circuit). The primary winding of the converter transformer becomes an inductor (second inductor) that forms a series resonance circuit (primary side series resonance circuit) in the class E switching converter.
In this way, the primary side switching converter is a class E switching converter, so that, for example, ZVS (Zero Voltage) under a load condition that is an intermediate load, such as an interaction between the primary side parallel resonant circuit and the secondary side series resonant circuit. Switching: Zero voltage switching) This eliminates the factors that prevent operation.
In addition, in order to improve the power factor, an alternating voltage (resonant voltage pulse) obtained in the primary side parallel resonant circuit is induced from the first inductor to the power factor improving choke coil, and the voltage is fed back to the smoothing capacitor. Is done. At the same time, the current obtained in the primary winding is regenerated as power and fed back to the smoothing capacitor via the first inductor.
Thus, for example, in configuring a power supply circuit having a power factor correction function, it is not necessary to provide an active filter.

As described above, the switching power supply circuit of the present invention can omit an active filter as a switching power supply circuit having a power factor correction function. By omitting the active filter, the power conversion efficiency characteristic of the switching power supply circuit is improved. This leads to omission and reduction of, for example, a heat sink. In addition, the number of parts is greatly reduced as compared with a configuration including an active filter, so that the circuit can be reduced in size and weight and the cost can be reduced.
Further, the active filter has a hard switching operation, whereas the switching converter of the present invention is based on a voltage resonance type converter and thus has a soft switching operation. Depending on this, since switching noise is greatly reduced, it is not necessary to reinforce the noise filter, which contributes to the reduction in size and weight and cost.
Furthermore, according to the configuration of the present invention, since different switching frequencies do not operate simultaneously, the problem of interference between the ground potential on the primary side and the secondary side is also eliminated, so that the ground potential is stabilized. As a result, reliability is improved and pattern design of the circuit board is facilitated.

Prior to describing the best mode for carrying out the present invention (hereinafter referred to as an embodiment), a switching converter (hereinafter referred to as a class E) that performs switching operation by a class E resonance type, which is the background art of the present embodiment. A basic configuration of the switching converter is also described with reference to FIGS.
FIG. 11 shows a basic configuration as a class E switching converter. The class E switching converter shown in this figure employs a configuration as a DC-AC inverter that operates in a class E resonance type.
The class E switching converter shown in this figure includes one switching element Q1. In this case, the switching element Q1 is a MOS-FET. In the switching element Q1 as the MOS-FET, a body diode DD is formed so as to be connected in parallel between the drain and the source. The forward direction of the body diode DD in this case is along the direction from the source to the drain.
Similarly, a primary side parallel resonant capacitor Cr is connected in parallel between the drain and source of the switching element Q1.

  The drain of the switching element Q1 is connected to the positive electrode of the DC input voltage Ein through a series connection of the choke coil L10. The source of the switching element Q1 is connected to the negative electrode of the DC input voltage Ein.

  Further, one end of the choke coil L11 is connected to the drain of the switching element Q1, and a series resonant capacitor C11 is connected in series to the other end. An impedance Z serving as a load is inserted between the series resonant capacitor C11 and the negative electrode of the DC input voltage Ein. Specific examples of the impedance Z here include a piezoelectric transformer and a high-frequency fluorescent lamp.

  The class E switching converter having such a configuration is formed by a parallel resonance circuit formed by the inductance of the choke coil L10 and the capacitance of the primary side parallel resonance capacitor Cr, an inductance of the choke coil L11, and a capacitance of the series resonance capacitor C11. It can be considered that this is a form of a composite resonance type converter including a series resonance circuit. In addition, it can be said that it is the same as a single-ended voltage resonant converter in that it is formed with only one switching element.

FIG. 12 shows the operation of the main part of the class E switching converter having the configuration shown in FIG.
The switching voltage V1 is a voltage obtained at both ends of the switching element Q1, and has a waveform that is a 0 level in the period TON in which the switching element Q1 is on and becomes a sine wave pulse in the period TOFF in which the switching element Q1 is off. This switching pulse waveform is obtained by the resonance operation (voltage resonance operation) of the parallel resonance circuit.

The switching current IQ1 is a current that flows through the switching element Q1 (and the body diode DD), and is 0 level in the period TOFF, and in the period TON, first, it flows negatively by flowing through the body diode DD for a certain period from the start time. Thereafter, it is inverted to become positive polarity, and flows from the drain to the source of the switching element Q1.
As the output of the class E switching converter, the current I2 that flows through the series resonant circuit includes the switching current IQ1 that flows through the switching element Q1 (and the body diode DD) and the current that flows through the primary side parallel resonant capacitor Cr. It becomes a composite and becomes a waveform including a sine wave component.

  Further, depending on the relationship between the switching current IQ1 and the switching voltage V1, it is also shown that the ZVS operation is obtained at the turn-off timing of the switching element Q1, and the ZVS and ZCS operations are obtained at the turn-on timing.

  The current I1 flowing into the class E switching converter from the positive terminal of the DC input voltage Ein through the choke coil L10 sets the relationship of L10> L11 for the inductances of the choke coils L10 and L11. As shown in the figure, a pulsating flow waveform having a predetermined average level is obtained. Such a pulsating waveform can be viewed as an approximate direct current.

In this embodiment, a class E switching converter based on the above basic configuration is applied to a power supply circuit. First, a configuration example of the power supply circuit according to the first embodiment is shown in the circuit diagram of FIG.
In the switching power supply circuit shown in this figure, first, a pair of common mode choke coils CMC and two across capacitors CL are inserted into the commercial AC power supply AC line as shown. The common mode choke coil CMC and the across capacitors CL and CL form a noise filter that removes common mode noise superimposed on the line of the commercial AC power supply AC.

The commercial AC power supply AC (AC input voltage VAC) is rectified by a bridge rectifier circuit Di formed by bridge-connecting four low-speed rectifier elements (diodes), and the rectified output is charged to the smoothing capacitor Ci. As a result, the rectified and smoothed voltage Ei is obtained as the voltage across the smoothing capacitor Ci. In this case, the rectified and smoothed voltage Ei is at a level corresponding to an equal magnification of the AC input voltage VAC. This rectified and smoothed voltage Ei becomes a DC input voltage for the subsequent switching converter.
However, in the present embodiment, the power factor correction circuit 10 is provided in a line between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. The configuration and operation of the power factor correction circuit 10 will be described later.

In this figure, the switching converter that performs the switching operation by inputting the rectified and smoothed voltage Ei as a DC input voltage is formed as a class E switching converter based on the basic configuration of FIG.
In this case, a high breakdown voltage MOS-FET is selected as the switching element Q1. In this case, the driving method of the class E switching converter is a separately excited type in which the switching element is switched by the oscillation / drive circuit 2.

In this case, the drain of the switching element Q1 is connected via a series connection of a choke coil winding N10 which is the main winding in the choke coil PCC1 and a power factor improving choke coil winding N11 wound around the same choke coil PCC1. Connected to the positive terminal of the smoothing capacitor Ci. Accordingly, the DC input voltage (Ei) in this case is supplied from the power factor improving choke coil winding N11 to the switching element Q1 through the series connection of the choke coil winding N10. The source of the switching element Q1 is connected to the primary side ground.
A switching drive signal (voltage) output from the oscillation / drive circuit 2 is applied to the gate of the switching element Q1.

  Since the MOS-FET is selected for the switching element Q1 in this case, a body diode DD is incorporated so as to be connected in parallel with the source and drain as shown in the figure. As the body diode DD, the anode is connected to the source of the switching element Q1, and the cathode is connected to the drain of the switching element Q1. The body diode DD forms a path for flowing a switching current in the reverse direction, which is generated by the on / off operation (switching operation) of the switching element Q1.

A primary side parallel resonant capacitor Cr is connected in parallel between the drain and source of the switching element Q1.
As can be seen from FIG. 11, the primary side parallel resonant capacitor Cr in this circuit configuration is adapted to the switching current flowing through the switching element Q1 by at least its own capacitance and the inductance L10 (leakage inductance) of the choke coil winding N10. A primary side parallel resonant circuit (voltage resonant circuit) is formed. However, in this configuration, the primary side parallel resonance circuit is formed by inserting the power factor improving choke coil winding N11 between the choke coil winding N10 and the positive terminal of the smoothing capacitor Ci. It can be considered that the inductance Lo includes the inductance Lo of the choke coil winding N11 for power factor improvement.
When the primary side parallel resonant circuit performs a resonance operation, a voltage resonance type operation is obtained as one of the switching operations of the switching element Q1. Accordingly, a sinusoidal resonance pulse waveform is obtained as an end-to-end voltage (drain-source voltage) V1 of the switching element Q1 during the off period.

  In the present embodiment, the choke coil PCC1 is additionally wound with a power factor improving choke coil winding N11 that functions as a power factor improving choke coil. As a result, the magnetic coupling between the choke coil winding N10 and the power factor improving choke coil winding N11 is obtained. In this embodiment, the degree of coupling between the choke coil winding N10 and the power factor improving choke coil winding N11 is set so as to be in a loosely coupled state with a predetermined coupling coefficient of about 0.8 or less.

  A series connection circuit composed of the primary winding N1 of the insulating converter transformer PIT and the primary side series resonance capacitor C11 is connected in parallel between the drain and source of the switching element Q1. In this case, the winding end end of the primary winding N1 is connected to the drain of the switching element Q1, and the winding start end is connected to the primary side series resonance capacitor C11. The pole terminal on the side not connected to the primary winding N1 of the primary side series resonance capacitor C11 is connected to the source of the switching element Q1 at the primary side ground potential.

  The oscillation / drive circuit 2 is a gate voltage for switching the MOS-FET based on the oscillation circuit and the oscillation signal obtained by the oscillation circuit in order to drive the switching element Q1 by, for example, separate excitation. A drive signal is generated and applied to the gate of the switching element Q1. Thereby, the switching element Q1 continuously performs on / off operation according to the drive signal waveform. That is, a switching operation is performed.

The insulating converter transformer PIT transmits the switching output of the primary side switching converter to the secondary side in a state where the primary side and the secondary side are galvanically insulated.
FIG. 2 is a cross-sectional view showing a structural example of an insulating converter transformer PIT included in the power supply circuit of FIG.
As shown in this figure, the insulating converter transformer PIT includes an EE type core (EE-shaped core) in which E-shaped cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other.
And the bobbin B formed with the shape which divided | segmented so that it might mutually become independent about the winding part of a primary side and a secondary side, for example with a resin etc. is provided. The primary winding N1 is wound around one winding portion of the bobbin B. Further, the secondary winding N2 is wound around the other winding portion.
By attaching the bobbin B on which the primary side winding and the secondary side winding are wound in this way to the EE-shaped core (CR1, CR2), the primary side winding and the secondary side winding are different from each other. By the winding area, the center magnetic leg of the EE-shaped core is wound. In this way, the structure of the insulating converter transformer PIT as a whole is obtained.

  In addition, a gap G having a gap length of about 2 mm or more is formed on the central magnetic leg of the EE-shaped core as shown in the figure. As a result, as the coupling coefficient k, for example, a loosely coupled state with about k≈0.7 or less is obtained. That is, as a conventional technique, the state is more loosely coupled than the insulating converter transformer PIT of the power supply circuit shown in FIG. The gap G can be formed by making the central magnetic legs of the E-type cores CR1 and CR2 shorter than the two outer magnetic legs.

  The primary winding N1 of the insulating converter transformer PIT is an element for forming a primary series resonance circuit in a class E switching converter formed on the primary side, as will be described later, and depends on the switching output of the switching element Q1. Alternate output is obtained.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2.
A secondary side series resonance capacitor C2 is connected to the secondary winding N2 by a serial connection relationship. Thus, a secondary side series resonance circuit is formed by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side series resonance capacitor C2. This secondary side series resonance circuit performs a resonance operation in accordance with a rectification operation of a secondary side rectifier circuit described later, and thereby the secondary winding current flowing in the secondary winding N2 becomes a sine wave. That is, a current resonance operation is obtained on the secondary side.

  The secondary side rectifier circuit in this case has two rectifier diodes Do1 and Do2 and one smoothing with respect to the secondary winding N2 to which the secondary side series resonant capacitor C2 is connected in series as described above. By connecting the capacitor Co, a double voltage half-wave rectifier circuit is formed. As a connection mode of the voltage doubler half-wave rectifier circuit, first, the anode of the rectifier diode Do1 and the rectifier diode Do2 are connected to the winding end end side of the secondary winding N2 via the secondary side series resonant capacitor C2. Connect the cathode. Further, the cathode of the rectifier diode Do1 is connected to the positive terminal of the smoothing capacitor Co. The winding start end of the secondary winding N2, the anode of the rectifier diode Do2, and the negative terminal of the smoothing capacitor Co are connected to the secondary side ground.

The rectification operation of the voltage doubler half-wave rectifier circuit thus formed is as follows.
First, in a half cycle corresponding to one polarity of the alternating voltage obtained in the secondary winding, a forward voltage is applied to the rectifier diode Do2, so that the rectifier diode Do2 conducts and the rectified current is An operation of charging the secondary side series resonant capacitor C2 is obtained. As a result, a voltage across the level corresponding to the same multiple of the alternating voltage level induced in the secondary winding N2 is generated in the secondary side series resonance capacitor C2. In the next half cycle corresponding to the other polarity of the alternating voltage obtained in the secondary winding, a forward voltage is applied to the rectifier diode Do2 to conduct. At this time, the smoothing capacitor Co is charged with a potential obtained by superimposing the potential of the secondary winding voltage V1 and the voltage across the secondary side series resonance capacitor C2.
As a result, the secondary side DC output voltage Eo having a level corresponding to twice the alternating voltage level induced in the secondary winding N2 is obtained as the voltage across the smoothing capacitor Co. In this rectification operation, the smoothing capacitor Co is charged only in one half cycle of the alternating voltage excited by the secondary winding N2. That is, a rectification operation as a double voltage half wave is obtained.
The secondary side DC output voltage Eo is supplied to the load. Further, it branches and is output as a detection voltage to the control circuit 1.

  The control circuit 1 supplies a detection output corresponding to the level change of the input secondary side DC output voltage Eo to the oscillation / drive circuit 2. In the oscillation / drive circuit 2, the switching element Q 1 is driven mainly by changing the switching frequency according to the input detection output of the control circuit 1.

  As described above, the switching frequency of the switching element Q1 is variably controlled, whereby the primary and secondary resonance impedances in the power supply circuit change, and the primary winding N1 to the secondary winding N2 from the primary winding N1 of the insulating converter transformer PIT change. The amount of power transmitted to the side and the amount of power to be supplied to the load from the secondary side rectifier circuit will change. As a result, an operation for controlling the level of the secondary side DC output voltage Eo is obtained such that the level fluctuation of the secondary side DC output voltage Eo is canceled. That is, the secondary side DC output voltage Eo is stabilized.

The switching converter (Q1, Cr, N10-N11, N1, C11) formed on the primary side of the power supply circuit of the present embodiment formed as described above, and the class E converter previously shown in FIG. The switching converter of the present embodiment omits the impedance Z as a load from the circuit of FIG. 10, and the choke coil L11 is replaced with the primary winding N1 (leakage inductance) of the insulating converter transformer PIT. It can be seen as a replacement for L1). As described above, in the primary side switching converter of the present embodiment, the primary side parallel resonant circuit is at least based on the inductance L10 of the choke coil winding N10 (first inductor) and the capacitance of the primary side parallel resonant capacitor Cr. The primary side series resonance circuit is formed by the leakage inductance L1 of the primary winding N1 (second inductor) of the insulating converter transformer PIT and the capacitance of the primary side series resonance capacitor C11.
From this, it can be said that the primary side switching converter of this embodiment is formed as a class E switching converter that performs a class E resonance type switching operation. Then, the switching output (alternate output) obtained by the switching operation of the primary side switching converter is passed through the magnetic coupling in the insulating converter transformer PIT, and the primary winding N1 corresponding to the choke coil L11 to the secondary winding N2. And rectified on the secondary side to obtain a DC output voltage Eo.
Further, the class E switching converter formed as in the present embodiment is a switching element Q1 (and body diode DD) that forms a voltage resonance type converter together with the choke coil winding N10 and the primary side parallel resonance capacitor Cr. On the other hand, it can be considered that the configuration is a composite resonance type converter and a soft switching power source in which a series connection circuit of a primary winding N1 and a primary side series resonance capacitor C11 forming a primary side series resonance circuit are connected in parallel.

Next, the power factor correction circuit 10 will be described.
The power factor improving circuit 10 is provided so as to be inserted into a rectified current path in a rectifying and smoothing circuit for obtaining a DC input voltage (Ei) from a commercial AC power supply AC. Use a structure for improvement.

  The power factor correction circuit 10 of the present embodiment includes, for example, as shown in the figure, a power factor improving choke coil winding N11 (corresponding to a winding as a power factor improving inductor), a switching diode D1 (for power factor improving). Switching element) and a filter capacitor CN. Among these, the power factor improving choke coil winding N11 is wound around the choke coil PCC1 by a predetermined number of turns so as to obtain a predetermined inductance Lo.

A structural example of the choke coil PCC1 in this case is shown in a sectional view of FIG.
As shown in this figure, the choke coil PCC1 also includes an EE type core (EE-shaped core) in which E-shaped cores CR11 and CR12 made of a ferrite material are combined so that their magnetic legs face each other. A bobbin B1 formed of, for example, a resin or the like in a shape in which the primary side and secondary side winding portions are divided independently from each other together with the wire N10 and the power factor improving choke coil winding N11 is provided. It is done. A choke coil winding N10 is wound around one winding portion of the bobbin B1, and a power factor improving choke coil winding N11 is wound around the other winding portion. In this case, the bobbin B1 in this case seems to have a larger volume in the winding portion of the choke coil winding N10 than in the winding portion of the choke coil winding N11 for power factor improvement, as shown in the figure. Formed. In this case, litz wires are used for the choke coil winding N10 and the power factor improving choke coil winding N11, respectively.
By attaching the bobbin B1 on which the primary side winding and the secondary side winding are wound in this way to the EE-shaped core (CR1, CR2), the primary side winding and the secondary side winding are EE-shaped. It will be in the state wound by the central magnetic leg of a core.
For the central magnetic leg of the EE-shaped core, a gap G of about 1.6 mm is formed as shown in the figure, so that the choke coil winding N10 and the power factor improving choke coil winding N11 are arranged. For example, a loosely coupled magnetic coupling state with a coupling coefficient k≈0.7 is obtained.

  In addition, as the power factor improving circuit 10, the anode of the switching diode D1 is connected to the positive output terminal of the bridge rectifier circuit Di, and the cathode of the switching diode D1 is the winding end of the choke coil winding N11 for power factor improvement. Connect to the part side. The winding start end of the power factor improving choke coil winding N11 is connected to a connection point with the positive terminal of the smoothing capacitor Ci. Further, the winding end of the choke coil winding N10 is connected to the connection point between the cathode of the switching diode D1 and the winding end of the power factor improving choke coil winding N11. The winding start end of the choke coil winding N10 is connected to the drain of the switching element Q1.

In such a connection form as the power factor correction circuit 10, in the path of the rectified output current between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci, the switching diode D1 (anode → cathode) -force It can be considered that a series connection circuit of the choke coil winding N11 for rate improvement is inserted.
A filter capacitor CN is connected in parallel to the series connection circuit of the switching diode D1 (anode → cathode) and the choke coil winding N11 for power factor improvement. The filter capacitor CN is provided to suppress normal mode noise.

The operation of the power factor correction circuit 10 formed in this way is as follows.
The primary side switching converter in the power supply circuit of FIG. 1 is a class E switching converter, and therefore includes a primary side parallel resonant circuit including at least a primary side parallel resonant capacitor Cr and a choke coil winding N10. For this reason, during the period in which the switching element Q1 is turned off, a voltage resonance operation in which a charge / discharge current flows through the primary side parallel resonance capacitor Cr is obtained. As a result, a sinusoidal pulse voltage (resonance pulse voltage) is generated as the voltage V1 serving as the voltage across the primary side parallel resonance capacitor Cr (switching element Q1) during the period when the switching element Q1 is turned off. A component of an alternating voltage waveform corresponding to the resonance pulse voltage is transmitted to the choke coil winding N10 forming the primary side parallel resonance circuit. In the choke coil PCC1, the power factor improving choke coil winding N11 is provided so as to be magnetically coupled to the choke coil winding N10 forming the primary side parallel resonant circuit, as described above. The resonance pulse voltage generated in the choke coil winding N10 is induced in the power factor improving choke coil winding N11. That is, an alternating voltage of the switching period is induced in the power factor improving choke coil winding N11.
The power factor improving choke coil winding N11 in this case is inserted in the path of the rectified output current between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. The induced voltage obtained in the choke coil winding N11 is superimposed on the rectified current path. In such an operation, the current and voltage as the switching output obtained in the primary winding N1 are converted to voltage by the power factor improving choke coil winding N11 and fed back to the smoothing capacitor Ci. You can see. That is, an operation as a voltage feedback system is obtained in which the switching output is fed back to the smoothing capacitor Ci (rectified current path) through the magnetic coupling between the choke coil winding N10 and the power factor improving choke coil winding N11. It has been.
Further, in the power factor correction circuit 10 in this case, the end of winding end of the choke coil winding N10 is at the connection point between the anode of the switching diode D1 and the end of winding end of the choke coil winding N11 for power factor improvement. Are connected to each other. In addition, the winding start end portion of the power factor improving choke coil winding N11 is also connected to the winding end end portion of the primary winding N1. Therefore, in the power factor correction circuit 10, the switching output of the switching element Q1 In response to this, the current obtained in the primary winding N1 is regenerated as electric power, and is fed back from the choke coil winding N10 to the smoothing capacitor Ci via the magnetic coupling of the power factor improving choke coil winding N11. It can be seen that That is, operation as a magnetic coupling type power regeneration system is also obtained.

In this way, in this embodiment, the feedback operation of the switching output as the two feedback methods, that is, the magnetic coupling type voltage feedback method and the same magnetic coupling type power regeneration method is generated simultaneously. Yes. In this way, the switching output is fed back to the rectified current path, whereby an alternating voltage corresponding to the switching frequency is superimposed on the rectified current path.
As a result, the voltage applied to the switching diode D1 is a potential obtained by further superimposing an alternating voltage component based on the switching period on the rectified and smoothed voltage Ei. In the fast recovery type switching diode D1, the positive / negative absolute value of the AC input voltage VAC is, for example, about 1/2 or more of the peak value by applying the alternating voltage component superimposed on the rectified and smoothed voltage Ei. At this time, the switching operation is performed to interrupt the rectified current that is going to flow into the smoothing capacitor Ci.
The conduction period of the envelope of the rectified current flowing as described above also flows during a period in which the rectified output voltage level output from the bridge rectifier circuit Di is lower than the voltage across the smoothing capacitor Ci. Then, the conduction period of the AC input current IAC that flows based on the AC input voltage VAC is also substantially coincident with the conduction period of the rectified current. That is, the conduction angle of the AC input current IAC is larger than that without the power factor correction circuit, and the waveform of the AC input current IAC approaches the waveform of the AC input voltage VAC. Yes. That is, the power factor is improved.

Further, in the configuration of the power factor correction circuit 10 of the present embodiment, the power factor value improved by the power factor correction circuit 10 can be changed by setting the number of turns of the power factor correction choke coil winding N11. .
As described above, the power factor improving choke coil winding N11 is magnetically coupled to the choke coil winding N10 in the choke coil PCC1, and an alternating voltage is induced by the choke coil winding N10. Therefore, by changing the number of turns of the power factor improving choke coil winding N11, the winding ratio between the choke coil winding N10 and the power factor improving choke coil winding N11 is changed. The voltage level induced in the choke coil winding N11 will change. The change in the induced voltage level of the power factor improving choke coil winding N11, that is, the change in the feedback amount of the switching output.

In the present embodiment, the winding end ends of the choke coil winding N10 and the power factor improving choke coil winding N11 are connected to each other so that a relationship of opposite polarities can be obtained. As a result, the power factor value to be improved is increased. That is, the power factor is efficiently improved.
The power factor correction circuit 10 includes an operation as a power regeneration system in which the current of the switching period obtained from the primary winding N1 to the choke coil winding N10 is regenerated as power and returned to the rectification current path. In FIG. 4, the magnetic coupling of the power factor improving choke coil winding N11 is used. At this time, due to the action of the inductance of the power factor improving choke coil winding N11, the alternating voltage component assumed to be superimposed on the rectified current path by the feedback operation as the power regeneration system includes the alternating voltage obtained in the choke coil winding N10. It will have a certain amount of phase difference with respect to the voltage. For this reason, if the alternating voltage induced in the choke coil winding N11 for power factor improvement is in phase with the choke coil winding N10, the alternating voltage component superimposed on the rectified current path is the power factor improvement. The alternating voltage induced by the choke coil winding N11 and the alternating voltage fed back by the power regeneration operation are close to opposite phases and cancel each other out. This causes a reduction in the feedback amount of the switching output. Therefore, in the present embodiment, the power factor improving choke coil winding N11 is magnetically coupled with the polarity opposite to that of the choke coil winding N10 so that the voltage of the choke coil winding N10 is reduced. An induced voltage at a timing that is almost reversed is generated in the choke coil winding N11 for power factor improvement. That is, phase compensation is performed. As a result, the action of canceling out the alternating voltage superimposed by the voltage feedback and the alternating voltage superimposed by the power regeneration is eliminated and suppressed, so that a high power factor can be obtained.

As the actual configuration of the power supply circuit of FIG. 1 having the above-described configuration, the main part was selected as follows in order to obtain the experimental results described later.
First, for the insulating converter transformer PIT, EER-35 was selected for the EE-shaped cores (CR1, CR2) based on the structure shown in FIG. 2, and a gap length of 2.2 mm was set for the gap G1. For the number of turns (number of turns) T of the primary winding N1 and the secondary winding N2, N1 = 60T and N2 = 30T were selected. Due to the structure of the insulating converter transformer PIT described above, for the coupling coefficient k between the primary side and the secondary side in the insulating converter transformer PIT itself, a value that is smaller than 0.7, for example, k = 0.67 is set.
For choke coil PCC1, EER-28 was selected under the structure shown in FIG. 3, and the gap length of gap G was set to 1.6 mm. In addition, L10 = 1 mH was set for the inductance L10 of the choke coil winding N10, and Lo = 14 μH was set for the inductance Lo of the choke coil winding N10 for power factor improvement.
As is well known, the core of the EER is one of the types and standards of the core as a product. It is also known that this type includes ER and EE. In the present application, in the case of E-shape, EE-shape, etc., the E-shape or EE-shape for any type of EER, ER, EE depending on whether the cross section is E-shape or EE-shape. Shall be treated as the core of
Further, regarding the respective capacitances of the primary side parallel resonance capacitor Cr, the filter capacitor CN, and the secondary side series resonance capacitor C2,
Cr = 5600pF
C11 = 0.022μF
CN = 1μF
C2 = 0.082μF
Was selected.
The corresponding load power is the maximum load power Pomax = 300 W, the minimum load power Pomin = 0 W (no load), and the rated level of the secondary side DC output voltage Eo is 175V.

As an experimental result of the power supply circuit of FIG. 1, the waveform diagram of FIG. 4 is given. FIG. 4 shows the operation of the main part of the power supply circuit of FIG.
First, as shown in the figure, the current I1 flowing through the switching diode D1 of the power factor correction circuit 10 is a smooth convex positive-polarity pulse in a period of a predetermined level or more near the peak of the AC input voltage VAC. As an alternating waveform.
The voltage V2, which is the potential between the connection point between the cathode of the switching diode D1 and the end of the choke coil winding N11 for power factor correction, and the primary side ground has a positive polarity peak at the rectified smoothing voltage Ei (140V). ), And in a period other than when the AC input voltage VAC is at a predetermined level or higher, an alternating waveform in which the DC component increases or decreases in a sine wave form is obtained.
The voltage V3, which is the potential between the positive output terminal of the bridge rectifier circuit Di and the primary side ground, is a rectified output voltage during a period in which the AC input voltage VAC is higher than a predetermined level with respect to the level of the rectified smoothing voltage Ei. The alternating voltage component of the waveform corresponding to is superimposed.
In this case, the AC input current IAC has a waveform in which the substantially convex envelope is inverted according to the polarity of the AC input voltage VAC without forming a non-conduction period as shown in the figure. Such a waveform is closer to a sine wave, for example, compared to a configuration in which the power factor improvement circuit 10 is omitted from the configuration shown in FIG. 1, and the power factor is improved accordingly. It can be seen as a thing.

  Further, in FIG. 4, the secondary side DC output voltage Eo is shown. The secondary side DC output voltage Eo is averaged at the rated level (175 V) of the stabilization target, and a ripple having a period substantially corresponding to the switching frequency is superimposed by the illustrated envelope. In this case, the ripple voltage level is 80 mVp-p.

FIG. 5 shows an experimental result of the power supply circuit shown in FIG. 1 in the range of maximum load power Pomax = 300 W to minimum load power Pomin = 0 W (no load) under the input voltage condition of AC input voltage VAC = 100V. A rectified smoothing voltage (DC input voltage) Ei, power factor (PF), and AC → DC power conversion efficiency (ηAC → DC) with respect to load fluctuations are shown.
FIG. 6 shows the experimental results for the power supply circuit shown in FIG. 1, with rectification and smoothing with respect to fluctuations in the range of AC input voltage VAC = 85 V to 144 V under a constant load condition with maximum load power Pomax = 300 W. Voltage (DC input voltage) Ei, power factor (PF), and AC → DC power conversion efficiency (ηAC → DC) are shown.

First, the rectified and smoothed voltage Ei, which is the DC input voltage of the switching converter, changes substantially proportionally in accordance with the AC input voltage VAC as shown in FIG. Further, as shown in FIG. 5, the load fluctuation tends to gradually increase as the load becomes light, but the load power Po = 100 W or more is about 130 V to 140. It is a characteristic that can be considered to be almost within the range. The fluctuation range (ΔEi) of the rectified smoothing voltage Ei with respect to the range of the maximum load power Po = 300 W to the minimum load power Pomin = 0 W was 24V.
Further, the power factor PF obtained in accordance with the operation of the power factor correction circuit 10 is a characteristic that tends to decrease as the light load tendency as a whole as shown in FIG. The load power range Pomax = 300 W to Po = 50 W is maintained at PF = 0.75 or more with respect to the load fluctuation range, and it can be said that a practically sufficient power factor value is obtained. As shown in FIG. 6, the fluctuation of the AC input voltage VAC decreases with a gradual slope as the AC input voltage VAC increases, but the characteristic may be regarded as almost constant. Become.

Further, as shown in FIGS. 5 and 6, the AC → DC power conversion efficiency (ηAC → DC) has a peak at around Po = 100 W with respect to load fluctuation, but it may be regarded as almost constant. Thus, the fluctuation of the AC input voltage has a characteristic that is substantially constant with respect to the fluctuation of the AC input voltage VAC. As a measurement result when the maximum load power Po = 300 W and the AC input voltage VAC = 100 V, a result of ηAC → DC = 91.8% was obtained. In addition, a good result was obtained that ηAC → DC = 90% or more was maintained in the range of maximum load power Pomax = 300 W? Po = 25 W.
As a comparison, in the power supply circuit shown in FIG. 16, the AC → DC power conversion efficiency at the same maximum load power Pomax = 300 W and AC input voltage VAC = 100 is 83%, which is an improvement of about 8.8%. It is illustrated. Accordingly, the AC input power is reduced by about 34.7 W in the present embodiment as compared with the power supply circuit of FIG.

When the power supply circuit of the embodiment described so far is compared with the power supply circuit shown in FIG. 16 which is a prior art for improving the power factor by providing an active filter, the following is possible. I can say that.
First, as can be seen from the description of the experiment shown in FIG. 5 and FIG. 6, the power conversion efficiency (ηAC → DC) is improved in the power supply circuit shown in FIG. Yes.
This is mainly because the active filter is not required because the power factor correction circuit using the voltage feedback method is provided. That is, in this embodiment, the total efficiency is not reduced by the product of the two power conversion efficiency values of the former stage and the latter stage as in the case where the active filter is provided.

Further, in the circuit shown in FIG. 1, the active filter is not required, so that the number of circuit components can be reduced.
In other words, the active filter constitutes a set of converters, and as can be seen from the description with reference to FIG. 16, in practice, there are many switching elements and ICs for driving them, and so on. Consists of the number of parts.
On the other hand, the power supply circuit shown in FIG. 1 includes at least a filter capacitor CN, a switching diode D1, and a power factor improving choke coil winding N11 (inductance Lo) as additional components necessary for improving the power factor. What is necessary is that the number of parts can be reduced as compared with the active filter.
Thereby, the power supply circuit shown in FIG. 1 can be manufactured at a much lower cost than the circuit shown in FIG. 16 as a power supply circuit having a power factor correction function. Further, since the number of parts is greatly reduced, the circuit board can be effectively reduced in size and weight. In particular, the inductor component such as the power factor improving choke coil is a large class, but in this embodiment, the rectified current flowing into the smoothing capacitor Ci is an alternating current due to the switching period. The inductance Lo that can be set to the power factor improving choke coil winding N11 that functions as an inductor for current is small, and actually, Lo = 14 μH. Thus, for example, when the choke coil PCC1 is formed by winding the choke coil N11 for power factor improvement as in the present embodiment, the core of the choke coil PCC1 can be suppressed to EE-28. Therefore, it does not prevent the circuit board from being reduced in size and weight. For comparison, for example, the power choke coil PCC shown in FIG. 16 requires about 140 μH, so EER-39 is selected.

In the power supply circuit shown in FIG. 1, the operation of the class E resonant converter and the power factor correction circuit 10 is a so-called soft switching operation. Therefore, the level of switching noise is significantly higher than that of the active filter shown in FIG. Reduced.
Therefore, as shown in FIG. 1, for example, if a single-stage noise filter comprising a pair of common mode choke coils CMC and two across capacitors CL is provided, it is possible to sufficiently satisfy the power disturbance standard. It is said. Further, as shown in FIG. 1, the normal mode noise of the rectified output line is taken by only one filter capacitor CN.
By reducing the number of components as a noise filter in this way, the cost reduction of the power supply circuit and the reduction in size and weight of the circuit board are promoted.

  In the case of the power supply circuit shown in FIG. 1, the number of switching elements forming the primary side switching converter is one stone, and the secondary side rectifier diode and the like perform switching operations in synchronization with the switching element Q1. Therefore, the primary side ground potential can be stabilized regardless of the change of the switching frequency without causing interference between the active filter side and the subsequent switching converter as in the power supply circuit of FIG. .

Note that the power factor PF obtained by the power supply circuit shown in FIG. 1 is as described in FIGS. 5 and 6. According to such power factor characteristics, for example, power supply harmonic distortion regulation is cleared. It can be said that a practically sufficient power factor is obtained.
In this way, the power supply circuit of the present embodiment shown in FIG. 1 obtains a power factor improving power supply that solves various problems of a power supply circuit including an active filter.

  Further, the power supply circuit according to the present embodiment includes a class E switching converter as the primary side switching converter. As described above, in the class E switching converter, the primary side parallel resonant circuit (Cr, N10 (L10)) and the primary side series resonant circuit (C11, N1 (L1)) are combined. Such a configuration can be viewed as a combination of a primary side series resonant circuit (C11, N1 (L1)) with a voltage resonant converter as a basic configuration.

In general, a power supply circuit having a voltage resonance type converter on the primary side has a narrow control range of load power and cannot maintain ZVS at light load, so it is considered impossible to put it to practical use as it is. ing. Therefore, a secondary side series resonant circuit is provided for the single-sided primary voltage resonant converter with a single switching element, and a power supply circuit (in this case, a double voltage half-wave rectifier circuit is formed as the secondary side rectifier circuit) The inventor of the present application conducted an experiment by configuring an experimental power supply circuit), and it was confirmed that characteristics closer to realization can be obtained than a power supply circuit having a voltage resonant converter so far.
However, in the above experimental power supply circuit, the current flows in the positive direction (in this case, the drain → source direction) through the switching element Q1 before the OFF period (TOFF) of the switching element Q1 is not completed at the intermediate load, and the operation of the ZVS An abnormal operation occurs that cannot be obtained. For this reason, even with the configuration of the experimental power supply circuit, it was still difficult to put into practical use.
On the other hand, in the power supply circuit of the embodiment shown in FIG. 1, it has been experimentally confirmed that the ZVS operation can be obtained over the entire range of the corresponding load power. That is, the abnormal operation at the intermediate load is eliminated. As a result, a single-ended voltage resonant converter including a secondary side series resonant circuit can be easily put into practical use.

It has been confirmed that abnormal operation at the time of intermediate load of the experimental power supply circuit is likely to occur when a complex resonance type converter having a secondary resonance circuit in the voltage resonance type converter is configured. This is mainly caused by the interaction between the primary side parallel resonant circuit forming the voltage resonant converter and the secondary side series resonant circuit (rectifier circuit) operating simultaneously.
Therefore, in the case of the present embodiment, the abnormal operation at the time of the intermediate load described above is due to the fact that the circuit configuration is a combination of the primary side voltage resonance type converter and the secondary side series resonance circuit. First, the primary side switching converter has a configuration in which a class E switching converter is applied instead of the voltage resonance type converter.
Further, the coupling coefficient k is set to about 0.65, and the degree of coupling between the primary side and the secondary side of the insulating converter transformer PIT is made lower than before. As a result, the interaction between the operation of the primary side switching converter and the rectification operation (switching operation) of the secondary side rectifier circuit is dilute to obtain an effect of suppressing abnormal operation at the intermediate load.

In the experimental power supply circuit, the current flowing into the switching converter from the smoothing capacitor Ci that generates the DC input voltage passes through the primary winding of the insulating converter transformer to the parallel connection circuit side of the switching element and the primary parallel resonant capacitor. Inflow. In this case, the current flowing from the smoothing capacitor Ci into the switching converter has a relatively high frequency due to the switching period. That is, a charging / discharging current flows to the smoothing capacitor Ci at a high frequency with respect to the commercial AC power supply cycle.
An aluminum electrolytic capacitor is often employed depending on the fact that a high breakdown voltage is required for the component element as the smoothing capacitor Ci. Compared to other types of capacitors and the like, aluminum electrolytic capacitors have the property that when operated at a high frequency, the capacitance decreases and the loss angle tangent tends to increase. For this reason, it is necessary to select a special product having a low ESR (equivalent series resistance) and a large allowable ripple current as the aluminum electrolytic capacitor used for the smoothing capacitor Ci. It is also necessary to select a correspondingly large value for the capacitance of the element as the smoothing capacitor Ci. For example, in the configuration of the experimental power supply circuit (primary side voltage resonance type converter), in order to correspond to the maximum load power Pomax = 300 W equivalent to the present embodiment, about 1000 μF is selected. Such an aluminum electrolytic capacitor is more expensive than a general-purpose aluminum electrolytic capacitor, and is disadvantageous in terms of cost, including an increase in component price in accordance with an increase in capacitance.

  On the other hand, in the power supply circuit of the present embodiment shown in FIG. 1, the current flowing from the smoothing capacitor Ci to the switching converter is switched through the choke coil winding N10 to the switching element Q1, the primary side parallel resonant capacitor Cr, the primary It flows through the side series resonance circuit (N1-C11). For this reason, the current flowing from the smoothing capacitor Ci into the switching converter becomes a direct current (pulsating flow) in the same manner as the input current I1 in FIG. In this way, since the current flowing from the smoothing capacitor Ci into the switching converter becomes a direct current, in the present embodiment, the above-described problems of a decrease in capacitance and an increase in tangent of the loss angle do not occur. Therefore, a general-purpose aluminum electrolytic capacitor can be selected as the smoothing capacitor Ci. Also, the capacitance of the element as the smoothing capacitor Ci can be selected to be lower than that of the experimental power supply circuit. As an actual circuit of FIG. 1, about 680 μF can be selected. In this way, in the present embodiment, it is possible to reduce the cost of the smoothing capacitor Ci.

FIG. 7 shows a configuration example of a power supply circuit according to the second embodiment. In this figure, the same parts as those in FIG.
In the power supply circuit shown in this figure, a power factor correction circuit 11 is provided. In the power factor correction circuit 11, a low-speed rectifier diode D1A is added to the configuration of the power factor correction circuit 10 previously shown in FIG. 1 as the first embodiment.
The rectifier diode D1A has an anode connected to the positive output terminal of the bridge rectifier circuit Di and a cathode connected to the positive terminal of the smoothing capacitor Ci. Therefore, in the power factor correction circuit 11, the diode D1A is connected in parallel to the series connection circuit of the switching diode D1 and the power factor correction choke coil winding N11. In this case, the filter capacitor CN is connected to the series connection circuit of the switching diode D1 and the power factor improving choke coil winding N11 and the rectifier diode D1A in parallel.

In the power supply circuit of the second embodiment including the power factor correction circuit 11 formed in this way, the rectified current obtained as the rectified output of the positive output terminal of the bridge rectifier circuit Di is the switching diode D1—power factor improvement. The choke coil winding N11 is connected in series to the series connection circuit and the rectifier diode D1A. Even in such an operation, the basic operation of the power factor correction circuit 11 is the same as that of the power factor correction circuit 10, and the resonance pulse voltage and current fed back by voltage feedback and power regeneration are used. The switching diode D1 switches the rectified current, and the power factor is improved by expanding the conduction angle of the AC input current IAC.
In addition, as described above, the rectified current from the bridge rectifier circuit Di also branches and flows to the rectifier diode D1A, so that the amount of rectified current on the side flowing to the switching diode D1 is reduced. Thereby, the switching loss in the switching diode D1 is reduced and the power conversion efficiency is improved. In particular, this effect becomes conspicuous as the current flowing through the power supply circuit increases as the load becomes heavy.

FIG. 8 shows a configuration example of a power supply circuit according to the third embodiment. In this figure, the same parts as those in FIG. 1 and FIG.
In this figure, the choke coil PCC1 has a configuration in which only the choke coil winding N10 is wound. The power factor improving choke coil winding N11 included in the choke coil PCC in each of the previous embodiments corresponds to the winding of the power factor improving choke coil Lo in the power factor improving circuit 12 here. The power factor improving choke coil Lo is formed by winding a core having a predetermined shape size with a predetermined number of turns. That is, in the third embodiment, a component element as a choke coil PCC that is an inductance that originally forms a primary side parallel resonance circuit and a power factor improving choke coil Lo that is an inductor for improving the power factor are used. These component elements are formed as separate components.
In addition, the power factor improvement circuit 12 in this case is configured to improve the power factor correction choke coil Lo (power factor improvement choke coil winding N11), the switching diode D1, and the filter capacitor CN, for example, as shown in FIG. It is configured to be connected in the same connection manner as the circuit 10.
Even in such a configuration, the power factor improving choke coil Lo and the choke coil PCC can obtain a loosely coupled state with a predetermined coupling coefficient. Therefore, similarly to the power factor correction circuits of the embodiments described so far, the switching output feedback operation by the voltage feedback method and the power regeneration method can be obtained, and the power factor can be improved.

Subsequently, as variations common to the first and second embodiments, variations of the secondary side rectifier circuit are shown in FIGS.
9 and 10, only the configuration of the secondary winding N2 and the secondary side rectifier circuit are extracted and shown, but other parts not shown are the parts of the insulating converter transformer PIT. The configuration as the embodiment described above may be adopted including the structure.

First, in the power supply circuit shown in FIG. 9, four rectifiers are used as the secondary side rectifier circuit connected to the series connection circuit (secondary side series resonant circuit) of the secondary winding N2 and the secondary side series resonant capacitor C2. A bridge rectifier circuit including diodes Do1, Do2, Do3, and Do4 and a bridge full-wave rectifier circuit including one smoothing capacitor Co are provided.
In this case, the end of winding end of the secondary winding N2 is connected to the connection point between the anode of the rectifier diode Do1 and the cathode of the rectifier diode Do2 via the secondary side series resonant capacitor C2. Further, the winding start end of the secondary winding N2 is connected to a connection point between the anode of the rectifier diode Do3 and the cathode of the rectifier diode Do4. The cathode of the rectifier diode Do1 and the cathode of the rectifier diode Do3 are connected to the positive terminal of the smoothing capacitor Co. The negative terminal of the smoothing capacitor Co is connected to the connection point between the anode of the rectifier diode Do2 and the anode of the rectifier diode Do4 at the secondary side ground potential.

Depending on the full-wave rectifier circuit formed as described above, a pair of rectifier diodes [Do1, Do4] of the bridge rectifier circuit is formed in one half cycle of the alternating voltage induced (excited) in the secondary winding N2. An operation of conducting and charging the rectified current to the smoothing capacitor Co is obtained. Further, in the other half cycle of the alternating voltage induced in the secondary winding N2, the operation of charging the rectified current to the smoothing capacitor Co by obtaining a set of rectifier diodes [Do2, Do3] conductive.
As a result, a secondary side DC output voltage Eo having a level corresponding to the same voltage as the alternating voltage level induced in the secondary winding N2 is generated as the voltage across the smoothing capacitor Co.

Further, the power supply circuit shown in FIG. 10 includes a voltage doubler full wave rectifier circuit as a secondary side rectifier circuit.
As a voltage doubler full-wave rectifier circuit in this case, first, a center tap is applied to the secondary winding N2, and the secondary tap is divided into two parts N2A and N2B with the center tap as a boundary. The same predetermined number of turns (number of turns) is set in the secondary winding portions N2A and N2B. The center tap of the secondary winding N2 is connected to the secondary side ground.
Further, a secondary side series resonance capacitor C2A is connected in series to the end of the secondary winding N2 on the secondary winding portion N2A side, and the end of the secondary winding N2 on the secondary winding portion N2B side is connected. The secondary side series resonant capacitor C2B is connected in series to the unit. As a result, the first secondary side series resonance circuit composed of the leakage inductance component of the secondary winding part N2A and the capacitance of the secondary side series resonance capacitor C2A, and the leakage inductance component and secondary side of the secondary winding part N2B A second secondary side series resonance circuit composed of the capacitance of the series resonance capacitor C2B is formed.

The end of the secondary winding N2 on the secondary winding N2A side is connected to the connection point between the anode of the rectifier diode Do1 and the cathode of the rectifier diode Do2 via the series connection of the secondary side series resonant capacitor C2A. Connect. Further, the end of the secondary winding N2 on the secondary winding N2B side is connected to the connection point between the anode of the rectifier diode Do3 and the cathode of the rectifier diode Do4 via the series connection of the secondary side series resonant capacitor C2B. Connect.
The cathodes of the rectifier diodes Do1 and Do3 are connected to the positive terminal of the smoothing capacitor Co. The negative terminal of the smoothing capacitor Co is connected to the secondary side ground. The connection point of each anode of the rectifier diodes Do2 and Do4 is connected to the secondary side ground.

In the above connection configuration, the first voltage doubler half including the first secondary side series resonance circuit including the secondary winding portion N2A, the secondary side series resonance capacitor C2A, the rectifier diodes Do1 and Do2, and the smoothing capacitor Co. A second voltage doubler half comprising a wave rectifier circuit, a secondary winding section N2B, a secondary side series resonant capacitor C2B, a rectifier diode Do1, Do2, and a smoothing capacitor Co. A wave rectifier circuit is formed.
In the first voltage doubler half-wave rectifier circuit, during the half-cycle period of one polarity of the alternating voltage induced in the secondary winding N2, [secondary winding portion N2A → rectifier diode Do2 → secondary series The rectification operation is performed by the rectification current path of the resonance capacitor C2A → secondary winding portion N2A], and the secondary side series resonance capacitor C2A is charged by the potential of the alternating voltage of the secondary winding portion N2A. In the period of the other half cycle, the rectification operation is performed by the rectification current path of [secondary winding portion N2A → secondary side series resonance capacitor C2A → rectifier diode Do1 → smoothing capacitor Co → secondary winding portion N2A]. Thus, the smoothing capacitor Co is charged by the superimposed potential of the both-ends voltage of the secondary side series resonance capacitor C2A and the alternating voltage of the secondary winding N2A.
Further, the second voltage doubler half-wave rectifier circuit [secondary winding portion N2B → rectifier diode Do4 → second] in the half cycle period of the other polarity of the alternating voltage induced in the secondary winding N2. The secondary series resonance capacitor C2B is charged by the potential of the alternating voltage of the secondary winding part N2A by performing a rectification operation by the rectification current path of the secondary side series resonance capacitor C2B → secondary winding part N2B]. During a half cycle period of one polarity, rectification operation is performed by a rectification current path of [secondary winding portion N2B → secondary side series resonance capacitor C2B → rectifier diode Do3 → smoothing capacitor Co → secondary winding portion N2B]. Thus, the smoothing capacitor Co is charged by the superimposed potential of the voltage across the secondary side series resonant capacitor C2B and the alternating voltage of the secondary winding N2B.

  According to the rectifying operation described above, with respect to the smoothing capacitor Co, the induced voltage of the secondary winding N2B and the secondary side series resonant capacitor in the half cycle of one polarity of the alternating voltage of the secondary winding N2 The rectified current is charged by the superimposed potential of the voltage at both ends of C2B. In the other half cycle, the rectified current by the superimposed potential of the induced voltage of the secondary winding N2A and the voltage at both ends of the secondary side series resonant capacitor C2A is charged. Will be charged. As a result, a level corresponding to twice the induced voltage level of the secondary winding portions N2A and N2B is obtained as the secondary side DC output voltage Eo that is the voltage across the smoothing capacitor Co. That is, a voltage doubler full wave rectifier circuit is obtained.

Note that the specific design examples of the power supply circuit according to the embodiment described so far are based on the assumption that an AC 100V commercial AC power supply is input. For example, the power supply circuit corresponds to an AC 200V commercial AC power input. Even in the case of design, the same effect can be obtained by adopting a configuration based on the present invention.
Further, the present invention is not limited to the configurations shown as the above embodiments. For example, the detailed circuit configuration of the primary side voltage resonance type converter and the configuration of the secondary side rectifier circuit formed including the secondary side series resonance circuit are also conceivable.
As the switching element, it may be considered to select an element other than the MOS-FET, such as an IGBT (Insulated Gate Bipolar Transistor) or a bipolar transistor. In each of the above embodiments, a separately excited switching converter is cited, but the present invention can also be applied to a case where it is configured as a self-excited type.

1 is a circuit diagram illustrating a configuration example of a switching power supply circuit according to a first embodiment of the present invention. It is sectional drawing which shows the structural example of the insulation converter transformer with which the switching power supply circuit of embodiment is provided. It is sectional drawing which shows the structural example of the choke coil with which the switching power supply circuit of embodiment is provided. It is a wave form diagram which shows operation | movement of the principal part in the power supply circuit of 1st Embodiment by a commercial alternating current power supply period. It is a figure which shows the characteristic of the rectification smoothing voltage with respect to load fluctuation | variation, a power factor, and AC-> DC power conversion efficiency about the power supply circuit of embodiment. It is a figure which shows the characteristic of the rectification | straightening voltage, power factor, and AC-> DC power conversion efficiency with respect to alternating current input voltage fluctuation | variation about the power supply circuit of embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit as the 2nd Embodiment of this invention. It is a circuit diagram which shows the structural example of the switching power supply circuit as the 2nd Embodiment of this invention. It is a circuit diagram which shows the structural example as a modification of the secondary side corresponding to the power supply circuit of embodiment. It is a circuit diagram which shows the other structural example as a modification of the secondary side corresponding to the power supply circuit of embodiment. It is a circuit diagram which shows the basic structural example of a class E switching converter. FIG. 12 is a waveform diagram showing an operation of the class E switching converter shown in FIG. 11. It is a circuit diagram which shows the basic circuit structure of an active filter. It is a wave form diagram which shows the operation | movement in the active filter shown in FIG. It is a circuit diagram which shows the structure of the control circuit system of an active filter. It is a circuit diagram which shows the structural example of the conventional power supply circuit which mounted the active filter. FIG. 17 is a waveform diagram showing waveforms of an AC input voltage and an AC input current obtained corresponding to the AC 100 V system in the power supply circuit shown in FIG. 16. FIG. 17 is a waveform diagram showing waveforms of an AC input voltage and an AC input current obtained corresponding to the AC 200 V system in the power supply circuit shown in FIG. 16. FIG. 17 is a characteristic diagram showing each characteristic of power conversion efficiency, power factor, and rectified smoothing voltage with respect to load fluctuation of the power supply circuit shown in FIG. 16. FIG. 17 is a characteristic diagram illustrating each characteristic of power conversion efficiency, power factor, and rectified smoothing voltage with respect to AC input voltage variation of the power supply circuit illustrated in FIG. 16.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 Control circuit, 2 Oscillation drive circuit 10, 11, 12 Power factor improvement circuit, Di bridge rectification circuit, Ci smoothing capacitor, Q1 switching element, PIT insulation converter transformer, Cr primary side parallel resonance capacitor, C2 secondary side series Resonant capacitor, N1 primary winding, N2 secondary winding, Do1 to Do4 (secondary side) rectifier diode, Co (secondary side) smoothing capacitor, CN filter capacitor, D1 switching diode, N11 choke coil winding for power factor improvement Wire, C11 primary side series resonant capacitor, PCC1 choke coil, N10 choke coil winding, Lo power factor improving choke coil

Claims (1)

A rectifying / smoothing circuit that is formed with at least a rectifying element and a smoothing capacitor, and generates a rectified smoothing voltage as a voltage across the smoothing capacitor by inputting a commercial AC power supply and rectifying and smoothing;
A switching element that performs switching by inputting the rectified and smoothed voltage;
Switching driving means for switching and driving the switching element;
A first inductor inserted in series with respect to a path through which the rectified and smoothed voltage is input to the switching element;
A primary side parallel resonant capacitor that is connected in parallel with the switching element and forms a primary side parallel resonant circuit by at least the inductance of the first inductor and its own capacitance;
A second inductor;
By being connected in series with the second inductor, a primary side series resonance circuit is formed by at least the inductance of the second inductor and its own capacitance, and the second inductor and itself A primary series resonant capacitor provided so that a series connection circuit is connected in parallel with the switching element;
The second inductor is wound as a primary winding, and is formed by winding a secondary winding in which an alternating voltage is induced by the switching output obtained in the primary winding. A converter transformer configured to obtain a coupling coefficient between the primary side and the secondary side of
An alternating voltage induced in the power factor improving inductor from the first inductor is fed back to the smoothing capacitor forming the rectifying and smoothing circuit, and obtained in the primary winding according to the switching operation of the switching element. The current is regenerated as electric power, and the operation of returning to the smoothing capacitor through the first inductor is performed, and the rectification operation in the rectifying and smoothing circuit is performed according to the switching output fed back by these operations. Power factor improving means configured to interrupt the obtained rectified current by the power factor improving switching element;
A switching power supply circuit comprising:

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