JP2006303691A - Mimo-receiving device, reception method, and radio communications system - Google Patents

Mimo-receiving device, reception method, and radio communications system Download PDF

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JP2006303691A
JP2006303691A JP2005119756A JP2005119756A JP2006303691A JP 2006303691 A JP2006303691 A JP 2006303691A JP 2005119756 A JP2005119756 A JP 2005119756A JP 2005119756 A JP2005119756 A JP 2005119756A JP 2006303691 A JP2006303691 A JP 2006303691A
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correlation matrix
correction coefficient
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Masayuki Kimata
昌幸 木全
Naomasa Yoshida
尚正 吉田
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NEC Corp
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<P>PROBLEM TO BE SOLVED: To provide a MIMO receiving device capable of sufficiently suppressing interference from other transmission antennas in the filter reception of MINO of DS-CDMA, using the same code set in each transmission antenna, and to provide a reception method and a radio communications system. <P>SOLUTION: A correction coefficient calculating section 41 inputs an orthogonal coefficient ρ of a transmission line calculated by a transmission line orthogonal coefficient calculating section 43, and calculates a correction coefficient β that considers the interfering power of a transmission antenna signal as being larger than an actual value, as compared with the power of a desired transmission antenna signal. A weight calculating section 40 inputs the impulse response of the transmission line converted into a frequency region by FFT sections 12-1-1 to 12-M-N, chip noise power estimated by a chip noise estimating section 16, and the correction coefficient β calculated by the correction coefficient calculating section 41, and calculates the weight of a filter, based on MMSE. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

本発明はＭＩＭＯ(Multiple Input Multiple Output)受信装置、受信方法および無線通信システムに関し、特に複数の受信アンテナを用いて、最小平均自乗誤差法（ＭＭＳＥ:Minimum
Mean Square Error)により、信号の復調を行うＭＩＭＯ受信装置、受信方法および無線通信システムに関する。 The present invention relates to a MIMO (Multiple Input Multiple Output) receiving apparatus, a receiving method, and a wireless communication system, and more particularly to a minimum mean square error method (MMSE) using a plurality of receiving antennas.
The present invention relates to a MIMO receiving apparatus, a receiving method, and a wireless communication system that perform signal demodulation by Mean Square Error).

次世代移動通信の無線通信方式では、高速データ伝送を実現することが重要である。高速データ伝送を実現する技術として、複数の送信アンテナから同一の周波数、時間を用いて信号を送信し、複数の受信アンテナを用いて信号の復調（信号分離）を行うＭＩＭＯ多重方式が注目されている。   It is important to realize high-speed data transmission in the next-generation mobile communication wireless communication system. As a technique for realizing high-speed data transmission, a MIMO multiplexing system that transmits signals from a plurality of transmitting antennas using the same frequency and time and demodulates signals (signal separation) using a plurality of receiving antennas has attracted attention. Yes.

図３は、送信アンテナの数をＭ（Ｍは１以上の整数）、受信アンテナの数をＮ（Ｎは１以上の整数）とした場合のＭＩＭＯ送受信装置を示す図である。送信側は送信アンテナ１−１〜１−Ｍおよび送信装置２で構成され、受信側は受信アンテナ３−１〜３−Ｎおよび受信装置４で構成される。複数の送信アンテナ１−１〜１−Ｍから異なる信号を同一の周波数、時間を用いて送信し、複数の受信アンテナ３−１〜３−Ｎを用いて信号を受信することにより、伝送帯域幅を増加せずに送信アンテナ数に比例した高速データ伝送が可能となる。受信側では複数の受信アンテナ３−１〜３−Ｎで受信した信号から複数の送信アンテナ１−１〜１−Ｍからの信号を復調する信号分離処理が必要となる。   FIG. 3 is a diagram illustrating a MIMO transmitting / receiving apparatus when the number of transmission antennas is M (M is an integer of 1 or more) and the number of reception antennas is N (N is an integer of 1 or more). The transmission side includes transmission antennas 1-1 to 1-M and the transmission device 2, and the reception side includes reception antennas 3-1 to 3-N and the reception device 4. By transmitting different signals from the plurality of transmission antennas 1-1 to 1-M using the same frequency and time and receiving signals using the plurality of reception antennas 3-1 to 3-N, the transmission bandwidth High-speed data transmission proportional to the number of transmission antennas is possible without increasing the number of transmission antennas. On the receiving side, signal separation processing is required to demodulate signals from the plurality of transmitting antennas 1-1 to 1-M from signals received by the plurality of receiving antennas 3-1 to 3-N.

ＭＩＭＯ多重信号の復調方法には種々の方法があるが、比較的簡易な方法に線形フィルタ受信がある。ＤＳ−ＣＤＭＡ(Direct Sequence-Code Division Multiple Access)信号にＭＩＭＯ多重方式を用いた場合、他の送信アンテナからの干渉に加え、希望の送信アンテナ信号のマルチパスも干渉となり、これらの干渉を同時に抑圧するフィルタ受信が有効である。この信号処理を周波数領域で簡易に行う周波数領域イコライザが提案されている（例えば、非特許文献１参照）。   There are various methods for demodulating a MIMO multiplexed signal, and linear filter reception is a relatively simple method. When MIMO multiplexing is used for DS-CDMA (Direct Sequence-Code Division Multiple Access) signals, in addition to interference from other transmission antennas, multipath of the desired transmission antenna signal also becomes interference, and these interferences are simultaneously suppressed. Filter reception is effective. A frequency domain equalizer that performs this signal processing easily in the frequency domain has been proposed (see, for example, Non-Patent Document 1).

図４は、非特許文献１に記載された周波数領域イコライザをＤＳ−ＣＤＭＡのＭＩＭＯ受信装置に用いた場合の構成の一例を示す。送信アンテナの数をＭ（Ｍは１以上の整数）、受信アンテナの数をＮ（Ｎは１以上の整数）とし、ＭＩＭＯ受信装置について説明する。従来のＭＩＭＯ受信装置は、伝送路推定部１０−１−１〜１０−Ｍ−Ｎ、Ｓ／Ｐ（直並列）変換部１１−１−１〜１１−Ｍ−Ｎ、１４−１〜１４−Ｎ、ＦＦＴ（高速フーリエ変換）部１２−１−１〜１２−Ｍ−Ｎ、１５−１〜１５−Ｎ、ＧＩ（ガードインターバル）除去部１３−１〜１３−Ｎ、チップ雑音推定部１６、ウエイト計算部１７、フィルタ処理部１８、ＩＦＦＴ（高速逆フーリエ変換）部１９−１〜１９−Ｍ、Ｐ／Ｓ（並直列）変換部２０−１〜２０−Ｍおよび逆拡散回路２１−１〜２１−Ｍで構成される。   FIG. 4 shows an example of a configuration when the frequency domain equalizer described in Non-Patent Document 1 is used in a DS-CDMA MIMO receiver. The MIMO receiving apparatus will be described assuming that the number of transmitting antennas is M (M is an integer of 1 or more) and the number of receiving antennas is N (N is an integer of 1 or more). Conventional MIMO receivers include transmission path estimators 10-1-1 to 10-MN, S / P (serial-parallel) converters 11-1-1 to 11-MN, 14-1 to 14-. N, FFT (Fast Fourier Transform) units 12-1-1 to 12-MN, 15-1 to 15-N, GI (guard interval) removing units 13-1 to 13-N, chip noise estimating unit 16, Weight calculation unit 17, filter processing unit 18, IFFT (fast inverse Fourier transform) units 19-1 to 19-M, P / S (parallel serial) conversion units 20-1 to 20-M, and despreading circuit 21-1 21-M.

伝送路推定部１０−１−１〜１０−Ｍ−Ｎは、受信アンテナ３−１〜３−Ｎで受信した信号を入力し、受信信号に含まれる既知のパイロット信号を用いて、送信アンテナ１−１〜１−Ｍと受信アンテナ３−１〜３−Ｎ間の伝送路推定値をパス毎に推定し、インパルス応答を求める。   The transmission path estimators 10-1-1 to 10-MN receive signals received by the receiving antennas 3-1 to 3-N, and use a known pilot signal included in the received signal to transmit the transmitting antenna 1 The transmission path estimation value between −1 to 1-M and the receiving antennas 3-1 to 3-N is estimated for each path, and the impulse response is obtained.

Ｓ／Ｐ変換部１１−１−１〜１１−Ｍ−Ｎは、伝送路推定部１０−１−１〜１０−Ｍ−Ｎで推定した伝送路のインパルス応答をＳ／Ｐ変換する。   The S / P conversion units 11-1-1 to 11-MN perform S / P conversion on the impulse response of the transmission path estimated by the transmission path estimation units 10-1-1 to 10-MN.

ＦＦＴ部１２−１−１〜１２−Ｍ−Ｎは、Ｓ／Ｐ変換部１１−１−１〜１１−Ｍ−Ｎで変換した伝送路のインパルス応答を入力し、周波数領域に変換する。   The FFT units 12-1-1 to 12 -MN receive the impulse responses of the transmission lines converted by the S / P converters 11-1-1 to 11 -MN and convert them to the frequency domain.

チップ雑音推定部１６は、受信アンテナ３−１〜３−Ｎで受信した信号および伝送路推定部１０−１−１〜１０−Ｍ−Ｎで推定した伝送路推定値を入力し、チップ雑音電力を推定する。   The chip noise estimator 16 receives the signals received by the receiving antennas 3-1 to 3-N and the transmission path estimation values estimated by the transmission path estimators 10-1-1 to 10-MN, and the chip noise power Is estimated.

ウエイト計算部１７は、ＦＦＴ部１２−１−１〜１２−Ｍ−Ｎで周波数領域に変換した伝送路のインパルス応答およびチップ雑音推定部１６で推定したチップ雑音電力を入力し、最小平均自乗誤差法（ＭＭＳＥ）により、フィルタのウエイトを計算する。   The weight calculation unit 17 receives the impulse response of the transmission path converted into the frequency domain by the FFT units 12-1-1 to 12 -MN and the chip noise power estimated by the chip noise estimation unit 16, and receives the minimum mean square error. The weight of the filter is calculated by the method (MMSE).

ＧＩ除去部１３−１〜１３−Ｎは、受信アンテナ３−１〜３−Ｎで受信した信号を入力し、受信パスタミングを基準にして、ＧＩに相当する部分の受信信号を除去する。   The GI removal units 13-1 to 13-N receive the signals received by the reception antennas 3-1 to 3-N, and remove received signals corresponding to the GI with reference to reception path timing.

Ｓ／Ｐ変換部１４−１〜１４−Ｎは、ＧＩ除去部１３−１〜１３−ＮでＧＩを除去した信号をＳ／Ｐ変換する。   The S / P converters 14-1 to 14-N perform S / P conversion on the signals from which the GI has been removed by the GI removers 13-1 to 13-N.

ＦＦＴ部１５−１〜１５−Ｎは、Ｓ／Ｐ変換部１４−１〜１４−Ｎで変換した受信信号を入力し、周波数領域に変換する。   The FFT units 15-1 to 15-N receive the reception signals converted by the S / P conversion units 14-1 to 14-N and convert them into the frequency domain.

フィルタ処理部１８は、ウエイト計算部１７で計算したウエイトおよびＦＦＴ部１５−１〜１５−Ｎで周波数変換した受信信号を入力し、周波数領域で受信信号のフィルタリング（等化）を行う。   The filter processing unit 18 receives the weight calculated by the weight calculation unit 17 and the reception signal frequency-converted by the FFT units 15-1 to 15-N, and performs filtering (equalization) of the reception signal in the frequency domain.

ＩＦＦＴ部１９−１〜１９−Ｍは、フィルタ処理部１８で等化した周波数領域の信号を入力し、時間領域に変換する。   The IFFT units 19-1 to 19-M receive the frequency domain signal equalized by the filter processing unit 18 and convert it into the time domain.

Ｐ／Ｓ変換部２０−１〜２０−Ｍは、時間領域に変換した信号をＰ／Ｓ変換する。   The P / S conversion units 20-1 to 20-M perform P / S conversion on the signal converted into the time domain.

逆拡散回路２１−１〜２１−Ｍは、Ｐ／Ｓ変換部２０−１〜２０−Ｍで変換した時間領域の信号を入力し、逆拡散を行い、送信アンテナ１−１〜１−Ｍからの送信信号を復調する。   The despreading circuits 21-1 to 21-M receive time domain signals converted by the P / S conversion units 20-1 to 20-M, perform despreading, and transmit from the transmission antennas 1-1 to 1-M. The transmitted signal is demodulated.

図５は、ＦＦＴ後のサブキャリアｆ（１≦ｆ≦Ｆ）におけるウエイト計算部１７の構成を示すブロック図である。サブキャリアｆにおける従来のウエイト計算部１７は、相関行列生成部３０−１〜３０−Ｍ、相関行列加算部３１、雑音加算部３２、逆行列演算部３３およびウエイト生成部３４−１〜３４−Ｍで構成され、各サブキャリアでは同一の構成を有する。   FIG. 5 is a block diagram illustrating a configuration of the weight calculation unit 17 in the subcarrier f after FFT (1 ≦ f ≦ F). The conventional weight calculation unit 17 in the subcarrier f includes correlation matrix generation units 30-1 to 30-M, a correlation matrix addition unit 31, a noise addition unit 32, an inverse matrix calculation unit 33, and weight generation units 34-1 to 34-34. M, and each subcarrier has the same configuration.

相関行列生成部３０−１〜３０−Ｍは、図４のＦＦＴ部１２−１−１〜１２−Ｍ−Ｎで周波数領域に変換した送信アンテナと受信アンテナ間の伝送路推定値を入力し、送信アンテナ毎に各送信アンテナから受信アンテナへの相関行列を生成する。   Correlation matrix generation units 30-1 to 30-M input transmission path estimation values between the transmission antenna and the reception antenna converted into the frequency domain by the FFT units 12-1-1-1 to 12-MN of FIG. A correlation matrix from each transmission antenna to the reception antenna is generated for each transmission antenna.

相関行列加算部３１は、相関行列生成部３０−１〜３０−Ｍで生成した送信アンテナ毎の相関行列を入力し、相関行列を加算する。   The correlation matrix adding unit 31 inputs the correlation matrix for each transmission antenna generated by the correlation matrix generating units 30-1 to 30-M, and adds the correlation matrix.

雑音加算部３２は、相関行列加算部３１で加算した相関行列および図４のチップ雑音推定部１６で推定したチップ雑音電力を入力し、相関行列に雑音成分を加算する。   The noise adder 32 receives the correlation matrix added by the correlation matrix adder 31 and the chip noise power estimated by the chip noise estimator 16 of FIG. 4 and adds a noise component to the correlation matrix.

逆行列演算部３３は、雑音加算部３２で雑音成分を加算した相関行列を入力し、逆行列演算を行う。   The inverse matrix calculation unit 33 receives the correlation matrix obtained by adding the noise components by the noise addition unit 32 and performs an inverse matrix calculation.

ウエイト生成部３４−１〜３４−Ｍは、逆行列演算部３３で演算した逆行列および周波数領域に変換した送信アンテナと受信アンテナ間の伝送路推定値を入力し、フィルタのウエイトを生成する。   The weight generation units 34-1 to 34-M input the inverse matrix calculated by the inverse matrix calculation unit 33 and the transmission path estimation value between the transmission antenna and the reception antenna converted to the frequency domain, and generate filter weights.

以上の処理に関して、数式を用いて詳細に説明する。ＦＦＴ部１２−１−１〜１２−Ｍ−Ｎで伝送路のインパルス応答を周波数領域に変換したサブキャリアｆにおける送信アンテナｍ（１≦ｍ≦Ｍ）と受信アンテナ間の伝送路ベクトルＨｍ（ｆ）は次式で定義される。
Ｈｍ（ｆ）＝［ｈｍ，１（ｆ），ｈｍ，２（ｆ），…，ｈｍ，Ｎ（ｆ）］Ｔ …（１） The above processing will be described in detail using mathematical expressions. Transmission path vector H m (1 ≦ m ≦ M) and the transmission path vector H m (in the subcarrier f) obtained by converting the impulse response of the transmission path into the frequency domain by the FFT units 12-1-1-1 to 12-MN. f) is defined by the following equation.
H m (f) = [h m, 1 (f), h m, 2 (f),..., H m, N (f)] T (1)

ここで、Ｔは転置を表す。また、ＦＦＴ部１５−１〜１５−Ｎで受信信号を周波数領域に変換したサブキャリアｆにおける受信信号ベクトルＸ（ｆ）は次式で定義される。
Ｘ（ｆ）＝［ｘ１（ｆ），ｘ２（ｆ），…，ｘＮ（ｆ）］Ｔ …（２）
ウエイト計算部１７で計算されるサブキャリアｆにおける送信アンテナｍのフィルタのウエイトベクトルＷｍ（ｆ）は、次式で表される。 Here, T represents transposition. Also, the received signal vector X (f) in subcarrier f obtained by converting the received signal into the frequency domain by FFT sections 15-1 to 15-N is defined by the following equation.
X (f) = [x 1 (f), x 2 (f),..., X N (f)] T (2)
The weight vector W m (f) of the filter of the transmission antenna m in the subcarrier f calculated by the weight calculation unit 17 is expressed by the following equation.

Figure 2006303691
ここで、Ｈは複素共役転置、Ｐはパイロット電力、Ｄはデータ電力、Ｋはデータのコード数、Ｎ０はチップ雑音電力、Ｉは単位行列を示す。
Figure 2006303691
 Here, H is a complex conjugate transpose, P is pilot power, D is data power, K is the number of data codes, N 0 is chip noise power, and I is a unit matrix.

フィルタ処理部１８で等化、信号分離されたサブキャリアｆにおける送信信号ベクトルＹ（ｆ）は、次式で表される。   The transmission signal vector Y (f) in the subcarrier f equalized and signal-separated by the filter processing unit 18 is expressed by the following equation.

Ｙ（ｆ）＝ＷＨ（ｆ）Ｘ（ｆ） …（４）
ここで、Ｗ（ｆ）、Ｙ（ｆ）は、次式で定義される。 Y (f) = W H (f) X (f) (4)
Here, W (f) and Y (f) are defined by the following equations.

Ｗ（ｆ）＝［Ｗ１（ｆ），Ｗ２（ｆ），…，ＷＭ（ｆ）］Ｔ …（５）
Ｙ（ｆ）＝［Ｙ１（ｆ），Ｙ２（ｆ），…，ＹＭ（ｆ）］Ｔ …（６） W (f) = [W 1 (f), W 2 (f),..., W M (f)] T (5)
Y (f) = [Y 1 (f), Y 2 (f),..., Y M (f)] T (6)

Ｘｕ Ｚｈｕ ａｎｄ Ｒｏｓｓ Ｄ．Ｍｕｒｃｈ，“Ｎｏｖｅｌ Ｆｒｅｑｕｅｎｃｙ−Ｄｏｍａｉｎ Ｅｑｕａｌｉｚａｔｉｏｎ Ａｒｃｈｉｔｅｃｔｕｒｅｓ ｆｏｒ ａ Ｓｉｎｇｌｅ−Ｃａｒｒｉｅｒ Ｗｉｒｅｌｅｓｓ ＭＩＭＯ Ｓｙｓｔｅｍ，” ＩＥＥＥ ＶＴＣ２００２−Ｆａｌｌ，ｐｐ．８７４−８７８，Ｓｅｐ．２００２．Xu Zhu and Ross D.D. Murch, “Novel Frequency-Domain Equalization Architectures for a Single-Carrier Wireless MIMO System,” IEEE VTC2002-Fall, pp. 196 874-878, Sep. 2002.

従来のＭＩＭＯ受信装置では、次のような問題点がある。ウエイト計算部１７におけるフィルタのウエイト計算では、相関行列生成部３０−１〜３０−Ｍで送信アンテナ毎に相関行列を生成し、相関行列加算部３１で全送信アンテナで加算する際にそれらの大小関係で送信アンテナ間の干渉の抑圧度合いが決まる。   The conventional MIMO receiver has the following problems. In the filter weight calculation in the weight calculation unit 17, the correlation matrix generation unit 30-1 to 30 -M generates a correlation matrix for each transmission antenna, and the correlation matrix addition unit 31 adds them to all transmission antennas. The degree of suppression of interference between transmission antennas is determined by the relationship.

非特許文献１に記載された周波数領域イコライザは非拡散方式（あるいは拡散方式でも、各送信アンテナで異なるコードセットを用いる場合）のフィルタのウエイトの計算式であり、各送信アンテナで同一のコードセットを用いるＤＳ−ＣＤＭＡのＭＩＭＯの場合にはウエイトの精度がずれ、特性が劣化するという問題がある。   The frequency domain equalizer described in Non-Patent Document 1 is a filter weight calculation formula of a non-spreading method (or a different code set for each transmitting antenna even in the spreading method), and the same code set is used for each transmitting antenna. In the case of DS-CDMA MIMO using the above, there is a problem that the weight accuracy is shifted and the characteristics are deteriorated.

その理由は、各送信アンテナで同一のコードセットを用いるＤＳ−ＣＤＭＡのＭＩＭＯの場合には、希望の送信アンテナ信号を復調する際に、他の送信アンテナからの干渉のうち、同一コードで同一パスの信号に対して逆拡散利得が得られないため、逆拡散回路２１−１〜２１−Ｍの出力で他の送信アンテナの干渉を十分に抑圧できなくなるからである。   This is because, in the case of DS-CDMA MIMO using the same code set at each transmission antenna, when demodulating a desired transmission antenna signal, the same path with the same code among the interference from other transmission antennas. This is because the despreading gain cannot be obtained with respect to the above signal, and interference from other transmitting antennas cannot be sufficiently suppressed by the outputs of the despreading circuits 21-1 to 21-M.

本発明の目的は、各送信アンテナで同一のコードセットを用いるＤＳ−ＣＤＭＡのＭＩＭＯのフィルタ受信において、逆拡散回路の出力で他の送信アンテナからの干渉が十分に抑圧されるように、伝送路の直交性に基づいて、希望の送信アンテナ信号の電力よりも干渉となる送信アンテナ信号の電力を実際より大きくみなす補正を行い、フィルタのウエイトを計算することで優れた干渉抑圧特性を実現できるＭＩＭＯ受信装置、受信方法および無線通信システムを提供することにある。   An object of the present invention is to provide a transmission path so that interference from other transmission antennas is sufficiently suppressed by the output of a despreading circuit in DS-CDMA MIMO filter reception using the same code set in each transmission antenna. MIMO that achieves excellent interference suppression characteristics by calculating the filter weight by correcting the transmission antenna signal power that causes interference more than the actual transmission antenna power based on the orthogonality of the desired To provide a receiving apparatus, a receiving method, and a wireless communication system.

上記課題を解決するため、本発明が提供する第一のＭＩＭＯ受信装置は、複数の送信アンテナから送信したＣＤＭＡ方式の信号を、複数の受信アンテナで受信する受信装置であって、伝送路の直交性に基づいて、希望の送信アンテナ信号の電力よりも干渉となる送信アンテナ信号の電力を実際より大きくみなす補正係数（１以上の定数）を計算する補正係数計算手段と、前記補正係数と全ての送受信アンテナ間の伝送路推定値とを用いて、最小平均自乗誤差法（ＭＭＳＥ）に基づいて、受信信号をフィルタリングするフィルタのウエイトを計算するウエイト計算手段と、前記ウエイトで受信信号をフィルタリングし、干渉となる送信アンテナ信号を抑圧し、希望の送信アンテナ信号を復調するフィルタ処理部とを有することを特徴とする。   In order to solve the above problems, a first MIMO receiving apparatus provided by the present invention is a receiving apparatus that receives a CDMA signal transmitted from a plurality of transmitting antennas using a plurality of receiving antennas, and is orthogonal to the transmission path. Correction coefficient calculation means for calculating a correction coefficient (a constant of 1 or more) that considers the power of the transmission antenna signal that causes interference more than the actual power than the power of the desired transmission antenna signal based on the characteristics; A weight calculation means for calculating a weight of a filter for filtering the received signal based on a minimum mean square error method (MMSE) using a transmission path estimation value between the transmitting and receiving antennas, and filtering the received signal with the weight; And a filter processing unit that suppresses a transmission antenna signal that causes interference and demodulates a desired transmission antenna signal.

例えば、前記補正係数計算手段は、干渉となる送信アンテナ信号の電力を、逆拡散利得が得られない同一コードで同一パスの干渉成分と逆拡散利得が得られる同一コードで異なるパスの干渉成分および異なるコードの干渉成分との和とみなして前記補正係数を計算する。   For example, the correction coefficient calculation means may calculate the interference component of the transmission antenna signal that causes interference, the interference component of the same path where the despread gain is obtained, and the interference component of the same path where the despread gain is obtained, The correction coefficient is calculated as a sum of interference components of different codes.

また、前記ウエイト計算手段および前記フィルタ処理部は、時間領域の信号処理で行うことができる。あるいは、前記ウエイト計算手段および前記フィルタ処理部は、周波数領域の信号処理で行うことができる。   The weight calculation means and the filter processing unit can be performed by time domain signal processing. Alternatively, the weight calculation unit and the filter processing unit can be performed by frequency domain signal processing.

前記ウエイト計算手段が、周波数領域の信号処理を行う場合には、例えば、周波数領域で表された希望の送信アンテナと受信アンテナとの間の伝送路推定値から相関行列を生成する第一の相関行列生成手段と、周波数領域で表された干渉となる送信アンテナと受信アンテナとの間の伝送路推定値から送信アンテナ毎に相関行列を生成する第二の相関行列生成手段と、前記第二の相関行列生成手段で生成した送信アンテナ毎の相関行列に前記補正係数計算手段で計算した前記補正係数をそれぞれ乗算する補正係数乗算手段と、前記補正係数を乗算した送信アンテナ毎の相関行列と前記第一の相関行列生成手段で生成した相関行列とを全て加算する相関行列加算手段と、前記相関行列加算手段で加算した相関行列に雑音成分を加算する雑音電力加算手段と、前記雑音電力加算手段で雑音成分を加算した相関行列の逆行列を演算する逆行列演算手段と、前記逆行列演算手段で演算した相関行列の逆行列と周波数領域で表された希望の送信アンテナと受信アンテナとの間の伝送路推定値からウエイトを生成するウエイト生成手段とを有する。   When the weight calculation means performs signal processing in the frequency domain, for example, a first correlation that generates a correlation matrix from a channel estimation value between a desired transmission antenna and reception antenna expressed in the frequency domain. Matrix generation means, second correlation matrix generation means for generating a correlation matrix for each transmission antenna from a transmission path estimation value between the transmission antenna and the reception antenna that causes interference expressed in the frequency domain, and Correction coefficient multiplication means for multiplying the correlation matrix for each transmission antenna generated by the correlation matrix generation means by the correction coefficient calculated by the correction coefficient calculation means; a correlation matrix for each transmission antenna multiplied by the correction coefficient; A correlation matrix addition means for adding all the correlation matrices generated by one correlation matrix generation means; and a noise power addition for adding a noise component to the correlation matrix added by the correlation matrix addition means. Means, an inverse matrix computing means for computing an inverse matrix of the correlation matrix obtained by adding the noise component by the noise power adding means, and an inverse matrix of the correlation matrix computed by the inverse matrix computing means and a desired matrix expressed in the frequency domain Weight generating means for generating a weight from a transmission path estimated value between the transmitting antenna and the receiving antenna;

本発明が提供する第二のＭＩＭＯ受信装置は、複数の送信アンテナから送信したＣＤＭＡ方式の信号を、複数の受信アンテナで受信する受信装置であって、受信信号を入力とし伝送路の遅延プロファイルを生成し、前記遅延プロファイルからレベルの大きな複数のパスを検出するパスサーチ部と、前記パスサーチ部で検出された複数のパスから伝送路の直交係数を計算する伝送路直交係数計算手段と、前記伝送路直交係数に基づいて、希望の送信アンテナ信号の電力よりも干渉となる送信アンテナ信号の電力を実際より大きくみなす補正係数（１以上の定数）を計算する補正係数計算手段と、前記補正係数と全ての送受信アンテナ間の伝送路推定値とを用いて、最小平均自乗誤差法（ＭＭＳＥ）に基づいて、受信信号をフィルタリングするフィルタのウエイトを計算するウエイト計算手段と、前記ウエイトで受信信号をフィルタリングし、干渉となる送信アンテナ信号を抑圧し、希望の送信アンテナ信号を復調するフィルタ処理部とを有することを特徴とする。   A second MIMO receiving apparatus provided by the present invention is a receiving apparatus that receives a CDMA signal transmitted from a plurality of transmitting antennas by a plurality of receiving antennas, and receives a received signal as an input and determines a delay profile of a transmission path. A path search unit that generates and detects a plurality of paths having a large level from the delay profile; a transmission path orthogonal coefficient calculation unit that calculates an orthogonal coefficient of a transmission path from the plurality of paths detected by the path search unit; Correction coefficient calculation means for calculating a correction coefficient (a constant of 1 or more) that considers the power of the transmission antenna signal causing interference more than the actual power based on the transmission path orthogonal coefficient, and the correction coefficient Is used to filter the received signal based on the minimum mean square error method (MMSE). A weight calculation means for calculating the filter weights to filter the received signal at the weight suppresses the transmission antenna signal serving as interference, and having a filter processing unit that demodulates a transmission antenna signal of the desired.

例えば、前記補正係数計算手段は、干渉となる送信アンテナ信号の電力を、逆拡散利得が得られない同一コードで同一パスの干渉成分と逆拡散利得が得られる同一コードで異なるパスの干渉成分および異なるコードの干渉成分との和とみなして前記補正係数を計算する。   For example, the correction coefficient calculation means may calculate the interference component of the transmission antenna signal that causes interference, the interference component of the same path where the despread gain is obtained, and the interference component of the same path where the despread gain is obtained, The correction coefficient is calculated as a sum of interference components of different codes.

また、前記ウエイト計算手段および前記フィルタ処理部は、時間領域の信号処理で行うことができる。あるいは、前記ウエイト計算手段および前記フィルタ処理部は、周波数領域の信号処理で行うことができる。   The weight calculation means and the filter processing unit can be performed by time domain signal processing. Alternatively, the weight calculation unit and the filter processing unit can be performed by frequency domain signal processing.

前記ウエイト計算手段が、周波数領域の信号処理を行う場合には、例えば、周波数領域で表された希望の送信アンテナと受信アンテナとの間の伝送路推定値から相関行列を生成する第一の相関行列生成手段と、周波数領域で表された干渉となる送信アンテナと受信アンテナとの間の伝送路推定値から送信アンテナ毎に相関行列を生成する第二の相関行列生成手段と、前記第二の相関行列生成手段で生成した送信アンテナ毎の相関行列に前記補正係数計算手段で計算した前記補正係数をそれぞれ乗算する補正係数乗算手段と、前記補正係数を乗算した送信アンテナ毎の相関行列と前記第一の相関行列生成手段で生成した相関行列とを全て加算する相関行列加算手段と、前記相関行列加算手段で加算した相関行列に雑音成分を加算する雑音電力加算手段と、前記雑音電力加算手段で雑音成分を加算した相関行列の逆行列を演算する逆行列演算手段と、前記逆行列演算手段で演算した相関行列の逆行列と周波数領域で表された希望の送信アンテナと受信アンテナとの間の伝送路推定値からウエイトを生成するウエイト生成手段とを有する。   When the weight calculation means performs signal processing in the frequency domain, for example, a first correlation that generates a correlation matrix from a channel estimation value between a desired transmission antenna and reception antenna expressed in the frequency domain. Matrix generation means, second correlation matrix generation means for generating a correlation matrix for each transmission antenna from a transmission path estimation value between the transmission antenna and the reception antenna that causes interference expressed in the frequency domain, and Correction coefficient multiplication means for multiplying the correlation matrix for each transmission antenna generated by the correlation matrix generation means by the correction coefficient calculated by the correction coefficient calculation means; a correlation matrix for each transmission antenna multiplied by the correction coefficient; A correlation matrix addition means for adding all the correlation matrices generated by one correlation matrix generation means; and a noise power addition for adding a noise component to the correlation matrix added by the correlation matrix addition means. Means, an inverse matrix computing means for computing an inverse matrix of the correlation matrix obtained by adding the noise component by the noise power adding means, and an inverse matrix of the correlation matrix computed by the inverse matrix computing means and a desired matrix expressed in the frequency domain Weight generating means for generating a weight from a transmission path estimated value between the transmitting antenna and the receiving antenna;

本発明の別の観点は、複数の送信アンテナからＣＤＭＡ方式で送信した信号を、複数の受信アンテナで受信する受信装置に適用される受信方法であって、伝送路の直交性に基づいて、希望の送信アンテナ信号の電力よりも干渉となる送信アンテナ信号の電力を実際より大きくみなす補正係数（１以上の定数）を計算し、前記補正係数と全ての送受信アンテナ間の伝送路推定値とを用いて、最小平均自乗誤差法（ＭＭＳＥ）に基づいて、受信信号をフィルタリングするフィルタのウエイトを計算し、前記ウエイトで受信信号をフィルタリングし、干渉となる送信アンテナ信号を抑圧し、希望の送信アンテナ信号を復調することを特徴とする。   Another aspect of the present invention is a receiving method applied to a receiving apparatus that receives signals transmitted from a plurality of transmitting antennas by a CDMA system using a plurality of receiving antennas, and is based on orthogonality of transmission paths. A correction coefficient (constant of 1 or more) that considers the power of the transmission antenna signal that causes interference to be greater than the actual power of the transmission antenna signal is calculated, and the correction coefficient and the transmission path estimation values between all transmitting and receiving antennas are used Then, based on the minimum mean square error method (MMSE), the weight of the filter for filtering the received signal is calculated, the received signal is filtered with the weight, the transmission antenna signal that causes interference is suppressed, and the desired transmission antenna signal Is demodulated.

本発明のさらに別の観点は、複数の送信アンテナを備え、各送信アンテナで同一コードセットを用いるＣＤＭＡ信号を送信する送信装置と、本発明の受信装置とを有することを特徴とする無線通信システムである。   According to still another aspect of the present invention, a wireless communication system includes a transmission device that includes a plurality of transmission antennas and transmits a CDMA signal using the same code set at each transmission antenna, and the reception device of the present invention. It is.

本発明のＭＩＭＯ受信装置では、各送信アンテナで同一のコードセットを用いるＤＳ−ＣＤＭＡのＭＩＭＯのフィルタ受信において、逆拡散回路の出力で他の送信アンテナからの干渉が十分に抑圧されるように、伝送路の直交性に基づいて、希望の送信アンテナ信号の電力よりも干渉となる送信アンテナ信号の電力を実際より大きくみなす補正を行い、フィルタのウエイトを計算することで優れた干渉抑圧特性を実現できる。   In the MIMO receiving apparatus of the present invention, in DS-CDMA MIMO filter reception using the same code set in each transmission antenna, interference from other transmission antennas is sufficiently suppressed by the output of the despreading circuit. Based on the orthogonality of the transmission path, a correction that considers the power of the transmitting antenna signal that causes interference to be greater than the power of the desired transmitting antenna signal is performed, and excellent interference suppression characteristics are achieved by calculating the filter weight. it can.

本発明の実施の形態について図面を参照して詳細に説明する。図１は、本発明のＭＩＭＯ受信装置の一実施例を示す構成図であり、図４と同等部分は同一符号にて示している。図１において、送信アンテナの数をＭ（Ｍは１以上の整数）、受信アンテナの数をＮ（Ｎは１以上の整数）とし、ＭＩＭＯ受信装置について説明する。本発明のＭＩＭＯ受信装置は、伝送路推定部１０−１−１〜１０−Ｍ−Ｎ、Ｓ／Ｐ変換部１１−１−１〜１１−Ｍ−Ｎ、１４−１〜１４−Ｎ、ＦＦＴ部１２−１−１〜１２−Ｍ−Ｎ、１５−１〜１５−Ｎ、ＧＩ除去部１３−１〜１３−Ｎ、チップ雑音推定部１６、フィルタ処理部１８、ＩＦＦＴ部１９−１〜１９−Ｍ、Ｐ／Ｓ変換部２０−１〜２０−Ｍ、逆拡散回路２１−１〜２１−Ｍ、ウエイト計算部４０、補正係数計算部４１、パスサーチ部４２および伝送路直交係数計算部４３で構成される。   Embodiments of the present invention will be described in detail with reference to the drawings. FIG. 1 is a block diagram showing an embodiment of a MIMO receiving apparatus according to the present invention. The same parts as those in FIG. 4 are denoted by the same reference numerals. In FIG. 1, the number of transmitting antennas is M (M is an integer of 1 or more), the number of receiving antennas is N (N is an integer of 1 or more), and the MIMO receiving apparatus will be described. The MIMO receiver of the present invention includes transmission path estimation units 10-1-1 to 10-MN, S / P conversion units 11-1-1 to 11-MN, 14-1 to 14-N, and FFT. Units 12-1-1-1 to 12-MN, 15-1 to 15-N, GI removal units 13-1 to 13-N, chip noise estimation unit 16, filter processing unit 18, IFFT units 19-1 to 19-19 -M, P / S conversion units 20-1 to 20-M, despreading circuits 21-1 to 21-M, weight calculation unit 40, correction coefficient calculation unit 41, path search unit 42, and transmission path orthogonal coefficient calculation unit 43 Consists of.

伝送路推定部１０−１−１〜１０−Ｍ−Ｎは、受信アンテナ３−１〜３−Ｎで受信した信号を入力し、受信信号に含まれる既知のパイロット信号を用いて、送信アンテナ１−１〜１−Ｍと受信アンテナ３−１〜３−Ｎ間の伝送路推定値をパス毎に推定し、インパルス応答を求める。   The transmission path estimators 10-1-1 to 10-MN receive signals received by the receiving antennas 3-1 to 3-N, and use a known pilot signal included in the received signal to transmit the transmitting antenna 1 The transmission path estimation value between −1 to 1-M and the receiving antennas 3-1 to 3-N is estimated for each path, and the impulse response is obtained.

Ｓ／Ｐ変換部１１−１−１〜１１−Ｍ−Ｎは、伝送路推定部１０−１−１〜１０−Ｍ−Ｎで推定した伝送路のインパルス応答をＳ／Ｐ変換する。   The S / P conversion units 11-1-1 to 11-MN perform S / P conversion on the impulse response of the transmission path estimated by the transmission path estimation units 10-1-1 to 10-MN.

ＦＦＴ部１２−１−１〜１２−Ｍ−Ｎは、Ｓ／Ｐ変換部１１−１−１〜１１−Ｍ−Ｎで変換した伝送路のインパルス応答を入力し、周波数領域に変換する。   The FFT units 12-1-1 to 12 -MN receive the impulse responses of the transmission lines converted by the S / P converters 11-1-1 to 11 -MN and convert them to the frequency domain.

チップ雑音推定部１６は、受信アンテナ３−１〜３−Ｎで受信した信号および伝送路推定部１０−１−１〜１０−Ｍ−Ｎで推定した伝送路推定値を入力し、チップ雑音電力を推定する。チップ雑音電力の推定方法は種々の方法があるが、本発明とは直接関係しないので、その詳細な構成および説明は省略する。   The chip noise estimator 16 receives the signals received by the receiving antennas 3-1 to 3-N and the transmission path estimation values estimated by the transmission path estimators 10-1-1 to 10-MN, and the chip noise power Is estimated. There are various methods for estimating chip noise power, but since they are not directly related to the present invention, the detailed configuration and description thereof will be omitted.

パスサーチ部４２は、受信アンテナ３−１〜３−Ｎで受信した信号を入力し、受信信号に含まれる既知のパイロット信号を用いて、伝送路の遅延プロファイルを生成し、遅延プロファイルからレベルの大きな複数のパスを検出する。   The path search unit 42 receives signals received by the reception antennas 3-1 to 3-N, generates a delay profile of a transmission path using a known pilot signal included in the reception signal, and determines a level from the delay profile. Detect large multiple paths.

伝送路直交係数計算部４３は、パスサーチ部４２で検出したレベルの大きな複数のパスを入力し、伝送路の直交係数ρ（０≦ρ≦１）を計算する。ρは大きいほど伝送路の直交性が高く、ρ＝１のとき完全直交した１パス伝送路を表し、ρ＝０のときパスが多く分離した非直交伝送路を表す。このように伝送路の直交係数ρは、受信信号を用いて適応的に計算することができるが、伝搬環境があらかじめわかっている場合には固定値を用いることもできる。   The transmission path orthogonal coefficient calculation unit 43 inputs a plurality of paths with a large level detected by the path search unit 42, and calculates the transmission path orthogonal coefficient ρ (0 ≦ ρ ≦ 1). The larger ρ is, the higher the orthogonality of the transmission line is. When ρ = 1, it represents a completely orthogonal one-path transmission line, and when ρ = 0, it represents a non-orthogonal transmission line with many paths separated. As described above, the orthogonal coefficient ρ of the transmission path can be adaptively calculated using the received signal, but a fixed value can also be used when the propagation environment is known in advance.

補正係数計算部４１は、伝送路直交係数計算部４３で計算した伝送路の直交係数ρを入力し、希望の送信アンテナ信号の電力よりも干渉となる送信アンテナ信号の電力を実際より大きくみなす補正係数β（βは１以上の定数）を計算する。   The correction coefficient calculation unit 41 receives the transmission channel orthogonal coefficient ρ calculated by the transmission channel orthogonal coefficient calculation unit 43, and corrects the transmission antenna signal power that causes interference more than the actual transmission antenna signal power. The coefficient β (β is a constant of 1 or more) is calculated.

ウエイト計算部４０は、ＦＦＴ部１２−１−１〜１２−Ｍ−Ｎで周波数領域に変換した伝送路のインパルス応答、チップ雑音推定部１６で推定したチップ雑音電力および補正係数計算部４１で計算した補正係数βを入力し、最小平均自乗誤差法（ＭＭＳＥ）により、フィルタのウエイトを計算する。   The weight calculation unit 40 calculates the impulse response of the transmission path converted into the frequency domain by the FFT units 12-1-1 to 12 -MN, the chip noise power estimated by the chip noise estimation unit 16, and the correction coefficient calculation unit 41. The correction coefficient β is input, and the weight of the filter is calculated by the minimum mean square error method (MMSE).

ＧＩ除去部１３−１〜１３−Ｎは、受信アンテナ３−１〜３−Ｎで受信した信号を入力し、受信パスタミングを基準にして、ＧＩに相当する部分の受信信号を除去する。   The GI removal units 13-1 to 13-N receive the signals received by the reception antennas 3-1 to 3-N, and remove received signals corresponding to the GI with reference to reception path timing.

Ｓ／Ｐ変換部１４−１〜１４−Ｎは、ＧＩ除去部１３−１〜１３−ＮでＧＩを除去した受信信号をＳ／Ｐ変換する。   The S / P converters 14-1 to 14-N perform S / P conversion on the reception signals from which the GI has been removed by the GI removers 13-1 to 13-N.

ＦＦＴ部１５−１〜１５−Ｎは、Ｓ／Ｐ変換部１４−１〜１４−Ｎで変換した受信信号を入力し、周波数領域に変換する。   The FFT units 15-1 to 15-N receive the reception signals converted by the S / P conversion units 14-1 to 14-N and convert them into the frequency domain.

フィルタ処理部１８は、ウエイト計算部４０で計算したウエイトおよびＦＦＴ部１５−１〜１５−Ｎで周波数変換した受信信号を入力し、周波数領域で受信信号のフィルタリング（等化）を行う。   The filter processing unit 18 receives the weight calculated by the weight calculation unit 40 and the reception signal frequency-converted by the FFT units 15-1 to 15-N, and performs filtering (equalization) of the reception signal in the frequency domain.

フィルタ処理部１８は、他の送信アンテナからの干渉を抑圧すると同時に希望の送信アンテナ信号のマルチパス干渉も抑圧する。   The filter processing unit 18 suppresses multipath interference of a desired transmission antenna signal at the same time as suppressing interference from other transmission antennas.

ＩＦＦＴ部１９−１〜１９−Ｍは、フィルタ処理部１８で等化した周波数領域の信号を入力し、時間領域に変換する。   The IFFT units 19-1 to 19-M receive the frequency domain signal equalized by the filter processing unit 18 and convert it into the time domain.

Ｐ／Ｓ変換部２０−１〜２０−Ｍは、時間領域に変換した信号をＰ／Ｓ変換する。   The P / S conversion units 20-1 to 20-M perform P / S conversion on the signal converted into the time domain.

逆拡散回路２１−１〜２１−Ｍは、Ｐ／Ｓ変換部２０−１〜２０−Ｍで変換した時間領域の信号を入力し、逆拡散を行い、送信アンテナ１−１〜１−Ｍからの信号を復調する。   The despreading circuits 21-1 to 21-M receive time domain signals converted by the P / S conversion units 20-1 to 20-M, perform despreading, and transmit from the transmission antennas 1-1 to 1-M. Is demodulated.

本発明の実施例の動作について、図を参照して説明する。ここでは特に図１に示されたウエイト計算部４０について詳細に説明する。図２は、本発明のＦＦＴ後のサブキャリアｆ（１≦ｆ≦Ｆ）における送信アンテナｍ（１≦ｍ≦Ｍ）のウエイト計算部４０の構成を示すブロック図である。サブキャリアｆにおける送信アンテナｍのウエイト計算部４０は、第一相関行列生成部５０、第二相関行列生成部５１−１〜５１−Ｍ、補正係数乗算部５２、相関行列加算部５３、雑音加算部５４、逆行列演算部５５およびウエイト生成部５６で構成され、各サブキャリアおよび各送信アンテナでは同一の構成を有する。   The operation of the embodiment of the present invention will be described with reference to the drawings. Here, the weight calculator 40 shown in FIG. 1 will be described in detail. FIG. 2 is a block diagram showing a configuration of the weight calculation unit 40 of the transmission antenna m (1 ≦ m ≦ M) in the subcarrier f (1 ≦ f ≦ F) after the FFT of the present invention. The weight calculation unit 40 of the transmission antenna m in the subcarrier f includes a first correlation matrix generation unit 50, second correlation matrix generation units 51-1 to 51-M, a correction coefficient multiplication unit 52, a correlation matrix addition unit 53, and noise addition. Unit 54, inverse matrix calculation unit 55, and weight generation unit 56. Each subcarrier and each transmission antenna have the same configuration.

第一相関行列生成部５０は、図１のＦＦＴ部１２−１−１〜１２−Ｍ−Ｎで周波数領域に変換した希望の送信アンテナｍと受信アンテナ間の伝送路推定値を入力し、送信アンテナｍから受信アンテナへの相関行列を生成する。   The first correlation matrix generation unit 50 inputs a transmission path estimation value between the desired transmission antenna m and the reception antenna converted into the frequency domain by the FFT units 12-1-1-1 to 12-MN of FIG. A correlation matrix from the antenna m to the receiving antenna is generated.

第二相関行列生成部５１−１〜５１−Ｍは、図１のＦＦＴ部１２−１−１〜１２−Ｍ−Ｎで周波数領域に変換した干渉となる送信アンテナと受信アンテナ間の伝送路推定値を入力し、干渉となる送信アンテナから受信アンテナへの相関行列を生成する。   The second correlation matrix generation units 51-1 to 51-M estimate the transmission path between the transmission antenna and the reception antenna that become interference converted into the frequency domain by the FFT units 12-1-1-1 to 12-MN of FIG. A value is input, and a correlation matrix from the transmitting antenna to the receiving antenna that causes interference is generated.

補正係数乗算部５２は、第二相関行列生成部５１−１〜５１−Ｍで生成した干渉となる送信アンテナの相関行列および図１の補正係数計算部４１で計算した補正係数βを入力し、補正係数βを送信アンテナ毎の相関行列にそれぞれ乗算し、希望の送信アンテナｍの相関行列よりも干渉となる送信アンテナの相関行列を実際よりも大きくみなす補正を行う。   The correction coefficient multiplication unit 52 inputs the correlation matrix of the transmission antenna that is the interference generated by the second correlation matrix generation units 51-1 to 51-M and the correction coefficient β calculated by the correction coefficient calculation unit 41 of FIG. A correction coefficient β is multiplied by the correlation matrix for each transmission antenna, and correction is performed so that the correlation matrix of the transmission antenna that causes interference is larger than the correlation matrix of the desired transmission antenna m.

相関行列加算部５３は、第一相関行列生成部５０で生成した希望の送信アンテナｍの相関行列および補正係数乗算部５２で補正係数βを乗算した干渉となる送信アンテナの相関行列を入力し、それぞれの相関行列を加算する。   The correlation matrix adding unit 53 inputs the correlation matrix of the desired transmission antenna m generated by the first correlation matrix generation unit 50 and the correlation matrix of the transmission antenna that becomes interference multiplied by the correction coefficient β by the correction coefficient multiplication unit 52, Add each correlation matrix.

雑音加算部５４は、相関行列加算部５３で加算した相関行列および図１のチップ雑音推定部１６で推定したチップ雑音電力を入力し、相関行列に雑音成分を加算する。   The noise adder 54 receives the correlation matrix added by the correlation matrix adder 53 and the chip noise power estimated by the chip noise estimator 16 of FIG. 1, and adds a noise component to the correlation matrix.

逆行列演算部５５は、雑音加算部５４で雑音成分を加算した相関行列を入力し、逆行列演算を行う。   The inverse matrix calculation unit 55 inputs the correlation matrix obtained by adding the noise component by the noise addition unit 54 and performs an inverse matrix calculation.

ウエイト生成部５６は、逆行列演算部５５で演算した逆行列および周波数領域に変換した希望の送信アンテナｍと受信アンテナ間の伝送路推定値を入力し、フィルタのウエイトを生成する。   The weight generation unit 56 inputs the inverse matrix calculated by the inverse matrix calculation unit 55 and the transmission path estimation value between the desired transmission antenna m and the reception antenna converted into the frequency domain, and generates the weight of the filter.

以上の処理に関して、数式を用いて詳細に説明する。ＦＦＴ部１２−１−１〜１２−Ｍ−Ｎで伝送路のインパルス応答を周波数領域に変換したサブキャリアｆにおける送信アンテナｍと受信アンテナ間の伝送路ベクトルＨｍ（ｆ）の定義を式（１）のように、ＦＦＴ部１５−１〜１５−Ｎで受信信号を周波数領域に変換したサブキャリアｆにおける受信信号ベクトルＸ（ｆ）を式（２）のように定義すると、図１のウエイト計算部４０で計算されるサブキャリアｆにおける送信アンテナｍのフィルタのウエイトベクトルＷｍ（ｆ）は、次式で表される。 The above processing will be described in detail using mathematical expressions. The definition of the transmission path vector H m (f) between the transmission antenna m and the reception antenna in the subcarrier f obtained by converting the impulse response of the transmission path into the frequency domain by the FFT units 12-1-1 to 12 -MN When the received signal vector X (f) in the subcarrier f obtained by converting the received signal into the frequency domain by the FFT units 15-1 to 15-N as defined in 1) is defined as in Expression (2), the weight shown in FIG. The weight vector W m (f) of the filter of the transmission antenna m in the subcarrier f calculated by the calculation unit 40 is expressed by the following equation.

Figure 2006303691
ここで、補正係数計算部４１で計算した補正係数βは、伝送路の直交性に依存した値であり、例えば、次式のように求められる。
Figure 2006303691
 Here, the correction coefficient β calculated by the correction coefficient calculation unit 41 is a value depending on the orthogonality of the transmission path, and is obtained, for example, by the following equation.

Figure 2006303691
ここで、ＳＦは逆拡散回路２１−１〜２１−Ｍの拡散率（データの拡散率）を表す。式（８）で分母は実際の干渉となる送信アンテナ信号の電力であり、分子は逆拡散回路２１−１〜２１−Ｍの出力で干渉となる送信アンテナ信号を十分に抑圧できるようにウエイト計算で大きくみなす干渉電力である。この電力は干渉となる送信アンテナ信号の電力を逆拡散利得が得られない同一コードで同一パスの干渉成分ＳＦρＤと、逆拡散利得が得られる同一コードで異なるパスの干渉成分（１−ρ）Ｄおよび異なるコードの干渉成分Ｐ＋（Ｋ−１）Ｄとの和とみなして求める。
Figure 2006303691
 Here, SF represents the spreading factor (data spreading factor) of the despreading circuits 21-1 to 21-M. In equation (8), the denominator is the power of the transmission antenna signal that becomes the actual interference, and the numerator calculates the weight so that the transmission antenna signal that causes the interference can be sufficiently suppressed by the output of the despreading circuits 21-1 to 21-M. This is the interference power that is considered to be large. This power is an interference component SFρD of the same path with the same code where despread gain is not obtained, and an interference component (1-ρ) D of a different path with the same code where despread gain is obtained. And the sum of interference components P + (K−1) D of different codes.

以上説明したように、ウエイト計算部４０では、伝送路の直交性に基づいた補正係数βを用いて、希望の送信アンテナ信号の電力よりも干渉となる送信アンテナ信号の電力を実際よりも大きくみなす補正を行い、フィルタのウエイトを計算するため、各送信アンテナで同一のコードセットを用いるＤＳ−ＣＤＭＡのＭＩＭＯにおいて、逆拡散回路２１−１〜２１−Ｍの出力で他の送信アンテナの干渉が十分に抑圧された送信アンテナ１−１〜１−Ｍからの信号を復調できる。   As described above, the weight calculation unit 40 uses the correction coefficient β based on the orthogonality of the transmission path and regards the power of the transmission antenna signal that causes interference more than the actual power of the transmission antenna signal. In DS-CDMA MIMO using the same code set for each transmission antenna to perform correction and calculate the filter weight, interference from other transmission antennas is sufficient at the output of the despreading circuits 21-1 to 21-M. It is possible to demodulate the signals from the transmission antennas 1-1 to 1 -M suppressed to.

また、上記の実施例では、ウエイト計算部４０およびフィルタ処理部１８は、周波数領域の信号処理を行っているが、時間領域の信号処理を行っても同様の効果を得ることができ、本発明は適用できる。   In the above-described embodiment, the weight calculation unit 40 and the filter processing unit 18 perform frequency domain signal processing. However, the same effect can be obtained by performing time domain signal processing. Is applicable.

また、本発明は移動通信システムの基地局無線装置および移動局無線装置のどちらにも適用できる。   Further, the present invention can be applied to both a base station radio apparatus and a mobile station radio apparatus of a mobile communication system.

本発明によれば、優れた干渉抑圧特性を有するＭＩＭＯ受信装置を実現することができる。   According to the present invention, a MIMO receiving apparatus having excellent interference suppression characteristics can be realized.

本発明のＭＩＭＯ受信装置の一実施例を示す構成図である。It is a block diagram which shows one Example of the MIMO receiver of this invention. 本発明のウエイト計算部の構成を示すブロック図である。It is a block diagram which shows the structure of the weight calculation part of this invention. ＭＩＭＯ送受信装置の構成を示す図である。It is a figure which shows the structure of a MIMO transmission / reception apparatus. 従来のＭＩＭＯ受信装置の一例を示す構成図である。It is a block diagram which shows an example of the conventional MIMO receiver. 従来のウエイト計算部の構成を示すブロック図である。It is a block diagram which shows the structure of the conventional weight calculation part.

符号の説明Explanation of symbols

１−１〜１−Ｍ 送信アンテナ
２ 送信装置
３−１〜３−Ｎ 受信アンテナ
４ 受信装置
１０−１−１〜１０−Ｍ−Ｎ 伝送路推定部
１１−１−１〜１１−Ｍ−Ｎ Ｓ／Ｐ変換部
１２−１−１〜１２−Ｍ−Ｎ ＦＦＴ部
１３−１〜１３−Ｎ ＧＩ除去部
１４−１〜１４−Ｎ Ｓ／Ｐ変換部
１５−１〜１５−Ｎ ＦＦＴ部
１６ チップ雑音推定部
１７ ウエイト計算部
１８ フィルタ処理部
１９−１〜１９−Ｍ ＩＦＦＴ部
２０−１〜２０−Ｍ Ｐ／Ｓ変換部
２１−１〜２１−Ｍ 逆拡散回路
３０−１〜３０−Ｍ 相関行列生成部
３１ 相関行列加算部
３２ 雑音加算部
３３ 逆行列演算部
３４−１〜３４−Ｍ ウエイト生成部
４０ ウエイト計算部
４１ 補正係数計算部
４２ パスサーチ部
４３ 伝送路直交係数計算部
５０ 第一相関行列生成部
５１−１〜５１−Ｍ 第二相関行列生成部
５２ 補正係数乗算部
５３ 相関行列加算部
５４ 雑音加算部
５５ 逆行列演算部
５６ ウエイト生成部 1-1 to 1-M transmitting antenna 2 transmitting apparatus 3-1 to 3-N receiving antenna 4 receiving apparatus 10-1-1 to 10-MN transmission path estimation units 11-1-1 to 11-MN S / P converters 12-1-1 to 12 -MN FFT units 13-1 to 13 -N GI removal units 14-1 to 14 -N S / P converters 15-1 to 15 -N FFT unit 16 Chip noise estimation unit 17 Weight calculation unit 18 Filter processing units 19-1 to 19-M IFFT units 20-1 to 20-M P / S conversion units 21-1 to 21-M Despreading circuits 30-1 to 30-M Correlation matrix generation unit 31 Correlation matrix addition unit 32 Noise addition unit 33 Inverse matrix operation units 34-1 to 34-M Weight generation unit 40 Weight calculation unit 41 Correction coefficient calculation unit 42 Path search unit 43 Transmission path orthogonal coefficient calculation unit 50 One correlation matrix generators 51-1 to 51-M Second correlation matrix Generating section 52 correction coefficient multiplication unit 53 correlation matrix addition unit 54 the noise adding section 55 inverse matrix calculator 56 weight generating unit

Claims (12)

複数の送信アンテナから送信したＣＤＭＡ方式の信号を、複数の受信アンテナで受信する受信装置であって、
伝送路の直交性に基づいて、希望の送信アンテナ信号の電力よりも干渉となる送信アンテナ信号の電力を実際より大きくみなす補正係数（１以上の定数）を計算する補正係数計算手段と、
前記補正係数と全ての送受信アンテナ間の伝送路推定値とを用いて、最小平均自乗誤差法（ＭＭＳＥ:Minimum Mean Square Error)に基づいて、受信信号をフィルタリングするフィルタのウエイトを計算するウエイト計算手段と、
前記ウエイトで受信信号をフィルタリングし、干渉となる送信アンテナ信号を抑圧し、希望の送信アンテナ信号を復調するフィルタ処理部と
を有することを特徴とするＭＩＭＯ受信装置。 A receiving device that receives signals of a CDMA system transmitted from a plurality of transmitting antennas by a plurality of receiving antennas,
Correction coefficient calculation means for calculating a correction coefficient (a constant of 1 or more) that considers the power of the transmission antenna signal that causes interference more than the actual power based on the orthogonality of the transmission path;
Weight calculation means for calculating a weight of a filter for filtering a received signal based on a minimum mean square error (MMSE) method using the correction coefficient and transmission path estimation values between all transmitting and receiving antennas. When,
A MIMO receiving apparatus comprising: a filter processing unit that filters a reception signal with the weight, suppresses a transmission antenna signal that causes interference, and demodulates a desired transmission antenna signal. 前記補正係数計算手段は、干渉となる送信アンテナ信号の電力を、逆拡散利得が得られない同一コードで同一パスの干渉成分と逆拡散利得が得られる同一コードで異なるパスの干渉成分および異なるコードの干渉成分との和とみなして前記補正係数を計算することを特徴とする請求項１記載のＭＩＭＯ受信装置。   The correction coefficient calculation means calculates the interference component of the different path in the same code that can obtain the interference component of the same path and the despread gain with the same code for which the despread gain cannot be obtained for the power of the transmission antenna signal that causes interference. The MIMO receiving apparatus according to claim 1, wherein the correction coefficient is calculated on the assumption that the correction coefficient is a sum of the interference component and the interference component. 前記ウエイト計算手段および前記フィルタ処理部は、時間領域の信号処理で行うことを特徴とする請求項１記載のＭＩＭＯ受信装置。   The MIMO receiving apparatus according to claim 1, wherein the weight calculation unit and the filter processing unit perform signal processing in a time domain. 前記ウエイト計算手段および前記フィルタ処理部は、周波数領域の信号処理で行うことを特徴とする請求項１記載のＭＩＭＯ受信装置。   The MIMO receiving apparatus according to claim 1, wherein the weight calculation unit and the filter processing unit perform signal processing in a frequency domain. 前記ウエイト計算手段は、周波数領域の信号処理であって、
周波数領域で表された希望の送信アンテナと受信アンテナとの間の伝送路推定値から相関行列を生成する第一の相関行列生成手段と、
周波数領域で表された干渉となる送信アンテナと受信アンテナとの間の伝送路推定値から送信アンテナ毎に相関行列を生成する第二の相関行列生成手段と、
前記第二の相関行列生成手段で生成した送信アンテナ毎の相関行列に前記補正係数計算手段で計算した前記補正係数をそれぞれ乗算する補正係数乗算手段と、
前記補正係数を乗算した送信アンテナ毎の相関行列と前記第一の相関行列生成手段で生成した相関行列とを全て加算する相関行列加算手段と、
前記相関行列加算手段で加算した相関行列に雑音成分を加算する雑音電力加算手段と、
前記雑音電力加算手段で雑音成分を加算した相関行列の逆行列を演算する逆行列演算手段と、
前記逆行列演算手段で演算した相関行列の逆行列と周波数領域で表された希望の送信アンテナと受信アンテナとの間の伝送路推定値からウエイトを生成するウエイト生成手段と
を有することを特徴とする請求項１記載のＭＩＭＯ受信装置。 The weight calculation means is frequency domain signal processing,
First correlation matrix generation means for generating a correlation matrix from a channel estimation value between a desired transmission antenna and reception antenna expressed in a frequency domain;
Second correlation matrix generation means for generating a correlation matrix for each transmission antenna from a transmission path estimation value between the transmission antenna and the reception antenna that causes interference expressed in the frequency domain;
Correction coefficient multiplication means for multiplying the correlation matrix for each transmission antenna generated by the second correlation matrix generation means by the correction coefficient calculated by the correction coefficient calculation means, respectively.
Correlation matrix addition means for adding all of the correlation matrix for each transmission antenna multiplied by the correction coefficient and the correlation matrix generated by the first correlation matrix generation means;
Noise power adding means for adding a noise component to the correlation matrix added by the correlation matrix adding means;
An inverse matrix calculating means for calculating an inverse matrix of a correlation matrix obtained by adding a noise component by the noise power adding means;
Weight generating means for generating a weight from an inverse matrix of a correlation matrix calculated by the inverse matrix calculating means and a transmission path estimation value between a desired transmitting antenna and a receiving antenna expressed in a frequency domain, The MIMO receiver according to claim 1. 複数の送信アンテナから送信したＣＤＭＡ方式の信号を、複数の受信アンテナで受信する受信装置であって、
受信信号を入力とし伝送路の遅延プロファイルを生成し、前記遅延プロファイルからレベルの大きな複数のパスを検出するパスサーチ部と、
前記パスサーチ部で検出された複数のパスから伝送路の直交係数を計算する伝送路直交係数計算手段と、
前記伝送路直交係数に基づいて、希望の送信アンテナ信号の電力よりも干渉となる送信アンテナ信号の電力を実際より大きくみなす補正係数（１以上の定数）を計算する補正係数計算手段と、
前記補正係数と全ての送受信アンテナ間の伝送路推定値とを用いて、最小平均自乗誤差法（ＭＭＳＥ）に基づいて、受信信号をフィルタリングするフィルタのウエイトを計算するウエイト計算手段と、
前記ウエイトで受信信号をフィルタリングし、干渉となる送信アンテナ信号を抑圧し、希望の送信アンテナ信号を復調するフィルタ処理部と
を有することを特徴とするＭＩＭＯ受信装置。 A receiving device that receives signals of a CDMA system transmitted from a plurality of transmitting antennas by a plurality of receiving antennas,
A path search unit that receives a received signal as input and generates a delay profile of a transmission path, and detects a plurality of paths having a large level from the delay profile;
A transmission path orthogonal coefficient calculating means for calculating an orthogonal coefficient of a transmission path from a plurality of paths detected by the path search unit;
Correction coefficient calculation means for calculating a correction coefficient (a constant of 1 or more) that considers the power of the transmission antenna signal that causes interference more than the actual power of the transmission antenna signal based on the transmission path orthogonal coefficient;
A weight calculating means for calculating a weight of a filter for filtering a received signal based on a minimum mean square error method (MMSE) using the correction coefficient and a transmission path estimation value between all transmitting and receiving antennas;
A MIMO receiving apparatus comprising: a filter processing unit that filters a reception signal with the weight, suppresses a transmission antenna signal that causes interference, and demodulates a desired transmission antenna signal. 前記補正係数計算手段は、干渉となる送信アンテナ信号の電力を、逆拡散利得が得られない同一コードで同一パスの干渉成分と逆拡散利得が得られる同一コードで異なるパスの干渉成分および異なるコードの干渉成分との和とみなして前記補正係数を計算することを特徴とする請求項６記載のＭＩＭＯ受信装置。   The correction coefficient calculation means calculates the interference component of the different path in the same code that can obtain the interference component of the same path and the despread gain with the same code for which the despread gain cannot be obtained for the power of the transmission antenna signal that causes interference. The MIMO receiving apparatus according to claim 6, wherein the correction coefficient is calculated by regarding the sum as an interference component. 前記ウエイト計算手段および前記フィルタ処理部は、時間領域の信号処理で行うことを特徴とする請求項６記載のＭＩＭＯ受信装置。   The MIMO receiving apparatus according to claim 6, wherein the weight calculation unit and the filter processing unit perform time-domain signal processing. 前記ウエイト計算手段および前記フィルタ処理部は、周波数領域の信号処理で行うことを特徴とする請求項６記載のＭＩＭＯ受信装置。   The MIMO receiving apparatus according to claim 6, wherein the weight calculation unit and the filter processing unit perform signal processing in a frequency domain. 前記ウエイト計算手段は、周波数領域の信号処理であって、
周波数領域で表された希望の送信アンテナと受信アンテナとの間の伝送路推定値から相関行列を生成する第一の相関行列生成手段と、
周波数領域で表された干渉となる送信アンテナと受信アンテナとの間の伝送路推定値から送信アンテナ毎に相関行列を生成する第二の相関行列生成手段と、
前記第二の相関行列生成手段で生成した送信アンテナ毎の相関行列に前記補正係数計算手段で計算した前記補正係数をそれぞれ乗算する補正係数乗算手段と、
前記補正係数を乗算した送信アンテナ毎の相関行列と前記第一の相関行列生成手段で生成した相関行列とを全て加算する相関行列加算手段と、
前記相関行列加算手段で加算した相関行列に雑音成分を加算する雑音電力加算手段と、
前記雑音電力加算手段で雑音成分を加算した相関行列の逆行列を演算する逆行列演算手段と、
前記逆行列演算手段で演算した相関行列の逆行列と周波数領域で表された希望の送信アンテナと受信アンテナとの間の伝送路推定値からウエイトを生成するウエイト生成手段と
を有することを特徴とする請求項６記載のＭＩＭＯ受信装置。 The weight calculation means is frequency domain signal processing,
First correlation matrix generation means for generating a correlation matrix from a channel estimation value between a desired transmission antenna and reception antenna expressed in a frequency domain;
Second correlation matrix generation means for generating a correlation matrix for each transmission antenna from a transmission path estimation value between the transmission antenna and the reception antenna that causes interference expressed in the frequency domain;
Correction coefficient multiplication means for multiplying the correlation matrix for each transmission antenna generated by the second correlation matrix generation means by the correction coefficient calculated by the correction coefficient calculation means, respectively.
Correlation matrix addition means for adding all of the correlation matrix for each transmission antenna multiplied by the correction coefficient and the correlation matrix generated by the first correlation matrix generation means;
Noise power adding means for adding a noise component to the correlation matrix added by the correlation matrix adding means;
An inverse matrix calculating means for calculating an inverse matrix of a correlation matrix obtained by adding a noise component by the noise power adding means;
Weight generating means for generating a weight from an inverse matrix of a correlation matrix calculated by the inverse matrix calculating means and a transmission path estimation value between a desired transmitting antenna and a receiving antenna expressed in a frequency domain, The MIMO receiver according to claim 6. 複数の送信アンテナからＣＤＭＡ方式で送信した信号を、複数の受信アンテナで受信する受信装置に適用される受信方法であって、
伝送路の直交性に基づいて、希望の送信アンテナ信号の電力よりも干渉となる送信アンテナ信号の電力を実際より大きくみなす補正係数（１以上の定数）を計算し、前記補正係数と全ての送受信アンテナ間の伝送路推定値とを用いて、最小平均自乗誤差法（ＭＭＳＥ）に基づいて、受信信号をフィルタリングするフィルタのウエイトを計算し、前記ウエイトで受信信号をフィルタリングし、干渉となる送信アンテナ信号を抑圧し、希望の送信アンテナ信号を復調することを特徴とするＭＩＭＯ受信方法。 A receiving method applied to a receiving apparatus that receives signals transmitted from a plurality of transmitting antennas by a CDMA system using a plurality of receiving antennas,
Based on the orthogonality of the transmission path, a correction coefficient (a constant greater than or equal to 1) that considers the power of the transmission antenna signal that causes interference to be greater than the power of the desired transmission antenna signal is calculated. Based on the minimum mean square error method (MMSE), the transmission weight between the antennas is calculated based on the minimum mean square error method (MMSE), and the weight of the filter for filtering the reception signal is calculated. A MIMO receiving method comprising suppressing a signal and demodulating a desired transmitting antenna signal. 複数の送信アンテナを備え、各送信アンテナで同一コードセットを用いるＣＤＭＡ信号を送信する送信装置と、
請求項１あるいは請求項６のいずれかに記載の受信装置と
を有することを特徴とする無線通信システム。 A transmission apparatus comprising a plurality of transmission antennas, each transmitting antenna for transmitting a CDMA signal using the same code set;
A wireless communication system comprising: the receiving device according to claim 1.
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Cited By (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2010527186A (en) * 2007-05-10 2010-08-05 アルカテル−ルーセント Method and apparatus for preprocessing data transmitted in a multiple input communication system
US8019031B2 (en) 2007-07-02 2011-09-13 Nec Corporation User selection method and user selection device for multiuser MIMO communication
KR101481070B1 (en) * 2013-04-04 2015-01-12 전북대학교산학협력단 MMSE Filtering Apparatus and Method with Information-Theoretic Limits in Quantity and Quality of Signal Transmission

Cited By (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2010527186A (en) * 2007-05-10 2010-08-05 アルカテル−ルーセント Method and apparatus for preprocessing data transmitted in a multiple input communication system
US8019031B2 (en) 2007-07-02 2011-09-13 Nec Corporation User selection method and user selection device for multiuser MIMO communication
KR101481070B1 (en) * 2013-04-04 2015-01-12 전북대학교산학협력단 MMSE Filtering Apparatus and Method with Information-Theoretic Limits in Quantity and Quality of Signal Transmission

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