JP2006250754A - Radar-type probing device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To obtain a phase change between the transmission wave and the reception wave with a simple configuration. <P>SOLUTION: A phase of a carrier signal is modulated by a lamp signal, and waveform is shaped in a waveform shaping circuit 8 so that its falling or rising is steeper, and thus generated higher-order harmonics wave is transmitted from a transmitting antenna 10. A reflection wave reflected by a target is received by a receiving antenna 11. Not only the microwave transmitted by the transmitting antenna, but also any of the reflection wave received by the receiving antenna is made to be the modulated wave by the lamp signal. Therefore, a base band signal including phase modulation can be obtained by a simple circuit configuration by a low-pass filter, and the phase change by the phase modulation of both of them can be obtained with the usage of a differential amplifier 23. A cut-off frequency of the low-pass filter is set to be the frequency corresponding to a ratio of the frequency of the carrier signal and the frequency of the lamp signal. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention can also be applied to an underground radar or an air radar capable of exploring underground objects such as landmines. In particular, the phase change between a transmitted wave and a received wave can be easily simplified. The present invention relates to a radar-type exploration device that can be obtained by configuration.

  For example, landmines such as antipersonnel mines are used in conflict areas around the world, and many remain buried in the ground even after the end of the conflict, and their exploration and removal are extremely difficult. As a method of searching for landmines buried in the ground, a method using a metal detector is the mainstream, but most of the above-mentioned antipersonnel mines use materials other than metal, for example, plastic, and antipersonnel mines are converted into metal. It is often difficult to detect only with a detector.

Therefore, the microwave from the transmitting antenna is propagated into the ground, the reflected wave is detected by the receiving antenna, and the detected reflected wave is image-processed to determine the presence or absence of the object and its embedded position (distance). Many proposals of ground penetrating radar to be verified have been made, for example, disclosed in Patent Documents 1 to 3 shown below.
JP-A-5-142343 JP 2002-228760 A JP 2002-228599 A

  By the way, in this type of radar type exploration device, the relative permittivity varies greatly depending on whether the main material of the above-mentioned landmine as a target is metal or plastic. There arises a problem that it must be selected according to the object.

  In other words, a dedicated ground penetrating radar for metal and a dedicated ground penetrating radar for plastic described above must be prepared and used separately. Therefore, not only is the inefficient exploration work forced, but there is also an economic burden due to the provision of dedicated exploration devices.

  Therefore, the present applicant has proposed a radar type exploration device having versatility that can be used for a wide range of targets regardless of the nature of the target. The present invention can be suitably applied to the search device having the above-described configuration, and in particular, to provide a radar-type search device that can acquire a phase change between a transmission wave and a reception wave with a simple configuration. It is a subject.

  The radar-type exploration device according to the present invention, which has been made to solve the above-mentioned problems, phase-modulates a carrier signal using a ramp signal as a modulated wave, and a microwave at the rising or falling timing of the phase-modulated carrier signal. A signal is transmitted from a transmitting antenna, and a reflected wave of the microwave signal is received by a receiving antenna, and a phase modulation component of the microwave signal on the transmission side and a phase modulation component of the microwave signal on the reception side A radar-type exploration device provided with a phase change acquisition means for acquiring a phase change between the first and second phase change acquisition means for extracting a phase modulation component of the microwave signal on the transmission side. A low-pass filter, a second low-pass filter for extracting a phase modulation component of the microwave signal on the reception side, and the first And a differential extraction means capable of acquiring a change in phase in the phase modulation component respectively obtained by the second low-pass filter, and the respective cut-off frequencies in the first and second low-pass filters are It is characterized in that it is set to a frequency corresponding to the ratio of the frequency of the carrier signal and the frequency of the ramp signal.

  In this case, the difference extracting means is preferably constituted by a differential amplifier that extracts a difference between a baseband signal obtained by the first low-pass filter and a baseband signal obtained by the second low-pass filter. In addition, the carrier signal and the scan signal are preferably configured to be obtained based on frequency-divided outputs obtained by dividing the reference signal from the reference signal generation unit.

  The radar type exploration apparatus according to the present invention further comprises waveform shaping means for shaping the rising or falling edge of the carrier signal phase-modulated by the ramp signal into a steeper form, and the waveform shaping means A microwave signal including high-order harmonics obtained by the above is configured to be transmitted by the transmission antenna.

  According to the radar type exploration device described above, the carrier signal is phase-modulated by the ramp signal, and the microwave signal is transmitted from the transmitting antenna at the falling timing of the phase-modulated carrier signal, for example. On the other hand, the carrier signal in the reflected wave received by the receiving antenna is phase-modulated by the ramp signal. Therefore, by verifying the phase modulation component without verifying the high-frequency carrier signal, The amount of phase change can be captured.

  In order to realize this, low-pass filters are used as detection means for transmission and reception signals, respectively, and the difference in the baseband signal is verified through the low-pass filter, thereby capturing the phase change during reception with respect to transmission. This makes it possible to calculate the position of the target. Therefore, the radar type exploration apparatus according to the present invention can contribute to greatly simplifying the circuit configuration.

  Hereinafter, a radar type exploration device according to the present invention will be described based on an embodiment shown in the drawings. FIG. 1 is a block diagram showing the overall configuration of the exploration device. The main part of the exploration device shown in FIG. 1 is preferably configured by ASIC (Application Specific Integrated Circuit), and the terminal indicated by a circle is between a CPU (Central Processing Unit) or an external circuit (not shown). The input / output terminals are shown.

  In this exploration device, for example, a reference signal generation unit 1 using a crystal is provided, and a reference signal from the reference signal generation unit 1 is used as a reference clock signal in the CPU and a first frequency division. The frequency is divided by the device 2 and supplied to the carrier signal generator 3. In this carrier signal generation unit 3, a rectangular wave signal of 2 MHz to 5 MHz, for example, is generated by the divided output from the first frequency divider 2, but in this embodiment, the carrier signal generation unit 3 Description will be made on the assumption that a 2 MHz rectangular wave signal is generated.

  On the other hand, the output from the first frequency divider 2 is supplied to the second frequency divider 4, and the output from the second frequency divider 4 is supplied to the scan signal generator 5 and the synchronization signal generator 6. Is done. 2A shows a rectangular carrier signal output from the carrier signal generator 3, and FIG. 2B shows a scan signal output from the scan signal generator 5. As shown in FIG. Show. This scan signal generates a step-like sawtooth wave (ramp signal) using the rectangular wave signal output from the second frequency divider 4 described above. It can also be called a ramp signal generation means.

  The scan signal (stepped sawtooth wave) shown in FIG. 2 (b) output from the scan signal generator 5 is shown by a thin solid line in FIG. 2 (b) through the low-pass filter 7 shown in FIG. As shown, the level is converted into a ramp signal that varies linearly. The frequency of the ramp signal is desirably set to about 40 Hz to 80 Hz, for example, but in this embodiment, the description will be made on the assumption that a 40 Hz ramp signal is generated.

  On the other hand, the synchronization signal generator 6 generates a synchronization signal using the rectangular wave signal output from the second frequency divider 4, which is preferably the ramp signal described above. A 40 Hz rectangular wave signal synchronized with the frequency is output. This synchronization signal is used in an external processing circuit together with an analog data signal output from a data output terminal 32 described later.

  The frequency division ratios in the first and second frequency dividers 2 and 4, the frequencies of the ramp signal and the synchronization signal generated by the scan signal generation unit 5 and the synchronization signal generation unit 6 are as follows. , And can be changed as appropriate according to the command from the CPU.

  The frequency of the carrier signal shown in FIG. 2 (a) and the frequency of the ramp signal shown in FIG. 2 (b) are greatly different from each other as described above. Therefore, the signals shown in FIGS. 2 (a) and 2 (b) are different. The time axis of the period in the waveform is not the same. In this embodiment, all of the transmission timing, the reception timing, the operation of the STC circuit, and the like are controlled using the two signals of the carrier signal and the scan signal as described later.

  As shown in FIG. 1, the ramp signal output through the low-pass filter 7 operates so that the carrier signal output from the carrier signal generator 3 is level-shifted by the ramp signal. As a result, the carrier signal undergoes phase modulation by the ramp signal. That is, the intersection point A between the output end of the low-pass filter 7 and the output end of the carrier signal generator 3 substantially constitutes a carrier signal phase modulation means using a ramp signal. FIG. 2C schematically shows the state of the carrier signal that has undergone phase modulation by the ramp signal. The carrier signal of 2 MHz has a degree of phase delay according to the level of the ramp signal of 40 Hz. Undergoes phase modulation.

  In FIG. 2 (c), the interval between arrows indicated by solid lines indicates the half cycle of the carrier signal, and the interval between arrows indicated by broken lines indicates, for example, the half cycle of the carrier signal when the maximum phase modulation is applied. The time position of is illustrated. The carrier signal shown in FIG. 2 (c) is drawn with a much larger period than the carrier signal shown in FIG. 2 (a) for convenience of drawing on the paper, but the period (frequency) of both is Are the same.

  Returning to FIG. 1, the carrier signal that has undergone phase modulation by the ramp signal is configured to be supplied to a waveform shaping circuit 8 as a waveform shaping means. The specific configuration and operation of the waveform shaping circuit 8 will be described in detail later. In this embodiment, the waveform shaping circuit 8 blunts the rise of the phase-modulated carrier signal and accelerates the fall. Waveform shaping processing is performed (steeply). Thereby, high-order harmonics including odd-numbered orders of the carrier signal are generated at the falling timing of the phase-modulated carrier signal.

  The harmonic output generated by the waveform shaping circuit 8 is current-amplified by a transmission amplifier 9 that amplifies a microwave of 200 MHz or higher, and is fed to the transmission cavity antenna 10. Therefore, the transmission cavity antenna 10 transmits a harmonic generated at the falling timing of the phase-modulated carrier signal as a transmission signal, which is propagated toward the ground facing the cavity antenna 10.

  FIG. 3 mainly shows a configuration example of the waveform shaping circuit 8 described above, and FIG. 4 is a timing diagram for explaining the operation of the waveform shaping circuit shown in FIG. In the waveform shaping circuit 8 shown in FIG. 3, a discharge resistor R1 is connected in parallel to a nonpolar capacitor C1, and one end (primary side) of this parallel connection body is subjected to phase modulation by the ramp signal. The carrier signal is configured to be supplied. The other end (secondary side) of the parallel connection body is connected to the base electrode of the npn-type bipolar transistor Q1.

  The transistor Q1 functions as the transmission amplifier 9 described above, has a load resistor R2 at its collector terminal, and is connected to the operating power supply Vcc via the load resistor R2. The collector terminal is connected to the transmitting cavity antenna 10 described above. The emitter terminal of the transistor Q1 is connected to the ground which is the reference potential point of the circuit.

  FIG. 4A shows a form of a carrier signal that has undergone phase modulation by the ramp signal (in the description of the waveform shaping circuit 8, this may be simply referred to as a rectangular wave pulse). FIG. 4B shows a signal after waveform shaping, that is, a base potential applied to the transistor Q1, and FIG. 4C schematically shows one of the signals after waveform shaping shown in FIG. 4B. This is a magnified view. Note that the rectangular wave pulse shown in FIG. 4 (a) has a slight slope at the rise and fall in order to clarify the difference from the signal after waveform shaping shown in FIGS. 4 (b) and 4 (c). Shown exaggerated to exist.

  Here, the rectangular wave pulse shown in FIG. 4A is supplied to the waveform shaping circuit 8 shown in FIG. When the pulse signal rises from 0 V to 5 V, for example, as shown in FIG. 3, the resistor R1 transmits the square wave pulse to the base electrode of the transistor Q1 while attenuating the peak value of the rectangular wave pulse. On the other hand, the capacitor C1 is charged with the polarity shown in FIG. Therefore, the base potential of the transistor Q1 gradually rises toward the peak value of the input pulse due to the charging operation of the capacitor C1. Therefore, at the rising timing of the rectangular wave pulse shown in FIG. 4A, the base potential operates so as to rise slowly as shown in FIG. 4B. This state is indicated by symbol h in FIG.

  The charging operation to the capacitor C1 ends with the passage of time, and the potentials at both ends thereof become equal. As a result, the base potential reaches 5 V, which is the peak value of the rectangular pulse. This state is indicated by symbol i in FIG.

  Next, when the rectangular wave pulse falls and changes from 5V to 0V, the potential of 0V is instantaneously transmitted to the base electrode side via the capacitor C1. At the same time, the resistor R1 connected between both ends of the capacitor C1 acts as a discharge resistor of the capacitor C1, and an operation of pulling the terminal on the base electrode side of the capacitor C1 to the 0V side through the discharge resistor R1 is executed. Therefore, the potential change on the base electrode side of the capacitor C1 is accelerated much more than the falling operation of the rectangular wave pulse, and the falling waveform is shaped so as to be steeper. The capacitor that performs such an operation is also called a speed-up capacitor.

  Therefore, at the falling timing of the rectangular wave pulse, a high-order harmonic signal is generated in which a large number of odd-order harmonics having a carrier wave of 2 MHz as a fundamental wave are superimposed. This state is indicated by a symbol j in FIG. In short, the capacitor C1 and the resistor R1 constituting the waveform shaping circuit 8 are time constants so that the above-described charge and discharge operations are repeated within a period corresponding to a frequency of 2 MHz. Is set.

  The signal waveform shown in FIG. 4C is applied to the base electrode of the transistor Q1 functioning as the transmission amplifier 9. Therefore, in the signal waveform shown in FIG. 4C, in the period in which the potential slowly rises as indicated by the symbol h, the transistor Q1 is turned on when the threshold voltage of the transistor Q1 is reached. When the state indicated by reference symbol i in FIG. 4C reaches the state indicated by reference symbol j, a harmonic signal is instantaneously applied to the base electrode, and the harmonic signal is converted into a current from the collector electrode. Power is supplied to the transmitting cavity antenna 10.

  Note that the output of the above-described harmonic signal is generated at the fall timing of the modulated carrier signal when the 2 MHz carrier signal is phase-modulated at 40 Hz as already described. Therefore, 2 million harmonic signals (microwaves) are transmitted from the antenna 11 per second. Moreover, since the 2 MHz carrier signal is phase-modulated by the 40 Hz ramp signal, the harmonic signal at the end of the 40 Hz period is compared with the output interval of the harmonic signal at the beginning of the 40 Hz period. The output interval increases.

  That is, the operation of gradually changing the output timing of the harmonic signal so as to become coarser from the dense according to the degree of modulation by the ramp signal is repeated within the period corresponding to the scanning period of 40 Hz. This operation will be described in detail in the description of the gate pulse signal generation operation described later.

  The cavity antenna 10 has a configuration in which a copper plate is pasted on a plate having a predetermined dielectric constant, and a high-frequency current is passed from the amplifier 9 at a predetermined point as a feeding point. The high-frequency signal includes extremely large distortion, but by supplying power to the cavity antenna 10 via the linear amplifier 9, the cavity antenna 10 acts as a so-called resonance box, and 2 MHz is used as a fundamental wave. The microwave including the odd-order sine wave is transmitted from the antenna 10. In this case, all the frequencies in the range of, for example, 200 MHz to 2 GHz transmitted from the antenna 10 are added with the above-described 40 Hz modulation.

  On the other hand, the carrier signal that has undergone phase modulation by the ramp signal is supplied to the time axis variable circuit 15 shown in FIG. The time axis variable circuit 15 and the phase shifter 16 are constituted by, for example, a semiconductor delay circuit using a gate IC, which is employed for the purpose of adjusting reception timing (Tx) in a gate pulse to be described later.

  FIG. 2D shows a state in which the time axis of the carrier signal is changed and the phase delay is received by the time axis variable circuit 15 and the phase shifter 16 described above. The time axis variable in the time axis variable circuit 15 and the phase shift amount in the phase shifter 16 are configured to be changed as appropriate according to the command from the CPU.

  Then, the carrier signal having undergone the change of the time axis and the phase delay by the time axis variable circuit 15 and the phase shifter 16 is taken into the CPU, and the signal processing is carried out in the CPU, and the gate shown in FIG. A pulse signal is generated. The gate pulse signal shown in FIG. 2 (e) is synchronized so that it is output at the crossing point between the falling edge of the 2 MHz carrier signal and the ramp signal as shown in FIG. 2 (f). At the output timing Tx of the gate pulse signal, control is performed so that the gate in the reception gate circuit described later is closed and the gate is opened at other times Rx to enter the reception state.

  The 2 MHz carrier signal is phase-modulated according to the level of the ramp signal. As a result, the generation timing of the gate pulse signal is modulated within a period corresponding to a 40 Hz scan period. Gradually changes from dense to coarse. That is, the generation interval of the gate pulse signal Tx shown in FIG. 2E is t1 <t2 <t3... As shown in FIG. This is the same as the output interval of harmonics (microwaves) in the cavity antenna 10 already described. Note that the rectangular wave in FIG. 2 (f) is the carrier signal shown in FIG. 2 (a), and the signal that crosses the rectangular wave obliquely is the ramp signal shown in FIG. 2 (b).

  The gate pulse signal shown in FIG. 2 (e) obtained by signal processing in the CPU is supplied to the reception gate circuit 18 shown in FIG. As a result, the reception gate circuit 18 is closed at the output timing Tx of the harmonics (microwaves) output from the cavity antenna 10, and the reception gate circuit 18 is opened at a timing other than the transmission output timing (reception state) Rx. It acts to receive the echo signal.

  Here, as schematically illustrated in FIG. 5, the microwave including high-order harmonics transmitted from the transmitting cavity antenna 10 is projected onto the target Ta, that is, a land mine buried in the ground in this example. The reflected wave is received by the receiving cavity antenna 11. In this case, any specific frequency included in the harmonics corresponding to the relative permittivity of the target Ta reacts with the target Ta, and the phase is inverted as schematically illustrated in FIG. It returns in the state and is received by the receiving antenna 11.

  The specific frequency signal received by the receiving antenna 11 is a carrier signal having an inverted phase, which is phase-modulated by the ramp signal described above. Here, as a function of the radar, the degree of delay between the transmitted wave and the received reflected wave, that is, the phase change is extracted.

  Therefore, generally, a technique such as Fourier transform must be adopted to capture the carrier signal having the inverted phase and extract the phase change from the frequency change. Therefore, to employ this, the circuit configuration becomes extremely complicated, and a received radio wave having a relatively high strength is required. Thus, in order to obtain a received radio wave having a high intensity, the output power during transmission must be increased.

  Therefore, when trying to capture the phase change at the time of reception with respect to the time of transmission, it is only necessary to verify whether or not the phase is moving at the sample point in 40 Hz. That is, since the modulation waveform is synchronized with the carrier signal, the phase change can be captured by verifying the modulation waveform of 40 Hz without verifying the carrier signal having a very high frequency.

  For this purpose, the transmission signal and the reception signal may be configured so as to remove carrier components through low-pass filters, extract the respective modulated waves, and see the change in phase between the two. That is, it is only necessary to extract a phase shift between the modulated wave at the time of transmission and the modulated wave at the time of reception (which is in reverse phase).

  Each of the low-pass filters described above functions as a means for detecting a transmission signal and a reception signal, so that baseband signals of the transmission and reception signals can be obtained. Each low-pass filter preferably has a cutoff frequency set to a frequency corresponding to the sampling period in order to capture information on whether or not the phase is moving at each sample point in 40 Hz. .

In this embodiment, the frequency corresponding to the sampling period described above can be expressed as a ratio of 2 MHz, which is the frequency of the carrier signal, and 40 Hz, which is the frequency of the ramp signal that performs phase modulation, as shown in Equation 1. be able to.
(2 × 10 ⁶ ) / 40 = 50000 (Formula 1)

  Therefore, by using the above-described low-pass filter having a cutoff frequency of 50 KHz, the low-pass filter functions as if it were a synchronous detector, and can effectively remove the carrier signal, A baseband signal capable of effectively grasping the phase movement at each sample point can be obtained. Then, as the phase change acquisition means for extracting the phase shift amount, the difference between the baseband signal of the transmission signal and the baseband signal of the reception signal is extracted by a differential amplifier, so that the phase shift amount is extracted. Can be acquired effectively and the presence of the target can be grasped.

  Each configuration of FIG. 1 indicated by reference numerals 18 to 23 described below is configured according to the technical viewpoint described above, and the degree of delay between the transmitted wave and the received reflected wave, that is, the amount of phase change. This constitutes a calculation means for verification. First, the reception gate circuit 18 is provided with gates G1 and G2, and the gates G1 and G2 are controlled by the gate pulse signal shown in FIG. Note that the states of the gates G1 and G2 shown in FIG. 1 indicate the reception state Rx in which the gate is open. When the above-described gate pulse signal (Tx) arrives, both the gates G1 and G2 are grounded and the gate is closed. To be made.

  From the common ground point of the transmitting cavity antenna 10 and the receiving cavity antenna 11, the gate G1 is supplied with a signal voltage corresponding to a transmission signal that wraps around the ground point, that is, a harmonic signal that is phase-modulated by a ramp signal. The This is supplied to the first low-pass filter 21 via the voltage amplifier 19. The gate G2 is supplied with a signal voltage corresponding to a signal received by the receiving cavity antenna 11. This is also supplied to the second low-pass filter 22 via the voltage amplifier 20.

  Each of the low-pass filters 21 and 22 is set to a cut-off frequency of 50 KHz as described above. Therefore, the carrier signal component in the transmission signal obtained through the gate G1 is removed, and the modulation component, that is, the scan of 40 Hz. A signal wave (baseband signal) corresponding to the signal can be obtained. This signal wave corresponds to the signal wave modulated by the ramp signal. An example of this baseband signal is shown in the form of a zero cross signal in FIG.

  On the other hand, the received signal received by the receiving antenna 11 is supplied to the low-pass filter 22 via the gate G2 and the voltage amplifier 20 as described above. The low-pass filter 22 also removes the carrier signal component received by the antenna 11 from the relationship set to the cut-off frequency (50 KHz), and as a result, obtains a modulation component, that is, a signal wave corresponding to the scan signal. be able to. This signal wave becomes a reverse phase component of the signal wave modulated by the ramp signal. An example of the signal wave of the reverse phase component is shown in the form of a zero cross signal indicated by a broken line in FIG.

  However, the signal wave of the anti-phase component indicated by the broken line in FIG. 6B shows a state in which no delay occurs with respect to the baseband signal in FIG. There is always a delay corresponding to the time until the transmitted microwave is projected and reflected on the target Ta and returned to the receiving antenna 11. The solid line in FIG. 6B shows an example of a signal waveform delayed with respect to the baseband signal obtained in the low-pass filter 22 described above.

  Here, the baseband signal shown in FIG. 6 (a) is supplied to the non-inverting input terminal of the differential amplifier 23 by the operational amplifier shown in FIG. 1, and the non-inverting input terminal is shown in FIG. 6 (b). The baseband component of the delayed received signal indicated by the solid line is supplied. Therefore, the differential amplifier 23 functions as a difference extracting means, and the difference is output to the output terminal. This differential signal can be shown as an example as shown in FIG. 6C, and a peak value p is generated in synchronization with one period of the baseband signal.

  Therefore, by measuring the time T from the start timing of one period of the baseband signal to the center of the peak value p, the transmission antenna 10 reaches the targate Ta, and the echo arrives at the reception antenna 11. The time, that is, the position of the target can be known. Further, the amplitude L of the peak p shown in FIG. 6 corresponds to the relative dielectric constant of the target Ta, and can be captured as information on which frequency included in the harmonic has reacted. That is, by knowing the amplitude L, it is possible to guess what the target Ta is.

  Here, in the radar type exploration apparatus of this type, the sensitivity of the search is increased when the position of the target Ta with respect to the transmission / reception antennas 10 and 11 is far, and the sensitivity of the search is decreased when the target Ta is close. Is the action. In order to realize the above-described operation, in this embodiment, a voltage control resistor element 29 that functions as a gain adjusting means together with an STC (Sensitive Timing Control) circuit 26 is used.

  That is, the scan signal from the scan signal generator shown in FIG. 1, that is, the ramp signal is configured to be supplied to the STC circuit 26 via the voltage amplifier 25. In this case, the voltage amplifier 25 incorporates an integration circuit as necessary. Therefore, the STC circuit 26 has a ramp signal whose level changes linearly as in the example shown by the thin solid line in FIG. Supplied. This ramp signal is converted into digital data by the A / D conversion circuit 27 and taken into the CPU. The STC circuit 26 is configured so that it can appropriately adjust the level of the ramp signal supplied to the STC circuit 26 in response to a command from the CPU.

  On the other hand, the digital data captured by the CPU is converted into an analog signal by the D / A converter 28 shown in FIG. 1 and supplied to the voltage control resistor element 29. The voltage control resistance element 29 is connected in series with the fixed resistance element R0, and is connected between the output terminal of the differential amplifier 23 using the operational amplifier and the reference potential point. A voltage amplifier 31 is connected to the midpoint of connection between the voltage control resistance element 29 and the fixed resistance element R0, and the voltage amplification output is output to the output terminal 32 as analog data. That is, the voltage control resistance element 29 and the fixed resistance element R0 constitute an electronic attenuator.

  According to the circuit including the STC circuit 26 and the voltage control resistor element 29, the resistance value of the voltage control resistor element 29 is varied according to the level of the ramp signal, and the output of the differential amplifier 27 is transferred between the resistor element R0. The voltage is divided and supplied to the voltage amplifier 31. In this case, when the level of the ramp signal shown in FIG. 2B is low, the resistance value of the voltage control resistance element 29 is increased as the level of the ramp control signal 29 is increased. A preferable gain adjustment can be realized by controlling the value to be low.

  That is, as schematically shown in FIG. 5, when the target Ta exists near the antenna, the echo (reflected wave) returns quickly, and the substantial level of the reflected wave in this case is large. Further, the farther the position of the target Ta with respect to the antenna is, the slower the return of the echo is. In this case, the substantial level of the reflected wave becomes smaller due to attenuation. Therefore, by using the ramp signal in the STC circuit as described above, it is possible to effectively adjust the reception sensitivity of the radar according to the position of the target Ta with respect to the antenna. Thereby, the level of the analog data output from the output terminal 32 can be adjusted to be in a certain range.

  The configuration shown in FIG. 1 is configured such that a synchronization signal of 40 Hz is output from the synchronization signal generation unit 6. Therefore, in an external circuit not shown in FIG. 1, the analog data signal corresponding to FIG. 6C output from the output terminal 32 is digitally converted by the A / D converter, and the synchronization signal generator 6 By using the synchronization signal from, it is possible to determine the property of the target (in other words, whether it is a landmine) based on the position of the target Ta and the relative dielectric constant.

  In the embodiment described above, the waveform shaping circuit 8 that generates the transmission signal executes the waveform shaping process that dulls the rising edge of the carrier signal and accelerates (fallier) the falling edge of the carrier signal. However, on the contrary, the waveform shaping may be performed so that the rising edge of the carrier signal is accelerated (steeper) and the falling edge becomes dull. An example of the signal waveform in that case is shown in FIG. In addition, (a)-(c) shown in FIG. 7 respond | corresponds to (a)-(c) of FIG. 4 already demonstrated, respectively.

  In this case, the gate pulse generation circuit including the time axis variable circuit 15 and the phase shifter 16 described above needs to be configured to output the gate pulse signal in synchronization with the rise of the carrier signal of 2 MHz. .

  As described above, according to the embodiment described above, since the carrier signal is phase-modulated by the ramp signal, not only the microwave transmitted by the transmitting antenna but also the reflected wave received by the receiving antenna The modulated wave is a ramp signal. Therefore, by using a low-pass filter having a specific cut-off frequency as described above, both phase modulation components can be captured, and the target position can be captured by the phase change in both phase modulation components. it can. Therefore, the radar type exploration device according to the present invention can contribute to greatly simplifying the circuit configuration.

  In the above-described embodiment, the time axis variable circuit 15 and the phase shifter 16 are provided to control the delay amount of the phase-modulated carrier signal, and the transmission and reception gate pulses shown in FIG. It is trying to generate. However, the transmission and reception timings can be predicted in advance based on the carrier signal frequency, the ramp signal frequency, the modulation level parameters, and the like. Therefore, transmission and reception gate pulses as shown in FIG. 2E can be generated by measuring time with a counter in the CPU. Therefore, when the above-described counter is used, the above-described time axis variable circuit 15 and phase shifter 16 are not necessary.

  In the above-described embodiment, the higher-order harmonic is generated by shaping the falling or rising of the carrier signal subjected to phase modulation into a steeper waveform. The present invention can also be used for a monopulse type exploration device that does not use the waveform shaping circuit 8 shown in FIG. In this case, the carrier signal generation unit 3 is controlled to generate a carrier signal of about 400 MHz, for example.

  Furthermore, in the above-described embodiment, for example, the underground radar for exploring landmines buried in the ground has been described as an example. However, this is not only for landmines, but for example for exploring water pipes and water leaking portions. The same effect can be enjoyed even if it is used in radar and airborne radar.

1 is a block diagram showing an embodiment of a radar type exploration device according to the present invention. It is a wave form diagram which shows the signal form produced | generated in each part of FIG. FIG. 2 is a circuit configuration diagram illustrating an example of a waveform shaping circuit in FIG. 1. It is a timing chart explaining the effect | action of the waveform shaping circuit shown in FIG. It is the schematic diagram which showed the relationship between a transmission / reception antenna and a target. It is a wave form diagram explaining the example which calculates the position etc. of a target. 4 is a timing chart for explaining another example of operation of the waveform shaping circuit shown in FIG. 3.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Reference signal generation part 2 1st frequency divider 3 Carrier signal generation part 4 2nd frequency divider 5 Scan signal generation part 6 Synchronization signal generation part 7 Low pass filter 8 Waveform shaping circuit 9 Transmission amplifier 10 Transmitting antenna 11 Reception Antenna 15 Time axis variable circuit 16 Phase shifter 18 Reception gate circuit 21 Low-pass filter (first low-pass filter)
22 Low-pass filter (second low-pass filter)
23 Differential Amplifier (Op Amp)
26 STC circuit 27 A / D converter 28 D / A converter 29 Voltage control resistance element 32 Analog data output terminal A Phase modulation means C1 capacitor (speed-up capacitor)
R1 Discharge resistance Q1 Transistor (Transmission amplifier)
Ta target

Claims (4)

The carrier signal is phase-modulated using the ramp signal as a modulated wave, and the microwave signal is transmitted from the transmitting antenna at the rising or falling timing of the phase-modulated carrier signal, and the reflected wave of the microwave signal is transmitted from the receiving antenna. A radar-type exploration provided with phase change acquisition means for receiving and acquiring a phase change between the phase modulation component of the microwave signal on the transmission side and the phase modulation component of the microwave signal on the reception side A device,
The phase change acquisition means includes a first low-pass filter that extracts a phase modulation component of the microwave signal on the transmission side, and a second low-pass filter that extracts a phase modulation component of the microwave signal on the reception side, Differential extraction means capable of acquiring a change in phase in the phase modulation component obtained by each of the first and second low-pass filters, and a cutoff in each of the first and second low-pass filters. A radar-type exploration device characterized in that the frequency is set to a frequency corresponding to a ratio between the frequency of the carrier signal and the frequency of the ramp signal.   The differential extraction means is configured by a differential amplifier that extracts a difference between a baseband signal obtained by the first low-pass filter and a baseband signal obtained by a second low-pass filter. The radar type exploration device according to claim 1.   The carrier signal and the scan signal are configured to be obtained based on frequency-divided outputs obtained by frequency-dividing the reference signal from the reference signal generation unit, respectively. Radar type exploration equipment.   Waveform shaping means for shaping the rising or falling edge of the carrier signal phase-modulated by the ramp signal into a steeper form, and a microwave signal containing higher harmonics obtained by the waveform shaping means, The radar type exploration device according to any one of claims 1 to 3, wherein the radar type exploration device is configured to transmit by a transmission antenna.

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