JP2006230056A - Estimation method of polarity of motor, and device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide an estimation method of the polarity of a motor that dispenses with a bandpass filter, does not need for constantly applying a high-frequency component, and can exactly estimate the position of the polarity even in low-speed driving and in a stop. <P>SOLUTION: Among three cycles to which triangle waves C are continued, the first cycle is set as T<SB>1</SB>, the next cycle is set as T<SB>2</SB>, and the last cycle is set as T<SB>3</SB>. Then, with respect to a u-phase 31u, currents measured at time points when the cycle T<SB>1</SB>is on a crest and the cycle T<SB>2</SB>is in a trough are set as iu-, iu+, and a difference between the currents is set as a harmonic content Iu. Similarly, with respect to a v-phase 31v, currents measured at time points when the cycle T<SB>2</SB>in on a crest and the cycle T<SB>3</SB>is in a trough are set as iv-, iv+, and a difference between the currents is set as a harmonic content Iv. Also similarly, with respect to a w-phase 31w, currents measured at time points when the cycle T<SB>3</SB>is on a crest and the cycle T<SB>1</SB>is in a trough are set as iw-, iw+, and a difference between the currents is set as Iw. Finally, the polarity position θ is obtained by a formula <A>. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a magnetic pole position estimation method and apparatus for an electric motor having saliency. Examples of the electric motor having saliency include an embedded permanent magnet synchronous motor and a reluctance motor.

  An embedded permanent magnet synchronous motor (hereinafter abbreviated as “IPMSM (interior permanent magnet synchronous motor)”) having a structure in which a permanent magnet is embedded in a rotor can use reluctance torque in addition to magnet torque, so that it has high efficiency. As an electric motor with a wide variable speed range, it is widely used for home appliances such as air conditioners, electric vehicles and general industries.

  Since IPMSM needs to control the current phase of the armature according to the magnetic pole position, generally, a mechanical sensor such as an encoder is attached to obtain magnetic pole position information. However, mechanical sensors are expensive and unreliable, and there is a problem that installation space increases. Accordingly, various sensorless control methods have been proposed in which magnetic pole position information is obtained using only a current sensor that measures an alternating current of each phase of IPMSM.

  Among them, Non-Patent Document 1 discloses a technique for estimating a magnetic pole position at a low speed including a stop time by using a carrier frequency component. In the technique of Non-Patent Document 1, electromagnetic noise is reduced by superimposing a carrier frequency component by making a carrier signal a three-phase triangular wave.

  Further, Non-Patent Document 2 discloses a technique for estimating a magnetic pole position based on a direct current supplied to an inverter.

  Furthermore, a method of adding a high-frequency component or a pilot voltage on a pulse is known in order to enable estimation of the magnetic pole position at the time of stopping and at a low speed. In IPMSM, the dq axis inductance differs due to the saliency. Therefore, since the inductance can be detected by each high frequency component, the magnetic pole position can be estimated. For example, Non-Patent Document 3 discloses a method of applying a pilot voltage intermittently and estimating a magnetic pole position from a current change rate.

Oyama, Higuchi, Abe et al. “Improvement of sensorless control accuracy of IPM motor using carrier frequency component of PWM inverter”, 2002 IEEJ Industrial Application Conference, No. 149 Kawabata, Endo, Takakura, “Study on position sensorless motor current sensorless permanent magnet synchronous motor control”, IEEJ Industrial Application Division Conference, No. 171 M. Scroedl, 1996 IEEE IAS Annual Meeting, PP.270-277

  However, the technique of Non-Patent Document 1 has drawbacks such as that current detection needs to be performed at a frequency about ten times that of a carrier wave, a band-pass filter is required, and a high-frequency component needs to be constantly applied.

  In addition, in the method of adding a sine wave having a frequency sufficiently lower than the carrier frequency as a high frequency component, current detection is performed once per sampling, as in normal PWM control, but a band for extracting a specific frequency component. There are drawbacks such as requiring a pass filter, generating electromagnetic noise, and constantly applying a high-frequency component.

  In the technique of Non-Patent Document 2, neither a mechanical sensor nor a Hall sensor is required, and a direct current can be measured with an inexpensive shunt resistor. However, since the speed electromotive force is used for estimating the magnetic pole position, it cannot be used in principle at low speeds or at a stop.

  On the other hand, the technique of Non-Patent Document 3 has drawbacks such that current detection at the time of applying a pilot voltage is complicated and current control cannot be performed at the time of applying a pilot voltage.

  SUMMARY OF THE INVENTION An object of the present invention is to provide a method and apparatus for estimating the magnetic pole position of an electric motor that does not require a bandpass filter, does not need to constantly apply a high-frequency component, and can accurately estimate the magnetic pole position even at low speeds and when stopped. Is to provide.

  The three-phase PWM inverter used in combination with the magnetic pole position estimation method and apparatus according to the present invention obtains a PWM signal using a carrier wave composed of a single-phase triangular wave, inputs a DC voltage from a DC voltage power source, and responds to the PWM signal. By turning on and off the switch element, a DC voltage is output as a three-phase AC voltage to the three-phase winding of the motor having saliency. Then, when obtaining the PWM signal, for each phase of the three-phase winding, the original command value is tripled in one-third of the three consecutive periods of the carrier wave, and the remaining 2/3 It has a function to superimpose high-frequency components in the period. As a result, a high frequency component having a frequency of 1/3 of the carrier wave is superimposed on the PWM signal. Therefore, a high frequency component is also generated in the current flowing in each phase.

  The magnetic pole position estimation method according to the present invention includes the following steps. First, for each phase, the current flowing in each phase at the time of the peak and valley of the triangular wave is measured, and the difference between these measured values is taken as the harmonic component. Subsequently, the magnetic pole position of the electric motor is estimated based on the harmonic component for each phase. The magnetic pole position estimation apparatus according to the present invention includes a current sensor and a calculation means. The current sensor measures the current of each phase. The calculation means measures the current flowing through each phase through the current sensor at the time of the peak and valley of the triangular wave for each phase, and uses the difference between these measured values as a harmonic component, The magnetic pole position of the motor is estimated based on the harmonic component.

  More specifically, for example, the following steps or operations are performed. The three-phase windings are u-phase, v-phase, and w-phase, and among the three periods of the carrier wave, the first period is the first period, the next period is the second period, and the last period is the third period. First, for the u phase, the currents measured at the first period peak and the second period valley are iu−, iu +, and the difference between them is the harmonic component Iu. Similarly, for the v phase, the current measured at the time of the peak of the second cycle and the valley of the third cycle is iv−, iv +, and the difference between them is the harmonic component Iv. For the w phase, the current measured at the peak of the third period and the valley of the first period is iw−, iw +, and the difference between them is the harmonic component Iw. Finally, Iu, Iv, and Iw are substituted into a predetermined arithmetic expression to obtain the magnetic pole position θ.

Or, the three-phase winding is u-phase, v-phase, and w-phase, and among the three periods of the carrier wave, the first period is the first period, the next period is the second period, and the last period is the third period. The periods before and after the three periods are defined as a zeroth period and a fourth period, respectively. First, the u-phase, iu + ₁ the current measured at the time of the peaks of the valley and the third period and the second period of the first cycle and a zero period, iu + _2, iu- _1, and Iu- _2, these differences {(Iu + _1- iu + ₂ )-(iu- ₁ -iu- ₂ )} is defined as a harmonic component Iu. Similarly, the v-phase, the current measured at the peak of the second period and valleys as well as the fourth period and the third period of the first cycle _{iv + 1, iv + 2,} iv- 1, and IV- _2, these the difference _{{(iv + 1 -iv + 2} ) - (iv- 1 -iv- 2)} is referred to as harmonic components iv. For w phase, the third period and iw + ₁ the measured current at the time of the second cycle of the trough and the second period and the first period of the mountain, iw + _2, Iw- _1, and Iw- _2, these differences {( _{iw + 1 -iw + 2) -} (iw- 1 -iw- 2)} it is referred to as harmonic components Iw. Finally, the magnetic pole position θ is obtained by substituting Iu, Iv, and Iw into a predetermined arithmetic expression.

  Here, the predetermined arithmetic expression is, for example, the following expression <A>.

θ = (1/2) tan ⁻¹ [{Iu− (1/2) (Iv + Iw)} / {(√3 / 2) (Iv−Iw)}]... <A>

  Further, the time points of the peaks and valleys of the triangular wave may be any time within a range of T / 2 centered on the peaks and valleys, for example, where T is the period of the triangular wave. The electric motor may be an embedded permanent magnet synchronous motor.

  In the present invention, the current is measured at the peaks and troughs of the triangular wave, and the magnetic pole position is obtained by calculation based on the current information of six points or eight points obtained in the three or four periods of the triangular wave. Since this method is based on the saliency, it can be used at a stop and at a low speed. Further, since the above formula <A> consists only of measured values, it is not affected by parameter errors.

  According to the present invention, the current supplied to the inverter at the time of the peak and trough of the triangular wave is measured, and the difference between these measured values is used as a harmonic component for each phase, and the magnetic pole position is determined based on these harmonic components. The estimation eliminates the need for a band-pass filter and does not require a high-frequency component to be constantly applied, and can accurately estimate the magnetic pole position even at low speeds and when stopped (see FIG. 5). Further, by using a simple arithmetic expression (formula <A>), the time required for processing can be shortened, so that unprecedented high-speed control can be realized. In addition, since it is not necessary to always apply a high frequency component, effects such as noise reduction, efficiency improvement, and torque ripple reduction can be achieved as compared with the case where a high frequency component is always applied.

  As a result, the speed control range of the low-cost variable speed drive can be greatly expanded. As this applied product, home appliances such as an air conditioner and automobile equipment such as an electric power steering can be considered. At this time, smooth operation is possible in a low speed range including when the vehicle is stopped.

  FIG. 1 is a block diagram showing a first embodiment of a magnetic pole position estimation apparatus according to the present invention. Hereinafter, description will be given based on this drawing. The magnetic pole position estimation method according to the present invention will be described as the operation of the magnetic pole position estimation apparatus of the present embodiment.

  The three-phase PWM inverter 10 used in combination with the magnetic pole position estimation device 40 of the present embodiment obtains a PWM signal using a carrier wave composed of a single-phase triangular wave and a signal wave composed of a three-phase sine wave, and from the DC voltage power supply 20. A DC voltage is input, and the switch element is turned on / off according to the PWM signal, so that the DC voltage is output to the three-phase winding of the IPMSM 30 as a three-phase AC voltage. At this time, when obtaining the PWM signal, the three-phase PWM inverter 10 sets the original command value in three-thirds of the three consecutive periods of the carrier wave for each phase of the three-phase winding. The high frequency component is superimposed in the remaining 2/3 period. As a result, a high frequency component having a frequency of 1/3 of the carrier wave is superimposed on the PWM signal. As a result, a high frequency component is also generated in the current flowing in each phase.

The three-phase PWM inverter 10 includes a control unit 11 mainly composed of a microcomputer or a DSP, and a switch unit 12 composed of switch elements 12u +, 12u−, 12v +, 12v−, 12w +, 12w−. The switch elements 12u +,... Are, for example, IGBT (insulated
gate bipolar transistor) and constitutes a three-phase bridge circuit. The control unit 11 has a general control function of the IPMSM 30 in addition to the function as a part of the magnetic pole position estimation device 40. The general control function is well known and will not be described.

  The DC voltage power supply 20 is a general one that includes a commercial AC power supply 21, a rectifier circuit 22, a smoothing capacitor 23, and the like. The IPMSM 30 is generally composed of a u-phase 31u, a v-phase 31v, and a w-phase 31w, which are three-phase windings, and a rotor 32 having a structure in which a permanent magnet is embedded.

  The magnetic pole position estimation device 40 includes current sensors 41 u and 41 v such as Hall sensors, and a calculation means 42 realized by software as one function of the control unit 11. Current sensors 41u and 41v measure currents flowing through u-phase 31u and v-phase 31v, respectively. Since the u-phase 31u, the v-phase 31v, and the w-phase 31w are connected by a Y connection, if the current flowing through the u-phase 31u and the v-phase 31v is known, the current flowing through the w-phase 31w is also known. The current sensors 41u and 41v made up of Hall sensors are inserted into the wiring between the switch unit 12 and the IPMSM 30 and output the electromotive force generated according to the current to the control unit 11.

  FIG. 2 is a waveform diagram showing the interrelationship between the single-phase triangular wave, the command value for each phase, the switch state of the upper and lower arms, and the instantaneous space voltage vector in this embodiment. 3 [1] is an instantaneous space voltage vector diagram in the present embodiment, and FIG. 3 [2] is a chart showing voltage commands for three periods of the carrier wave in the present embodiment. Hereinafter, description will be given with reference to FIGS.

The control unit 11 obtains a PWM signal based on FIG. 3 [2] while comparing a carrier wave made up of a single-phase triangular wave C and signal waves made up of three-phase sine waves Su, Sv, Sw (not shown). At the same time, the switch elements 12u +,... Are turned on / off according to the PWM signal. As a result, phase voltages vu, vv, vw and line voltages vuv, vvw, vwu (not shown) are obtained. In FIG. 3 [2], α of the command value is a modulation rate, that is, −1 or more and 1 or less. When “α = 1”, the upper arm is always on, and when “α = −1”, the lower arm is always on. The reason why a value three times the original command value (vu ₀ ^* , vv ₀ ^* , vw ₀ ^* ) is input is that “α” + “− α” = 0 before and after that.

At this time, the calculating means 42 operates as follows. Of the three consecutive cycles of the triangular wave C, the first cycle is the cycle T ₁ , the next cycle is the cycle T ₂ , and the last cycle is the cycle T ₃ . For the u phase 31u, the current measured at the time point of the peak of the cycle T ₁ and the valley of the cycle T ₂ is iu−, iu +, and the difference between them is the harmonic component Iu. Similarly, for the v phase 31v, the current measured at the time point of the peak of the period T ₂ and the valley of the period T ₃ is iv−, iv +, and the difference between them is the harmonic component Iv. Similarly, for the w phase 31w, the current measured at the peak of the period T ₃ and the valley of the period T ₁ is iw−, iw +, and the difference between them is the harmonic component Iw. Finally, the magnetic pole position θ is obtained by the above-described formula <A>.

The carrier wave for generating PWM is a single-phase triangular wave C, and the switching elements 12u +,... Are turned on and off based on FIG. In the present embodiment, the current of each phase is measured at the peaks and valleys of the triangular wave C, and six points of current information iu−, iu +, iv−, iv +, iw− obtained in the three periods T _{1 to} T ₃ of the triangular wave C are obtained. , Iw + to obtain the magnetic pole position θ by calculation. Since this method is based on saliency, it can be used at low speeds and when stopped. Further, since the above formula <A> consists only of measured values, it is not affected by parameter errors.

  FIG. 4 [1] is a graph showing the magnetic pole position θ in the α-β coordinate system, and FIG. 4 [2] is a waveform diagram showing the single-phase triangular wave and the current measurement timing in this embodiment. Hereinafter, a method for deriving the above formula <A> will be described with reference to FIGS. In FIG. 4 [2], the three-phase triangular waves Cu, Cv, and Cw are superimposed on the single-phase triangular wave C for reference.

  A general voltage equation of IPMSM can be expressed by the following equation <1> in the α-β coordinate system.

ただし、Ｌ0＝（Ｌd＋Ｌq）／２、Ｌ1＝（Ｌd−Ｌq）／２、Ｌd：ｄ軸インダクタンス、Ｌq：ｑ軸インダクタンス、Ｒ：電機子巻線抵抗、Ψ：永久磁石による界磁磁束である。

However, L0 = (Ld + Lq) / 2, L1 = (Ld−Lq) / 2, Ld: d-axis inductance, Lq: q-axis inductance, R: armature winding resistance, Ψ: field magnetic flux by permanent magnet .

  Here, a sufficiently large carrier angular frequency ωh is set with respect to the motor rotational angular frequency ω1, and a carrier frequency component is considered. Then, the voltage drop due to the armature winding resistance in the first term on the right side of Equation <1> is sufficiently smaller than the reactance voltage drop of the armature winding due to the high-frequency current and can be ignored. The voltage drop due to the change in inductance in the third term on the right side can be ignored because the change in inductance is sufficiently small with respect to the change in applied voltage. The speed electromotive force of the fourth term on the right side is negligible because the change in the rotor position is sufficiently small.

  Therefore, the voltage equation for the high-frequency voltage of the carrier frequency in the α-β coordinate system can be expressed by the following equation <2>.

ここで、添え字ｈは搬送波周波数成分であることを示す。

Here, the suffix h indicates a carrier frequency component.

  Then, assuming that the harmonic voltage of each phase is a symmetric wave and solving Equation <2> for the current, the current solution for each phase can be obtained. Subsequently, the current necessary for the magnetic pole position estimation is obtained by taking the difference between the measured values of each phase from the obtained current and removing the fundamental wave component. This magnetic pole position estimation method is not affected by parameter errors. This will be described in detail below.

  Expression <2> can be rewritten as the following expression <3>.

ただし。Δ＝Ｌ0^２−Ｌ1^２である。

However. Δ = L0 ² −L1 ²

  Here, FIG. 4 [2] shows a carrier wave composed of a single-phase triangular wave and a current measurement timing. The vicinity of the measurement point ↓ is within a range of T / 2 (that is, T / 4 each on the left and right) centered on a peak and valley of the triangular wave, where T is one period of the triangular wave.

  Here, it is assumed that the high-frequency voltage of each phase is a symmetric wave. This is true as the fundamental wave is smaller. That means

となる。そして、ｉαhは、式<3>，<4>から次のように表せる。

It becomes. And iαh can be expressed as follows from the formulas <3> and <4>.

diαh / dt = Vh {(L0−L1cos2θ) cosωht−L1sin2θ · sinωht} <5>
∴iαh = (Vh / ωhΔ) {(L0−L1cos2θ) sinωht + L1sin2θ · cosωht} <6>

The measurement point ↓ in FIG. 2 [2] is ωht = 0 and π of the corresponding phase. Therefore, in the u phase, substituting ωht = 0, π into the formula <6>
iuh = (Vh1 / ωhΔ) (± L1sin2θ) (7)
It becomes. However, +: ωht = 0, −: ωht = π, and Vh1 are carrier frequency components of the phase voltage.

Subsequently, taking the difference between the two measured values of ± shown in Equation <7>,
2Iuh = (2Vh1 / ωhΔ) L1sin2θ ... <8>
Is obtained. Considering v phase and w phase in the same way, according to the measured value at the measurement point ↓,
2Ivh = (2Vh1 / ωhΔ) L1sin2 (θ-2π / 3) ... <9>
2Iwh = (2Vh1 / ωhΔ) L1sin2 (θ + 2π / 3) ... <10>
Is obtained.

And by formulas <8>-<10>
Iuh + (− 1/2 + j√3 / 2) Ivh + (− 1 / 2−j√3 / 2) Iwh = (3/2) (Vh1 / ωhΔ) (sinδ + jcosδ) (11)
Is obtained. However, δ = 2θ.

Therefore, by equation <11>
δ = 2θ = tan ⁻¹ (real part) / (imaginary part) ・ ・ ・ <12>
As a result, the magnetic pole position θ is obtained.

  That is, the formula <12> is as follows.

θ = (1/2) tan ⁻¹ [{Iu− (1/2) (Iv + Iw)} / {(√3 / 2) (Iv−Iw)}]... <A>
However, in the expression <A>, the subscript h is omitted, and the expression is concise.

  Below, some supplementary explanations will be given for this embodiment.

(1). The harmonic component of each phase is in the range of ω1 << ωh: | vuh | = (2E / π) cos (πvu / 2E) (13)
| Vvh | = (2E / π) cos (πvv / 2E) ... <14>
| Vwh | = (2E / π) cos (πvw / 2E) ... <15>
It becomes. However, E: Edc / 2. vu, vv, and vw are instantaneous values, and the sign is also taken into consideration. Therefore, the error range of 5% is
| Vu / E | ≦ 0.202 ... <16>
It becomes. The 10% error range is
| Vu / E | ≦ 0.287 ・ ・ ・ <17>
It becomes.

(2). To add the voltage according to the original voltage command as an average value,
| Vu | <E / 3 = 0.333E ... <18>
It is. Alternatively, the amplitude of the single-phase triangular wave C shown in FIG. 4 [2] is made larger than three times the amplitude of the three-phase sine wave.

  (3). Since ωh is in the denominator as in the formulas <8> to <10>, the detection accuracy decreases when ωh is too large.

  (4). The current (fundamental wave component) of each phase takes an average value of two corresponding points.

  (5). In the range where the voltage increases, the induced voltage information is used.

  Next, a second embodiment of the magnetic pole position estimation method and apparatus according to the present invention will be described. Since the part different from the first embodiment is only the operation of the calculation means, it will be described with reference to FIGS.

The calculation means 42 of this embodiment operates as follows. As shown in FIG. 2, among the three consecutive cycles of the triangular wave C, the first cycle is the cycle T ₁ , the next cycle is the cycle T ₂ , and the last cycle is the cycle T _3. Let them be period T ₀ and period T ₄ , respectively. Then, the u-phase 31u, the period T ₁ and period T ₀ of the trough as well as the period T ₃ and period T ₂ of the current measured at the mountain _{iu + 1, iu + 2,} iu- 1, and Iu- _2, these the difference _{{(iu + 1 -iu + 2} ) - (iu- 1 -iu- 2)} is referred to as harmonic components Iu. Similarly, the v-phase 31v, period T ₂ and the period T ₁ of the valley and the period T ₄ and the period T current iv + ₁ measured at the time of the mountain _{3, iv + 2, iv- 1} , and IV- _2, these difference _{{(iv + 1 -iv + 2} ) - (iv- 1 -iv- 2)} is referred to as harmonic components iv. For w phase 31w, the current measured at the point of the mountain of the period T ₃ and valleys as well as the period T ₂ and the period T ₁ of the cycle _{T 2 iw + 1, iw +} 2, iw- 1, and Iw- _2, these differences { Let (iw + _1- iw + ₂ )-(iw- ₁ -iw- ₂ )} be the harmonic component Iw. Finally, the magnetic pole position θ is obtained by the above-described formula <A>. This embodiment also has the same effect as the first embodiment.

  Next, the present invention will be described once again from another viewpoint using different expressions.

  The present invention provides a magnetic pole position estimation method that eliminates the need for a band-pass filter by superimposing a 1/3 frequency component of a carrier wave signal so that only current detection at the peaks and valleys of the carrier wave is required. Since no bandpass filter is used, high frequency components can be added intermittently.

  1. PWM waveform and current detection for magnetic pole position estimation

In order to superimpose the 1/3 carrier frequency component, one set of 3 carrier wave periods is divided into 6 sets. FIG. 3 [2] shows voltage commands for each phase for three periods of the carrier wave. For each phase, the original command value (vu ₀ ^* , vv ₀ ^* , vw ₀ ^* ) is tripled in the 1/3 period, and the remaining 2/3 period is a high frequency component to be superimposed. The superimposition component has a rectangular wave shape of size α.

  FIG. 2 shows a carrier wave waveform and a PWM signal when the voltage command of each phase is zero and α is Edc / 4 (modulation factor 0.5). Instantaneous space vectors 0 to 7 are shown in FIG. A to H indicated by circles in the carrier wave signal in FIG. 2 indicate current detection timings, and are detected at peaks and troughs of the carrier wave signal as in the case of current control by general PWM.

  2. Magnetic pole position estimation method

  A method for estimating the magnetic pole position from the PWM signal and the current measured at the detection timing will be described. There are two types of estimation methods depending on whether the high frequency component is considered as a rectangular wave or a sine wave.

  2-1. Estimation method when a high-frequency component is considered as a rectangular wave (second embodiment)

  The superimposed component is considered as a rectangular wave, and a magnetic pole position estimation method based on the current change rate when applying a pulse voltage (Non-Patent Document 3) is referred to. In Non-Patent Document 3, current detection is performed twice while a single space vector is being applied, and the current change rate is obtained from the difference between two consecutive detection values. On the other hand, in the present invention, current detection is performed at the peak and valley of the carrier wave, and the average current change rate of the carrier wave is used. Using the current detected from point A to point H shown in FIG. 2, the magnetic pole position is estimated as follows.

f ₁ (2θ) = K ₁ {cos (−2θ) + jsin (−2θ)}
= ΔΔiu + ΔΔive ^{j2 / 3π} + ΔΔiwe ^{-j2 / 3π} ... <21>
_{However, ΔΔiu = (iu C -iu A} ) - (iu F -iu D)
ΔΔiv = (iv _E −iv _C ) − (iv _H −iv _F )
ΔΔiw = (iw _G −iw _E ) − (iw _D −iw _B )
θ is the magnetic pole position, and the subscripts A to H indicate the current detection timing in FIG. The magnetic pole position can be obtained from the ratio of the real part to the imaginary part of the formula <21>.

  2-2. Estimation method when a high frequency component is considered as a sine wave (first embodiment)

  Considering the superimposed component as a sine wave, the magnitude of the high-frequency current contained in the detected current at the peak of the sine wave voltage of the 1/3 frequency component of the carrier wave superimposed on each phase is obtained, and the magnetic pole position is estimated. In order to extract the high-frequency current, the difference between the currents at the positive and negative peaks of the high-frequency voltage is taken. Using the current detected from point B to point G shown in FIG. 2, the magnetic pole position is estimated as follows.

f ₂ (2θ) = K ₂ {sin (2θ) + jcos (2θ)}
= Δiu + Δive ^{j2 / 3π} + Δiwe- ^{j2 / 3π} ... <22>
However, Δiu = iu _B -iu _E
Δiv = iv _D −iv _G
Δiw = iw _F −iw _C
The magnetic pole position can be obtained from the ratio of the real part and the imaginary part of the formula <22>.

  If α is Edc / 2 and the modulation factor at the time of applying the high frequency component is 1, the current of the phase to be detected can be grasped only by the inverter DC section current (idc in FIG. 1).

  3. Summary

  According to the magnetic pole position estimation method of the IPMSM according to the present invention, by superimposing the 1/3 frequency component of the PWM carrier wave signal, it has the following features.

1) No band pass filter is required.
2) Current detection is performed only at the peaks and valleys of the carrier signal.
3) There is no need to constantly apply high frequency components.
4) The present invention can also be applied to a system that detects only the inverter DC section current.

  In the present invention, since the maximum value of the voltage command is 1/3 of the conventional value, the method may be switched to another magnetic pole position estimation method such as a method using speed electromotive force in the middle / high speed range.

  Next, the actual machine experiment result of the magnetic pole position estimation method and apparatus according to the present invention will be described as a first embodiment.

  FIGS. 5 [1] and [2] show the magnetic pole position estimation results by the methods described in the first and second embodiments, respectively. As a sample machine, 1.5 kW, 90 Hz IPMSM (GNA152GA1-M2G) manufactured by Fuji Electric was used. In the experiment, the motor was driven by open loop control by V / f control, and the magnetic pole position was estimated. The fundamental frequency was operated at 5 Hz with no load, and the magnitude α of the high frequency component was Edc / 4. It was confirmed that both methods were able to estimate the magnetic pole position.

It is a block diagram which shows 1st and 2nd embodiment of the magnetic pole position estimation apparatus which concerns on this invention. It is a wave form diagram which shows the interrelationship of the single phase triangular wave in this embodiment, the command value to each phase, the switch state of an upper and lower arm, and an instantaneous space voltage vector. FIG. 3 [1] is an instantaneous space voltage vector diagram in the present embodiment, and FIG. 3 [2] is a chart showing voltage commands for three periods of the carrier wave in the present embodiment. FIG. 4 [1] is a graph showing the magnetic pole position θ in the α-β coordinate system, and FIG. 4 [2] is a waveform diagram showing the single-phase triangular wave and the current measurement timing in this embodiment. It is a wave form diagram which shows the actual machine experiment result in 1st and 2nd embodiment.

Explanation of symbols

DESCRIPTION OF SYMBOLS 10 Three-phase PWM inverter 11 Control part 12 Switch part 12u +, 12u-, 12v +, 12v-, 12w +, 12w- Switch element 20 DC voltage power supply 21 Commercial AC power supply 22 Rectifier circuit 23 Smoothing capacitor 30 IPMSM
31u u phase 31v v phase 31w w phase 32 rotor 40 magnetic pole position estimation device 41u, 41v current sensor 42 computing means θ magnetic pole position

Claims (14)

A three-phase winding of an electric motor having saliency is obtained by obtaining a PWM signal using a carrier wave composed of a single-phase triangular wave, inputting a DC voltage from a DC voltage power supply, and turning on and off the switch element according to the PWM signal. Is used in combination with a three-phase PWM inverter that outputs the DC voltage as a three-phase AC voltage, and estimates the magnetic pole position of the motor,
When obtaining the PWM signal, for each phase of the three-phase winding, the original command value is tripled in one third of the three consecutive periods of the carrier wave, and the remaining 2 / Used in combination with the three-phase PWM inverter having a function of superimposing high-frequency components in three periods;
For each phase, measure the current flowing in each phase at the time of the peak and valley of the triangular wave, the difference between these measurements as a harmonic component,
Estimating the magnetic pole position of the motor based on the harmonic components for each of these phases,
A method for estimating the magnetic pole position of an electric motor. The three-phase windings are u-phase, v-phase and w-phase,
Among the three periods, when the first period is the first period, the next period is the second period, and the last period is the third period,
For the u phase, the current measured at the time of the peak of the first cycle and the valley of the second cycle is iu−, iu +, and the difference between these is the harmonic component Iu,
For the v phase, the current measured at the time of the peak of the second cycle and the valley of the third cycle is iv−, iv +, and the difference between these is the harmonic component Iv,
For the w phase, the current measured at the peak of the third period and the valley of the first period is iw−, iw +, and the difference between these is the harmonic component Iw,
Substituting the Iu, the Iv, and the Iw into a predetermined arithmetic expression to obtain the magnetic pole position θ.
The method for estimating a magnetic pole position of an electric motor according to claim 1. The three-phase windings are u-phase, v-phase and w-phase,
Of the three cycles, the first cycle is the first cycle, the next cycle is the second cycle, the last cycle is the third cycle,
When the periods before and after these three periods are the zeroth period and the fourth period, respectively,
For the u-phase, the first period and the first zero-cycle valley and the third period and the current measured at the peak of the second period _{iu + 1, iu + 2,} iu- 1, and Iu- _2, these differences _{{(iu + 1 -iu + 2} ) - (iu- 1 -iu- 2)} was used as a harmonic components Iu,
For the v phase, the second period and the current measured at the time of the peaks of valleys and the fourth period and the third period of the first cycle _{iv + 1, iv + 2,} iv- 1, and IV- _2, these differences _{{(iv + 1 -iv + 2} ) - (iv- 1 -iv- 2)} was used as a harmonic components iv,
For the w-phase, the third period and the second period of the trough and the second period and the current measured at the time of the mountain of the first period _{iw + 1, iw + 2,} iw- 1, and Iw- _2, these differences _{{(iw + 1 -iw + 2} ) - (iw- 1 -iw- 2)} was a harmonic component Iw,
Substituting the Iu, the Iv, and the Iw into a predetermined arithmetic expression to obtain the magnetic pole position θ.
The method for estimating a magnetic pole position of an electric motor according to claim 1. The predetermined arithmetic expression is the following expression <A>.
θ = (1/2) tan ⁻¹ [{Iu− (1/2) (Iv + Iw)} / {(√3 / 2) (Iv−Iw)}]... <A>
The method of estimating a magnetic pole position of an electric motor according to claim 2 or 3. The time point of the peak and valley of the triangular wave is any time within a range of T / 2 centered on the peak and valley, where T is the period of the triangular wave.
The method for estimating a magnetic pole position of an electric motor according to any one of claims 1 to 4. A sawtooth wave instead of the triangular wave,
The method for estimating a magnetic pole position of an electric motor according to claim 1. The electric motor is an embedded permanent magnet synchronous motor;
The method for estimating a magnetic pole position of an electric motor according to claim 1.
A three-phase winding of an electric motor having saliency is obtained by obtaining a PWM signal using a carrier wave composed of a single-phase triangular wave, inputting a DC voltage from a DC voltage power supply, and turning on and off the switch element according to the PWM signal. Is used in combination with a three-phase PWM inverter that outputs the DC voltage as a three-phase AC voltage, and estimates the position of the magnetic pole of the motor,
When obtaining the PWM signal, for each phase of the three-phase winding, the original command value is tripled in one third of the three consecutive periods of the carrier wave, and the remaining 2 / Used in combination with the three-phase PWM inverter having a function of superimposing high-frequency components in three periods;
A current sensor for measuring the current of each phase;
For each phase, the current flowing through each phase is measured via the current sensor at the time of the peak and trough of the triangular wave, and the difference between these measured values is used as a harmonic component. Arithmetic means for estimating the magnetic pole position of the electric motor based on a wave component;
An apparatus for estimating a magnetic pole position of an electric motor. The computing means is
The three-phase windings are u-phase, v-phase and w-phase,
Among the three periods, when the first period is the first period, the next period is the second period, and the last period is the third period,
For the u phase, the current measured at the time of the peak of the first cycle and the valley of the second cycle is iu−, iu +, and the difference between these is the harmonic component Iu,
For the v phase, the current measured at the time of the peak of the second cycle and the valley of the third cycle is iv−, iv +, and the difference between these is the harmonic component Iv,
For the w phase, the current measured at the peak of the third period and the valley of the first period is iw−, iw +, and the difference between these is the harmonic component Iw,
Substituting the Iu, the Iv, and the Iw into a predetermined arithmetic expression to obtain the magnetic pole position θ.
The apparatus for estimating a magnetic pole position of an electric motor according to claim 8. The computing means is
The three-phase windings are u-phase, v-phase and w-phase,
Of the three cycles, the first cycle is the first cycle, the next cycle is the second cycle, the last cycle is the third cycle,
When the periods before and after these three periods are the zeroth period and the fourth period, respectively,
For the u-phase, the first period and the first zero-cycle valley and the third period and the current measured at the peak of the second period _{iu + 1, iu + 2,} iu- 1, and Iu- _2, these differences _{{(iu + 1 -iu + 2} ) - (iu- 1 -iu- 2)} was used as a harmonic components Iu,
For the v phase, the second period and the current measured at the time of the peaks of valleys and the fourth period and the third period of the first cycle _{iv + 1, iv + 2,} iv- 1, and IV- _2, these differences _{{(iv + 1 -iv + 2} ) - (iv- 1 -iv- 2)} was used as a harmonic components iv,
For the w-phase, the third period and the second period of the trough and the second period and the current measured at the time of the mountain of the first period _{iw + 1, iw + 2,} iw- 1, and Iw- _2, these differences _{{(iw + 1 -iw + 2} ) - (iw- 1 -iw- 2)} was a harmonic component Iw,
Substituting the Iu, the Iv, and the Iw into a predetermined arithmetic expression to obtain the magnetic pole position θ.
The apparatus for estimating a magnetic pole position of an electric motor according to claim 9. The predetermined arithmetic expression is the following expression <A>.
θ = (1/2) tan ⁻¹ [{Iu− (1/2) (Iv + Iw)} / {(√3 / 2) (Iv−Iw)}]... <A>
The apparatus for estimating a magnetic pole position of an electric motor according to claim 9 or 10. The time point of the peak and valley of the triangular wave is any time within a range of T / 2 centered on the peak and valley, where T is the period of the triangular wave.
The motor magnetic pole position estimation apparatus according to any one of claims 8 to 11. A sawtooth wave instead of the triangular wave,
The magnetic pole position estimation apparatus for an electric motor according to any one of claims 8 to 12. The electric motor is an embedded permanent magnet synchronous motor;
The motor magnetic pole position estimation apparatus according to any one of claims 8 to 13.

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