JP2006217267A - Submerged communication system - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To accurately conduct a submerged communication by ultrasonic waves even under a Doppler effect and the effect of noises. <P>SOLUTION: A receiver 1 receives a synchronous signal as a linear FM signal, a reference signal as a known sine-wave signal of a frequency and an information signal as the sine-wave signal of the frequency corresponding to the value of a transmission information. A matched filter 24 carries out a complex matched filter processing corresponding to the synchronous signal to an input signal, and outputs an absolute-value system D [n]. A gate generator 25 uses a time when the maximum value of D [n] as a synchronous-signal receiving time, and generates gates for the reference signal and the information signal while using the time as a reference. A frequency estimator 27 carries out a discrete Fourier transform to the input signals in the gates, and estimates the frequency corresponding to the maximum value of a frequency spectrum as the frequencies of the reference signal and the information signal. A frequency corrector 28 corrects the estimated frequency of the information signal on the basis of the known frequency of the reference signal and the estimated frequency. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a technique for preventing underwater communication performed by a fishnet depth meter or the like from being affected by the Doppler effect.

  In net fishing, a fishnet depth meter is used to know the depth of a fishnet (for example, Patent Document 1). The fishnet depth meter is composed of a transmitter attached to the bottom of the fishnet, a receiver attached to the bottom of the fishing boat, and a receiver installed in the fishing boat. The transmitter repeatedly transmits a synchronization pulse, which is an ultrasonic signal, and a frequency modulation pulse. The synchronization pulse is a signal for synchronizing communication between the transmitter and the receiver. The frequency modulation pulse is a signal whose frequency changes according to the magnitude of the water pressure detected by the transmitter. The receiver obtains the depth at which the transmitter is located from the water pressure by synchronizing with the synchronization pulse received by the receiver and further detecting the frequency of the frequency-modulated ultrasonic pulse. In addition, since fish nets do not necessarily settle uniformly, a plurality of sets of transmitters and receivers are generally used.

  However, the transmitter moves in the same way when the fishnet is submerged or raised, or the fishnet rises or falls due to the influence of the current. Since fishing boats pitch and roll with waves, the receivers also oscillate. As a result, a relative speed is generated between the transmitter and the receiver, that is, since the ultrasonic signal transmitted from the transmitter is affected by the Doppler effect, the frequency of the received signal is transmitted from the transmitter. Shifting from the frequency, the correct depth cannot be obtained at the receiver. Accordingly, it has been proposed to obtain the depth by correcting the Doppler shift (frequency change due to the Doppler effect) as follows (for example, Patent Document 1).

  FIG. 6 is a diagram showing a waveform in a fishnet depth meter. (A) shows the received signal of the receiver. (B) shows an enlarged waveform of the pulse for synchronization. (C) shows a gate signal generated by the receiver. The time width of the gate signal is one period of the synchronization pulse. (D) shows a clock pulse generated by the receiver. The clock pulse is a signal having a frequency much higher than the frequency of the synchronizing pulse. First, the number of clock pulses during the period when the gate signal is high is counted, and the frequency of the synchronization pulse is calculated from the count value. The frequency of the frequency modulation pulse is also calculated by counting clock pulses. Next, based on the calculated frequency of the synchronization pulse, the known frequency of the synchronization pulse, and the underwater sound speed, the relative speed calculation unit calculates the relative speed. Furthermore, based on the underwater sound speed and the relative speed, the calculated frequency of the frequency modulation pulse is corrected by the correction calculation means. In this way, the depth is obtained by correcting the Doppler shift.

  Further, a method of transmitting the water pressure value after converting it into a 1/0 bit string is also used. Patent Document 2 below shows a circuit for modulating the bit string by MSK (Minimum Shift Keying). In this method, since the digital value of the water pressure is transmitted, correction of the Doppler shift is unnecessary. However, a relatively long communication time is required to reduce the influence of indirect waves generated on the sea surface and the sea floor. In Patent Document 3 below, a correlation process is performed by comparing a digital signal representing phase information generated from a received signal with a rectangular reference signal, and the received signal is properly received based on the degree of correlation between the two signals. It is shown that it is determined whether it is a signal.

JP-A-6-341838 (paragraphs 0001 to 0024) Japanese Patent Laid-Open No. 5-268276 (paragraphs 0001 to 0011) Japanese Patent Laid-Open No. 2003-194921 (paragraphs 0022 to 0035, FIGS. 4 to 9)

  By the way, in what is shown by patent document 1, the pulse for a synchronization has a function which synchronizes communication, and a function as a reference signal for calculating said relative speed. On the other hand, the frequency of the frequency modulation pulse changes depending on the water pressure. Therefore, in order to distinguish between the synchronization pulse and the frequency modulation pulse that are repeatedly received, the frequency of the synchronization pulse must be assigned outside the variable range of the frequency modulation pulse. In addition, it must be assigned with a certain margin in consideration of the Doppler shift. For this reason, since the frequency band of the received signal is widened, the interference waves generated by the ultrasonic waves transmitted from the scanning sonar of other fishing boats operating nearby, and the noise generated by waves and propellers (hereinafter collectively referred to as these) Transmission errors caused by simply “noise”).

  Further, since the frequency of the synchronization pulse and the frequency of the frequency modulation pulse are calculated based on the number of clock pulses for one period of the synchronization pulse and the frequency modulation pulse, it corresponds to the gate signal of both pulses. If the waveform of the portion is distorted by noise, a frequency calculation error that the frequency is not correctly calculated occurs. When such a transmission error or frequency calculation error occurs, the water pressure value cannot be accurately transmitted from the transmitter to the receiver.

  An object of the present invention is to enable accurate underwater communication using ultrasonic waves even under the influence of the Doppler effect and noise.

In the underwater communication system according to the first aspect of the invention, a transmitter that transmits an ultrasonic signal to the water, a receiver that receives the ultrasonic signal, and a signal process for the received signal received by the receiver. In an underwater communication system including a receiver, a transmitter modulates a synchronization signal that has been modulated in a predetermined manner so that it can be distinguished from subsequent reference signals and information signals, a reference signal having a known frequency, and modulation according to transmission information The transmitted information signal is transmitted into the water as an ultrasonic signal.
Here, the information signal modulated according to the transmission information is a signal that is frequency-modulated with the transmission signal.

  As described above, in the transmission signal, the synchronization signal for synchronizing communication and the reference signal having a known frequency are separate signals, and the synchronization signal is predetermined modulation so that it can be distinguished from the reference signal and the information signal. For example, the receiver estimates the frequency of the received reference signal, obtains the Doppler shift amount from the estimated frequency and the known frequency of the reference signal, and further uses the Doppler shift amount to obtain the information signal. By removing the influence of the Doppler shift from the information, information can be extracted from the received information signal. Further, since the relationship among the synchronization signal, the reference signal, and the information signal is as described above, a common frequency band can be used for the synchronization signal, the reference signal, and the information signal. In this way, the frequency band of the received signal can be narrower than that shown in Patent Document 1, and the above-described transmission error can be reduced. That is, information can be extracted from the received information signal even under the influence of the Doppler effect or noise. That is, it is possible to accurately perform underwater communication using ultrasonic waves. Furthermore, if the receiver detects a plurality of information signals based on the synchronization signal, a plurality of pieces of information (for example, water pressure and water temperature shown in the embodiment) can be transmitted.

  In the underwater communication system according to the second aspect of the invention, a transmitter for transmitting an ultrasonic signal into the water, a receiver for receiving the ultrasonic signal, and a signal processing for the received signal received by the receiver. In the underwater communication system including the receiving receiver, the transmitter includes a synchronization signal that is modulated in a predetermined manner so as to be distinguishable from the subsequent reference signal and information signal, a reference signal that is a sine wave signal having a known frequency, and a transmission An information signal, which is a sine wave signal having a frequency corresponding to the information value, is transmitted into the water as an ultrasonic signal. The receiver detects the received reference signal and the information signal based on the received synchronization signal, estimates the detected reference signal frequency and the detected information signal frequency, and estimates the known frequency of the reference signal The frequency of the information signal estimated based on the frequency of the reference signal is corrected.

  In this way, an accurate frequency of the information signal can be obtained even under the influence of the Doppler effect or noise. That is, underwater communication using ultrasonic waves can be performed accurately. Specifically, the synchronization signal for synchronizing communication and the reference signal having a known frequency are set as separate signals, and the synchronization signal is a signal that has been modulated in a predetermined manner that can be distinguished from the reference signal and the information signal. By correcting the estimated frequency of the information signal based on the known frequency of the signal and the estimated frequency, the frequency of the information signal from which the influence of the Doppler effect is removed can be obtained. Further, since the relationship among the synchronization signal, the reference signal, and the information signal is as described above, a common frequency band can be used for the synchronization signal, the reference signal, and the information signal. In this way, the frequency band of the received signal can be narrower than that shown in Patent Document 1, and the above-described transmission error can be reduced. Furthermore, by detecting a plurality of information signals based on the synchronization signal and estimating the frequencies of the plurality of information signals, a plurality of pieces of information (for example, water pressure and water temperature shown in the embodiment) can be transmitted. .

In the underwater communication system according to the third aspect of the invention, a transmitter for transmitting an ultrasonic signal into the water, a receiver for receiving the ultrasonic signal, and a signal processing for the reception signal received by the receiver. In the underwater communication system including the receiving receiver, the transmitter includes a synchronization signal that is modulated in a predetermined manner so as to be distinguishable from the subsequent reference signal and information signal, a reference signal that is a sine wave signal having a known frequency, and a transmission An information signal, which is a sine wave signal having a frequency corresponding to the information value, is transmitted into the water as an ultrasonic signal. The receiver detects the synchronization signal based on the correlation processing means for performing correlation processing between the waveform of the received signal and the assumed waveform of the synchronization signal, and the correlation output signal representing the degree of correlation obtained by the correlation processing. Gate generating means for generating a gate of the reference signal and information signal based on the reception time of the synchronized signal, and the frequency and information signal of the reference signal based on the reference signal and information signal in the gate generated by the gate generating means Frequency estimation means for estimating the frequency of the information signal, and frequency correction means for correcting the frequency of the information signal estimated by the frequency estimation means based on the known frequency of the reference signal and the frequency of the reference signal estimated by the frequency estimation means And comprising.
Here, the correlation processing means corresponds to the matched filter section shown in the embodiment, the gate generation means corresponds to the gate generation section shown in the embodiment, and the frequency estimation means uses the frequency shown in the embodiment. This corresponds to the estimation unit, and the frequency correction means corresponds to the frequency correction unit shown in the embodiment. The assumed waveform of the synchronization signal corresponds to the waveform of the synchronization signal transmitted from the transmitter, not the waveform of the synchronization signal received by the receiver.

  In this way, an accurate frequency of the information signal can be obtained even under the influence of the Doppler effect or noise. That is, underwater communication using ultrasonic waves can be performed accurately. Specifically, the synchronization signal for synchronizing communication and the reference signal having a known frequency are set as separate signals, and the synchronization signal is a signal that has been modulated in a predetermined manner that can be distinguished from the reference signal and the information signal. By correcting the estimated frequency of the information signal based on the known frequency of the signal and the estimated frequency, the frequency of the information signal from which the influence of the Doppler effect is removed can be obtained. Further, since the relationship among the synchronization signal, the reference signal, and the information signal is as described above, a common frequency band can be used for the synchronization signal, the reference signal, and the information signal. In this way, the frequency band of the received signal can be narrower than that shown in Patent Document 1, and the above-described transmission error can be reduced. In addition, the synchronization signal is not detected by frequency as in Patent Document 1, but the synchronization signal is detected based on the degree of correlation between the waveform of the reception signal and the assumed waveform of the synchronization signal. The synchronization signal can be detected even if a part is affected by noise. Furthermore, the reference signal and the information signal gate are generated based on the synchronization signal reception time, and the reference signal frequency and the information signal frequency are estimated based on the reference signal and the information signal in the gate. The head portion can be excluded from the frequency estimation target signal. As a result, it is possible to avoid estimating the frequency using a signal in which a synchronization signal that has arrived after reflection at the seabed or the sea surface and a reference signal that has arrived directly from the transmitter can be avoided, so that the frequency of the reference signal can be estimated accurately. be able to. The same applies to the estimation of the frequency of the information signal. Moreover, since the frequency of a plurality of information signals can be estimated by generating a gate for each information signal, a plurality of pieces of information (for example, water pressure and water temperature shown in the embodiment) can be transmitted.

  In an embodiment of the third invention, the correlation processing means generates a correlation output signal by performing correlation processing using a matched filter, and the gate generation means determines the time when the maximum value of the correlation output signal is generated as the synchronization signal. Reception time. In this way, an optimum correlation output signal is generated for obtaining the time at which the maximum value occurs, so that the reception time of the synchronization signal without time lag, that is, the reference time of the gate is obtained, and the reference signal It is possible to generate an optimum gate used for estimating the frequency of the information signal.

  In the embodiment of the third invention, the filter coefficient updating means for updating the filter coefficient of the matched filter based on the difference between the known frequency of the reference signal and the frequency of the reference signal estimated by the frequency estimation means. Prepare. Here, the filter coefficient updating means corresponds to the filter coefficient calculation unit shown in the embodiment. By doing so, the influence of the Doppler shift is reflected in the filter coefficient, and the optimum matched filter processing adapted to the Doppler distortion is performed, so that the synchronization signal can be reliably detected even under the influence of the Doppler effect.

  Further, although not described in the claims, in the embodiment of the third invention, the frequency estimation means sets the frequency corresponding to the maximum value of the frequency spectrum of the reference signal and the information signal, respectively. Estimate the frequency. By doing so, even if a part of the reference signal or the information signal is affected by noise, the influence on the frequency spectrum is very small, so that the above-described frequency calculation error can be prevented. In addition, the frequency spectrum is not affected even if a reference signal that arrives directly from the transmitter and a reference signal that arrives after being reflected at the sea floor or the sea surface are mixed, so there is no frequency estimation (calculation) error due to the mixture of signals. Does not occur. The same applies to information signals.

  Furthermore, in the embodiment of the third invention, the frequency estimation means obtains a frequency spectrum by performing a discrete Fourier transform on the digitized reference signal and information signal. In this case, a discrete frequency corresponding to the maximum value of the frequency spectrum may be used as the estimated frequency, or a frequency obtained by performing interpolation processing on the discrete frequency and the preceding and succeeding discrete frequencies may be used as the estimated frequency. If interpolation processing is performed, the frequency can be estimated more accurately.

  Furthermore, in the embodiment of the third invention, the frequency band of the synchronization signal and the frequency variable range of the information signal overlap, and the known frequency of the reference signal is included in the frequency band or the frequency variable range. By doing in this way, since the frequency band of a received signal becomes narrower than what is shown by patent document 1, the above-mentioned transmission error can be reduced.

  According to the present invention, underwater communication using ultrasonic waves can be accurately performed even under the influence of the Doppler effect and noise.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings. FIG. 1 shows a configuration of a fishnet depth meter which is an example of an underwater communication system. FIG. 2 shows a transmission signal transmitted from the transmitter. FIG. 3 shows the configuration of the receiver. The transmitter 3 is attached to the bottom of the fish net and converts information such as water pressure into an ultrasonic signal for transmission. This ultrasonic signal is received by the receiver 2 attached to the bottom of the fishing boat and converted into an electrical signal. And a received signal is processed with the receiver 1 with which the fishing boat was equipped, and information, such as a fishnet depth, is displayed on the display part 30 of the receiver 1. FIG.

  Next, the configuration and operation of the transmitter 3 will be described. In FIG. 1, a transmitter 3 is a water temperature gauge 12, a water pressure gauge 13, a transmitter / receiver 14 that transmits an ultrasonic wave toward the seabed and receives an echo, and a transmitter that transmits an ultrasonic signal toward the wave receiver 2. A wave detector 15, a battery 17 that is a power source of the transmitter 3, a voltage detection circuit 16 that detects the voltage of the battery 17, and a control unit 11 that generates an ultrasonic signal and the like. By measuring the time from when the ultrasonic wave is transmitted to when the echo is received, the bottom distance, which is the distance from the fishnet to the seabed, is obtained. Information such as the water pressure obtained from the water pressure gauge 13, the water temperature, the bottom distance, and the battery voltage is converted into an ultrasonic signal and transmitted from the transmitter 15.

The transmission signal shown in FIG. 2 includes a synchronization signal, a reference signal, and two information signals, and is transmitted periodically. A synchronization signal is placed at the head, followed by a reference signal and an information signal. Here, the information signal follows the reference signal, but the order may be reversed. Although the number of information signals is two, the number may be one or three or more. The information signal is a sine wave signal having a frequency corresponding to the value of information such as water pressure (hereinafter referred to as transmission information). That is, the signal is frequency-modulated with transmission information. Here, the variable range of the frequency is 72 kHz to 74 kHz, and the time width is 100 ms. Moreover, the frequency (represented by finf) of the information signal and the value of the transmission information are associated with each other by the following equation (1), for example.
finf = 72 kHz + (Iv−Imin) / (Imax−Imin) · (74 kHz−72 kHz) (1)
Here, Iv is a value of transmission information, Imin is a minimum value of transmission information, and Imax is a maximum value of transmission information.

  The synchronization signal is a signal for synchronizing communication between the transmitter 3 and the receiver 1, and is a linear FM signal (LFM signal) whose instantaneous frequency continuously changes. Here, it is assumed that the time width (represented by Tkp) of the synchronization signal is 50 ms, and the instantaneous frequency continuously changes from 72 kHz to 74 kHz within the time width Tkp. This frequency sweep width (2 kHz) is represented by Bkp. The reference signal is a signal whose frequency is referred to when the influence of the relative velocity is removed, and is a sine wave signal whose frequency is known. Here, the frequency (expressed by fplt) is 73 kHz, and the time width is 100 ms. From the above, the frequency band (also referred to as transmission band) of the transmission signal is 72 kHz to 74 kHz. The center frequency (73 kHz) of the transmission band is represented by fc. The above 72 kHz to 74 kHz is an example of numerical values such as a transmission band.

  As described above, a synchronization signal for synchronizing communication and a reference signal having a known frequency are set as separate signals, and the synchronization signal is a modulated signal (linear FM) that can be distinguished from the reference signal and the information signal. Signal), a common frequency band can be used for the synchronization signal, the reference signal, and the information signal. Thereby, since the frequency band of the received signal can be narrower than that shown in Patent Document 1, the above-described transmission error can be reduced. In order to reduce transmission errors, it is desirable to match the variable range of the frequency of the information signal and the frequency band of the synchronization signal as described above. However, if the two overlap (for example, the variable frequency of the information signal If the range is 72.2 kHz to 74.2 kHz and the frequency sweep range of the linear FM signal as the synchronization signal is 71.8 kHz to 73.8 kHz), the same effect can be obtained.

  Next, the configuration and operation of the receiver 1 will be described. In FIG. 3, the frequency converter 21 converts the center frequency of the received signal output from the receiver 2 from fc (73 kHz) to an intermediate frequency (represented by f0) of 10.24 kHz by a known method. Also, signal components outside the effective band of the receiver 2 are removed and the received signal is amplified. The A / D converter 22 samples the reception signal output from the frequency conversion unit 21 at a constant sampling rate (expressed by fs) and converts it into a digital signal. Here, the sampling rate fs is set to 40.96 kHz, which is four times the intermediate frequency f0. In addition, when a plurality of sets of transmitters 3 and receivers 2 are used as described above, and ultrasonic signals having different center frequencies are used in each set, signal processing in the receiver 1 is shared. For this purpose, frequency conversion is performed, but the received signal itself may be sampled without performing frequency conversion.

  The quadrature detection unit 23 generates a complex envelope sequence of the received signal from the output signal of the A / D converter 22, and the sampling rate (data rate) of the generated complex envelope sequence at a predetermined ratio (decimation ratio). Reduce and output. The real part of the output complex envelope sequence is represented by I [n], and the imaginary part is represented by Q [n]. Further, the thinning ratio is represented by rdec, and the sampling rate after the thinning is represented by fsdec. Here, since rdec is 8, fsdec is 5.12 kHz (fs / rdec = 40.96 kHz / 8). Since the sampling rate is reduced in this way, the subsequent signal processing load is reduced. The above processing includes frequency conversion, and the subsequent processing is performed in a frequency range with 0 Hz as the center frequency. Hereinafter, the operation of each unit of the quadrature detection unit 23 will be described.

  The real part code conversion unit 23a performs “no conversion, replacement with 0, sign inversion, 0” with respect to the output sequence x [n] (n = 0, 1, 2,...) Of the A / D converter 22. The operation of “Replace” is periodically performed. Therefore, the output sequence of the real part code conversion unit 23a is {x [0], 0, −x [2], 0, x [4], 0, −x [6], 0,. . The imaginary part code conversion unit 23b periodically performs an operation of “substitution to 0, sign inversion, substitution to 0, no conversion” on the output series x [n]. Therefore, the output sequence of the imaginary part code conversion unit 23b is {0, −x [1], 0, x [3], 0, −x [5], 0, x [7],. . As described above, since the sampling rate fs is selected to be four times the intermediate frequency f0, the multiplication of x [n] and the cosine sequence of the frequency f0 and the multiplication of x [n] and the sine sequence of the frequency f0 are performed. By performing the above simple operation, a complex envelope sequence of the received signal can be obtained.

ここで、Ｍdecは間引きフィルタ２３ｃ、２３ｄのフィルタ長、ｈdec（ｍ＝０，１，２，・・・，Ｍdec−１）は間引きフィルタ２３ｃ、２３ｄのフィルタ係数、ｒdecは上記の間引き比である。 The decimation filters 23c and 23d perform low-pass filter and resample (decimation) operations on the output sequences of the real part code conversion unit 23a and the imaginary part code conversion unit 23b, respectively. When the input sequence and output sequence of the decimation filters 23c and 23d are represented by p [n] and q [n], respectively, the operation of the decimation filters 23c and 23d is represented by the following equation (2).

Here, Mdec is the filter length of the decimation filters 23c and 23d, hdec (m = 0, 1, 2,..., Mdec-1) is the filter coefficient of the decimation filters 23c and 23d, and rdec is the above decimation ratio. .

  An example of the amplitude response of the decimation filters 23c and 23d is shown below. The amplitude response in the frequency range of ± 1.25 kHz (1 kHz + 250 Hz) is only 1 dB lower than the amplitude response at 0 Hz. Here, 0 Hz is the center frequency of the processing signal, and corresponds to the center frequency fc (73 kHz) of the transmission band. ± 1 kHz corresponds to the upper limit / lower limit frequency of the transmission band. 250 Hz is a Doppler shift amount generated in a 73 kHz signal when the relative speed between the transmitter 3 and the receiver 2 is 10 knots. In addition, the amplitude response at the frequency component (frequency component of 2.56 kHz or more or −2.56 kHz or less) of the aliasing distortion caused by the resampling is 40 dB or more lower than the amplitude response at 0 Hz. That is, by using the thinning filters 23c and 23d, it is possible to secure a passband width of a received signal that allows for a Doppler shift corresponding to a relative speed of 10 knots, and to suppress aliasing distortion caused by resampling.

  The matched filter unit 24 subjects the complex envelope sequence (I [n], Q [n]) output from the quadrature detection unit 23 to complex matched filter processing corresponding to the above-described synchronization signal, The absolute value is calculated, and the calculated absolute value series D [n] is output. Here, the real part sequence generation unit 24a and the imaginary part sequence generation unit 24b constitute a complex matched filter. The real part sequence generation unit 24a generates the real part sequence Imf [n] of the complex matched filter output, and the imaginary part sequence generation unit 24b generates the imaginary part sequence Qmf [n] of the complex matched filter output. The absolute value calculation unit 24c generates an absolute value series D [n] from the real part series Imf [n] and the imaginary part series Qmf [n]. These generation operations are represented by the following formulas (3) to (5), respectively.

ここで、Ｍmfはマッチドフィルタのフィルタ長であり、ｆsdec・Ｔkpで定義される。すなわち、同期信号にかかる複素包絡線系列（Ｉ［ｎ］，Ｑ［ｎ」）のデータ数と等しくなるように、マッチドフィルタのフィルタ長Ｍmfが決められており、Ｍmfは２５６（５．１２ｋＨｚ×５０ｍｓ）となる。

Here, Mmf is the filter length of the matched filter and is defined by fsdec · Tkp. That is, the filter length Mmf of the matched filter is determined so as to be equal to the number of data of the complex envelope series (I [n], Q [n]) related to the synchronization signal, and Mmf is 256 (5.12 kHz × 50 ms).

  Hc [m] and hs [m] are filter coefficients supplied from a filter coefficient calculation unit 26 described later. These hc [m] and hs [m] are filter coefficients for detecting the synchronization signal, and all (256) complex envelope sequences (I [n], Q related to the synchronization signal in the matched filter). [N]) is a filter coefficient determined so that the value of the absolute value series D [n] is maximized. That is, the matched filter unit 24 performs correlation processing between the waveform of the received signal and the assumed waveform of the synchronization signal, and outputs an absolute value series D [n] that is a correlation output signal indicating the degree of correlation. Here, it can be said that the filter coefficients hc [m] and hs [m] are equivalent to the assumed waveform of the synchronization signal.

  As described above, since correlation processing is performed using a matched filter, a correlation output signal (absolute value series D [n]) that is optimal for obtaining the time at which the maximum value occurs is generated. The time at which the maximum value of the absolute value series D [n] occurs is detected by the gate generation unit 25 described later, and this time is set as the reception time of the synchronization signal. The width of the peak including the maximum value of the absolute value series D [n] is inversely proportional to the sweep frequency width Bkp of the synchronization signal, and the S / N ratio of D [n] is proportional to the time width Tkp of the synchronization signal. Therefore, in order to detect the synchronization signal, it is desirable to increase Bkp and Tkp. However, since this causes problems such as an increase in the bandwidth of the transmission signal and a longer communication time, The optimum values of Bkp and Tkp are determined by comprehensively judging the points.

  The gate generation unit 25 detects a synchronization signal based on the absolute value series D [n] output from the matched filter unit 24, and further, a frequency estimation unit 27 described later based on the reception time of the synchronization signal causes the reference signal and The time range of both signals (more precisely, the complex envelope series (I [n], Q [n]) concerning both signals) used when estimating the frequency of the information signal is determined. Hereinafter, this time range is referred to as a gate.

  The buffer memory 25a temporarily stores a part of the absolute value series D [n]. The maximum value detection unit 25b searches for the maximum value of the absolute value series D [n] within a predetermined range of the buffer memory 25a, and when the maximum value is equal to or greater than the predetermined value, the discrete time (1 / fsdec is generated) when the maximum value is generated. (Time value in units) is output. This discrete time is hereinafter referred to as synchronization signal reception time. The synchronization signal reception time is a time corresponding to the boundary between the synchronization signal and the reference signal based on the characteristics of the matched filter unit 24 described above (see FIG. 4). Then, the gate calculation unit 25c calculates the start time and end time of the gate of the reference signal and the gate of the information signal based on the synchronization signal reception time, and outputs the calculated values.

  FIG. 4 shows an example of the time relationship between the received signal and the gate. Here, the time when 50 ms elapses from the synchronization signal reception time is set as the gate start time of the reference signal, and the time when 50 ms elapses from the start time is set as the gate end time, but when 40 ms elapses from the synchronization signal reception time. May be the start time, and the end time may be when 50 ms has elapsed since the start time. In this way, if the gate end time side is also provided with a margin, even if the synchronization signal is affected by noise and the determined synchronization signal reception time is not accurate, the gate of the reference signal is Since it is located within the reception time zone, it does not adversely affect the estimation of the reference signal described later.

  Further, since the synchronization signal that arrives after being reflected on the sea bottom or the sea surface is mixed in the head part (first half part) of the reference signal, the head part is out of the gate range. On the other hand, a reference signal that has arrived directly from the transmitter 3 and a reference signal that has arrived after being reflected at the seabed and the like are mixed in a portion other than the head portion. This affects the frequency spectrum of the reference signal described later. Don't give. The same applies to information signals.

  As described above, the synchronization signal is not detected by frequency as in Patent Document 1, but based on the degree of correlation between the waveform of the received signal and the assumed waveform of the synchronization signal, specifically, the maximum value of the degree of correlation. By detecting this, the synchronization signal is detected. Therefore, even if a part of the received synchronization signal is affected by noise, the synchronization signal can be detected. However, although there may be some time lag in the synchronization signal reception time when affected by noise, there is a margin for time lag between the gate start time (front end) and end time (rear end) as described above. By providing this, inconvenience due to time lag can be avoided.

  The frequency estimator 27 performs DFT (discrete Fourier transform) and spectral interpolation on the complex envelope sequence (I [n], Q [n]) in the gate of the reference signal and information signal output from the gate generator 25. To estimate the frequencies of the reference signal and the information signal. This estimated frequency is a Doppler shifted frequency, that is, a reception frequency. Hereinafter, the operation of each unit of the frequency estimation unit 27 will be described.

FIG. 5 shows a configuration example of the frequency estimation unit 27. The buffer memories 27a and 27b temporarily store a part of the real part series I [n] and the imaginary part series Q [n] of the complex envelope series, respectively. The data selection unit 27c selects a portion corresponding to the gate of the reference signal or the gate of the information signal among the signals in the buffer memories 27a and 27b. The window coefficient memory 27d is a memory for storing a Hanning window coefficient series w [m] expressed by the following equation (6).
w [m] = {1-cos (2π · m / Mdft)} / 2 (m = 0,1,2, ..., Mdft-1) (6)
Here, Mdft is the number of DFT data points and is defined by fsdec · Tgate (gate width). Therefore, Mdft is 256 (5.12 kHz × 50 ms).

  The multiplier 27e outputs a real part sequence which is the product of the Mdft real part sequences I [n] selected by the gate and the window coefficient series w [m]. The multiplier 27f outputs an imaginary part sequence that is a product of the Mdft imaginary part series Q [n] selected by the gate and the window coefficient series w [m]. The DFT calculation unit 27g performs DFT on a complex sequence having Mdft real part sequences and imaginary part sequences output from the multipliers 27e and 27f as real parts and imaginary parts, respectively, using a fast Fourier transform algorithm. Apply. With this DFT, a DFT sequence G [k] (k = 0, 1, 2,..., Mdft−1) composed of Mdft real part sequences and imaginary part sequences is calculated. Then, the absolute value calculator 27h calculates the absolute value | G [k] | (the square root of the sum of the square of the real part and the square of the imaginary part of G [k]) of each G [k]. This | G [k] | is a frequency spectrum (amplitude spectrum) for Mdft discrete frequencies.

  Since the center frequency of the frequency is 0 Hz and the frequency resolution is fsdec / Mdft (5.12 kHz / 256 = 20 Hz), | G [k] | is a frequency spectrum in 20 Hz increments in the frequency range of 0 Hz ± 2.56 kHz. It becomes. Since this frequency range is sufficiently wider than the bandwidth (2 kHz) of the transmission signal, even if a Doppler shift occurs in the information signal, for example, even if the reception frequency of the 74 kHz information signal becomes 74.25 kHz, the information signal Can be estimated. Of course, the frequency of the reference signal having the same frequency as the center frequency fc (73 kHz) of the transmission band can be estimated.

  The spectrum shifter 27i cyclically shifts the sequence of | G [k] | in the right direction (direction in which k increases) by Mdft / 2. When the shifted sequence is represented by Gshift [k], | G [0] | becomes Gshift [Mdft / 2] and | G [Mdft-1] | becomes Gshift [0]. The reason why the frequency shift is cyclic is that the frequency spectrum | G [k] | is not in ascending order of frequency, and Gshift [k] whose frequency is in ascending order is obtained by this cyclic shift. That is, Gshift [0] is the frequency spectrum for the lowest frequency, and Gshift [Mdft-1] is the frequency spectrum for the highest frequency.

The spectrum interpolation unit 27j calculates a frequency estimation value (represented by fest) of the reception frequency of the reference signal and the information signal in the following procedure. In the first step, k (this k is represented by kmax) corresponding to the maximum value of Gshift [k] is obtained. In the second step, the ratio r (Gshift [kmax-1] / Gshift [kmax]) and the ratio s (Gshift [kmax + 1] / Gshift [kmax]) are calculated. In the third step, when r ≧ s, the frequency estimated value fest is calculated by the following equation (7). In the case of r <s, it is calculated by the following formula (8).
fest = fc + Δf · {kmax− (2r−1) / (1 + r) −Mdft / 2} (7)
fest = fc + Δf · {kmax + (2s−1) / (1 + s) −Mdft / 2} (8)
As described above, fc is the center frequency of the transmission band, and Δf is the DFT frequency resolution. In addition, the interpolation of the discrete frequency is performed by subtraction of (2r-1) / (1 + r) and the addition of (2s-1) / (1 + s), and after interpolation for the center frequency (0 Hz) by subtraction of Mdft / 2. The displacement amount of the discrete frequency is corrected.

  The reason why the interpolation processing is performed is that the probability that the frequency spectrum is maximized between kmax and kmax-1 or between kmax and kmax + 1 is higher than the probability that the frequency spectrum is maximized at kmax. . By this interpolation processing, the frequency estimation value fest is estimated more accurately. By applying the above-described processing to the complex envelope sequence (I [n], Q [n]) selected by the gate of the reference signal and the gate of the information signal, the frequency estimation value (represented by fplt_est) and information of the reference signal A frequency estimation value (represented by finf_est) of the signal is calculated and output from the frequency estimation unit 27.

  As described above, since the frequency corresponding to the maximum value of the frequency spectrum of the reference signal and the information signal is estimated as the frequency of the reference signal and the information signal, respectively, even if a part of the reference signal or the information signal is affected by the noise The influence on the frequency spectrum is very small. That is, the frequency calculation error described above can be prevented. In addition, as described above, even when a reference signal (or information signal) that arrives directly from the transmitter and a reference signal (or information signal) that arrives after being reflected at the sea floor or the sea surface are mixed, the frequency spectrum has an effect. Since it is not received, the reference signal (or information signal) can be estimated.

The frequency correction unit 28 corrects the frequency estimation value finf_est of the information signal based on the known frequency fplt of the reference signal and the frequency estimation value fplt_est. When this corrected frequency is represented by finf_corr, finf_corr is defined by the following equation (9).
finf_corr = finf_est · fplt / fplt_est (9)
This frequency finf_corr is the frequency of the information signal from which the influence of the Doppler effect is removed, and ideally becomes equal to the transmission frequency finf of the information signal. In this way, finf_corr is obtained without using the relative speed detected by the relative speed detection means disclosed in Patent Document 1 and the underwater sound speed.

  The numerical value converter 29 converts the corrected frequency finf_corr of the information signal into a value of transmission information. This conversion is performed by the inverse conversion of the above equation (1). The value of the transmission information (water pressure, water temperature, etc.) is displayed on the display unit 30. Actually, the fishnet depth converted from the water pressure is displayed on the display unit 30.

The filter coefficient calculation unit 26 calculates the filter coefficients hc [m] and hs [m] of the above-described matched filter based on the known frequency fplt of the reference signal and the frequency estimation value fplt_est (10 ) And (11).
hc [m] = cos (ω · m + μ · m ² ) (m = 0,1,2,... Mmf−1) (10)
hs [m] = sin (ω · m + μ · m ² ) (m = 0,1,2,... Mmf−1) (11)
Here, ω and μ are defined by the following equations (12) and (13), respectively.
ω = (2π / fsdec) · {(fplt_est−fplt) −Bkp / 2} (12)
μ = (π / fsdec ² ) · (Bkp / Tkp) (13)
As described above, Mmf is the filter length of the matched filter (fsdec · Tkp = 256), Bkp is the frequency sweep width (2 kHz) of the synchronization signal, and Tkp is the time width (50 ms) of the synchronization signal.

  The filter coefficients hc [m] and hs [m] calculated by the equations (10) and (11) are stored in the memory in the filter coefficient calculation unit 26 and supplied to the matched filter unit 24. In the matched filter processing at the first reception, fplt_est is unknown, so a coefficient calculated using the value of fplt is used for fplt_est. The coefficient calculated using the calculated fplt_est is used. As described above, the filter coefficients hc [m] and hs [m] are updated based on the difference between fplt_est and fplt corresponding to the Doppler shift amount, and optimal matched filter processing adapted to Doppler distortion is performed. The synchronization signal can be reliably detected even under the influence of the effect.

  In the embodiment described above, the case where the linear FM signal whose instantaneous frequency continuously increases is used as the synchronization signal, but other signals that can be distinguished from the reference signal and the information signal can be used as the synchronization signal. . For example, a linear FM signal whose instantaneous frequency continuously decreases, or a sine wave whose frequency is constant and whose phase changes with time, for example, a phase modulation signal whose phase changes by 180 °, is also used as the synchronization signal. be able to.

  In the above embodiment, the matched filter processing is used to detect the synchronization signal. However, other methods, for example, as long as the correlation processing between the waveform of the reception signal and the assumed waveform of the synchronization signal is performed. The present invention can also be implemented using the method disclosed in Patent Document 3 above. Furthermore, in the said embodiment, although the frequency of a reference signal and an information signal was estimated using DFT (discrete Fourier transform), if the frequency spectrum of each signal can be calculated | required, another method will be used. Even if the frequency is estimated by using the present invention, the present invention can be implemented.

  Furthermore, in the above embodiment, the detection of the synchronization signal and the estimation of the frequency are performed using the complex envelope sequence (I [n], Q [n]) output from the quadrature detection unit 23. The present invention can also be implemented by processing as a real part sequence. Furthermore, although the case where the underwater communication system is a fishnet depth meter has been described in the above embodiment, the present invention can also be applied to an underwater communication system other than a fishnet depth meter.

  Furthermore, in the above embodiment, the time width of the two information signals is the same, but the time width of the first information signal related to the transmission information that requires high accuracy is set to 200 ms, and the transmission information that does not require high accuracy is used. The time width of the second information signal may be set to 100 ms. Furthermore, in the above-described embodiment, the synchronization signal, the reference signal, and the information signal are continuously transmitted. However, a non-transmission section may be provided between the signals.

It is a figure which shows the structure of the fishnet depth meter which is an example of an underwater communication system. It is a figure which shows the transmission signal transmitted from a transmitter. It is a figure which shows the structure of a receiver. It is a figure which shows an example of the time relationship between a received signal and a gate. It is a figure which shows the structural example of a frequency estimation part. It is a figure which shows the waveform in the conventional fishnet depth meter.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Receiver 2 Receiver 3 Transmitter 23 Quadrature detection part 24 Matched filter part (correlation processing means)
24a Real part series generation part (matched filter)
24b Imaginary sequence generator (matched filter)
25 Gate generator (gate generator)
26 Filter coefficient calculation unit (filter coefficient updating means)
27 Frequency estimation unit (frequency estimation means)
27g DFT calculation unit 27j Spectral interpolation unit fest Frequency estimation value finf Information signal frequency finf_est Information signal frequency estimation value finf_corr Information signal corrected frequency estimation value fplt Reference signal frequency estimation value hc [m] , Hs [m] Filter coefficient of matched filter Mdft Number of discrete Fourier transform data Mmf Filter length of matched filter

Claims (5)

In an underwater communication system including a transmitter that transmits an ultrasonic signal into water, a receiver that receives the ultrasonic signal, and a receiver that performs signal processing on a reception signal received by the receiver ,
The transmitter uses a synchronization signal modulated in a predetermined manner so as to be distinguishable from the subsequent reference signal and information signal, a reference signal having a known frequency, and an information signal modulated according to transmission information as the ultrasonic signal. The underwater communication system characterized by transmitting to. In an underwater communication system including a transmitter that transmits an ultrasonic signal into water, a receiver that receives the ultrasonic signal, and a receiver that performs signal processing on a reception signal received by the receiver ,
The transmitter includes a synchronization signal that is modulated in a predetermined manner so as to be distinguishable from the subsequent reference signal and information signal, a reference signal that is a sine wave signal having a known frequency, and a sine wave signal having a frequency corresponding to a value of transmission information. And transmitting the information signal as the ultrasonic signal underwater,
The receiver detects the received reference signal and the information signal based on the received synchronization signal, estimates the detected reference signal frequency and the detected information signal frequency, the reference signal known frequency and the information signal An underwater communication system, wherein the frequency of the estimated information signal is corrected based on the estimated frequency of the reference signal. In an underwater communication system including a transmitter that transmits an ultrasonic signal into water, a receiver that receives the ultrasonic signal, and a receiver that performs signal processing on a reception signal received by the receiver ,
The transmitter includes a synchronization signal that is modulated in a predetermined manner so as to be distinguishable from the subsequent reference signal and information signal, a reference signal that is a sine wave signal having a known frequency, and a sine wave signal having a frequency corresponding to a value of transmission information. And transmitting the information signal as the ultrasonic signal underwater,
The receiver is
Correlation processing means for performing correlation processing between the waveform of the received signal and the assumed waveform of the synchronization signal;
A gate generating means for detecting a synchronization signal based on a correlation output signal representing a degree of correlation obtained by the correlation processing, and generating a gate for the reference signal and the information signal based on the detected reception time of the synchronization signal;
Frequency estimation means for estimating the frequency of the reference signal and the frequency of the information signal based on the reference signal and the information signal in the gate generated by the gate generation means;
Frequency correction means for correcting the frequency of the information signal estimated by the frequency estimation means based on the known frequency of the reference signal and the frequency of the reference signal estimated by the frequency estimation means, Underwater communication system. The underwater communication system according to claim 3,
The correlation processing means generates the correlation output signal by performing the correlation processing using a matched filter,
The underwater communication system characterized in that the gate generation means sets the time when the maximum value of the correlation output signal occurs as the reception time of the synchronization signal. The underwater communication system according to claim 4,
An underwater communication system comprising: filter coefficient updating means for updating a filter coefficient of the matched filter based on a difference between a known frequency of the reference signal and a frequency of the reference signal estimated by the frequency estimation means. .

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