JP2005536314A - Electrosurgical generator - Google Patents

Abstract

The present invention relates to an electrosurgical generator suitable for a cutting arc configuration at an activated electrode that provides constant voltage and variable power output, and in particular exhibits a dynamically activated surface area that changes shape. In principle, constant voltage-based control is achieved by using a level of d.c. link voltage that functions to achieve amplification of the output of the RF resonant inverter. Dual loop feedback control indicates that output voltage based control signals are introduced late at low gains while link motor voltage based control is applied early. Improved control of the d.c. link voltage is achieved through the use of an input network that includes a power element correction stage.

Description

  The use of electrotherapy by medical researchers dates back to the 18th century. At that time, electrotherapy static generators were the real topic. In the 20th century, the application of high-frequency current to living tissue began with d'Arsonal, which was first devised to use high-frequency current therapy. The use of high frequency currents to perform electrosurgical cutting and similar acts was promoted by the Cushing and Bovie in the 1920s. In the 1970s, solid state electrosurgical generators were introduced, and a number of such generators are currently available in many medical settings.

  When high frequency currents are used for cutting and cauterization, the tissue at the surgical site suffers controlled damage. Cutting is accomplished by splitting or removing the tissue by momentarily attaching to the activated cutting electrode, i.e., slightly away so that the formation of a cutting arc is achieved. Usually, a continuous sine wave is employed to perform a cutting function that evaporates tissue cells in proximity to the electrode. The advantage of this electrosurgical cutting procedure over the use of a cold scalpel is that it is easy to cut, in addition to reducing tissue damage and reducing the area in addition to reducing the area. In both. In this regard, the cells that are close to the arc of the cutting electrode are evaporated, and the cells that are several layers deep are basically not damaged. These cutting systems are generally used in a monopolar type, where the cutting electrode is considered active, the surgical current is a large dual component, and the current is in a remote location Return from the dispersive electrode connected to the patient's skin.

Coagulation by using a high frequency generator device is also feasible, and is achieved by denaturation of tissue proteins due to heat damage. The disturbing and discontinuous waveform is typically used to perform clotting. Coagulation is generally
(1) Electric breakdown in which tissue is carbonized by arc strike (2) Drying in which cells are dehydrated (3) White coagulation that heats tissue more slowly to coagulate is included. Intermittent waves of the solidification procedure have been performed by both monopolar and bipolar systems.

  To perform a cut while performing hemostasis, today's electrosurgical generators may be controlled to mix the cutting and coagulation waveforms. To achieve this mixing, for example, a lower intensity continuous sine wave is mixed with a higher intensity coagulation pulse with a power amplification procedure or similar procedure prior to output voltage rise. .

Electrosurgical cutting response has been a subject of much research. In this regard, several investigators have found that as the electrical conduction of the heat of cutting takes place, the tissue rises in temperature to the boiling point and the cells basically rupture as a result of the phase change. Apart from that, the parallel mechanism by which the electromagnetic wave hits the uptake tissue and the change is generated by the heat shrinkage property of the tissue became clear. The cause of the pressure wave is the lack of ability to maintain thermal equilibrium during rapid heating in the tissue. In this regard, the following document:
"Electrosurgery" JA Pierce, John Wiley & Sons New York, NY
Please refer to.

  Overcoming the cutting procedure is arc formation in a guided gaseous state. When cutting is performed, the cutting electrode is not in mechanical contact with the tissue, but rather travels over the vapor membrane as it moves through the tissue. As such, a separation between the cutting electrode and the tissue may occur, which may allow arc formation during cutting. Due to the presence of such an arc, an electric current is confined in a narrow space and becomes an electrode due to the property of being highly electroded both in place and in time, and electric discharge at a high current density is performed in a very short time.

  Typically, an electrosurgical generator is configured to obtain the requisite arc configuration with a fixed shape active electrode. For example, the active electrode may take the form of a rod or spade surgical knife. Arcing requires a technique from the surgeon, and generally the electrode moves gradually toward the target tissue until the impedance is suitable for arc striking. The energy that creates the arc is typically generated by a resonant inverter that operates at the RF frequency. Control in such an inverter involves the problem that the arc exhibits negative dynamic impedance. Generally, the voltage supplied to the RF inverter is adjusted, but overall output control is performed by selecting a power level. Inverter control by feedback of output voltage was avoided mainly by the above-mentioned load characteristics of the essential arc. Control of such attempts often leads to vibration instability. Power control is therefore used at the limit of medically viable output performance.

  However, at present, the development of medical instruments that are performed electrosurgically requires an active cutting electrode with a precision structure that has a shape that changes in the active surface area during the operating procedure. Generators that exhibit a relatively constant power output cannot withstand arcing under such operating conditions. In this regard, the power output is variable and may follow changes in the shape and size of the active electrode. In conclusion, this requires an electrosurgical generator capable of providing an RF disconnected output under conditions of constant voltage control and variable power conditions.

  Another requirement for the operation of the electrosurgical generator relates to initial arc formation. The application of systems that have been considered in recent years requires an arc start-up when the active electrode is incorporated inside and contacts the tissue to be cut. A preliminary impedance, such as defining a time interval, is not available to achieve the initial arc configuration unless it is achieved by physician skill.

  The present invention relates to an electrosurgical generator capable of forming and maintaining a cutting arc (electric arc) in an active electrode exhibiting dynamic activated surface area characteristics. In achieving this sustained arc formation, the generator is controlled and tuned at a level that functions to control the arc generating inverter. C. Achieve link voltage. The characteristics of constant voltage and variable power are: d. c. Realized by outer loop output voltage feedback control, indicating lower gain or slower input control over the link voltage. This outer loop control is d. c. Mixed with inner loop feedback control, which has a quick and high gain characteristic on the link voltage. The latter control facilitates an increase in power at the start or restart of the procedure. This increase works to produce the required cutting arc under conditions such that the active electrode is implanted in the tissue. As soon as the normal cutting power level is drawn to generate startup power, d. c. Through adjustment of the link voltage to the boost voltage level for a resist interval sufficient to generate an arc d. c. The link voltage is increased.

  The electrosurgical generator has an input treatment network that includes a control stage for the power element to function to combine external currents and voltages with the associated conventional advantages. However, this input stage allows the use of a universal generator, despite the variant embodiment, and is controlled d. c. Introduce link voltage d. c. Establish a provisional regulated voltage level that is preferably used for link inverters.

  Other objects of the invention will become apparent and will be apparent from the following description. Accordingly, the present invention comprises methods, systems, and apparatus, and includes structures, component combinations, component arrangements, and work steps, which will become apparent in the detailed examples that follow. Is done. A more complete understanding of the nature and objects of the invention can be obtained by reference to the detailed description when taken in conjunction with the accompanying drawings.

  As will be described in detail below, the electrosurgical generator of the present invention with an associated boost voltage feature is described in connection with a topology selected for use with an electrosurgically supported tissue retrieval instrument. Has been. An embodiment of the instrument uses an electrosurgical cutting current sine wave, which is an ablation characteristic that is not part of the discussion. However, a device including a modified embodiment that requires shochu is described in Eggers, et al., US patent application Ser. No. 09 / 472,673 (December 27, 1999) “Minimally Invasive Intact Recovery of Tissue” It is also described in US Patent Application (Attorney Docket No. NET-2-039-3) of Eggers, et al. The electrosurgical generator of the present invention is initially configured with a variable power output suitable for an active cutting electrode that changes its surface area range during a given procedure, and a configuration based on constant voltage characteristics. It is described in. In addition to those descriptions, a generator configuration with power monitoring control for use with a conventional active electrode having a fixed region is shown.

  Referring to FIG. 1, the electrosurgical generator of the present invention is described as a component of an electrosurgical intact tissue retrieval system, generally indicated at 10. The system 10 is a tissue retrieval device, usually as indicated at 12, which has a reusable component 14, a disposable component 16 having a rear portion slidably mounted on the component 14. Have. The reusable component 14 has a polymeric housing 18.

  The disposable component 16 includes a longitudinal delivery cannula or elongate delivery cannula, generally indicated at 22, extending along the instrument axis 24. The distal end of the delivery cannula 22 extends through a rotatable threaded connector 26 that can be screwed onto the housing 18 as well as through a freely rotating suction manifold 28 held in place by a collar 30. To do. The forward region of the cannula 22 as indicated at 32 extends to the tip or tip as indicated generally at 34. A flexible intake pipe provides a smoke / steam exhaust function and, as indicated at 36, is pushed by the connector 38 from the manifold 28 and passes through the connector 40, as well as the housing or console of the vacuum system 46. Reusable connector 42 extends toward the air input 44 of the vehicle. The housing 46 has an on / off switch 48 and is activated to provide smoke / steam / body fluid cleaning exhaust in the pipe 36 by using a foot switch 50 connected to the console 46 via a cable 52. The Smoke / steam evacuation from the tip 34 is required to avoid thermally damaging the tissue due to the backward steam movement along the outer surface of the cannula 22. The vacuum system extends to the tip region 32. In this regard, five smoke / steam collection or intake intake ports are provided at the end 32 as indicated at 35. Slightly behind these ports 35 are provided blocking ribs or rings 37 that function to prevent steam or smoke from moving along the outer surface of the delivery cannula 22.

  A grip connector as indicated at 38 is disposed on each side portion of the housing 18 and is additionally provided to support a stabilizer hand grip such as a deformed grip as indicated at 54. Function. Located on the front portion of the housing 18 are three button switches 56-58, each of which is an arm / disarm switch, an energyized position switch, and a tissue capture start switch. There is a linear array of LED-based indicators or cue lights on the side of each side of the housing 18 slightly above the switches 56-58, and one such array is indicated at 60. The visual cues provided by the indicator 60 are described from front to back, starting / reset cues that are green lights, tissue capture completion cues provided with green lights, and tissue capture start provided with yellow lights. Cue (above switch 58), energized position cue provided by yellow light (above switch 57), and arm / disarm tissue capture cue provided by green light (above switch 56) . Energization and control is provided to the instrument 12 via a multi-strand cable 62 connected to a composite control assembly and electrosurgical generator console, typically indicated at 64. The connection is shown via multi-lead connector 66, which is connected to console connector 68. The electrosurgical activation electrode assembly of the instrument 12 operates monopolar. As such, a relatively large distributed return electrode assembly of the prior art is indicated at 70 and is placed near the patient's skin. The assembly 70 is configured with two electrode components 72 and 74 that are connected to a console connector 80 via a cable 76 and a connector 78. Alternatively, a return electrode may be placed on the near-tip surface of the delivery cannula 22, which is the desired location for use of the illustrated return electrode 70.

  When the on / off switch 82 is activated, power is supplied to the circuit at the console 64. When switch 82 is “on”, the green visual indicator LED located on the switch shines. When the connector 66 having the console connector 68 and the cable are properly connected, the green LED 86 provided on the connector 68 is indicated by light. This connection test is performed by passing a current through the coding resistor inside the housing 18. Three pedal foot switches 88 are connected to the rear panel of the console 64 via a cable 90. The three pedals 88a-88c of the switch 88 mimic and provide an alternative to the button switches 56-58.

  A visual cue corresponding to the LED array of housing 18 as shown at 60 is also provided at console 64. In this regard, the start / reset switch 92 has an operational association with the LED indicator light 94, where the indicator light 94 glows green when the switch is activated. The yellow position mode visual cue LED indicates the energization of the aforementioned precursor electrode, which is indicated at 96. This LED provides a yellow output during electrosurgical advancement of the delivery cannula tip 34 opening in close proximity to the targeted tissue volume. Next, a visual cue in arm capture mode is provided by LED 98 to represent the arming of the tissue capture feature of instrument 12. Once the arm / disarm switch as shown at 56 or 88a is depressed, the energized position switch as shown at 57 or 88b can no longer be activated. However, the physician may return to position mode by pressing the arm / disarm switch again. A yellow capture mode visual cue is provided by LED 100, indicating the start and execution of a tissue capture procedure, and when such a capture is finished, a green capture end mode visual cue is provided by LED 102 in green. The pause mode state is indicated by turning on the green LED 104. Generally, this pause mode is executed by opening the capture switch 58 or the foot switch 88c.

  Since steam movement occurs as described above, it is desirable for the system 10 to compensate for activation of a vacuum system such as that shown by the housing or console 46. Desirably, the control assembly of the console 46 functions to be allowed to begin the procedure only at the start of the system 46. Such monitoring of the system 46 is accomplished by a vacuum activation switch shown in block 51 attached to the tube 36. The output of this monitoring for console 64 is indicated by arrow 53.

  When the connector 78 of the return electrode 70 is connected to the console connector 80 and the switch 82 is turned on, the patient's circuit safety monitor circuit (PCSM) performs a self test. If the start / reset switch 94 is depressed continuously, a defect test for the two electrode components 72 and 74 is performed. If the latter test is not passed, a warning cue on visual and audio pulses is emitted, and the visual cue is provided at the red LED 106 located proximate to the connector 80.

  With reference to FIG. 2, the disposable component 16 of the instrument 12 is shown prior to being inserted into the housing 18 of the reusable component 14. In the figure, it can be seen that the delivery cannula 22 extends further forward than the support housing 108 which has a cylindrical shape again. A front region of the support housing 108 supports the rotatable connector 26. In this regard, it can be seen that the connector 26 is configured with an outer screw 110 that is rotatably fixed with the knurled flange 112. At the rearward end of the support housing 108 is a protruding index pin 114 that extends inwardly along a long receiving cavity 118 inside the housing 18 when the disposable component 16 is inserted. It is slidably received within an elongated slot 116 provided upward. The thread 120 inside the cavity 118 is threaded with the thread 118 outside the connector 26 when the disposable component 16 is inserted into the reusable component 14.

  Located on the opposite side of the index pin 114 on the support housing 108 are two spaced apart electrical contacts 122 and 124 that are used to insert the support housing 108 into the receiving cavity 118. , Directed to form lighting contacts with corresponding electrical terminals disposed within the housing 18. Contacts 122 and 124 selectively receive electrosurgical cutting current applied to the cable for electrosurgical cutting and purging associated with the precursor electrode assembly and capture component at tip 32, respectively. The cables extend from a capture component within the delivery cannula 22 to a cable terminator component having a guidance tab or ear, one of which is indicated at 126 and is provided parallel to the axis 24. The long stabilizer slot 130 is slidably mounted. Corresponding guidance tab and slot combinations are also found on the opposite side of the support housing 108. Located at the front of the slot as seen at 130 are two additional long drive slots, one of which is shown at 134 and is configured in parallel with the axis 24. ing. Extending out of the slot are ears or guide tabs that extend outwardly of the drive components of the drive assembly and are indicated by reference numerals 138 and 140. These ears or tabs 138 and 140 support a rearwardly disposed drive surface that is used to transmit forward motion to the drive assembly. This forward movement functions to deploy the capture component from the delivery cannula 22. When the support housing 108 is inserted within the receiving cavity 118 of the housing 18, their tabs 138 and 140 are disposed on opposite sides as indicated by reference numerals 142 and 144, respectively, provided in the front portion of the housing 18. Insert the notch. At the same time, the notch 146 is positioned in front of the reusable housing 18 to allow the electrical contacts 122 and 124 to pass through. Clearly, the procedure for inserting the disposable component 16 within the reusable component 14 includes the process of inserting the disposable support housing 108 within the receiving cavity 118 and the indented portion 112 of the connector 26. The thread 110 is screwed into the thread 120 with the process of rotating. The figure also shows the vacuum closure plug 148 used following the procedure for the blocking connector 42 of the hose or tube 36, with some fluid in the mechanically connected front component of the rear tube. To capture. Finally, it can be seen that the tab 150 extends through the forward portion of the drive slot 134. This tab is a component of the plug assembly complete stop 304 (FIG. 9), which shows the forward travel distance allowed by the drive parts having ears 138 and 140 in relation to the size of the diameter of the preselect capture component. It works to limit.

  Referring to FIG. 3, a cross-sectional view illustrating the operating environment of the drive features held within the reusable component 14 and the drive features of the disposable component 16 is shown. In this figure, the motor assembly is typically indicated at 160. The assembly 160 is composed of a DC electric motor 160A, which is integrated with the planetary gear assembly 160B. The assembly 160 provides a rotational output in a stainless steel bellows-type somewhat flexible coupler 162 that is disposed within the motor mount chamber 164. Inside the chamber 164, the motor assembly 160 is allowed to operate in a somewhat self-aligning manner, however, the rotational movement is limited by the cork stop component 166. In this embodiment, coupler 162 extends through a torus type fluid seal 168 disposed within seal chamber 170. It can be seen that component 172 extends forward and is rotatable and fixed with thrust bearing 174. The bearing 174 provides support for all of the driving force provided by the motor assembly 160. In this regard, a rod-type threaded conversion component 172 is typically screwed with a transfer assembly as indicated at 176. The transfer assembly 176 has a ball screw or nut component 178 that is threadedly engaged with the component 172 and the threads of the normally Y-shaped yoke 180. The yoke 180 is aligned to driveably engage the drive component tabs or ears 138 and 140 (FIG. 2) when the support housing 108 is initially inserted into the receiving cavity 118, but extends to a distant position. It is configured with spaced apart drive components configured to be present. In order to ensure the unconstrained performance of the drive components as described above, it is necessary to avoid the axial creep phenomenon that is manifested as compression of the bellows 162. Generally, a sleeve is provided beyond the output drive shaft of assembly 160, while the corresponding step-down diameter in component 172 provides a shoulder for coupler 162 to strike.

  Electrosurgical cutting current as well as control inputs and outputs are introduced from cable 62 into housing 18. Two of the multi-lead type components, one of which is indicated at 180 and extends to the contact clamp 182, which are electrosurgical to the contacts 122 and 124 of the disposable component 16. Two contacts are held to provide surgical cutting energy.

  FIG. 3 also shows details of the tip of the delivery cannula 22. The tip 34 is depicted as being used to contain a relatively small tissue volume, for example with a diameter of about 10 mm. The tip includes four precursor electrode components, which are arranged symmetrically across the longitudinal axis 24. Two of the electrosurgical cutting portions of the precursor electrode are designated 184 and 185 and are located slightly in front of a hot-shaped ceramic (alumina) protective tip 190 with the tip cut off. Behind the ceramic tip 190 are polymeric compound chip components 192 and 194 that are connected to the delivery cannula 22. It can be seen that the rear component 194 holds the intake port 35 and smoke / steam block ring 37 as described above. The front component 192 provides a beveled structure for an array of five thin stainless steel leaves of the capture component, the capture component tips holding a braided stainless steel purging cable They are electrosurgically excited for cutting and perform a cutting purging operation to form a birdcage-like structure around the targeted tissue volume. Alternatively, the precursor electrode, leaf, purging cable, and cannula may be composed of a non-magnetic material (eg, titanium or nitinol), and this device is used for guidance by magnetic resonance imaging of the biopsy procedure. Make it possible. The drive applied to these capture component leaves originates from the yoke 180 and the drive component ears 138 and 140. Each of these leaves terminates with an eyelet at the leading edge, one of which is indicated at 196. Polymeric chip components 192 and 194 work together to form a guidance assembly, typically as shown at 198, during the capture process to direct the leaves at the proper distance and the appropriate attack angle. Function. The attack angle in this embodiment is 45 degrees.

  The delivery cannula 22 has a relatively small diameter, for example about 5 mm. In the forward portion 32, the previously described capture arrangement consisting of a long leaf structure made of stainless steel in the shape of a pentagon with a leading edge formed with two eyelets holding five purging cable assemblies The element is placed. Referring to FIG. 4, the capture component is indicated generally at 200 in a manufacturing stage prior to mounting the described purging cable with a polymeric guide tube. As revealed in the normal cross-sectional view of FIG. 5, the capture component 200 typically has a pentagonal cross-sectional structure and is initially chemically milled from a flat stainless steel stock, The front portion 202 is formed as a series of five leaves that are 0.003 inches thick and 0.080 inches wide. Five leaves are indicated by reference numerals 210 to 214 in the figure and extend from the pentagonal base portion 218 to the two eyelet tips 196 described above. Each leaf 210-214 is chemically milled in a longitudinally extending groove located somewhat centrally over its length. In this groove, as shown in FIG. 5, a flexible guide tube made of polyamide is fixed. The guide tubes are very small, for example, having a diameter of about 0.020 inches and a wall thickness of about 0.0015 inches. Guide tubes are indicated at 220-224 in FIG. 5 and are each bonded to the leaves 210-214. Each guide tube 220-224 slidably guides a purging cable, each as indicated at 230-234. These multi-strand stainless steel cables have a diameter of about 0.005 inches. The guide tubes 220 to 224 made of polyamide are initially attached to the grooves described above. The tube is then brought into the corresponding groove within the chemically milled groove by the process of achieving the insulating coating material and adhesion and providing the required electrical insulation for the complete capture component assembly 200. And glued. Here, the coating is about 0.001 inch thick and is a vapor phase polymerization equivalent coating such as that marketed under the name “Parylene”. Parylene is a general name for polymers. One of them, called parylene C, is polyparaxylene, a completely linear and highly crystalline material. Such coatings are available from parylene coating service providers such as Specialty Coating Systems, (SCS) of Indianapolis, IN. In FIG. 4, the structure of the eyelet at the leading edge of the capture component 200 is revealed. The leading edge, including the eyelet, is curved outwardly from the direction shown prior to cable attachment therethrough. Further, the capture component 200 is welded and attached to a drive tube or drive rod 236 that extends rearwardly into the support housing 108 and is associated with tabs or ears 138 and 140. It is connected to the drive component (FIG. 2).

  Referring to FIG. 6, the delivery cannula anterior region 32 and tip 34 are disclosed as cross-sectional views. In this figure, it can be seen that the delivery cannula 22 extends forward to the previously described polymeric compound (polyether rimide) chip component 194. The delivery cannula 22 is electrically insulated by a 5 mil thick polyolefin shrink tube 238 extending toward the border 240 at component 194. Next to the inner surface of the delivery cannula 22 are five capture component leaves in the form of a pentagon, two of which are indicated at 210 and 212. Next on the inside is a stainless steel support tube 242, which is mounted in the rear portion of the support housing 108 of the disposable component 16 and is attached to the polymeric compound tip via the delivery cannula 22. Extending forward to an outwardly extending region portion 244 coupled with component 192. This spread is useful for the support tube to withstand the further continuous forward effect that occurs during the forward deployment movement of the capture component leaf and cable. Inside the support tube 242 is an electrosurgical precursor electrode tube 246, behind the support housing 108 for the purpose of receiving and holding electrosurgical cutting energy transmitted via the electrical contacts 122. It extends towards the part (FIG. 2). As the precursor electrode tube 246 extends rearward, the support tube 242 is insulated by the polymer compound shrink wrap 248.

  The precursor electrodes are mounted as subassemblies of four stainless steel electrode wires, usually having an elongated L shape, two of which are indicated by electrodes 184 and 185. In this regard, the elongated components of the electrodes 184 and 185 are indicated by the numerals 250 and 251 respectively extending into the subassembly tube 252. Four such electrode assemblies are crimped inside this tube 252, which is then configured in a cross-shaped pattern that is crimped at the front portion of the precursor electrode tube 246. It can be seen that the use of four cutting surfaces for the electrodes results in a suitable instrument positioning. Such a configuration of opposing electrode surfaces is clarified in FIGS. 7 and 8, for example. Typically, the cutting portion of the precursor electrode may extend perpendicular to the longitudinal axis of the instrument and is cut during insertion and positioning of the instrument in an opposing relationship to a complex tissue volume. It can be configured to face tissue directly. The degree of opposing size of such precursor electrode cuts is selected to provide an effective distance that is less than the corresponding maximum diameter size developed by the capture component. In FIG. 6, an operation line 254 in which such a range is indicated by a locus on a dotted line is clarified.

  FIG. 6 also discloses a polymeric chip component 194 that serves as a guide for the leaves 210-214. At the same time, the polymeric chip component 192 constitutes five slopes that are angled 45 degrees with respect to the instrument axis 24. One of these slopes is indicated at 258 in connection with leaf 210. These slopes provide a 45 degree attack angle for the leaves 210-214 to appear during the capture procedure.

  Generally, the precursor electrodes 184-187 have a tissue cut and opposing length of about 6.5 mm to 7.0 mm for use in a maximum effective capture design of a 10 mm capture component 200. At the same time, when the effective diameter is expanded to 20 mm, the precursor electrodes or their longitudinal opposing lengths may be about 10 mm. When configured with one of the longer longitudinal lengths, the electrodes are slightly inclined toward the front and have elasticity so that electrosurgically activated purging It is possible to bend forward when the cable physically contacts the precursor electrode. In this procedure, the precursor electrode is open circuit and may be reconfigured when aligned with the capture component leaf. It can be seen that this temporary re-energization of the longer precursor electrode is effective when the electrode contracts or bends against a larger tissue sample to be captured.

  7 and 8 show front views of the tip 34 of the delivery cannula 22, with the leaf and cable shown in FIG. 7 as well as the leaf and cable expanded in FIG. In particular, the arrangement of the precursor electrodes is illustrated. In the arrangement at the start of the procedure shown in FIG. 7, the activation area indicated by the electrosurgical cuts of cables 230-234 is somewhat smaller, but that at maximum purging at the completion of the procedure is Slightly larger. In FIG. 7, the leaf tips of the five leaves 210-214 are represented in relation to a portion of the purging cables 230-234. In this configuration, the precursor electrodes 184-187 may be activated and form an arc, while the instrument 12 is handled oriented with the tip 34 facing the targeted tissue volume. . The precursor electrode structure is then deployed in conjunction with an arc that stops activation (circuit is opened) and the capture component 200 forms the activation of purging cables 230-234 with electrosurgical cutting energy. . However, in order for the cable to be embedded in tissue, the boost voltage must be sufficient to form a cutting arc between the activated portion of cables 230-234 and the opposing tissue during the boost time. Is done.

  FIG. 8 shows the deployment of the leaves 210-214, with the purging cables 230-234 "played out" and the effective diameter of the capture component is the target tissue volume. It extends to the surrounding boundaries where it can be recovered. To provide this expansion and subsequent purging configuration, it should be noted here that the cable 230 is slidable through the guide tube 220 and attached to the tip of the leaf 214. . The cable 231 is slidable through the guide tube 221 and is attached to the tip of the leaf 213. The cable 232 is slidable through the guide tube 222 and is attached to the tip of the leaf 212. The cable 233 is slidable through the guide tube 222 and is attached to the tip of the leaf 211. The cable 234 can slide through the guide tube 224 and is attached to the tip of the leaf 210.

  Referring to FIG. 9, a partial cross-sectional view of the support housing 108 of the disposable component 16 is provided. In this view, it can be seen that the support tube 242 extends to couple with the bulkhead 270 in the rear portion of the support housing 108. Tube 270 is held in place by a collar 272. Extending through the support tube 242 is the previously described precursor tube 246 that does not have an insulating shrink wrap cover 248. It can be seen that the precursor electrode tube 246 is in contact with the electrical contact 122. In this configuration, electrosurgical cutting energy may be transmitted into the tube 246 from the contact 122 and may be transmitted to the precursor electrodes 184-187. It can be seen that the rear portion of the drive assembly of the capture component is generally designated 274 and has the drive tube 236 and drive component 276 described above. The drive component 276 is shown in cross-section where the three-dimensional molded ears 138 and 140 (FIG. 2) are not represented. However, it should be noted that it is coupled to the end of the drive tube 236 so that both the tube 236 and the drive component 276 are slidably moved along the instrument axis 24 through the support tube 242. It is to let you. The yoke 180 is shown in FIG. 3, but is coupled to the ears 138 and 140 and moves the drive assembly 274 forward by impactably coupling with the ears or tabs 138 and 140 (see FIGS. 2 and 2). FIG. 3).

  Purging cables 230-234 extend rearward into the internal cavity 278 of the support housing 108 and outboard of the drive tube 236. Two of these purging cables are shown symbolically with reference numerals 230 and 231. These cables extend slidably through the corresponding five channels extending through the drive component 276, one of which is indicated at 280. Cables 230-234 extend further until they are fixedly connected to the polymeric cable terminator component 282. Component 282 is slidably mounted on support tube 242 and includes a forward annulus or collar 284 that is pressed against cables 230-234. These cables then extend through the central flange portion 286 of the component 282 to have a rigid electrical connection with the rear annulus or collar 288. The collar 288 is then coupled to a flexible electrical cable 290 that is connected to an electrical connector 124 where it continues to the cable terminator component 282 to slide forward. Accordingly, an electrosurgical cutting current is supplied from the connector 124, cable 290, and metal ring 288 to the cables 230-234. The cable terminator component 282 is stabilized by two outwardly extending ears or tabs, one of which is described in FIG. 2 as a tab 126 incorporated within the stabilizer slot 130. Is. A cable stop 292 is provided in front of the cable terminator component 282. This collar-type stop 292 is adhered to the support tube 242 at a location that defines the maximum diameter range developed by the leading edge of the leaf of the capture component 200. This maximum diameter range extends beyond the tissue volume and is symbolically shown in the figure passing through the targeted tissue volume indicated by the dotted line 294 near the middle. In the arrangement of the capture component 200 as shown, the cable terminator component 282 may begin to strike and couple to the cable stop 292, and the motor assembly 160, conversion component 172, and transfer assembly 176 (FIG. 3). ), The purging cables 230 to 234 are pulled when the drive assembly 274 continues to be driven forward. Note that when the drive assembly 274 moves forward and the placement of the capture components as shown in the figure is achieved, a latch such as that shown generally at 296 is used. Passing through resilient latches located on two opposite sides of the assembly. Here, the latch assembly includes resilient latches 298 and 300. The assembly 296 limits the degree of contraction of the capture component post means for the basic release of the leaf shown in the figure to access and facilitate the movement of the retrieved tissue specimen. Eventually, a drive safety stop mechanism comprised of stop piece 304 is secured within cavity 278, beyond the position showing maximum purging or contraction of capture component 200 for the selected maximum diameter size of the capture. All the drive assemblies 274 are restricted from moving forward. Such unnecessary movement may cause, for example, breakage of the cable stop 292 to interrupt the forward movement of the cable terminator component 282. As drive assembly 274 continues to be driven forward and drive component 276 is proximate to safety stop component 304, the leaves of capture mechanism 200 can be purged inward from one another to define the surrounding structure surrounding the tissue volume to be removed. At this time, the relative length of the activated electrode cutting components of the purging cable begins to decrease to ultimately determine a very small activated cutting area. In FIG. 10, the positioning of the components following the placement for the procedure shown in FIG. 9 is shown. It can be seen that drive component 276 and its associated drive tube or lot 236 of drive assembly 274 are driven further forward, and drive component 276 is positioned proximate to drive safety stop mechanism 302. Cable terminator component 282 strikes cable stop 292. This tensions the five cables 230-234, purges the target tissue 294 surround, and the capture volume of the capture component 200 surrounds the tissue volume.

  Here, the pulling force performed by the cables 230-234 is such that the control feature of the control assembly can be aware of a forward stall condition in a portion of the motor 160A. When this happens, the cutting energy for the cables 230-234 is terminated and the motor assembly 160 is energized to drive in the reverse direction. Accordingly, the yoke 180 moves away from the joint that freely contacts the tabs or ears 138 and 140 of the drive component 276 and returns to the home position. As such, the reusable component 14 of the instrument 12 is reconfigured for subsequent use. Typically, the physician captures fluid by releasing the coupler 42 of the inspiratory line 36 and inserting a plug 148 therein. The disposable component 16 is then removed by unscrewing the front screw connection at the connector 26 and the ears or tabs 138 and 140 are manually returned to strike the latch assembly 296. This generally returns the capture component 200 to the arrangement as shown in FIG. 9, allowing the physician to easily access the collected tissue sample.

  Referring again to FIG. 1, the procedure performed by the system 10 includes the process of administering a local anesthetic at the skin level in the area of the initially intended biopsy. Switch 82 is activated to power on console 64 and cable 62 is attached to connector 68. If the connection is successful as a result of the test, the green LED 86 is lit. The physician depresses the start / reset button 92 on the console 64, which results in the patient safety circuit monitor test being performed, and if this test fails, the red LED 106 and audio cue provide an output on the pulse. The removable component 16 is mounted inside the reusable component 14 and skin incision is achieved using a cold scalpel to a width of 2 mm and a depth of about 4 mm, which is greater than the maximum width of the tissue volume to be removed. Executed. Smoke / steam evacuator 46 is switched on by foot switch 50 and tip 34 of delivery cannula 22 extends to the incision so that the precursor electrode on the tip is at least 3 mm below the skin surface. As such, these electrodes are initially implanted within the skin. The positioning mode is then initiated by depressing and continuing to depress the energy / position of the foot switch 88b or the button switch 57 of the housing 18 and affects the initial boost. The LED 96 corresponding to the array 60 is turned on as well as the corresponding LED. Voice cues are provided with regular tones. The tip 34 of the delivery cannula 22 is advanced to a position that closely opposes the tissue volume to be removed. When such a position is reached, the positioning mode is terminated (foot switch 88b or button switch 57 is opened), the arm / disarm tissue capture button or switch 56 or foot switch 88A is momentarily depressed, and LED 98 is Of course, the LEDs on these switches are lit and the system 10 enters arm capture mode. During this mode, switches 57 and 88b are not available. The start capture button or switch 58 or foot switch 88c is then depressed and continues to be depressed, the LED 100 as well as the LEDs on the switch 58 are lit, the capture mode is initiated, and the motor 160a (FIG. 3) is activated. Initially, the yoke assembly 180 is advanced forward for 1.5 seconds while the motor current is monitored to ensure proper operation. When the yoke 180 engages the ears 138 and 140 of the drive component 276, the motor assembly 160 stops moving. The electrosurgical generator applies a first boost and then normal cutting energy is applied to the purging cables 230-234 (FIG. 7) and after 1.5 seconds the motor assembly 160 begins to deploy the capture component 200. Energized as much as possible. During the operation of the purging cables 230-234, the above described regular sound is provided from the console 64. This capture mode is continued until the capture component 200 reaches the arrangement described in FIG. In that case, when the front stall condition is confirmed in the motor 160a, the energy generation in the forward direction of the motor assembly 160 is finished, the motor rotates backward, and the transfer assembly 176 is pulled back to the initial home position. The LED 102 on the console 64 illuminates, as well as the output of the corresponding LED in the array 60, ending the sound indicating that the electrosurgical current has been applied. Delivery cannula 22 is removed from the patient, plug 148 is attached to vacuum connector 42 and connector 26 is rotated to allow removal of disposable unit 16. Upon removal of the removable unit, the ears or tags 138 and 140 may be manually retracted and coupled with the latch assembly 296 so that the capture component 200 can access the tissue sample as shown in FIG. Makes it possible to become.

  Electrosurgical operation at the aforementioned boost voltage level at the start of a procedure associated with both the positioning mode and capture mode or at the resumption following a pause activity that occurs in connection with the opening of the finger switch 58 or foot switch 88c and the lighting of the LED 104 Energy is applied. This degree of application is a boost interval until the construction of the arc for enabling cutting is started at least, and is a predetermined boost interval based on experience such as 3/8 second.

  Control over the cutting energy supplied to the parsing cables 230-234 from the electrosurgical generator function is, among other things, a conventional design approach where the generated power needs to be durable and effective in cutting the arc. At the same time, it is based both on the fact that the tissue, instrument, or recovered tissue specimen in the vicinity of the cut is not excessively damaged. However, in the case of the system 10, additional criteria arise. When manifested as a tissue impact portion of the cables 230-234, the probe electrode changes the area of the surface area during the procedure. It is initially excited under a boost voltage and has a shape somewhat similar to a point light source. Then its peripheral area is increased to resemble a gradually expanding line light source, after which it returns to a shape resembling a point light source as well. That is, the system 10 requires an increase in power output during the initial expansion of the capture component 200, and subsequently reduces the power output characteristics when shrink parsing envelop occurs. In addition, at the beginning of this procedure, a probe electrode assembly, either a precursor electrode or a capture component cable, is implanted in the tissue and extends between the cut portions of the cables 230-234 and the tissue to be cut. The boost voltage is invoked during the start of the existing arc and the boost time interval appropriate for steam generation. In effect, it is this arc that causes the cut, not the cable. The active electrode portion is simply slid into the vapor generated from the adjacent tissue cell layer. Thus, the control can withstand the arc throughout the procedure.

  Conventional electrosurgical generators are designed to be implemented with probe electrodes of a fixed configuration or shape, such as blades or rods. Depending on the surgeon's skill and experience to produce the initial arc formation or arc generation by moving the probe electrode up to the target tissue, e.g., up to about 1 millimeter, the necessary cutting arc Production is achieved. Referring to FIGS. 11-A and 11-B, this technique is depicted. A patient is shown at 310 with a large dispersive electrode 312 attached to the back that provides a return to the electrosurgical generator 314 as shown at line 316. The generator 314 supplies cutting energy to the fixed shape exploration electrode 318 as indicated by line 312.

To achieve arc initiation, the output of the electrosurgical generator should address an appropriate range of impedances, for example, 1300 to 1500 ohms. This impedance is inherently resistive and has a resistance R _tissue as indicated by the body of the patient 310 as indicated by the distance from B to C, and by a probe electrode spaced from the tissue 322 of the patient 310. Combined with the impedance or resistance that is generated, its value ranges from about 300 ohms to about 500 ohms. In FIG. 11-B, the distance L _g is shown as an enlarged distance from A to B. Referring to FIG. 12, this resistance R _AB is plotted with dashed lines 324 and 326 against changes in distance L _g . As indicated by the dotted line 324, where the value of L _g becomes a value greater than about 2 millimeters, like resistance is an electrical arc approaching infinite be noted that not generated. However, as the fixed configuration probe electrode approaches a distance L _g of about 1 millimeter, the resistance can be combined with the resistance R _tissue (B to C) to form an arc of about 500 ohms to 1000 ohms. Shown to be ohms. Using the inherent resistance R _total indicating A to C, the cutting arc is maintained as in the general formula: R _total = R _tissue + R _arc . Using the above arrangement, a conventional electrosurgical generator is operated with a fixed output power and a variable applied voltage. Therefore, this output power level is kept in a safe range, for example, about 80 watts to about 100 watts.

Methods other than occur at electrode separation, depending on the surgeon's technique achieved using system 10, even when the probe electrode is implanted in the tissue at initial and restart and there is no space available Thus, the equivalent of arc formation is performed. A tissue solid adjacent to the probe (active portion) or precursor electrode of the cable 230-234 that was initially exposed and applied to the tissue by application of a boost voltage (V _boost ) for a short time (t _boost ). Vaporization of the structure occurs. This causes the equivalent of initial separation and achieves the necessary impedance for the start of the arc. The time interval for applying the boost voltage may be, for example, a fixed duration of less than about 500 milliseconds, or may be defined by the generation of an arc after applying this boost voltage. The impedance change R _total at the time of the arc formation shows a sudden change, and a corresponding sudden drop occurs in the flow of the output current. Therefore, the formation of the electric arc is easily detected so as to end the boost voltage application.

Referring to FIG. 13, the performance of the system 10 for an experiment carried out using a 10 mm capture maximum diameter range and plate bacon is depicted. In this figure, the calculated total resistance in ohms is plotted against time. Furthermore, the applied maximum amplitude voltage is plotted with respect to time. In addition, the current being witnessed by the DC motor 170a is shown. At the start of the procedure, prior to applying the boost voltage, the total resistance is equal to the tissue resistance R _issue as described above with respect to the distance B-C in FIG. 11-A. The 500 ohm level is indicated by the dashed area 328. The boost voltage is applied to the cable electrode to initiate a boost time interval with a boost voltage with a maximum amplitude of 1400 volts, as shown by line 330. The boost voltage is imposed for a time interval T _boost of 500 milliseconds, after which the applied voltage drops rapidly, as indicated by the solid line portion 334. During the boost time interval, as indicated by the dashed line portion 336 and the vertical dashed line portion 338, after about 200 milliseconds, an arc is formed and the total resistance rises abruptly, causing the applied voltage at the line portion 334 to increase. Reaches 1500 ohms near the end of the fixed boost time interval, as is shown to drop to the normal cut voltage level indicated by the horizontal solid line portion 342. It can be seen that this normal cutting voltage applied has a maximum amplitude of 1000 volts. As indicated by the vertical dashed portion 344, the motor 160a is energized substantially simultaneously after the head start time interval identified as t _hs from application of the boost voltage. The expansion of the leaves 210-214 is initiated when the application of voltage to the motor assembly 160 initiates the deployment of the cables 230-234 toward the maximum diameter peripheral area. In order for this to occur, the length of the cable coupled to the active cutting of the tissue and, as a result, the area of the surface is expanded to correspond as indicated by the dashed curve portion 346. The total resistance drop starts. When the maximum perimeter region of the leaf tip and the length of the active cable cut reach the maximum value, the resistance has reached its minimum value, as shown by the vertical dashed line 348, and the output is increased. With this, the applied current reaches the maximum value.

If the time interval of the procedure is continued beyond the time indicated by the vertical dashed line 348, the usable surface of the cables 230-234 used to cut the tissue will continue to be parsed afterwards. Decreasing and, as indicated by the dashed curve portion 350, the total resistance again increases, thus reducing the effective cable length associated with tissue cutting. During this time interval, the DC motor current initiated at line portion 344 gradually increases as indicated by dashed line portion 352 until a motor stall threshold as indicated by current level 353 is reached. The motor drive voltage and the normal cut voltage end abruptly as indicated by the broken line portions 354 and 356, respectively. After this procedure, the total resistance R _total returns to the _tissue resistance R _tissue as indicated by line portions 358 and 360.

  Referring to FIG. 14, a plot of peak current output occurring during the time interval shown by the procedure performed in connection with FIG. 13 is illustrated. In this drawing, the voltage application time at the aforementioned boost voltage is indicated by a vertical dotted line 362. The voltage is applied at the boost level and continues as indicated by the horizontal dotted control line 364, which ends after a 500 millisecond boost interval and is indicated by the vertical dotted line 366. It becomes like this. The applied voltage is then maintained at a constant voltage level as shown by the horizontal dotted line 368. During the application of the boost voltage as indicated by control lines 362 and 364, during a very short time interval of approximately 200 milliseconds, a sharp rise as indicated by curve 370, the peak level indicated by reference numeral 372 is achieved. A very rapid increase in current is shown, followed by a very rapid decrease. During this 200 milliseconds, an effective initial separation is performed by tissue cell vaporization. Furthermore, the plot can be considered consistent with the power dissipation during the procedure. Curve portion 372 may be detected for the purpose of boost voltage termination, so the boost time corresponds to the arc formation time.

Returning to FIG. 13 and recalling it with the system 10, the power applied from the electrosurgical generator is changing according to the application of the boost voltage and the change in the cable electrode shape, and as a handy example, estimate the power dissipation. It is possible. The start of the application of the boost voltage is indicated by line 330 and faces a tissue resistance of 500 ohms. Thus, a power loss of about 500 watts has occurred under an applied boost voltage with a maximum amplitude of 1400 volts until the arc is formed. However, the power is generated in a very limited area, for example, at a time interval of about 200 milliseconds, for a very short time interval before the arc is formed, as shown by dashed line 338. . As soon as the arc is formed as shown by the dashed line 338, the impedance indicated by the arc is added to the impedance of the 500 ohm tissue, and the power dissipation drops to about 167 watts. This 167 watts is a bit high, but only maintained until the boost voltage is removed as shown by the vertical line 334. Thereafter, a normal cutting voltage with a maximum amplitude of 1000 volts with a power loss of about 85 watts occurs. However, now that the exploration electrode expansion has begun, the cable length has been expanded and the maximum peripheral area of the leading edge of the capture device has been achieved, as shown by the dashed positioning line 348, When the total resistance drops to about 800 ohms, the power rises again. Thus, the power will increase from about 85 watts to about 159 watts. However, this 159 watt power value is associated with the very wide expenditure of linear source electrodes being in the maximum linear range. Next, a parsing action occurs and its linear range decreases towards the point value, as does the power loss, again approaching 85 watts at the end of capture. As is apparent from the foregoing, electrosurgical energy at a boost voltage level (eg, a maximum amplitude of 1100 volts) can be applied continuously throughout the procedure. In short, this boost time interval t _boost is extended to encompass the entire time of the procedure, regardless of either the placement of the precursor electrode or the capture by the parsing cable. However, as a result of this extended boost time interval, excessive power is generated during the biopsy procedure that can cause significant depth of thermal damage to the biopsy specimen and the surrounding healthy tissue. There is a possibility that.

The boost current is applied for a time interval between 100 milliseconds and 1000 milliseconds, preferably for a time t _boost between about 250 milliseconds and about 750 milliseconds. The boost voltage V _boost has a maximum amplitude in the range of about 1000 volts to about 2000 volts, and preferably has a maximum amplitude in the range of about 1100 volts to about 1300 volts. At the end of the boost time interval, the electrosurgical energy drops to the normal cutting voltage level V _cut as shown by dashed line 1446. The cutting voltage V _cut is selected from a maximum amplitude in the range of about 700 volts to about 1200 volts, and preferably a maximum amplitude in the range of about 800 volts to about 1000 volts.

  In order to achieve the above-described variable output power performance required to maintain the cutting arc, the electrosurgical generator uses constant voltage control. Such a control method typically results in an unstable vibration output since the necessary cutting arc exhibits a negative dynamic impedance. However, stable output is possible if the generator of the present invention is used.

  Referring to FIG. 15, a schematic block diagram illustrating an electrosurgical generation mechanism and a control assembly incorporating a console 64 is illustrated. In general, the electrosurgical inputs to the parsing cables 230-234 and the instrument precursor electrodes are provided at an operating frequency of about 350 KHz. However, this operating frequency may be selected in the range of about 250 KHz to about 10 MHz. As described above in U.S. patent application Ser. No. 09 / 472,673, now U.S. Pat. In the case of such a bipolar or quasi-bipolar instrument format, the operating frequency may be as low as possible, on the order of about 100 KHz. The maximum diameter values of various capture devices, and the associated longitudinal capture dimensions, are based solely on the cable stop 292 (see FIGS. 9 and 10). Depending on the configuration, the motor assembly 160 can be operated along with other load characteristic states indicative of fault conditions, as well as detection control of forward and rear stall conditions. In the drawing, the input of a conventional AC line is shown as a line 380 that extends to an electromagnetic interference (EMI) filter shown at block 382. As indicated by line 384 and symbol 386, the filtered output is then passed through a fuse to a front panel output on / off switch function indicated by block 388. This switching function is described with respect to 82 in FIG. The filtered function 388 passes this filtered input to the power factor correction boost converter indicated by line 390 and block 392. Converter 392 rectifies the AC input to a DC current to boost the DC voltage level to a stable provisional level and simultaneously generates a sine wave input current waveform that matches the sine wave input voltage waveform. This provides a high power factor to reduce line current harmonics. Converter 392 provides the provisional voltage as a 380 voltage DC bus as indicated by lines 394 and 396. Providing power factor correction characteristics at block 392 results in various useful attributes. Less current is drawn compared to conventional electrosurgical generators, and the device can be used universally with standard power supplies worldwide. More importantly, the converter 392 produces a pre-regulated provisional voltage on line 394 that allows the electrosurgical generator function to optimize the following link inverter.

  Line 396 functions to provide a DC input to a main low voltage power supply (LVPS) and an auxiliary low voltage power supply shown in blocks 468 and 470, respectively, in relation to lines 402 and 404. In view of the criticality of the control system associated with the device 12, a redundant low voltage power supply is used. In this context, in the absence of such redundancy, a failure of the low voltage power supply can cause the entire control system to shut down at a specific point in time in the critical time interval of the procedure in the near future. There is sex.

  A 380 voltage DC regulated at lines 394 and 396 is similarly shown at block 406 which functions to provide a very specific motor voltage to the motor drive circuit as shown at line 408 and block 410. Directed to low voltage power supply. In controlling the motor voltage, for example, a level around 10 volts is important as long as it is the voltage level that provides the appropriate forward speed of the capture device leaf and cable components. In this context, the deployment of the leaf and electrosurgically excited cable is measured in terms of millimeters / second. If a drive that is too fast is applied, the excited cable will be pressed against the tissue and will not be properly cut, resulting in a capture failure stall based on part of the control system response. It may occur. The control system, for example, determines the completion of the parsing operation based on detection of forward stall current and operates the motor drive 410 so that the control system moves forward before the capture device operates properly. This is adapted to an abnormality in the motor drive device caused by a catching phenomenon in which the stall is detected. Since the leaf and associated parsing cable advance speeds are carefully controlled, for example, stall conditions detected before a predetermined initial test time interval initiated using initial motor operation all indicate a malfunction. It is known that A device 14 or “handle connector” 68 is shown in simplified block 412, in which communication motor drive input communication indicated by an arrow 414 connected to a motor drive function in block 480. The situation is illustrated. The term “handle” refers to the reusable component 14. Further, control to the motor drive shown by block 410 is provided from a control arrangement that includes a control circuit and a drive circuit, as shown by block 416 and bi-directional arrow 418.

  Returning to line 394, the converter further turns into a 100 KHz link inverter, indicated by block 420, which is shown to be under control of the control and drive circuit board of block 416, as indicated by the double arrow 422. A 392 stabilized 380 volt DC output is introduced. This control is required for the effect of constant voltage control of electrosurgical output energy, thereby achieving arc maintenance, non-vibration performance while adapting to the negative dynamic impedance of the cutting arc. Has been. Also shown is the AC (square waveform) output of the inverter 420 indicated by line 424 to one side of the isolation transformer indicated by block 426. Transformer 426 provides an output as shown at line 428, which is rectified and output as shown at block 430 to generate a DC link voltage having a value of about 100 volts at line 502. Filtered. The amplitude of this link voltage on line 432 is very effectively controlled and functions to change the amplitude of the output of the system. Line 432 is directed to two relay disconnectors as indicated by block 434. These relay disconnectors are controlled from a control and drive circuit board 416 as indicated by arrow 436. The DC link voltage is then directed to the RF inverter as indicated by block 440 as indicated by arrow 438. Inverter 440 operates in a controlled relationship with a control and drive circuit board indicated by block 416 as indicated by arrow 442. Note that placing this relay disconnect 434 in front of the RF inverter 440 disconnects the input to the RF inverter 440 itself in the event of a failure or other anomaly. Inverter 440 is a type of conventional resonant or tank circuit that is tuned to a specific frequency. The output peak-to-peak voltage amplitude is controlled by the amplitude of the DC link voltage. That is, for boost and normal cutting performance, the output voltage amplitude for the negative dynamic impedance arc drive is held constant and its frequency is held constant as well.

  The output of inverter 440 is the output of a high voltage transformer that increases its amplitude to a maximum amplitude of up to about 800 volts to about 1000 volts for normal (non-boost) disconnection, as indicated by line 444 and block 446. Oriented on one side. The output of this transformer output stage 446 at line 448 is an arc-generated electrosurgical disconnect output, which is eventually a series of relays at the high voltage output stage indicated by block 450 and indicated by arrow 452. Directed to either the precursor electrode input or the capture device cable indicated by arrow 454. Control above the stage 450 indicated by block 520 is indicated by arrow 456.

  The connector 80 of the console 64 that is electrically associated with the dispersive electrode 70 is indicated by block 458. In addition to providing a return to the high voltage output stage 450 as indicated by line 460, the connector is connected to a personal circuit safety monitor (PCSM) as indicated by block 462. Each of the distributed electrodes 72 and 74 is connected to the monitor circuit 462 as shown by lines 464 and 466 and is controlled and provided to the control and drive board 16 as shown by arrow 468. The As described above with respect to return electrode 70 in FIG. 1, the system operates unipolar and utilizes a dual component dispersion pad as the return electrode. A small high frequency current is applied along the patient from one pad at 72, for example 74, as indicated by the double arrows 464 and 466, respectively directed to the leads of the return electrode connector RE1 and RE2. May be directed to another pad in to verify the resistance of the tissue between those pads. The PCSM circuit 462 applies a signal of about 10 volts to the two return electrode pads at 50 KHz to verify the appropriateness of the resistance. Only after such verification is made, the system is allowed to allow the performer to continue this procedure by proceeding to a ready state. If the PCSM test is not satisfied or does not pass the PCSM test, the system will not proceed and both a visual pulse alarm and an audio pulse alarm will be generated. The PCSM circuit 462 also performs a self test when the on / off switch shown at block 388 is on.

  Front panel controls, such as those described for console 64 in FIG. 1, are shown at block 470. These controls and block 474, shown at line 472, are associated with the front panel circuit board, which is shown at line 476, from the drive board closed at control and block 416. Provide input and output. Both the control and drive board receive input from the foot switch 88 as indicated by block 478 and switching line bus arrow 480. Input from button switches 56-58 of component 14 is indicated by arrow 482, while output to the LED array at 60, for example, is indicated by arrow 484. Finally, vacuum switch 51 is indicated by the same numbered block as arrow 53 previously described as extending to block 416. An arrow 53 indicates two lead inputs.

By using the circuit arrangement thus described, a main circuit is created between the AC input on line 380 and the isolation transformer 426. A second low voltage circuit is derived from the output of the isolation transformer 426 providing the aforementioned DC link voltage. This second circuit extends to the high voltage transformer indicated by block 446. With this circuit arrangement, a high voltage circuit with a system for generating the electrosurgical cutting output is achieved. These three different circuit areas are incorporated into the system using various insulating barriers. In this connection, some components are placed under the control of safe additional low voltage circuits (SELV), while all other circuits are completely isolated from potential contact. . In the case of a medical device attached to a patient, more stringent consideration is required to ensure that no current flows from one device to another device attached to the patient, for example. Referring to FIG. 16, there is shown an isolation and isolation diagram that may be associated with the system diagram of FIG. In FIG. 16, circled insulation cords 1 to 7 are arranged. Each of these codes corresponds to an insulation type: BI, BOP, RI, RI, BI, RI, and OP. These insulation types can be further identified as follows:
“OP” —functional insulation “BOP” —basic insulation between different polarity components “BI” —basic insulation providing first level protection against electric shock “RI” —reinforced insulation Referring to FIG. Shows the conductive enclosure of the console 64. A patient contacted by an exploration electrode (AE) as indicated by arrow 504 and a return electrode (RE) as indicated by lines 506 and 508 is illustrated using symbols at 502. The non-conductive handle of the instrument 12 is indicated by block 510 and the cable and connector cover 62 is indicated by 512. The non-conductive front panel of console 64 is also indicated by block 514.

The AC input to the control system is indicated by the lines illustrated by lines 516-518, neutral line, and ground line, respectively. Thereby, the main circuit described above is started. Note that insulation cord 1 extends between line 516 and chassis 500. Next, it is shown that the main circuit extends to the transformer function indicated with the symbol at 520 and has a boundary code 3 which is a high voltage insulation boundary. In addition, a conversion to an approximately 100 volt DC link voltage as indicated by line 522 with an insulated boundary cord 4 is provided. The system extends through the RF inverter shown at block 524 and previously described 440 to the high voltage transformation function shown generally at 526 with an insulating barrier cord 5. This conversion function is described in block 446 of FIG. This conversion function 526 produces a high voltage output indicated by line 528 along with the insulation cord 6. The system then extends from there to the patient 502 through a cable 512 to the instrument 12 that includes a blocking capacitor 530, a front panel 514, and a probe electrode 564. It can be seen that these return electrodes, indicated by lines 506 and 508, are now associated with the PCSM circuit indicated by block 532. The PCSM circuit is further insulated with an insulation barrier 5 before being functionally associated with the low voltage control circuit shown at block 536. 536 and the like are shown insulated with code 4 for these low voltage control circuit chassis 500. The predetermined input to and output from this low voltage control is indicated by a double-headed arrow 538 that extends across the front panel 514, cable assembly 512, and equipment housing assembly 510. The footswitch function 88 shown in FIG. 1 and described here in the dotted block illustrated in connection with the bus arrow 540 is shown with a transformer function 544, isolated from the circuit 536, along with the insulation cord 3. . Similarly, the vacuum switch 51 is identified by a dashed block with an arrow 53 extending to the transformer function 592. The +12 volt DC input to circuit 536 as shown by lines 546 and 548 is isolated as shown by the transformer function 550 associated with insulation cord 3. The DC link converter function indicated by block 552 is isolated from the low voltage control circuit 536 as indicated by the transformer function 558 with the isolation cord 3. Note that the link converter circuit 552 is connected between the line input at line 516 and the natural input at 517. Each line is indicated at 560 and 562. The control output to the RF inverter function in block 524 is indicated by line 564 and extends from the low voltage control circuit 536. The function shown at block 524 may operate in conjunction with a lower level dc link voltage as shown at lines 566 and 568. Note that finally, the insulation cord 7 is associated with the interface between the cable assembly 512 and the instrument 12 as indicated by block 510.

  The console 64 has a sequence of circuit boards, certain of which are identified in connection with FIG. 15 as control and drive boards and front panel boards. In general, a circuit board is a motherboard identified as a daughter board or a main power circuit board.

  Next, the discussion shifts to functions and components associated with the power circuit board (board). These components are described with respect to FIGS. 17-A, 17-B through 23-A, 23-B, and 23-C as follows. Figures 17-A and 17-B need to be interpreted as labeled on it. Referring to FIG. 17-A, a line input is provided to the EMI filter 382 shown again in this figure. This device 382, designated "Rear Panel Power Entry Module", is provided with a line filter with AC input model number 511100.33.3 marketed by Schurer, Inc. of Endigen 79343, Germany. It's okay. This filtered output from device 382 is shown as a line illustrated at 580-582, a neutral line, and a ground line, respectively. Lines 580 and 582 are directed to fuses f1 and f2, as well as components that provide additional EMI filtering. These components include capacitors C1 to C3, a double inductor type device L1, an inductor L2, and a discharge resistor R1. Additional protection is provided by varistors 584, 586 and capacitor C4. This filtered input is then extended across the front panel power switch as shown at 82 in FIG. 1 and block 388 in FIG. Closing switch 82 energizes lines 588 and 590, respectively. The inrush current caused by the presence of a very large hold-up capacitor in the system is controlled by a negative temperature coefficient thermistor 592 that extends across contact K1: B of relay K1 in line 588. Looking briefly at FIG. 18, the solenoid operating component of the relay K1 is designated K1: A. This solenoid operating component is performed in line 596 in connection with the RELAY_IL control input. Any inductive spike caused by solenoid control is controlled by diode D1. Referring again to FIG. 17-A, a diode D2 extending from line 598 within line 598 and a diode D3 extending from line 590 to line 600 are illustrated in FIG. It functions to produce a rectified AC_SENSE signal with an internal resistor R2. This AC_SENSE signal on line 602 is used to generate an indication to the control that the input is at a sufficiently high voltage amplitude to operate the system.

  FIG. 17-B shows that lines 588 and 590 extend to a rectifier 604 that produces a semi-positive waveform at lines 598 and 606. Small filter capacitors C5 and C6 extend between these lines. Rectifier 604 may be provided as Model No. D25X360 marketed by Schindengen America, Inc., Westlake Village, California. The aforementioned power factor corrected boost transformer composed of transistors Q1 and Q2 which are implemented with the main components including inductor L3 and diodes D4 and D5 under switching control of the drive for driving the controller represented by block 608. A full-wave rectified AC voltage is applied across the capacitor at the input of the generator (shown schematically at 392). In this regard, a control line 610 extends from the output A of driver 608 to the gate of transistor Q1, thereby providing peripheral components including resistors R3 and R4, diode D6, capacitor C6, and bead B1. Note that its switching control in relation to is acted upon. In a similar manner, switching control at the gate of transistor Q2 is performed via line 612 in relation to resistors R5 and R6, diode D7, capacitor C8, and bead B2 by output B of driver 608. Device 608 is controlled by a DRV_PFC signal on input line 692 and is adapted to receive the main circuit low voltage input, + 12V_PRI, on line 616 and is configured in conjunction with capacitors C9 through C11 and resistor R7. ing. Device 608 may be provided, for example, as a model No. MIC4424 BiCMOS / DMOS buffer / driver / MOFSET driver marketed by Micrel, Inc. of San Jose, California. The pre-stabilized 380 volts across lines 618 and 620, described above, is a very large hold-up that functions to protect the system from unpredictable changes such as transient drops and rises induced by line inputs. • Applied across capacitors C12 and C13. In essence, these capacitors provide energy storage to “get through” such anomalies. A temporary voltage of 380V is tapped on line 626.

  Further shown in the figure is an AC current sense signal (AC_l) on line 622 extending from line 598 associated with parallel resistors R8 and R9. This signal is used in connection with the power factor control (FIG. 24-B) associated with the corresponding AC voltage sense signal (AC_V) on line 624 extending from line 606. Thus, the detailed circuit configuration shows the main circuit passing through the main transformer function 426 and subsequently extending to the second circuit.

  Referring to FIG. 19, an overtemperature switch attached to the heat sink inside the console 64 is shown at 628. A low logic TRUE signal TEMP is generated on line 630 when an overtemperature condition exists.

  Referring to FIG. 20, an adjustment device for generating important motor voltage inputs is shown at 636. Device 636 may be, for example, a Model # LM2941 Low Dropout Adjustable Regulator marketed by National Semiconductor Corp. of Santa Clara, California. This device functions in conjunction with the + 12V input on line 638 and is configured with capacitors C14 through C16 and resistors R10 and R11 to provide a motor voltage output V_MOTOR on line 640.

  As described above with respect to FIG. 15, the control system includes two low voltage power sources as described in connection with blocks 398 and 400. These redundant power supplies provide a logical ORed output. FIG. 21 clarifies the phase of these same circuits by the two numbers described above. A low voltage power supply circuit connects (+ ap) a high voltage output of + 380V to a line 704 that incorporates fuse f3, which is directed to one end of the primary side of transformer T1. The opposite end of the primary side is connected to the ground of the main circuit that is ultimately supplied from line 650. The input switched to the input side of the transformer 1 is executed by a control device or control device 652 configured with capacitors C17 and C18, resistors R12 to R14, and diodes D8 to D10. The switching control 652 is called a “smart power switch” in which an adjustment circuit including a power transistor is incorporated together with a PWM control circuit and the like. This device may be provided by Model No. TOP234Y Integrated Off-Line Switcher marketed by Power Integrations, Inc. of Sunnyvale, California. Galvanic isolation is provided by transformer T1 and the secondary side of transformer T1 is connected so that there is a + 12V low power supply to ORing diode D11 on lines 654 and 656. This output is rectified by diode pair D12 and filtered by inductor L4 and capacitors C19 to C21.

  Feedback control to the switching controller 652 extends on the secondary side of the transformer T1 to a secondary side input network, indicated generally at 660, and includes resistors R15-R18, capacitors C12 and C23, and a diode D13. Drawn on line 658 consisting of: Network 660 provides a voltage proportional signal to the input diode of opto-isolator 662. This output of the opto-isolator 662 is the primary power supply by changing the input from the connection with the second part of the secondary side of the transformer T1 in which the line 664, the diode D14 and the capacitor C24 are incorporated. On the roadside, a feedback signal indicating the voltage level on line 658 is returned. This signal is varied by opto-isolator 662 and directed to the control input of controller 652 via line 666.

  A wide variety of relays are used for motor operation, safety, and control purposes in dual electrosurgical cutting procedures and the like. Referring to FIG. 22, a relay controller 670 is shown with five series of relay input control signals at its IN1 to IN5 input terminals. These input signals are provided by a programmable logic circuit (PLD) described below. Controller 670 may be a High-Voltage, High Current Darington Array, model number ULN2004, marketed by Micro Systems, Inc. of Worcester, Massachusetts. This device 670 consists of a + 12V input and a capacitor C25 and functions to provide a drive output to a series of relay solenoid components. In this connection, relay solenoid components K6: A and K7: A are connected to terminal OUT1 and line 672 and from there to + 12V. Solenoid components K2: A and K3: A are connected by line 673 to output terminal OUT2 (between + 12V) and from there to + 12V. Relay solenoid components K4: A and K5: A are connected by line 674 to output terminal OUT3 and from there to + 12V. Relay solenoid K8: A is connected to output terminal OUT4 via line 675 and is further connected to + 12V therefrom, and relay solenoid K9: A is connected to device 670 via line 676. Is connected to the terminal OUT5, and is further connected to + 12V from there. The latter two solenoid actuators function to selectively actuate or drive each dual relay contact K8: B, K8: C, and K9: C, K9: C, directional to motor 160a Provide complete control. The inputs to the contacts K8: C and K9: C are connected to the V_MOTOR input described above on line 678, and the corresponding inputs of contacts K9: B and K8: C are connected to line 680. It will be seen that line 680 is connected to the secondary circuit ground with resistor R19 and filter capacitor C26. It can be seen that a positive motor drive output MOTOR + is provided on line 682 and a negative or heteropolar motor drive output MOTOR− is provided on line 684. Note that MOTOR + is connected to one side of relay contact K9: B by line 688, and line 684 is connected to one side of relay contact K9: C by line 686. Accordingly, forward motor drive is provided by energization of relay K8: A, while backward motor drive is provided by energization of relay K9: A. Motor current is monitored on lines 681 and 680 to provide a “MOTOR_I” signal that is used to evaluate instantaneous motor current draw or load characteristics.

  FIG. 23-A and FIG. 23-B need to be interpreted as labeled on it. Referring to FIG. 23-A, a more detailed view of the 100 KHz link inverter described in block 420 in connection with FIG. 15 is shown. Inverters are shown with generally the same number notation. The inverter 420 induces what it can call “soft” switching to drive the primary side of the main isolation transformer T6 previously described in block 426 of FIG. As long as it is an “inverter”, it is implemented in a unique way for electrosurgical applications. Furthermore, with this transformer, the numerical notation is considered the same as before. Inverter 420 is formed of MOSFET transistors Q3 to Q6. Of these transistors, transistors Q3 and Q4 are switched in a complementary manner, similar to transistors Q5 and Q6. Since these switching transistors are implemented in the primary circuit domain with respect to the existing 380V on line 690 including fuse f4, during the control input to inverter 420 and its switching components, 1 It is necessary to provide secondary and secondary circuit isolation. In this regard, a pair of transistors Q3 and Q4, Q5 and Q6 are connected between line 690 with fuse f4 and line 692 which is initially grounded. In the drawing, it can be seen that transistors Q3 and Q4 are connected in line 694. The transistor Q3 is configured with resistors R20 and R21 and a capacitor C27. Similarly, complementary transistor Q4 is implemented by resistors R22 and R23 and capacitor C28. Capacitor C29 is connected between lines 690 and 692. The secondary side of pulse transformer T4: B is connected to the gate of transistor Q3. Similarly, the secondary side of T4: C of the same pulse transformer is connected to the gate of transistor Q4. A node is provided between transistors Q3 and Q4 by a line 696 that extends to one end of the primary side of isolation transformer T6. Transistors Q3 through Q6 may be provided as model number IRF460, Repetitive Avalanche and d / v / dt Rated HEXFET® transistors marketed by International Rectifier, Inc. of El Segundo, California. . Transformer T6 is described in block 426 in connection with FIG. 15, and is schematically illustrated using the same numbers in the simplified drawing. The pulse output at line 696 is monitored for control purposes by current transformer T7, thereby providing control output signals CT- (line 698) and CT + (line 700). These signals are used in connection with the phase shift resonance control device that controls the inverter 420 (see FIG. 26).

  The transistor Q5 is configured with resistors R24 and R25 and a capacitor C30. Similarly, the transistor Q6 is configured with a resistor R26, a resistor R27, and a capacitor C31. Transistors Q5 and Q6 are connected in series in line 748, and the node between them is connected to lines 704-706 connected to the other end of the primary side of isolation transformer T6. ing. Complementary transistors Q5 and Q6 are switched by inputs T5: B and T5: C to the secondary side of the transformer, respectively.

  Next, referring to the primary side control inputs of these three-winding transformers, the primary side T4: A of this transformer T4 is connected via a line 708 incorporating a resistor R28. And connected to the output terminal of the drive component 712 via a line 710. The device 712 may be provided with model number MIC4424, for example. This device, which functions in connection with the + 12V input and consists of capacitors C37 to C40 and resistors R29 and R30, connected to the ground line 717, is now connected via the respective lines 714 and 716. Responding to inputs DRV_A and DRV_B arising from the drive circuit board shown connected to 712.

  Corresponding switching to transistors Q5 and Q6 is obtained from the primary side of the 3-winding transformer T5 at T5: A. This primary side is connected via a line 718 including a resistor R37 and a line 720 to the output terminal of the drive component 722, which can also be provided in model number MIC4424 or the like. Device 722 functions in conjunction with + 12V and consists of capacitors C41-C43, resistors R38 and R39, provided on each line 724 and 726 to perform complementary switching of transistors Q5 and Q6. In response to the control inputs DRV_C and DRV_D. Further, these inputs are obtained from a controller for inverter 420 (see FIG. 26).

  Looking briefly at FIG. 23-C, for example, a schematic representation of a square wave generated at the switching node between transistors Q5 and Q6 is shown schematically at 728. FIG. The corresponding square wave generated by switching node intermediate transistors Q3 and Q4 is shown schematically at 730. If these square waves are in phase, there is no voltage difference between them, so there is no voltage applied across the isolation transformer T6. However, the voltage output of the isolation transformer T6 may change the phase between the square wave arrays 728 and 730, for example, to produce a composite square wave as symbolically to the right of the resulting wave 772. Controlled by.

  Referring again to FIG. 23-A, when this inverter switching is performed, the secondary output of transformer T6 is directed to each half of the full-wave bridge rectifier described in block 430 with respect to FIG. It is done. A phenomenon called “resonant transition” couples capacitors C30 and C31, and capacitors C27 and C28 with the leakage inductance of transformer T6, creating a soft-switching resonant transition on the two-switch node. Thus, the transistor pairs Q3 and Q4 and Q5 and Q6 are switched in a very “soft” manner with low stress and high efficiency.

  The second side of the isolation transformer T6 is connected to the line 736 via a line 734 in which the relay contact K6: B is incorporated. Correspondingly, the opposite end of the transformer T6 on the secondary side is connected to a line 776 via a line 738 incorporating a relay contact K7: B. The relay corresponding to block 434 is described in FIG. In this regard, the system may be switched on and off at the step downlink voltage level. Relay contacts K6: B and K7: B are selectively actuated from the relay solenoids described in FIG. 22 as K6: A and K7: A, respectively. It has the aforementioned full wave rectifier implemented in combination with resistors R40 and R41 and capacitors C44 and C45 to produce a DC link voltage across lines 742 and 744. Further, rectified DC link voltage filtering is provided by inductor L5 and capacitor C46. Capacitor C46 holds the DC link voltage monitored on line 746 as a “LINK_V” signal used for high gain controller feedback and other control purposes. Resistor R43 at line 744, R44 at line 748, and R45 at line 750 are used to obtain, among other things, current proportional monitor signals IFB- and IFB + utilized by the inverter 420 controller (see FIG. 26). It is done. The controller applies a peak-to-peak normal disconnect voltage to the precursor electrode and capture component cable to provide a boost level voltage at the beginning of the disconnect operation, particularly to control the link voltage level on lines 746, 748. , And 750, use the signal. Such control is effectively performed by the phase shift control mechanism of the network 420.

  The voltage strength control link voltage across capacitor C46 has been applied to the RF inverter previously described in block 440 with respect to FIG. 15, and is shown in the same general number notation in FIG. 23-B. The RF inverter 440 is configured as a resonant tank circuit including capacitors C47 and C48 and an inductor L6. In this regard, note that capacitors C47 and C48 are located within lines 752 and 754 between lines 742 and 756. At the same time, inductor L6 is connected by lines 758 and 760 between lines 742 and 756. The four MOFSET transistors Q7-Q10 are selectively gated to connect line 756 with DC link voltage line 742 to excite or induce oscillations in this tank circuit. The gate of transistor Q7 is configured with resistors R46 and R47 and one output of the driver or buffer 764, line 764 extending to OUTA. The drive 764 is composed of capacitors C49 and C50, a resistor R48, and + 12V, and responds to the DRV_RF signal at its input line 766 to perform gating. This device 764 may be provided with model number MIC4424 or the like. The second output OUTB of device 764 is connected to the gate of transistor Q8 via line 768. The connection is configured in connection with resistors R49 and R50.

  In a similar manner, the gate of transistor Q9 is comprised of line 770 and resistors R51 and R52. Line 770 extends to the output terminal OUTA of the drive or buffer 772. The driving device 772 includes capacitors C51 to C53, a resistor R53, and + 12V, and receives a control input DRV_RF through its input line 774. Device 772 may also be model number MIC4424 as described above. The second output terminal OUTB of device 772 is connected via line 776 to the gate of transistor Q10 configured in connection with resistors R54 and R55. A SYNC signal is generated from line 756 at line 778 configured in conjunction with resistors R56-R58 and capacitor C54.

  The stable frequency sine wave generated by the RF inverter 420 is applied to the primary side of the step-up transformer T3 previously described in FIG. 15 at block 426 and is represented by the same number notation in the simplified drawing. It is generally regarded as the same thing. A step-up output from transformer T3 is provided on lines 780 and 782. A smoothing of the sinusoidal output is provided by an inductor L7 at the probe electrode line 780. The output on line 780 is directed through relay contacts K2: B and K3: B, and coupling capacitor C55, resulting in a cut output HV_PRECURSOR directed to the precursor electrode. Correspondingly, probe electrode line 784 extending from line 780 carries relay contacts K4: B and K5: B and provides an electrosurgical cutting output HV_CAPTURE supplied to parsing cables 230-234. As such, it is extended in combination with the coupling capacitor C56. Line 784 corresponds to line 454 described in FIG. Relay contacts K2: B through K5: B are controlled from the solenoid components described above in connection with FIG. 22 and function as components of output stage 450 (see FIG. 15). The return line 782 is connected to two corresponding pads or surfaces of the return electrode. In this regard, this line is connected to coupling capacitor C57 and is connected on line 786 to provide the RE2 signal. Line 782 is coupled to line 788 and coupling capacitor C58 to provide a second return for the opposite return electrode pad. Line 788 is connected to line 790 and extends to PCSM circuit 528 as depicted in FIG. The previously described signal identifier RE1 is represented again in the subsequent drawings in relation to the line 464. A current monitoring current transformer T9 is connected to line 782, and high voltage current monitoring signals HV_I− and HV_I + occur on lines 794 and 796, respectively.

  Similarly, a voltage monitoring transformer T10 is connected in line 798 between lines 780 and 782. The secondary side of transistor T10 is configured with diodes D23-D26 defining a rectifier, resistor R59, and capacitor C59, whereby a voltage monitoring signal HV_V is provided on line 800. A specially processed version of that signal provides an outer loop slow or low gain program input to the control of the link inverter 420.

  24-A and 24-B need to be interpreted together as labeled on it. These drawings relate, among other things, to the components mounted on the drive board that have control functions and monitoring for the PFC boost converter 392 described in connection with FIGS. 15 and 17-B. is there.

  Looking briefly at FIG. 24-A, the provisional voltage level present at capacitors C12 and C13, with a 380 volt provisional voltage level, provides a monitoring input + 380V, indicated by line 830. This provisional voltage level is divided by resistor groups R60 through R62, filtered by capacitor C60, and conveyed to one input of comparator 804 via line 802. The reference input of comparator 834 is derived from + 5REF on line 806, which incorporates level adjustment resistors R63 and R64, and provides a reference input on line 808. If the 380V input on line 626 is of the proper amplitude, it is generally represented at 812 and is presented to resistor R65, which is submitted to an RC timing network having resistor R66 and capacitor C61 in line 842. The output at the incorporated line 810 is provided by the comparator 804. The time constant selected for the network 812 provides for adaptation to any unpredictable variations and the like. Thus, a slightly delayed signal is introduced to one input of buffer 818 via line 816 and the other input to buffer 818 is provided from line 820. The output on line 822 of buffer 818 extends to line 824 connected to the + 12V primary power input at line 826 via resistor R67. Line 824 is connected to the gate of transistor Q11 via line 828. Transistor Q11 is connected in line 830, which incorporates resistor R68, between line 826 carrying + 12_PRI and the primary ground at line 827. Transistor Q11 is powered off in response to a logic TRUE low on line 828 and then voltageed from the + 12V primary supply (+ 12V_PRI) to the diode of opto-isolator 834 via lines 836 and 838. Is added. The resulting output from opto-isolator 834 provides a low logic true high voltage OK signal HVOK so that it can be used by the control circuitry on the low voltage secondary side. See FIG. 26 for the latter.

  The 380V DC output itself is not enabled until it is guaranteed that the AC input described in line 380 with respect to FIG. 15 is at the proper level. The detection of this value is provided from line 602 as described in connection with FIG. 17-A. As can be seen, AC_SENSE is monitored on line 602, which incorporates resistors R69 and R70, and capacitor C62, connected to line 827 and tapped on line 844. Resistor R71 is incorporated between lines 844 and 827.

  Referring to FIG. 24-B, it can be seen that line 844 extends to one input of comparator 846. The other input to comparator 846 is + 5REF, which leaves line 848 adjacent to resistor R72 and diode D30. The reference (+ REF) at line 848 is tapped at line 852, which incorporates resistor R73, and is connected to line 827 via a filtering capacitor C63. A line 844 carrying the adjusted AC_SENSE signal extends to the other input of the comparator 846 and provides an output on line 854 by the comparator 846 in the presence of an appropriate voltage level. Line 854 incorporates resistor R74 and extends to line 856, where the output is subject to the time constant established by resistor R75 and capacitor C64. The output from that RC network is then directed to one input of comparator-buffer 862 via line 860. The other input to buffer 862 extends from line 864, which extends from line 864 and incorporates resistors R76 and R77. The filtering capacitors are shown as C65 and C66, and it can be seen that the low logic TRUE output of comparator 862 on line 866 is directed to the gate of transistor Q12. Transistor Q12 is typically held from line 868 in which resistor R78 is incorporated. The source of transistor Q12 is connected to line 827, and its drain is connected to line 870, in which resistor R79 is incorporated. Line 870 is connected to line 872 which is filtered by capacitor C67 and extends to the VREF terminal of controller 874 for PFC boost converter 392. Further note that line 870 is connected to the available input terminal EN / SYNC of device 874 via line 876. Accordingly, transistor Q12 is turned off to enable controller 874 by applying the voltage from line 898, resistor R79, and line 876 in the presence of the AC_SENSE signal of the proper amplitude. Device 874 may be provided with a power factor controller, model number LT1248, commercially available from Linear Technology Corp. of Milpitas, California.

  In addition, line 866 is connected to line 826 via line 878 and resistor R80, and it can be seen that line 862 extends to the source of transistor Q13. The gate of transistor Q13 is connected to line 878 by line 880. Thus, a low TRUE signal on line 866 further functions to turn on transistor Q13, thereby providing a voltage application of the TRUE signal on line 569 by a solenoid. In this regard, the signal on line 596 provides a RELAY_IL signal that serves to apply a voltage to the relay solenoid K1: A described in connection with FIG. This relay closes the contact K1: B and short-circuits the thermistor 592 (see FIG. 17A) activated to prevent inrush current.

  Controller 874 functions to generate a control input DRV_PFC that is applied to line 614 of driver 608 described in connection with FIG. 17-B. Line 614 is protected by diode D61. Device 874 detects 380V level output: detection of AC current AC_I described in connection with line 622 in FIG. 17-B; and ACAC voltage AC_V described in connection with line 624 in FIG. 17-B. It works in conjunction with detection. 380V monitoring is shown on line 882, which incorporates resistors R81-R83, and capacitor C68. The adjusted voltage signal level is then introduced to the voltage detection terminal (VSENSE) of controller 874 via line 884 that is incorporated into resistor R84. This signal level on line 884 is also extended to the CVP terminal of device 874 via line 886. AC current level signal AC_I is provided from line 888 and exits from line 622 as described in connection with FIG. 17-B. It can be seen on line 888 that this signal extends through line 890 and resistor R85 to the MOUT terminal of controller 874. Further, line 888 incorporates resistor R86 and extends to line 892 which extends to the PKLIM terminal of controller 874. It can be seen that line 872 extends to line 892 including resistor R87. AC voltage signal AC_V is provided from line 894 and exits from line 624 in FIG. 17-B. Line 894 incorporates resistors R88 and R89 and extends to line 896 connected to the IAC terminal of controller 874. The controller 874 functions in conjunction with the primary circuit power supply + 12V_PRI, as shown from the introduction from line 898, where diode D62 is incorporated. Furthermore, this device is configured with capacitors C72 to C81 and resistors R91 to R97.

  As previously mentioned herein, the power factor correction generated in connection with the controller 874 allows the electrosurgical generator to be used universally with a wide variety of global utility line inputs. As well as pre-stabilized provisional voltage output that allows optimization of the link inverter stage, enabling the generation of persistent cuts in the presence of probing electrodes exhibiting dynamic surface area and shape It is also possible to perform constant voltage based control.

  Referring to FIG. 25, a low voltage primary circuit power floating bias supply is shown. The 380 VDC level (see FIG. 17-B) is tapped as shown by line 900 incorporating fuse f5 and is filtered by capacitor C85. Line 900 incorporates diodes D63 and D64 and extends to line 902 which extends to the D (drain) terminal of regulator 934. The regulator 904 may be provided with a model number TOP221P Three-terminal / Off-line PWM Switch marketed by Power Integration, Inc., Sunnyvale, Calif. Component 934 is called a “smart power supply” and combines a power transistor and a PWM control circuit. Its source terminal is connected to ground with respect to line 933. Line 902 is connected across the primary side of step-down transformer T12, where the chop input is asserted under the control of device 904. The secondary side of transformer T12 is connected to line 938 by line 908 and diode D65, which incorporates rectifier diodes D66 and D67 and via resistor R98, C of device 904. Connected to (control) input. This serves as feedback to device 904. In addition, a primary circuit power supply + 12V_PRI is present via resistor R99. Filtering capacitors are provided as indicated at C86-C88.

  In addition, a resonant transition control integrated circuit for generating DRV_A to DRV_D control signals to be provided to the inverter 420 described with reference to FIG. Referring to FIG. 26, the controller extends from the drive signal identified in association with each line 714, 716, 724, and 726 as repeatedly described from FIG. 23-A. Is shown at 920. The controller 920 may be a BiCMOS Advanced Phase Shift PWM Controller, model number UCC3895, marketed by Unidode Corp., Merrimack, New Hampshire. The value of the link voltage LINK_V is submitted to the EAN terminal and EAOUT terminal of the device 920 from each line 922 and 924 configured with resistors R100 to R102 and capacitors C91 and C92. Line 922 reappears with respect to the derivation of the LINK_V signal in FIG. 23-A. This link voltage input at resistor R100 shows a very fast or high gain control feedback loop to link voltage controller 920, which is, inter alia, very slow or low gain in nature. It works in conjunction with some external feedback loop program control. The link voltage current related signal IFB− or IFB + is input from the line 926 and 928 in which the resistors R103 and R104 are incorporated, respectively, to the amplifier 930 of the model No. LP1215 configured with the resistors R105 and R106 and the capacitor C93. Applies to These signals are obtained from the lines in FIG. 23-A. The output of amplifier 930 is provided to the CS terminal of device 920 via line 932.

  The current signals CT− and CT + of the inverter 420 are submitted to rectifier diode pairs D70 and D71 and D72 and D73, respectively, which are configured inside the network including the capacitor C94 and the resistor R107. The derivation of these signals is described in connection with FIG. 23-A. From this network, a corresponding signal is submitted to the RAMP terminal of device 920 via line 940 and resistor R108. Similarly, this signal is submitted to the ADS terminal via resistor R109 and to the line 932 and the CS terminal of device 920 via resistor R110. The system selection link voltage for derivation of the constant system output voltage, and the resulting control, is determined by the signal identified as “VPROG” (see FIG. 27-A) submitted to the EAP terminal of device 920 via line 940. It is determined. Line 940 is configured with resistor R111 and capacitor C95 and is connected to 5VREF via pull-up resistor R112. 5VREF is described in connection with FIG. 24-B. As mentioned above, the external feedback control loop that is eventually responsive to the level of the system output voltage is tied to the high gain inner loop. This arrangement allows a constant voltage base so that, in the absence of this arrangement, the unstable vibration tendency caused by the negative dynamic impedance of the required cutting arc, as well as the cable when operating in capture mode Is adapted to the impedance change caused by. Thus, the external feedback loop signal VPROG applied on line 940 selects a very high capacitance value for capacitor C95, such as 4.7 microfarad, etc. from which a time constant of about 35 milliseconds is derived. Device 920 is programmed in a very slow manner. This achieves stable and constant voltage control on the output of the RF inverter 440.

  In addition, device 920 is selectively enabled or disabled in response to three signal inputs. One of these signal inputs is the aforementioned logic low TRUE HVOK signal generated from the provisional voltage-responsive optical isolator 834 described in connection with FIG. It can be seen that this active low signal HVOK is introduced via line 942, which is connected to + 12V via pull-up resistor R113. Line 942 extends through the steering diode D74 and lines 944 and 946 to the gate of MOSFET transistor Q14. Line 946 is connected to ground and transistor Q14 through resistor R114. Transistor Q13 is connected between the ground and lines 948 and 950 to the soft start / disable terminal of device 920. Line 948 extends to ground through capacitor C96. Thus, if the signal on line 942 is a logic high value indicating an inappropriate provisional voltage level, transistor Q14 is turned on to bring line 950 to a logic low state. This disables device 920 until a logic TRUE low condition occurs on line 942, thereby turning off transistor Q14 to remove the low signal on line 950, and the internal circuitry of device 920. Is made possible to act on its availability.

  When the practitioner activates the voltage application / position switch 57 or foot switch 88b on the device 12, a high voltage is invoked to apply a voltage to the precursor electrode. A DC link voltage needs to be generated before that condition occurs. The PLD-based control system therefore provides a logic high DC_LINK_ENABLE input, as shown by line 952, which incorporates resistor R115 and is configured with filter resistor R116 and filter capacitor C97. Line 952 extends to an inverter buffer 954 that has an output at line 956 that extends to line 944 via steering diode D75. That is, lines 956, 944, and 946 are held at a logic high level, thereby turning on transistor Q14 and setting line 952 to a high logic level in response to an enable setting command DC_LINK_ENABLE from PLD-based control. Until the device 920 becomes unavailable. Thus, device 920 does not provide link control if there is no appropriate link availability signal or HVOK signal. Device 954 may be a CMOS Schmitt trigger, model number CD40106B, commercially available from Texas Instrument, Inc. of Dallas, Texas. By using such a component, its filtering hysteresis characteristic is effectively utilized.

  The detected link overvoltage fault condition derives a logic high TRUE “DISABLE” signal (see FIG. 39) present on line 946 via steering diode D76. Therefore, if such a failure occurs due to the presence of the BOOST_MASK signal (see FIG. 27-A), the system is shut down. It is from this position through diode D76 that by turning on transistor Q14, such outage activity occurs. Device 920 can also be seen configured with capacitors C98 through C102 and resistors R117 through R121, and is commercially available by Unidode Corp. of Merrimack, NH. A BiCMOS Advanced Phase Shift PWM Controller with model number UCC3895 may be provided.

  Referring to FIG. 27-A, control system output outer loop monitoring characteristics performed on the drive board are illustrated. This high voltage output monitoring signal, depicted as HV_V on line 800 in FIG. 23-B, is filtered as depicted in connection with FIG. 36 to provide the signal VOUT introduced on line 960. . Line 960 incorporates input register R 125 and extends to one input of error amplifier 962. This reference input to device 962 is shown schematically at 964 and is derived from a potentiometer that incorporates a resistor component R126 and a capacitor C107. Resistor component R126 is connected to a 7.5V reference input.

  Referring briefly to FIG. 27-B, the derivation of this criterion is illustrated. In the drawing, line 966 incorporating resistor R127 and diode D78 is tapped to provide a 7.5REF signal on line 968 that reappears in FIG. 27-A. The extended input of the wiper arm to device 662 is shown on line 970. The output of comparator 962 at line 972 is the output voltage directed to lines 974, 976 and input resistor R128 to the IN1, V-, GND, and IN4 terminals of analog switching device 978. Indicates an error signal. As the switch 978, there is provided a CMOS analog switch of the model number MAX4665 marketed by Maxim Integrated Products of Sunnyvale, California. Line 974 extends from input line 960 to the COM2 terminal of analog switch 978 and incorporates blocking diode D79 and resistor R129. In addition, switch 978 responds to a logic high TRUE “BOOST_MODE” signal generated from control board PLD and shown to be present on line 980. The boost mode provides an increase in output voltage, ie, an increase in the power output of the precursor electrode and the parsing cable for about 3/8 seconds at any start or restart. Line 980 is configured with resistors R130 and R131, and capacitor C108, and extends to the input of buffer inverter 982. The device 982 may be provided with a model number CD40106B Schmitt trigger (supra). Thus, a logic high TRUE signal on line 984 is converted to a logic low on line 984 and directed via lines 986 and 988 to the IN2 and IN3 terminals of device 978 to generate boost mode operation. The

  Since the control system includes a DC link overvoltage fault condition, it is necessary to simultaneously generate a “BOOST_MASK” signal to overcome the faulty fault condition during boost mode. Thus, line 984 incorporates a steering diode D80 that includes a resistor R132 and a capacitor C109 extending between + 12V and the second ground and is located in front of the RC network, generally indicated at 984. I can see that. Network 990 provides a normally high input to comparator 994 to establish a normally logic low on its output line 998. The opposite input of device 994 carries on line 996 the 7.5REF signal depicted in connection with FIG. 27-B. Comparator 994 provides a logic high or active high BOOST_MASK output on line 998 in response to a boost mode indicating a logic low on line 984. This BOOST_MASK active high output on line 998 is present during the BOOST_MODE command. However, as a safety feature, after the BOOST_MODE command signal ends, a logic high BOOST_MAST state is maintained on line 998, generally for the RC network 990 time constant. In this connection, capacitor C109 immediately discharges, assuming an active low state on line 984. At the end of boost mode, diode D80 is back biased and capacitor C109 is gradually charged through resistor R131, eventually boost mask comparator 994 returns its output to a logic low level. A voltage level is established that results in the removal of the BOOST_MASK signal.

  During the boost mode, the analog switch device 978 reacts to the situation on lines 986 and 988 and provides the signal level of the boost voltage value at the output output N02 terminal, line 1000, resistor R133 and line 1002. In boost mode, power is increased by a factor of two. Therefore, since power is proportional to the square of the voltage, the link voltage can be increased by √2 times VPROG. Generally, the boost voltage level is larger in the range of about 1.2 times to about 1.5 times than the normal cutting voltage level. Alternatively, device 978 provides a lower level normal disconnect voltage value signal on line 1002, as established by the resistance value of resistor R134. These resistors R133 and R134 substantially form a voltage divider with the pull-up resistor R112 described in connection with FIG. In addition, device 978 is comprised of a + 12V source and capacitor C110 on line 1004, such as 5-ohm, SPST, CMOS Analog Switch model no. MAX4465 sold by Maxim Integrated Products, Sunnyvale, CA May be provided.

  Referring to FIG. 27-C, the power derivation circuit characteristics of the control system held by the drive board are illustrated. For the purposes of the present invention, it is used as a monitor to determine the presence of any excess power, so that a typical electrosurgical application can be performed using a fixed shape active electrode. The power monitor circuit may be used as described later, and here, the control may be based on basically a constant power output.

  The total power is determined by monitoring the output voltage and output currents that derive the signals VOUT and IOUT respectively indicated by lines 1010 and 1012 extending to the solid state multiplier 1014. For example, the device 1014 may be an Analog Multiplier, model number AB633JN, available from Analog Devices, Norwood, Mass. The multiplier 1014 includes a power supply input of + 12V and −10V, and capacitors C111 and C112. Since it forms a component of the power derivation network, the product output of multiplier 1014 on line 1016 is sent to an integrating resistor R136. Further, line 1016 extends to lines 1018 and 1020, the latter line incorporating an integrating capacitor C113. Further, line 1018 extends to diode D81 and extends to amplifier 1022. Using the arrangement shown, in the end, power is a conventional formula:

Calculated according to

  That is, the capacitor C113 has a monitored power signal that is proportional to the output power. This signal is passed to an amplification stage 1022 composed of resistors R137 and R138 so that the amplitude of the signal is doubled. This provides on line 1024 a power value signal utilized by the system, identified as “PWR_OUT” (see FIG. 37) for monitoring the excess output power situation.

In an embodiment associated with a fixed shape active electrode and that can be used under fixed output power control point conditions, it is preferred for the user to use an electrosurgical generator, but only minor changes are required. The major changes are shown in FIG. 27D, which corresponds to the previously described content in the network combination of FIGS. 27A and 27C. Thus, the components, signals, and network conductors are the same and are designated by the same name and reference in FIG. 27D, but are labeled with a “′”. Referring to FIG. 27D, the output voltage signal VOUT It can be seen that the output current signal IOUT is again separated into the multiplier 1014 ′, where the multiplication is performed and transmitted to the amplifier stage 1022 ′. The power voltage signal is used as a control point and is represented by line 1024 ', which now incorporates resistor R125' and is introduced into amplifier 962 '. The amplified power value on line 1024 'is tapped on line 1025 to provide a PWR_OUT' signal that is used for the purpose of monitoring excess power output. The power reference input to amplifier 962 'is typically derived from a voltage divider, indicated generally at 964', which incorporates resistor R126 'and capacitor C107'. Resistor R126 'is coupled to a 7.5 volt reference input. Thus, line 970 is tapped at line 971, providing a PWR_REF ′ reference. The output of device 962 'on line 972' represents the output signal and is directed to lines 974 'and 976', connecting resistor R128 'to the IN1, V-, GND, and IN4 terminals on analog switch device 978' To do. The analog switch 978 ′ is a MAX4665 CMOS analog switch as described above. Line 974 'extends from line 1024' to the COM2 terminal of switch 978 'and conducts resistor R129' and block diode D79 '. Such a configuration ensures unidirectional input to the device 978 '. As previously mentioned, switch 978 ′ is responsive to a logic high active “BOOST_MODE” signal generated by a control PLD as shown in line 980 ′. Line 980 'is configured in conjunction with resistors R130' and R131 'and capacitor C108' and extends to the input of buffer inverter 982 '. Thus, a logic high true signal on line 980 'is reversed to a logic low signal on line 984' and sent via lines 986 'and 988' to the IN2 and IN3 terminals of analog switch 978 'to perform boost mode. Affects the switching associated with

  At the same time that operation in boost mode is provided, an active or logic high signal BOOST_MASK is emitted on line 998 '. This output comes from the logic low true on line 984 'and continues to exist beyond the resulting boost time of the RC network 990' as described above.

  During boost mode, the analog switching device 978 'reacts to the situation on lines 986' and 988 'and provides the signal level of the boost voltage value at its output NO2 terminal, line 1000', capacitor 133 ', and line 1002'. . In this boost mode, the power increases twice. Therefore, the link voltage increases with the square of VPROG, and the power is proportional to the square of the voltage. Alternatively, device 978 'provides a lower level normal disconnect voltage value signal on line 1002' from terminal NO3, which is probable by the value of resistor R134 '. Indeed, resistors R133 'and 134' form a voltage divider with a full up resistor R112 'as shown in FIG. Device 978 'further comprises a +12 volt power supply and is configured with a capacitor C110' on line 1004 'and may be provided as a MAX4465 type analog switch (existing).

  Referring to FIG. 28, a network for providing a control input DRV_RF applied to devices 764 and 772 in FIG. 43-B for the RF resonant inverter 440 is illustrated. In this drawing, the fundamental frequency is derived by the oscillator integrated circuit 1030. This oscillator integrated circuit 1040 configured with capacitors C114 through C116 and resistors R139 and R140 is provided by Model No. LMC555 Timer, etc., marketed by National Semiconductor Corp. of Santa Clara, California. Good. Frequency adjustment may be provided by the manufacturer in connection with a potentiometer, indicated at 1032. The frequency output of device 1030 is shown along line 1034 to the trigger input of another model number LMC555 device 1036 that establishes the pulse width. Device 1036 is configured with capacitors C117 and C119, and resistor R142. The pulse width is adjusted by the manufacturer with a potentiometer shown at 1038 including a resistor R143. Both devices 1030 and 1036 are enabled simultaneously by the PLD and start reset derived ENABLE inputs provided on lines 1040 and 1042, respectively. In this context, an available setting is provided if a series of signals from the PLD is provided as it goes through, but the RF inverter is not allowed to be enabled during initial system startup. Has been. Thus, for safety characteristics, a logic high or active high ENABLE signal is not provided until after the power-on reset (PWR_ON_RST, see FIG. 34) time interval. The final control signal DRV_RF is provided from device 1036 on line 1044 that incorporates resistor R144. Note the simplicity of the control input for the RF resonant inverter, which is consequently derived by using the DC link voltage as the highest and lowest voltage that controls the input to the standard inverter activation circuit.

  FIGS. 29-33 illustrate circuitry associated with the logic used for applying voltage to the motor 160a of the motor assembly 160. FIG. In this context, the motor current identified as “MOTOR_I” is monitored for the execution of this logic. This monitored current is generally too low to be useful and its derivation is described in connection with FIG. Thus, it is first amplified to produce an augmented signal identified as “MOTOR_CURR”. FIG. 29 shows the amplification of these signals. In this regard, an initial current signal is introduced into amplifier 1052 via resistor R146 and line 1050. Amplifier 1052 is configured with resistors R147 through R149, and C121 and C122, and provides an increased MOTOR_CURR signal on output line 1054.

  FIGS. 29-33 provide an analysis of the changing threshold of the motor current utilized by the PLD logic device of the system. FIG. 30 illustrates an initial threshold test for first determining the voltage applied to the motor to determine if it is really operating. For this purpose, a small amount of free movement of the yoke 180 is allowed prior to contact with the ears, as shown at 138 and 140 (see FIG. 2) of the drive component 276. Yes. During this very short test interval (about 0.5 seconds), the motor current is very low but recognizable, for example, having a threshold value of at least 10 milliamps. If the motor 160a is not turned on at the time it should be turned on, it is immediately put into a system failure with proper shutdown and visual cueing. FIG. 30 shows that the MOTOR_CURR signal is introduced on line 1056 to one input of the comparator 1058. The reference input to the comparator 1058 is the aforementioned 7.5REF disclosed in connection with FIG. 27-B. This reference voltage is adjusted by resistors R151 through R153 and introduced into device 1058 via line 1060. The output of device 1058 is provided on line 1062 which is connected to a + 12V source through pull-up resistor R154. If a properly executed motor current level is detected, the “MOTOR_ON” signal is generated by turning off transistor Q16.

  Referring to FIG. 31, the MOTOR_CURR signal is introduced into the comparator 1068 from along the line 1070. Comparator 1068 is configured with a 7.5 REF reference signal and resistors R156-R158 to react to the threshold provided on line 1072, which indicates, for example, a motor current draw of approximately 23 milliamps. Because the yoke 180 is coupled with the ears 138 and 140 (see FIG. 3), the motor begins a more complex operation and typically exhibits a current draw of about 45 milliamps. If this condition is evidenced by comparator 1068 and the threshold provided for this motor is exceeded, comparator 1068 acts to turn off transistor Q17 on its output line 1074. That is, a “MOTOR_ENGAGED” signal is generated for the logic of the control system. As before, line 1074 is connected to + 12V via pull-up resistor R159.

  The network of FIG.30 and FIG.31 is performed simultaneously. During the initial 0.5 second test time interval, if the motor current is above a low threshold of, for example, 5 milliamps, generation of the MOTOR_ON signal occurs as determined by the network of FIG. However, if the motor current exceeds the network threshold of FIG. 31 during this same test time interval, it will fail in the initial test and a fault condition will occur.

  After passing the initial 1/2 second test, the network of FIG. 31 detects whether it meets its threshold of, for example, 23 milliamps. Thereby, the proper relationship between the yoke 180 and the ear portions 138 and 140 is shown. If the threshold of the network 31 is not maintained during the forward movement of the drive component 276, a fault condition with system shutdown and visual cueing occurs.

  Referring to FIG. 32, for example, as illustrated in connection with FIG. 29, when tissue capture is complete, the motor enters a forward stall condition and the current is rapidly spiked to approximately 130 milliamps. The Referring to FIG. 32, the MOTOR_CURR signal is again introduced to the comparator 1080 via line 1082. Comparator 1080 reacts to the threshold on line 1084 and then provides a logic low true output on output line 1086 if a forward stall current level is indicated, 7.5REF and resistor R161. To R163. As described above, the line 1084 is connected to the + 12V source via the pull-up resistor R164, and is connected to the gate of the transistor Q18. Therefore, the “MOTOR_STALL” signal is generated by turning off transistor Q18.

  In response to detecting a forward motor stall condition, the system logic reverses the polarity of the drive to the motor 168 and the moving assembly 146 butt-couples with its drive components 276, ears 138 and 140. (Abutting engagement), thereby driving it back to the “home” position described in connection with FIG. The resulting reverse stall current is detected with a lower amplitude than the forward stall current.

  Referring to FIG. 33, the MOTOR_CURR signal is introduced into the comparator 1094 on line 1092. The reference level for comparator 1094 is set for detection of reverse stall current level and is provided in connection with resistors R166-R168 at line 1096 from 7.5REF. The output of comparator 1094 on line 1098 is connected to the gate of transistor Q19 as well as pull-up resistor R169 up to 12 volts. Thus, when a reverse stall condition is detected, a low true condition occurs at line 1098, turning off transistor Q19 and providing a “MOTOR_REV_STALL” signal or condition at line 1100. As the comparators 1058, 1068, 1080, and 1094, for example, Low Power, Low Offset Voltage Comparators of model number LM339 marketed by National Semiconductor Corp. (supra) may be provided.

  Referring to FIG. 34, there are shown circuits that provide an “ENABLE” signal and a “RESET” signal in response to the generation of the RF_INV_ENABLE signal and the PWR_ON_RST signal, respectively. The latter reset signal is generated from the control system PLD. In the drawing, the former logic high TRUE input signal is introduced via R171 to the input of buffer 1094, which incorporates a Schmitt trigger on line 1104, and its logic low inverted output is connected to an ORing diode on line 1108. Through D82, it extends to the input of the second inverter 1110, which provides an active or logic high "ENABLE" signal on the output line 1112. Filtering resistor R172 and filtering capacitor C124 are connected to line 1108. A power on reset (PWR_ON_RST) signal is introduced through resistor R173 and line 1114 to the input of the buffer 1106 where the Schmitt trigger is incorporated, and its logic low output is directed to the input of the inverter 1120. Provided on line 118. The logic high output of inverter buffer 1120 on line 1122 carries the RESET signal and overrides the ENABLE signal previously described by a wire O-ring configuration including line 1124, diode D83, line 1126, and resistor R174. In this regard, line 1126 provides a signal to the input of inverter buffer 1110. Filtering resistor R175 and filtering capacitor C125 are connected between line 1114 and ground. As described above, as a safety characteristic, the RF inverter operation at the start of the system that occurs during the power on reset time interval is blocked. This is achieved, among other things, by the aforementioned ORing arrangement derived from diodes D82 and D83, which functions to remove the ENABLE signal during the initial time interval.

  Referring to FIG. 35, monitoring of the comparator circuit for high voltage overcurrent conditions is shown. In the drawing, the current signals HV_I + and HV_I− as generated in the high voltage output stage 450 are rectified as described with respect to FIG. 23-B. In this regard, a positive current is introduced from line 796 to intermediate diode pair D84 and D85, and a negative current signal is introduced from line 820 to diode pair D86 and D87. These rectifier diode pairs are located between lines 1130 and 1132. The signal “IOUT” is generated from line 1114 and is indicated by line 134 (see FIGS. 27-C, D). Capacitor C127 and resistor R162 provide a filtering function, while diode D89 functions as a clamp. Line 1130 extends to one input of a comparator 1136 with an output connected to line +138 to a + 12V source through pull-up resistor R178. Comparator 1136 is configured to establish a high voltage overcurrent threshold reference at line 1142 with a + 12V source and resistors R179-R181. Thus, a low TRUE output at comparator 1136 generates a corresponding overcurrent state “HV_OC” on line 1144 by turning off transistor Q20 (see FIG. 61-A, where this line is again shown). ).

  Referring to FIG. 36, a comparator circuit is shown that determines the presence of an overvoltage condition at the generator output. The HV_V signal is introduced as described by line 800 in FIG. Line 800 reappears in this figure and is provided to apply a high voltage to one input of comparator 1148 via input resistor R183. It can be seen that line 800 is coupled to filter capacitor 129 and clamping diode D90. This filtering configuration is provided for signal VOUT, where VOUT is shown on line 1150 and is shown in conjunction with FIGS. 27A, C, and D. The overvoltage reference input to device 1148 is provided on line 1152, provided on resistors R184 through R186 and connected to a + 12V source. The output of comparator 1148 on lines 1154 and 1156 is connected to the + 12V source via pull-up resistor R187, and then to the gate of transistor Q21 via line 1158. Accordingly, transistor Q21 is turned off such that the low TRUE output of comparator 1148 produces an overvoltage state “HV_OV” that reappears in FIG.

  Referring to FIG. 37A, a comparator circuit is illustrated that determines the presence of an overpower condition at the generator output for embodiments of the present invention associated with various surface area or shape activated electrodes. To do. Therefore, this monitoring is performed on the PWR_OUT signal and its derivation is described in connection with FIG. 27-C. This signal is incorporated into resistor R189 and introduced at one input of comparator 1164. The reference input to the comparator 1164 is derived in connection with a potentiometer network incorporating a potentiometer resistor R191 associated with resistor R172 and a reference 7.5REF directed to capacitor C130. The output on line 1168 of device 1164 is connected to the + 12V source and the gate of transistor Q22 via pull-up resistor R193. Therefore, transistor Q22 is turned off so that the low TRUE output of device 1164 derives the “OVER_POWER” state on line 1170 (which reappears in FIG. 41-A). Filtering resistor R190 is connected between line 1024 and ground.

  Referring to FIG. 37B, a comparator circuit is shown, which keeps the activation electrode constant in the surface area and the output control is asserted at a constant power reference as described in FIG. 27D. Under the conditions of such an embodiment, the presence of an overpower condition at the generator output is determined. In the latter figure, the PWR_OUT signal is introduced on line 1025. This signal is incorporated into resistor R195 and introduced at one input of comparator 1164. Line 1025 is filtered by resistor R195. A reference input to the comparator 1172 has been introduced as described by line 971 in FIG. 27 and is identified as “WPR_REF ′”. Line 971 reappears in the drawing and provides a reference input for the opposite input of comparator 1172 associated with resistors R197 and 198. In general, the reference level of this embodiment is closer to the value for a constant generator output power control point than the embodiment of FIG. 37A. The output of the comparator 1172 on this line 1174 is coupled via a pull-up resistor R199 up to + 12V as well as the gate of transistor Q22 '. Thus, a low true output at comparator 1172 turns off transistor Q22 ', introducing "OVER_POWER'" that reappears in FIG. 41A, similar to the corresponding situation in line 1170 described in FIG. 37A.

  With reference to FIG. 38, an overtemperature circuit is depicted. A temperature signal TEMP having a low TRUE state when the monitored temperature is exceeded is described in connection with FIG. Line 630 from the temperature sensitive device described in that drawing, incorporating resistor R201, is connected to the + 12V source through pull-up resistor R202 and extends to the gate of transistor Q23. ing. The filter capacitor C132 is then grounded. With the illustrated arrangement, a low TRUE “OVER_TEMP” signal is derived on line 1176 and reappears in FIG. 41-A in the presence of excessive hardware temperature.

  The DC link voltage is described in connection with FIG. 23-A as being monitored on line 746. This monitoring signal is specified as “LINK_V”. The control system determines whether this voltage is either above or below the window of acceptable operation. Of course, such a window can be reduced to a decimal value.

  Referring to FIG. 39, line 746 is again shown, introducing the LINK_V input to the positive input terminal of link overvoltage comparator 1180. Line 746 is configured with resistors R204 and R205. Additional connected to line 746 is a line 1182 that extends to the input of the link undervoltage comparator 1184. The reference input for both comparators 1180 and 1184 is derived from a + 12V source on line 1186. In this regard, a + 12V source is introduced into line 1186 via resistor R206, and its reference value is then directed to device 1180 via line 1188. Line 1186 further incorporates resistors R207 and R208 so that the link undervoltage threshold reference input to comparator 1184 is established on line 1190. Line 1186 is filtered by capacitor C134.

  The output on line 1192 of comparator 1180 is connected to the + 12V source through pull-up resistor R209; connected to filtering capacitor C135; ) It extends to the terminal. Device 1194 may be a CMOS Dual “D” type Flip-Flop, model number 4013B, marketed by Texas Instruments, Inc. of Dallas, Texas. If the monitored link voltage level on line 746 exceeds the threshold set on line 1188, output line 1192 goes to a logic high state so that latch 1194 sets the state. As a result, the DISABLE signal turns on MOSFET transistor Q14 (see FIG. 26) so that its Q output on line 946 changes to a logic high level and disables link voltage controller 920. Generated. Complementary low TRUE output occurs at the Q terminal on line 1196. Line 1196 is connected to the gate of MOSFET transistor Q24, and its drain and source terminals are connected to line 1198 and ground, respectively. This turns off transistor Q24 to derive a link overvoltage signal “DC_LINK_OV” that is sent to the control PLD and further generated there.

  A logic high TRUE BOOST_MASK signal is generated on line 998 during the increased link voltage based boost mode as described in connection with FIG. 27-A. Line 998 reappears in the concise drawing and extends to line 1200 which incorporates resistor R210 through ORing diode D32 and extends to the reset (R) terminal of latch 1194. ing. Thus, during boost mode, latch 1194 is held in a reset state, at which time its Q terminal is held to block any DISABLE signal on line 946 and on that line 1196. The Q terminal is held at a logic high level that turns on transistor Q24. Thus, the DC_LINK_OV signal is blocked for the duration of boost mode.

  Another characteristic is that the system holds the latch 1194 in the reset state during the power up reset time interval to ensure that the overvoltage base signal as described above does not appear on lines 946 and 1196. Accordingly, the active high level RESET signal generated on line 1122 as described in connection with FIG. 34 is transmitted to the line 1200 and the R of the reset terminal latch 1194 via the ORing diode D93. Recall from FIG. 34 that the presence of the RESET signal negates the ENABLE signal and disables the function of the RF inverter 420.

  Looking at the DC link undervoltage comparator 1184, the output of this device is provided on line 1202. Line 1202 is connected to the + 12V source by pull-up resistor R212, and is connected to input line 1182 via resistor R211. The output line 1202 extends to the gate of the MOSFET transistor Q25. The drain of the transistor Q25 is connected to the line 1204, transmits the DC_LINK_UV signal, and its source is grounded. Thus, in the presence of undervoltage on the DC link, the output of the comparator 1184 exhibits a low logic TRUE condition on line 1202 so that transistor Q25 is turned off and carried to the PLD on the control board. An undervoltage signal “DC_LINK_UV” is generated. As such, the link voltage level is monitored as a function of overvoltage and undervoltage conditions.

  Referring to FIG. 40, a power converter and isolation circuit is depicted that uses a network to respond to the operation of foot switch 88 and vacuum switch 51 (see FIG. 1). This circuit is designed to be compatible with foot and vacuum switch devices that do not have built-in electrical shielding properties. That is, optical isolator characteristics are provided. In the figure, a + 12V source is applied to T13: A of the primary side isolation transformer T13 via resistor R214 and line 1210. Line 1210 is filtered with capacitors C137 and C138. The opposite side of the primary side of the transformer is connected by line 1212 to the source terminal of transistor Q26. Blocking diode D93 extends across the drain and source terminals of the transistor. The source of transistor Q26 is grounded through line 1214. The gate of transistor Q26 is connected to the OUT terminal of power converter 1218 by line 1216. Line 1216 is connected to filter resistor R215 and clamping diode D98. For example, a device of model number UC3845 sold by Unnitrode Corp. of Merrimack, New Hampshire is provided, and converter 1218 is configured with resistor R216 and capacitors C139 and C140 to selectively turn on transistor Q26. And by turning it off, it functions to chop the input to the primary side T13: A of the transformer. One of the secondary sides of transformer T13, indicated by T13: B, derives a -10V output and is shown to function with rectifier diode D95, resistor 217, and filter capacitor C121. This -10V source is used in line 1017 by multiplier 1014 (see FIG. 27-C). The next secondary side of the transformer T13 is shown at T13: C and provides electrical insulation of the foot switch 88 and the vacuum switch 51. A pair of input leads from each of the vacuum switch 51 and the foot switches 88a to 88c is shielded from light and connected to the secondary side T13: C. One side of the secondary side T13: C is connected to a line 1220 in which a rectifier diode D96 and a resistor R218 are incorporated. The opposite side of the secondary side T13: C is connected to the line 1224. Capacitor C142 and resistor R220 extend between lines 1220 and 1224, essentially creating a node across resistor R220 that is utilized by four identical isolation networks. For example, the first of these networks associated with foot switch 88a incorporates line 1220 and resistor R219 extending to the anode input of opto-isolator 1222. The cathode input of the optical isolator 1222 extends to one side of the foot switch 88a and is connected to a line 1226 labeled "FOOTSWITCH_1A". Line 1224 extends to the opposite side of switch 86a and is labeled "FOOTSWITCH_1B". The low voltage output side of the optical isolator 1222 is connected to the gate of the transistor Q27 by a line 1228, and the output on the opposite side is connected to the source terminal and the ground of the secondary circuit via the line 1230. Line 1228 is connected to the + 12V source through pull-up resistor R222, and therefore, activation of foot switch 88a generates signal “FOOTSWITCH_1” on line 1232 in a low logic TRUE fashion. This network, incorporating resistors R219 and R221, opto-isolator 1222, and transistor Q27, is repeated and connected across resistor R220 to the remaining foot switches 88b and 88c and vacuum switch 51. Accordingly, the same network identification number, but with a prime symbol, is used to describe these networks. In this regard, the footswitch 88b network is identified with a single prime symbol in combination with the switch labels “FOOTSWITCH_2A” and “FOOTSWITCH_2B” and provides a low logic TRUE output signal “FOOTSWITCH_2”. Yes. The foot switch 88c is identified with a double prime symbol in combination with the switch label "FOOTSWITCH_3". Similarly, the vacuum switch 51 network is identified with a triple prime symbol in combination with the switch labels “VACSWITCH_A” and “VACSWTICH_B” and provides a low logic TRUE output signal “VACSWITCH”.

  As explained earlier, the control board components of the control unit are mainly characterized by the implementation of a programmable logic gate (PLD), which is traditionally a programmable hardware for logic gate compilation. It is done. This gate compilation, in response to continuous logic, immediately produces a series of states that affect the control of the system. This device may be, for example, a Programmable Logic Device Model No. EPM192SQC160-15 marketed by Altera, Inc. of San Jose, California. This device is shown at 1240 in FIG. 41-A. In addition, the substrate incorporates filtering and logic supporting pull-up functions. In general, when a transistor is depicted as being turned off, the associated line is generally drawn to a logic high on the control board. FIG. 41-A uses a method labeled thereon to reveal the connection of those characteristics that have been drawn to connect seamlessly with this logic center. B to 41-E need to be considered in connection with FIG. In FIG. 41-A, a stabilized + 5V and associated ground is shown being introduced from each line array 1242 and 1244 to the VCC and GND display terminals at the corresponding terminals of device 1240. . The + 5V input is shown being filtered by a six capacitor array 1246.

  Further, referring to FIG. 41-B, a clock network is indicated generally at 1248. Network 1248 includes a crystal oscillator 1250 responsive to a _RESET input applied to line 1252 (eg, model number 74302 marketed by M-Tron Industries of Yankton, South Dakota may be provided). It is out. Network 1248 is configured with inductor L10 and capacitors C130-C132 to provide a 1 KHz input on line 1254 to PLD 1240.

  Referring to FIG. 41-C, a reset network is shown generally at 1258 that functions to hold the system low for a specific amount of time to ensure power supply stabilization. Recall that during a reset time interval, the function of the RF inverter 420 is disabled due to a safety failure (see FIG. 34). Network 1258 functions when the system is powered on or when the regulated + 5V power supply to the instantaneous circuit is reduced to a predetermined range. This network is concentrated around a reset device 1260 (which may be, for example, model number DS1233DZ-5 marketed by Dallas Semiconductor, Inc. of Dallas, Tex.), Capacitor C153 and C154 and resistor R223 are included. The A_RESET output is provided on line 1262 which is depicted as being introduced into oscillator device 1250 via resistor R213 and line 1252 in connection with FIG. 41-B. The same signal is directed to the _RESET terminal of PLD 1240 via line 1264. In addition, PLD 1240 provides a logic high TRUE PWR_ON_RST signal on line 1266, as depicted in connection with FIG.

  Referring to FIG. 41-A, shown generally at 1268 is an externally accessible jumper or connector that provides a four line input to the I / O port of the PLD 1240, as generally indicated at 1270. Three of these four line arrays 1270 are pulled to + 5V through a pull-up resistor array generally indicated at 1270.

  Extending from PLD 1240 is a four line array, generally designated 1274, that provides an output for controlling the four relays of PCSM circuit 462 (see FIG. 15). These arrays correspond to lines 468 in the figure. Below the array 1274 is a line 1276 that provides a PCSM circuit enable signal PCSM_ENBL. Below line 1276 is an input line 1278 that carries the PCSM circuit valid input signal PCSM_VALID, which indicates to PLD the appropriate path for the previously described PCSM test.

  The d.c. link monitoring feature shown in FIG. 34 as being input to PLD 1240 is illustrated in input lines 1204 and 1198 described above. The link relay 434 control RELAY_LINK is provided on line 1280, and on line 952, the DC_LINK_EN d.c. link enable signal as previously described reappears in the drawing. Representing in the figure is an array of inputs and outputs 1282 for the PLD 1240 associated with the high voltage function that includes the boost mode signal BOOST_MODE as previously described in line 980 in FIG. The RELAY_CUT or high voltage precursor electrode disconnect signal previously described in FIG. 22 is shown on line 1284, and the corresponding continuously activated RELAY_CAPTURE signal previously described in FIG. 22 is transmitted on line 1286. Has been. In line 140, the R inverter enablement signal RE_IND_EN described above in connection with FIG. 34 reappears with the same reference. The high voltage overvoltage signal HVOV input previously described in line 1160 in FIG. 36 reappears in line 1160 and is the corresponding high voltage overcurrent signal HV_OC, as previously described in FIG. 35 on line 1144. Appear again with the same number.

  Below line array 1282 is another array 1288 of inputs and outputs to PLD 1240. In this array 1288, the front stall signal (MOTOR_STALL) signal and the reverse motor stall signal (MOTOR_REV_STALL) of the motor 160a reappear with the same numbers on lines 1088 and 1100, respectively, in conjunction with FIGS. . The forward drive signal (RELAY_FWD) of the motor described in connection with FIG. 22 is shown on line 1290, while the forward drive signal (RELAY_REV) of the motor is shown on line 1292. The inputs on lines 1064 and 1276 are respectively signals MOTOR_ON (monitoring initial motor voltage application for rotation and monitoring conditions), MTR_ENGAGED (actuated when yoke 180 is coupled to drive component 276). Carry. The functions of these motors are described with respect to FIGS. 30-33, as specified with respect to lines 1088, 1100, 1064, and 1076, while lines 1290 and 1292 reappear in FIG.

  The input of the overtemperature OVER_TEMP (see FIG. 38) to the PLD 1240 is shown on line 1254, whereas the low voltage power supply undervoltage condition signal, LVRS_UN (see FIG. 43) is inputted on line 1294. Thus, as described with respect to FIG. 37A or B, an overpower status signal, OVER_POWER, is input to the PLD 1240 as indicated by line 1170 or 1170 ′.

  Referring to the opposite side of PLD 1240, a 13 line array is shown generally at 1296. Certain of the lines in array 1296 carry signals in response to external switching and interlock tests, and optionally light up light emitting diodes (LEDs) on both the front panel of console 64 and instrument 12. To provide output. Above line array 1296 is shown a line array 1302 labeled corresponding to the light-blocked input signals from foot switch 88 and vacuum switch 51. These input signals are described above in connection with FIG.

  Further, referring to FIG. 41-D, the line array 1296 reappears, and the inputs and outputs shown thereby extend to the three connectors 1298-1300 via an intermediate signal processing mechanism. I can see that. Connector 1298 is connected to a printed circuit board located on the upper part of the front panel of console 64; lower panel assembly in which connector 1299 provides functionality to the lower part of the front panel of console 64 And the connector 1300 is operatively associated with a connector that functions with the device 12. A line 1303 carrying a start switch signal identified as “START_SW” is at the top of the array 1296, as initially derived by actuation of switch 92 on console 64 (see FIG. 1). This is the only switch attached to the console that has an input to the PLD 1240. This switch needs to be activated to initiate any procedure, and the signal of this switch is used for the initial setup of the device's motor drive components to initiate the PCSM return electrode test. The start / reset signal provided by this switch is derived with respect to the regulated + 5V voltage associated with PLD 1240, as shown by line 1304 and filter capacitor C156. In addition, line 1302 is incorporated with a protection network generally indicated at 1306 comprised of clamping diodes D98 and D99, resistors R224 and R225, and capacitor C157. In such a configuration, the diode of network 1306 provides a clamp that limits the signal on line 1302 from fluctuating between + 5V and the RC filter incorporated with ground. This protective arrangement ensures proper signal transmission without interference.

  Output lines 1308-1312 provide outputs that affect the voltage application of the four LED illuminators at the top of the console 64 front panel. Still referring to FIG. 1, the READY_LED signal on line 1303 affects the lighting of the LED lighting device 94, and the CAPTURE_LED signal on line 1309 affects the lighting of the lighting device LED 100. The ENGZ / POS LED signal affects the lighting of the lighting device LED 96, the ARM_LED signal on the line 1310 affects the lighting of the lighting device LED 98, and the line 1311 carrying the COMPLETE_LED signal is connected to the lighting device LED 102. The lighting is affected, and the PAUSE_LED signal on the line 1314 affects the lighting of the lighting device LED104. These signals are buffered in buffer 1320 and filtered by connections to six resistors within resistor array 1322 that work with the filter-related six capacitors of capacitor array 1324.

  The PAUSE LED 104 is lit under the control of the PLD 1240, such as when the surgeon releases the foot switch 88 when the parsing cable is electrosurgically energized and the operation is in capture mode. During the stop time interval, the application of voltage to the motor assembly 160 and such excitation of the parsing cable is terminated, and these re-applied voltages are applied to the deployment / housing of the arm on the housing assembly 12. This can be done only after the switch 54 is activated and the foot switch 88 is recoupled and the capture switch 56 is activated. During such a restart operation, the control assembly again generates a boost voltage mode of operation to ensure the generation of a cutting arc in the purging cable executing the activated capture electrode.

  If the PCSM test performed by the PCSM circuit 462 fails, 1240 generates a PCSM_LED pulsation signal on the PLD output line 1313, which is buffered in the device 1320 and has an internal resistance in the device 1322 that is operatively connected to the capacitors in the array 1324. Filtered by As a result, the buffered pulse pulsates a low true signal on line 1326. Line 1326 is routed to front panel LED 92 and the return is provided at line 1328 and resistor 227 coupled to + 5V via line 1330.

  The handle interlock check LED 80 on the console 64 is lit in response to the presence of a HANDLE_LED signal at the terminal of the PLD 1240 connected to the line 1315. As shown in FIG. 41E, line 1315 is buffered and associated with an array 1324 of capacitors to provide a filtered and buffered illumination input on line 1334 extending to the prompt panel connector 1299 of console 64. Inside the working device 1322, it returns as the _LED_DRVOUT signal shown for filtering by resistance.

  The power LED 84 lights up upon activation of the switch 82 and then provides a value of + 5V on line 1330. This provides an input for line 1340 filtered at capacitor C158. The corresponding return at line 1342 is connected to line 1344 and is also grounded and filtered at capacitor C159.

  Once buffered and filtered, lines 1308-1313 are shown as lines 1308 a-1313 a, respectively, and are oriented to connector 1298 for application to the upper front panel of console 64. Lines 1308a through 1312a are additionally oriented to connector 67 and are associated with connector 1300 directed to housing 14 of instrument 12, as shown by lines 1308b through 1312b, respectively, as shown in lines 1308a. Thru 1312a are additionally tapped.

  Lines 1316-1319 of line array 1296 extending from PLD 1240 carry switching signals and interlock data from device 12. In this regard, the above-described interlock signal, INTERLOCK_ID, is a current through a coding resistor mounted within the housing assembly 14 to ensure proper interconnection with the connector 68 (see FIG. 1). One of the things that provides the passage of. A protection network, typically indicated at 1346, is provided in connection with line 1316, which extends to connector 1300 that is operatively associated with connector 68. In this regard, it is configured identically to network 1268, which is implemented with clamping diodes D102 and D103, resistors R231 and R232, and capacitor C162. It can be seen that the filter capacitor C163 is connected to the connector 1300.

  Line 1317 carries a signal indicating the actuation of voltage application / alignment switch 57 mounted on instrument 12. That signal, identified as “ENGZ / POS_SW”, is submitted to PLD 1240 via a protection network generally indicated by connectors 1300 to 1348. Network 1348 is identical to network 1346 and includes clamping diodes D104 and D105, resistors R233 and R234, and capacitor C164. Next, below line 1317 is line 1318 that carries the output signal “ARM_SW” of arm switch 56 mounted on instrument 12. This signal is identical to network 1346 and extends to network 1346 through a protection network identified generally at 1350 and includes clamping diodes D106 and D107, resistors R235 and R236, and capacitor C165. . Line 1319 extends through a protection network 1352 that is configured identically to network 1346, carrying the output of capture switch 56 at instrument 12 identified as "CAPTURE_SW". In this connection, the network 1352 is composed of clamping diodes D108 and D109, resistors R237 and R238, and a capacitor C166.

  In addition, provided to housing assembly 14 via connector 1300 is a + 5V regulated power supply on line 1354 filtered by capacitor C163.

  Looking briefly at FIG. 41-E, the 4-line array 1274 reappears and extends to the input terminal of the buffer circuit 1356. Further extending to the input of device 1356 is the previously described line 1315 that provides the signal “_LED_DRVOUT” on line 1332 as described in connection with FIG. 41D. The remaining four outputs of device 1356, shown as buffered signals, from array 1274 are shown as line array 1358, which are directed to relays in the PCSM circuit.

  The + 5V stabilized power supply described above in connection with FIGS. 41-A through 41-E is derived at the control board 624 by the circuit illustrated in FIG. Referring to the drawing, a regulator of model number LM2940CT-5.0, marketed by National Semiconductor Inc. of Sunnyvale, Calif., Is connected to the + 12V input on line 1364, and capacitors C168 to C170 and diodes. Configured with D111, shown at 1362 is providing the stabilized + 5V supply on line 1366. This + 12V input is derived as described in connection with FIG.

  Referring to FIG. 43, a network is shown for determining the presence of a low voltage power supply undervoltage condition, as is present at line 1294 in PLD 1240. Looking at this figure, a passive operational amplifier with the output of a feedback configuration at line 1370 where the + 12V power supply described above is directed to one input of resistor R240, capacitor C172, diode D113, and 1372. Processed and reduced by a network including 1368. Comparator 1372 may be model number LM358D sold by National Semiconductor, Inc. of Sunnyvale, California. The reference input to comparator 1372 is derived from a split network connected to the + 12V supply and configured with resistors R241 through R243 and capacitor C173 to provide a reference input on line 1374. Device 1372 is configured with a + 5V input and capacitor C174 to provide a low logic TRUE output on line 1294 in the event of an undervoltage condition. In this regard, note that line 1294 is connected to the + 5V supply via pull-up resistor R224.

  Referring to FIG. 44, input and output RC filtering associated with the PLD 1240 is provided and their filtering signals are provided along the power supply input to the connector 1378, thereby causing the control board to the power board. A filtering network providing a connection to is shown. In the drawing, a high voltage overvoltage signal, a DC link voltage overvoltage signal, and a DC link voltage undervoltage signal are received from connector 1378 on lines 1160, 1198, and 1204, respectively, and each pull-up resistor R246-R248 is connected. Connected to a + 5V source. In addition, the signal so received is filtered by the multi-resistor component 1380 and each filter capacitor 176-C178.

  Line 1280 carrying the high voltage precursor voltage application command signal carrying the reset output; and line 952 carrying the high voltage capture command signal are each processed by a separate resistor within the multi-resistor component 1380. In addition, lines 952, 1284, and 1286 are connected to a + 5V source through a pull-up resistor within multi-resistor component 1382, as provided by 3-line array 1384. A split voltage is provided from resistor array 1386 to connector 1378, with + 12V source and ground inputs presented to the connector from both sides of capacitor C179.

  High voltage overcurrent signal on line 1144; overtemperature signal on line 1176; motor forward stall signal on line 1088; and foot switch and vacuum switch generally indicated by arrow 1302 and labeled "OPTO_SW" The actuation signal is filtered by a separate resistor within multi-resistor component 1388 and each capacitor C180-C183. Of this group of lines, the foot and vacuum switch lines generally indicated by lines 1144, 1176, and 1302 are connected to the + 5V source through separate pull-up resistors within component 1382.

  RF inverter enable command; boost mode command; motor forward direction command; and motor rearward direction command are processed by separate resistors within multi-resistance component 1388. Of this population, lines 1290 and 1292 are connected to a + 5V source through a pull-up resistor within multi-resistor component 1328.

  Motor On Input; Motor Coupled Input; Motor Back Stall Command, and Overpower Input are processed by separate resistors inside multi-resistance component 1390. Of these lines, lines 1064 and 1076 are connected to a + 5V source through separate pull-up resistors within multi-resistor component 1382. Line 1076 is grounded through filter resistor R249 and filter capacitor C184. Lines 1064, 1100, and 1170 are each grounded through filter capacitors C185 through C187.

  Referring to FIG. 45, a circuit that drives a speaker inside the console 64 and has a potentiometer near the volume is clarified. A pair of lines from this speaker (not shown) is connected to a connector as indicated by reference numeral 1394. Similarly, a potentiometer (not shown) from the volume control is applied to the reverse connector 1396.

  The PLD-derived sound signal line 1398 (FIG. 41A) reappears in this simplified drawing and is asserted to the potentiometer via the resistor R251 in combination with the line 1344 via the connector 1396 and the resistor R252. ing. The volume input filtered by capacitor C158 is then provided on line 1400. Line 1400 is directed to an amplification stage that includes an operational amplifier 1402 comprised of a + 5V stabilized supply power, a capacitor C190, and a feedback line 1404. The output is provided on line 1406, which incorporates resistor R253 and extends to the oscillator network indicated generally at 1408. The oscillator network includes an amplifier component 1410 of model number LM386N-1, consisting of resistors R254 and R255, capacitors C191 through C195, and + 12V supply power, providing audio output on line 1412. . The audio output is provided whenever an electrosurgical cut is made, either by a precursor electrode or a parsing cable. Further, if a failure occurs in the PCSM test of the distributed return electrode 70, this sound is pulsed. Amplifiers such as 1410 are commercially available from Analog Device, Inc. of Norwood, Massachusetts.

  46-A to 46-C need to be interpreted together according to the label above. These drawings illustrate test signal generation and switching related to fault testing and self-testing for the distributed return electrode 70. The circuit shown is a component of the PCSM circuit described in connection with block 462 of FIG. This PCSM test is performed when the procedure is just started, and if it fails, the procedure is suppressed and both audible and visual types of pulsed alarm signals are generated. A visual alarm is shown in red LED 92 (FIG. 1). Normally, the self-test is performed when the instantaneous circuit is initially turned on with the operation of the switch 82. Later, as the switch 92 is activated, a test of the distributed electrode 70 is also performed.

  Referring to FIG. 46-B, a connector 1416 is provided that is connected to line pairs 464 and 466 as illustrated in connection with FIG. In FIG. 46-B, connection RE1 is illustrated by line 1418 connected to ground through resistor R257. Connection RE2 is shown by line 1419 connected to ground through resistor R258. This circuit, generally indicated by RE1 and RE2, extends from electrode pads 72 and 74 back to high voltage output stage 450 and is tapped for instantaneous testing purposes. The PCSM circuit 462 functions to apply an approximately 50 KHz low voltage signal across the pads 72 and 74 to verify that the distributed return electrode 68 is properly connected to the patient. Generally, this test is evaluated for a resistance tolerance between, for example, about 20 ohms and about 80 ohms. A resistance smaller than the former value indicates a short circuit state, and a resistance greater than the latter value indicates a disconnected state. These resistance values may be changed as desired by the designer.

  Referring to FIG. 46-A, an oscillator network that derives the 50 KHz frequency described above is shown generally at 1420. Network 1420 includes operational amplifiers 1422 configured with resistors R259 through R263; capacitors C200 through C203; resistors R264 through R267, capacitor C204, and complementary amplifier 1424 configured with potential difference frequency adjustment network 1426; power supply input 1428 A transistor Q30 and a diode D115. The potentiometer 1426 is configured with a capacitor C205 and resistance components R258 to R270. The input device 1428 may be a model number REF-02C / AD available from Analog Devices, Inc. of Norwood, Massachusetts. The 50 KHz output generated by network 1420 is provided on line 1430 through an input resistor R271 to an amplification stage, indicated generally at 1432, which functions to adjust to an RMS value of about 7V or 12V peak-to-peak. Directed. This 1432 is mounted using an operational amplifier 1434 configured with resistors R272 to R275 and capacitors C206 to C208. This processed 50 KHz output is provided on line 1436 which is filtered by resistor R276 and capacitor C209. Referring again to FIG. 46-B, it can be seen that line 1436 has been tapped at line 1438 to provide an “OSC_OUT” signal. After the tap at line 1238, line 1436 includes a resistor R277 having a value of about 50 ohms and extends to the tap located on the opposite side labeled "50KHz", identified at 1440. Be present. Extending between taps 1438 and 1440 is a series of four relay-implemented networks generally indicated at 1441-1444.

  Referring to network 1441, it can be seen that relay K12 is connected between lines 1446-1447. As it goes down, resistors R278 and R279 are incorporated and activated by PLD 1240 using the signal generated on line 1448 extending to the gate of pnp transistor Q31. Transistor Q31 is configured with diode D117 and resistor 280 and applies a voltage to the solenoid component of relay K12 in response to a signal applied from line 1448. This functions to connect the 50 KHz signal on line 1436 and ground to 1418 and 1419, respectively, to perform the PCSM test. This test is performed in response to the surgeon activating the start / reset switch 92 (see FIG. 1).

  Referring to relay network 1442, relay K13 is connected between lines 1450 and 1451, the latter extends to ground, and the former incorporates a 200 ohm resistor R281. As soon as it is pushed, the relay K13 is closed in response to the operation signal imposed by the PLD 1240 on the line 1452. Line 1452 incorporates resistors R282 and R283 and is connected to the gate of pnp transistor Q32. Transistor Q32, configured with diode D118 and resistor R284, affects the voltage application of the solenoid component of relay K13, closes it, and connects the 50 KHz signal to ground through resistor R281 on line 1436. Provide a high resistance self-test. Referring to relay network 1443, relay K14 is connected to a 50 KHz signal at line 1436 by line 1454 and is connected to ground via line 1455. Line 1454 incorporates a 49.9 ohm resistor R288. The solenoid component of relay K14 is energized so that as it squeezes, the relay is closed in response to the signal from PLD 1240 present on line 1456. Line 1456 incorporates resistors R285 and R286 and extends to the gate of pnp transistor Q33. Transistor Q33 is configured with diode D119 and resistor R287 so that when turned on in response to a signal on line 1456, a voltage is applied to the solenoid component of relay K14. This redirects the 50 KHz signal across the 49.9 ohm resistor at R288 from line 1436 to ground.

  Referring to relay network 1444, it can be seen that relay K15 is connected between line 1396 connected to line 1436 and line 1459 connected to ground. The solenoid component of relay K15 is derived from PLD 1240 as it is pushed in, and is energized as the signal asserted on line 1460 is generated. In line 1398, resistors R289 and R290 are incorporated and connected to the gate of pnp transistor Q34. Transistor Q34 is configured with diode D120 and resistor R291, and in response to being turned on from line 1460, a voltage is applied to the solenoid component of relay K15. This connects line 1436 to ground via lines 1458 and 1459, providing a self-test corresponding to a short circuit.

  Referring to FIG. 46-C, it can be seen that operating lines 1448, 1452, 1456, and 1460 are connected to the collector output stage of each optocoupler 1462-1465. The emitter components at the outputs of couplers 1462 through 1465 are connected to ground via lines 1466, and each coupler is connected to a + 12V source via respective resistors R293 through R296 and line 1468. . The anode input to optocouplers 1462 through 1465 is connected to the + 5V source at line 1470 via respective resistors R297 through R300, while its cathode side input is connected to each input line 1472 through 1475. It is connected to the. These input lines 1472 through 1475 are components of the array of lines 1358 described in connection with FIG. 41E and provide a buffered output of the lines in the line array 1274 extending from the PLD 1240. Shown as a thing. That is, similar to the PCSM self test, the return electrode 70 test is executed under the command of the PLD 1240. For example, even though relay K12 is open, the relay K15 of network 1444 is energized so that the signal is shorted on line 1436 during the time interval that the test is asserted from networks 1380-1444. Please keep in mind.

  Referring to FIG. 67, an isolated power supply used to generate the + 12V is shown. The power supply is configured around a supply component 1478 and may be provided with a model number NMS1212 device, etc., marketed by Newport Components, Inc. of Milton Keynes, England. In effect, device 1478 converts + 12V to + 12V and -12V. It is configured with inductors L16 through L19 and capacitors C211 through C216 to provide + 12V isolated at output 1480 and -12V isolated at output 1481. Device 1478 is provided with a + 12V input on line 1482 from power transistor Q36, its source is connected to + 12V from lines 1484 and 1485, and its gate terminal using line 1486. Connected to line 1484. Resistors R310 and R311 are incorporated in line 1484, connected to the collector of NPN transistor Q37, and its emitter is connected to ground. Transistor Q37 is gated on such that power supply 1478 is enabled by the PCSM_ENBL signal asserted from PLD 1240 on line 1276 via base register R312. Line 1276 is connected to ground through resistor R313 and can be seen extending from PLD 1240 in FIG. 41-A.

  Referring to FIGS. 48-A to 48-B that need to be interpreted as labeled above, a detection or comparison circuit defined by a window is illustrated, whereby the network 1442 to 1444 is self-explanatory. Similar to the test, an evaluation of the actual PCSM test from network 1441 (FIG. 46B) is performed. In general, an ohmic window showing an effective distributed electrode 70 connection belongs between about 20 ohms and about 80 ohms. Referring to FIG. 48-A, taps 1338 and 1440 as described in connection with FIG. 46-B are shown extending to the input of differential amplifier 1488. FIG. For example, the amplifier 1488 may be a device of model number AMP02FS sold by Analog Devices, Inc. of Norwood, Mass., And is implemented with + 12V and −12V and capacitors C218 and C219. . As configured, device 1488 provides a single-ended signal to ground on output line 1490 in response to a floating signal at resistor R277 (see FIG. 46-B). The AC signal is then sent on line 1490 via the input resistor R317 to a precision rectifier, indicated generally at 1426. It can be seen that this rectifier 1492 includes an operational amplifier 1494 configured with resistor R318, diodes D122 and D123, and capacitors C220 and C221, providing rectification without the phenomenon of diode voltage drop. The DC signal on output line 1496 is then proportional to the current in the return electrode or the test evaluation from networks 1442 to 1444 and is applied across capacitor C222. Resistor R319 extends between line 1476 and ground and functions with respect to the selective discharge of capacitor C222.

  The DC signal on line 1496 is directed to the positive input of comparator 1498 and to the negative input of the corresponding comparator 1433 via line 1500. The reference inputs for these comparators 1498 and 1502 are provided from lines 1504 and + 12V incorporating reference defining resistors R320 through R322. It can be seen that this reference input is further connected to filtering capacitors C223 and C225, while the + 12V input to comparator 1498 is filtered by capacitor 224. Comparators 1498 and 1502 may be provided as a model number LM319M device marketed by National Semiconductor, Inc. of Sunnyvale, California.

  If the current shown by line 1496 corresponds to a lower threshold of, for example, 20 ohms and an upper limit of, for example, 80 ohms, a positive voltage signal is applied from resistor R283 on line 1506. Referring to FIG. 48-B, it can be seen that line 1506 extends to the input-side anode of optocoupler 1508. The collector component at the output of optocoupler 1508 is connected to + 12V via resistors R324 and R325, while its emitter output is on line 1278 connected to ground via resistor R286. Is provided. Line 1278 functions to apply a signal therein indicating the validity of the test “PCSM_VALID” to PLD 1240 as shown in FIG. 41-A.

  Since some modifications may be made to the above methods, systems, and apparatus without departing from the scope of the present invention as it relates to the present application, they are included in the above description or accompanying drawings. It is intended that all of the content presented in is illustrative and is not to be construed as limiting.

1 is a system perspective view of the present invention showing a handheld device, a control console, a return electrode, a foot switch, and a suction system component. FIG. FIG. 2 is a perspective view of the instrument shown in FIG. 1 including a disposable component removed from the reusable housing. FIG. 3 is an enlarged view of the reusable housing shown in FIG. 2. FIG. 3 is a plan view of a leaf assembly used in the disposable component shown in FIG. 2. FIG. 5 is a cross-sectional view of a capture device leaf assembly and drive lot. FIG. 3 is a partial cross-sectional view of the front region of the instrument of FIG. 2. FIG. 2 is a front view of the distal end portion of the instrument shown in FIG. 1 showing the arrangement of the capture device leaf before deployment. FIG. 2 is a front view of the distal portion of the instrument of FIG. 1 showing an arrangement in which the leaves of the capture device are deployed. FIG. 2 is a partial cross-sectional view of the disposable component of the device of FIG. 1 schematically showing the capture device leaf deployed to its maximum diameter. FIG. 10 is a schematic leaf view of a capture device showing the completion of capturing a tissue volume, and is a partial cross-sectional view of the instrument of FIG. 1 is a schematic diagram illustrating an electrosurgical system and patient provided to demonstrate tissue impedance and total impedance. FIG. It is the schematic showing the part shown by FIG. 11-A. It is the schematic which shows the arc structure by the electrosurgical activation electrode of the conventional system which fixed the shape and magnitude | size. 4 is a graph of time against applied voltage and overall resistance of an electrosurgical system including an electrosurgical generator of the present invention. Figure 6 is a graph showing application of boost voltage and resulting current. 2 is a block diagram of electrosurgical generation and control features of the system of the present invention. FIG. It is an insulation diagram of the control system shown in FIG. FIG. 16 is a schematic circuit diagram showing the EMI filter, front panel switch, and PFC boost converter shown in the block of FIG. 15. FIG. 16 is a schematic circuit diagram showing the EMI filter, front panel switch, and PFC boost converter shown in the block of FIG. 15. FIG. 18B is an electrical schematic diagram illustrating relay solenoid components applied at the contacts shown in FIG. 17-A. FIG. 2 is an electrical schematic of a temperature responsive component used with the console shown in FIG. FIG. 3 is an electrical schematic diagram of a power source for providing input power to a motor included in the reusable housing of the instrument shown in FIG. 2. FIG. 16 is an electrical schematic diagram of one low-voltage power supply shown in the block diagram of FIG. 15. FIG. 16 is an electrical schematic of the motor drive shown in the block schematic of FIG. 15 and further shows the solenoid components of the relay used in the system of the present invention. FIG. 16 is an electrical circuit diagram of the high voltage transfer and high voltage output, RF inverter, relay disconnection, LC filter, rectifier, isolation transformer, and 100 KHz inverter shown in the block diagram of FIG. 15. FIG. 16 is an electrical circuit diagram of the high voltage transfer and high voltage output, RF inverter, relay disconnection, LC filter, rectifier, isolation transformer, and 100 KHz inverter shown in the block diagram of FIG. 15. FIG. 24 is a schematic pulse showing the operation of the resonant change phase shift converter shown in FIG. FIG. 3 is an electrical schematic diagram of a power factor modified boost converter controller and link voltage evaluation circuit including associated enablement circuits. FIG. 3 is an electrical schematic diagram of a power factor modified boost converter controller and link voltage evaluation circuit including associated enablement circuits. It is the electrical schematic which shows a 1st side power supply. FIG. 3 is an electrical schematic diagram of a control circuit for providing phase shift resonant movement control. It is an electrical schematic diagram of the control circuit for DC link voltage adjustment. FIG. 3 is an electrical schematic diagram of a reference voltage deriving circuit. FIG. 3 is an electrical schematic diagram of an output power monitoring signal acquisition multiplier circuit. FIG. 5 is an electrical schematic for providing constant power control for the electrosurgical generator of the present invention. 2 is an electrical schematic diagram of a control circuit used with an RF inverter. FIG. It is an electrical schematic diagram of a motor current amplifier circuit. It is an electric circuit schematic diagram of a motor current monitoring circuit. FIG. 3 is an electrical schematic diagram of a motor monitoring electrical circuit. FIG. 3 is an electrical schematic diagram of a motor monitoring electrical circuit. FIG. 3 is an electrical schematic diagram of a motor monitoring electrical circuit. FIG. 6 is an electrical schematic diagram illustrating induction of a reset signal and an enable signal. FIG. 2 is an electrical schematic diagram of a circuit for obtaining an overcurrent state. FIG. 2 is an electrical schematic diagram of a circuit for monitoring an overvoltage condition. FIG. 2 is an electrical schematic diagram of a circuit for monitoring power levels. FIG. 4 is a schematic diagram of a power level monitor circuit used in the constant power embodiment of the electrosurgical generator of the present invention. FIG. 2 is an electrical schematic diagram of a circuit for monitoring excessive temperature conditions. FIG. 3 is an electrical schematic diagram of a DC link voltage level monitoring circuit. FIG. 2 is an electrical schematic diagram illustrating a circuit for obtaining an input for operating a foot switch and a suction switch. FIG. 6 is a programmable logic device circuit diagram with associated output buffering and filtering. FIG. 6 is a programmable logic device circuit diagram with associated output buffering and filtering. FIG. 6 is a programmable logic device circuit diagram with associated output buffering and filtering. FIG. 6 is a programmable logic device circuit diagram with associated output buffering and filtering. FIG. 6 is a programmable logic device circuit diagram with associated output buffering and filtering. It is an electrical schematic diagram of a power supply. It is an electrical schematic diagram of a low voltage power supply monitoring circuit. It is an electric circuit diagram showing PLD signal input and output processing. It is an electric circuit diagram of a voice control circuit. FIG. 6 illustrates frequency generation and test switching components of a patient circuit safety monitor (PCSM) circuit. FIG. 6 illustrates frequency generation and test switching components of a patient circuit safety monitor (PCSM) circuit. FIG. 6 illustrates frequency generation and test switching components of a patient circuit safety monitor (PCSM) circuit. It is an electrical schematic diagram of a power supply. It is a figure which shows the circuit which performs the window analysis of a return electrode. It is a figure which shows the circuit which performs the window analysis of a return electrode.

Claims (65)

An electrosurgical generator connectable to a power input,
An input treatment network responsive to the power input to provide a first output;
A frequency generator responsive to the first output and the frequency control input to derive an output having a predetermined waveform;
An output voltage control circuit responsive to a voltage level control input to derive an electrosurgical voltage output and an electrosurgical energy output at the electrosurgical frequency;
An output stage responsive to the electrosurgical energy output of the output voltage control circuit and electrically connectable to an electrosurgical instrument;
In order to derive the voltage level control input to provide a boost electrosurgical voltage level during the boost interval and then derive a normal cutting electrosurgical voltage level lower than the boost electrosurgical voltage level. An electrosurgical generator including a responsive control assembly. The electrosurgical generator of claim 1, wherein the boost electrosurgical voltage level is 1.2 to 1.5 times greater than the normal cutting electrosurgical voltage level. The electrosurgical generator of claim 1, wherein the boost interval is approximately 100 to 1000 milliseconds. The electrosurgical generator of claim 1, wherein the boost interval is approximately 250 to 750 milliseconds. The electrical of claim 1, wherein the control assembly derives the voltage level control input to provide the boost electrosurgical voltage level at a maximum amplitude in a range of about 1000 volts to about 2000 volts. Surgery generator. The electrical of claim 1, wherein the control assembly derives the voltage level control input to provide the boost electrosurgical voltage level in a range of about 1100 volts to about 1300 volts at a maximum amplitude. Surgery generator. 6. The voltage level control input of claim 5, wherein the control assembly derives the voltage level control input to provide the normal cutting electrosurgical voltage level in a range of about 700 volts to about 1200 volts with a maximum amplitude. Electrosurgical generator. 7. The control assembly of claim 6, wherein the control assembly derives the voltage level control input to provide the normal cutting electrosurgical voltage level in a range of about 800 volts to about 1000 volts with a maximum amplitude. Electrosurgical generator. The input treatment network is
A boost converter network responsive to a converter control input to derive the first output at a first value provisional voltage level;
The electrosurgical procedure of claim 1, including a converter control network responsive to the provisional voltage level and the power input to effectively provide power factor correction and derive the converter control input. generator. The output voltage control circuit includes a relay switch responsive to a relay control input to terminate the electrosurgical energy output;
The electrosurgical generator of claim 1, wherein the control assembly is responsive to a fault condition to derive the relay control input. A high voltage monitor responsive to the electrosurgical energy output to derive a high voltage monitor signal;
The electrosurgical generator of claim 10, wherein the control assembly is responsive to derive the relay control input when the high voltage monitor signal exceeds a high voltage threshold. The electrosurgical generator of claim 11, wherein the control assembly is responsive in the presence of the voltage level control input providing a boost electrosurgical voltage level to disable the relay control input. . A method for generating an electrosurgical cutting arc at an electrode facing an animal tissue comprising:
Providing an input treatment network responsive to an applied source of electrical power to derive a first output;
Providing a link inverter containing network responsive to the first output to derive a controllable amplitude link voltage;
A predetermined electrosurgical cutting frequency R.D. F. Responsive to the link voltage to produce an output. F. Providing an inverter network and indicating an inverter voltage level corresponding to the link voltage controllable amplitude;
Increasing the inverter voltage level to derive an electrosurgical cutting output at an electrosurgical cutting voltage level;
Starting and continuing to apply the electrosurgical output to the electrode;
Monitoring selected electrical parameters of the electrosurgical output to provide an output monitor signal;
Comparing the monitor signal with a reference value indicative of a target value of the selected electrical parameter to derive a program control signal;
Controlling the link inverter-containing network by applying the program control signal. Monitoring the selected electrical parameter monitors the electrosurgical cutting voltage to provide the output monitor signal as a high voltage monitor signal;
Comparing the monitor signal with a reference value performs the comparison using a predetermined electrosurgical cutting voltage level as the target value;
The step of controlling the link inverter-containing network is performed by being applied to the program control signal at a slow rate effective to avoid oscillation of the electrosurgical cutting output. The method of claim 13. 15. The method of claim 14, wherein the process for controlling the link inverter applies the program control signal under low bandwidth conditions. Said d. To provide a link voltage control feedback signal; c. The process of monitoring the link voltage amplitude;
15. The method of claim 14, further comprising controlling the link inverter-containing network by applying the feedback signal to the link inverter-containing network at a higher speed than the slow speed. The method of claim 16, further comprising applying the feedback signal at a high gain in the process for controlling the network including the link inverter. The electrosurgery to influence the derivation of the link voltage at a boost level during a boost interval effective to cause generation of the electrosurgical cutting arc when the electrode is in contact with the tissue. 14. The method of claim 13, wherein the step of controlling the link inverter-containing network applies the program control signal when initiating the application of surgical output. The method of claim 18, wherein the step of controlling the link inverter containing network provides the boost level during the fixed boost interval. 20. The method of claim 19, wherein the fixed boost interval is about 0.5 seconds. The method of claim 19, wherein the fixed boost interval is approximately three-eighth seconds. Controlling the link inverter containing network is applied to the program control signal to derive the link voltage at the boost level during the boost interval, and then the link voltage at a disconnect level lower than the boost level. 19. The method of claim 18, wherein the program control signal is applied so as to be effective in deriving and maintaining arc formation at the electrodes. 23. The method of claim 22, wherein the cutting level corresponds to the applied power value of the electrosurgical output that is approximately half of the power value of the electrosurgical output at the boost level. Monitoring the selected electrical parameter monitors the electrosurgical cutting voltage level and a corresponding electrosurgical current to provide the output monitor signal as a power monitor signal;
The process of comparing the monitor signal and a reference value performs the comparison using a predetermined power value as the target value,
The method of claim 13, wherein the step of controlling the link inverter-containing network is performed by applying the program control signal. The electrosurgery to influence the derivation of the link voltage at a boost level during a boost interval effective to cause generation of the electrosurgical cutting arc when the electrode is in contact with the tissue. 26. The method of claim 24, wherein the step of controlling the link inverter containing network applies the program control signal when initiating the application of surgical output. 26. The method of claim 25, wherein controlling the link inverter containing network provides the boost level during the fixed boost interval. 27. The method of claim 26, wherein the fixed boost interval is about 0.5 seconds. 27. The method of claim 26, wherein the fixed boost interval is about 3/8 second. Controlling the link inverter containing network is applied to the program control signal to derive the link voltage at the boost level during the boost interval, and then the link voltage at a disconnect level lower than the boost level. 26. The method of claim 25, wherein the program control signal is applied so as to be effective in deriving and maintaining arc formation at the electrode. 30. The method of claim 29, wherein the cutting level corresponds to the applied power value of the electrosurgical output that is approximately half of the power value of the electrosurgical output at the boost level. Said step of providing an input treatment network provides a power factor modification for said applied source of electrical power and adjusted d. c. The method of claim 13, wherein the first output is derived as a voltage. The process of providing a link inverter-containing network includes an inverter control network that influences the resonant transformation phase shift control of the link inverter, and further d. c. 14. The method of claim 13, wherein the link inverter-containing network is provided as including a rectifier for providing the link voltage as a link voltage. A method for generating an electrosurgical cutting arc at an electrode configured to cut tissue comprising:
Providing an input treatment network in response to an applied source of electrical power to derive a first output;
Providing a frequency generator-containing network in response to the first output and the control input to derive a second output having a tissue cutting waveform;
More than the first voltage level effective to maintain the generated arc and subsequently the generated arc in response to the second output and effective to generate the arc. Providing an output stage electrically connectable to said electrode for applying electrosurgical energy at a second level of low voltage;
Deriving the first level of voltage at the beginning of application of the electrosurgical energy to the electrode during a boost interval effective to generate the cutting arc, followed by the second of voltage Controlling the frequency generator-containing network for deriving a level of. 34. The process of controlling the frequency generator-containing network provides the first voltage level approximately 1.2 to 1.5 times compared to the second voltage level. the method of. 34. The method of claim 33, wherein the step of controlling the frequency generator containing network provides a fixed boost interval of about 0.5 seconds. 34. The method of claim 33, wherein the step of controlling the frequency generator containing network provides the boost interval fixed at an interval of about 3/8 second. 34. The electrosurgical generator of claim 33, wherein the step of controlling the frequency generator containing network provides the first level in a range of about 1000 volts to about 2000 volts at a maximum amplitude. 34. The electrosurgical generator of claim 33, wherein the step of controlling the frequency generator containing network provides the first level in a range of about 1100 volts to about 1300 volts at a maximum amplitude. 38. The electrosurgical generator of claim 37, wherein the step of controlling the frequency generator containing network provides the second level in a range of about 700 volts to about 1200 volts at a maximum amplitude. 38. The electrosurgical generator of claim 37, wherein the step of controlling the frequency generator containing network provides the second level in a range of about 800 volts to about 1000 volts at a maximum amplitude. An electrosurgical generator connectable to a power input,
An input treatment network responsive to the power input to derive a first value provisional voltage output;
A first inverter network responsive to the provisional voltage and the first inverter control input to derive a first AC voltage output having a second value smaller than the first value at the first inverter output;
A first inverter control network connected to the first inverter network and deriving the first inverter control input;
Corresponding to the second value of the first AC voltage output; d. c. A rectifier network responsive to the first alternating voltage output to derive a link output at a voltage level;
The link output d. c. A second inverter network having a voltage amplitude established by the voltage level and responsive to the link output to derive a second AC voltage output at an electrosurgical frequency value;
A second inverter control network connected to the second inverter network to influence the derivation of the second alternating voltage output electrosurgical frequency;
A high voltage transformer having a first side responsive to the second alternating voltage output and a second side for deriving an electrocution energy input at an electrosurgical voltage level and the electrosurgical frequency;
An electrosurgical generator including an output stage connected to the second side of the high voltage transformer and electrically connectable to an electrosurgical instrument. 42. The electrosurgical generator of claim 41, wherein the first inverter control network derives the first control input and affects a resonance transform phase shift of the first inverter. The first inverter control network is
A power monitoring circuit responsive to the electrocutting energy input to derive a program signal;
42. The electrosurgical generator of claim 41, further comprising a controller network responsive to the program signal to derive the first inverter control input. A high voltage monitor responsive to the electrocutting energy input to derive a high voltage monitor signal;
A high voltage current monitor responsive to the electrical cutting energy input to derive a high voltage current monitor signal;
The first inverter control network is
A power derivation network responsive to the high voltage monitor signal and the high voltage current monitor signal to derive a monitored power signal;
A power reference value for deriving a program signal and a comparator network responsive to the monitored power signal;
42. The electrosurgical generator of claim 41, further comprising a controller network responsive to the program signal to derive the first inverter control input. The power derivation network is
A multiplier circuit responsive to the high voltage monitor signal and the high voltage current monitor signal to derive a product output;
45. The electrosurgical generator of claim 44, further comprising an integration network responsive to the product output to derive the monitored power signal. A control assembly for deriving a boost voltage signal during the boost interval;
The first inverter control network is responsive to the boost voltage signal for deriving the first inverter control input that affects derivation of the second value of the first AC voltage output at a boost voltage value. And responsive thereafter to derive the first inverter control input that affects the derivation of the second value of the first AC voltage output at a normal disconnect voltage value lower than the boost voltage value. 42. The electrosurgical generator of claim 41. The electrosurgical generator of claim 46, wherein the boost voltage value is 1.2 to 1.5 times greater than the normal cutting voltage value. 42. A separation transformer comprising a first side connected to the first AC output and a second side providing the first AC voltage output to the rectifier network. An electrosurgical generator as described in. 42. The electrosurgical generator of claim 41, wherein the second inverter network includes a resonant tank circuit. The electrosurgical generator of claim 46, wherein the boost interval is approximately 100 to 1000 milliseconds. The electrosurgical generator of claim 46, wherein the boost interval is approximately 250 to 750 milliseconds. The electrosurgical generator of claim 46, wherein the boost voltage value affects the derivation of the electrosurgical voltage level in a range of about 1000 volts to about 2000 volts with a maximum amplitude. The electrosurgical generator of claim 46, wherein the boost voltage value affects the derivation of the electrosurgical voltage level in a range of about 1100 volts to about 1300 volts at a maximum amplitude. 53. The electrosurgical generator of claim 52, wherein the normal cutting voltage value affects the derivation of the electrosurgical voltage level in a range of about 700 volts to about 1200 volts with a maximum amplitude. The electrosurgical generator of claim 53, wherein the normal cutting voltage value affects the derivation of the electrosurgical voltage level in a range of about 800 volts to about 1000 volts with a maximum amplitude. The input treatment network is
A boost converter network responsive to the converter control input to derive a first value provisional voltage level;
42. The electrosurgical procedure of claim 41, comprising: a provisional voltage level and a converter control network responsive to the power input to effectively provide power factor correction and derive the converter control input. generator. A relay switch connected between the input of the rectifier network and the second inverter network and responsive to a relay control input to effect transmission or termination of the link output to the second inverter network;
42. The electrosurgical generator of claim 41, further comprising a control assembly responsive to a fault condition to derive the relay control input that terminates transmission of the link output to the second inverter network input. The first inverter control network comprises a power monitoring circuit responsive to the electrical cutting energy input to derive a power signal corresponding to a level of power indicated by the electrical cutting energy input;
58. The electrosurgical generator of claim 57, wherein the control assembly is responsive to derive the relay control input that terminates the transmission of the link output when the power signal exceeds a power threshold. . A high voltage monitor responsive to the electrical cutting energy input to derive a high voltage monitor signal;
58. The electrosurgery of claim 57, wherein the control assembly is responsive to derive the relay control input that terminates the transmission of the link output when the high voltage monitor signal exceeds a high voltage threshold. Surgery generator. A high voltage current monitor responsive to the electrical cutting energy input to derive a high voltage current monitor signal;
58. The electrosurgery of claim 57, wherein the control assembly is responsive to derive the relay control input that terminates the transmission of the link output when the high voltage monitor signal exceeds a high voltage threshold. Surgery generator. The link output d. c. A link voltage monitor responsive to the rectifier network link output to derive a link monitor signal corresponding to the voltage level;
The link output where the control assembly exceeds a link over voltage threshold; d. c. 58. The electrosurgical generator of claim 57, responsive to deriving the relay control input to terminate the transmission of the link output when the link monitor signal corresponds to a voltage level. The control assembly is configured to cause the link output d. c. 62. The electrosurgical generator of claim 61, responsive to deriving the relay control input to terminate the transmission of the link output when corresponding to a transmission level. Having a high voltage monitor corresponding to the electrocutting energy input for deriving a high voltage monitor signal;
A comparator network in which the first inverter control network is responsive to a predetermined electrosurgical cutting voltage level and the high voltage monitor signal to derive a program signal;
42. The system of claim 41, further comprising a controller network responsive to the program signal to derive the first inverter control input. 64. The system of claim 63, wherein the controller network is configured to derive the first inverter control input as the program signal applied late. The first inverter control network is
A link voltage monitor responsive to the link output to provide a link voltage to control the feedback signal;
The system of claim 64, wherein the controller network is further responsive to the link voltage control feedback signal to derive the first inverter control input.

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