JP2005323400A - Variable amplifier, and mobile terminal - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a current switching circuit which is a circuit for switching a plurality of current paths in an analog manner and is capable of easily designing control characteristics and a control section and operating even with a low power supply voltage, and provide an amplifier using the same and a mobile terminal including the same. <P>SOLUTION: Differential pairs D1-D3 are stacked on multiple stages, one side of each of the differential pairs D1-D3 is connected to each of reference voltages Vr1-Vr3 of a reference voltage generating circuit RVC, and another side is connected to a control voltage generating circuit CVC, so that a current switching circuit is configured. In such a circuit configuration, switching voltage setting can be uniquely determined by the reference voltage generating circuit RVC and it can also be set to a constant voltage. Furthermore, the control voltage generating circuit CVC can be controlled only by a unipolar operation such as just discharging or absorbing a current. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a variable amplifier using a current switching circuit used for a variable gain amplifier, a variable attenuator, a power source, and the like, which uses a characteristic of continuously switching a plurality of circuit blocks by an analog signal, and a portable terminal having the variable amplifier It is.

  There are many cases where a plurality of circuit blocks must be switched by an analog signal. For example, a plurality of amplifiers having different gains are continuously switched by an analog signal to configure a variable gain amplifier, or a plurality of attenuators having different attenuation amounts are continuously switched by an analog signal to configure an electronic volume. In many fields, a continuous switching method using analog signals is useful. Examples of the field include mobile terminals such as mobile phones, notebook personal computers, and PDAs.

  As one of such switching methods, Non-Patent Document 1 proposes a circuit for switching an amplifier by combining a bipolar transistor and a current source. A circuit diagram described in Non-Patent Document 1 is shown in FIG.

  In this circuit format, by changing two control voltage values of the control voltage generation circuit indicated by VCTRL1 and VCTRL2, the current value flowing into the terminals indicated by I1, I2, I3, and I4 can be changed. For example, when the voltage value of VCTRL2 is fixed to 1.5V and the voltage value of VCTRL1 is changed from 0.5V to 2.5V, the base voltage value of each transistor changes as shown in FIG.

Due to the change in the base voltage value, the current value flowing into the terminals indicated by I1, I2, I3, and I4 continuously changes as shown in FIG. Further, since the sum of the current values of all the transistors is determined by the current value of the current source connected to the emitter of the transistor, it is always a constant value.
“A Low-Noise Wideband Variable-Gain Amplifier Using an Interpolated Ladder Attenuator”, ISSCC Digest of Technical Papers, pp.280-281, Feb. 1991

  In the case of the circuit shown in FIG. 11, the control voltage generation circuit VCTRL1 has a maximum current of about 500 μA flowing from the switching circuit to the control voltage generation circuit when the control voltage is less than 1.83 V, and a maximum of about 250 μA when the control voltage is 1.83 V or more. Current flows from the control voltage generation circuit to the current switching circuit.

  In addition, when the control voltage of the control voltage generation circuit VCTRL1 is less than 0.75V, the control voltage generation circuit VCTRL2 has a maximum current of about 100 μA flowing from the control voltage generation circuit to the current switching circuit. When the voltage is 0.75 V or higher, a maximum current of about 650 μA flows from the current switching circuit to the control voltage generation circuit VCTRL2. As described above, the control voltage generation circuit must perform a bipolar operation that allows both a current to flow and a current to be sucked by the control voltage.

  In addition, the transistor used in this simulation cannot completely switch the current unless the base voltage difference is about 200 mV, but as can be seen from FIG. 12, it is difficult to ensure the base voltage difference. For this reason, it is difficult to increase the number of switching circuits. For example, when the number of switching circuits is increased to eight paths, the switching characteristics are uniform as shown in FIG. Difficult switching characteristics.

  Further, as a result of the study, it has been found that the conventional circuit not only requires a bipolar operation in the control voltage generator, but also has the following problems.

  In the conventional circuit, the current value of the current source connected to the emitter, each current value of the current source connected to the base, each resistance value connected between the bases, etc. are switched with respect to the control voltage. There are many parameters that determine the characteristics, and it is difficult to design switching characteristics for the control voltage.

  In order to make the switching characteristics steep, it is necessary to increase the difference between the respective base voltages, but in order to create such a state, a considerably high voltage is required as the control voltage, There is a problem that it is difficult to use with a power supply voltage of about 3V as in a portable terminal in a portable device and the use range is narrowed.

  An object of the present invention is to provide a variable amplifier having a current switching circuit that operates stably even at a low voltage of about 3 V, and a portable terminal including the same.

  In order to solve the above problem, a variable amplifier according to the present invention includes a differential pair having two control terminals, two current switching terminals, and one common terminal, and a constant current unit, The common terminal of the pair is connected to the constant current portion, the common terminal of the second differential pair is connected to the second current switching terminal of the first differential pair, and thereafter, the common terminal of the nth differential pair is the first ( n-1) Connected to the second current switching terminal of the differential pair (n is an integer of 2 or more), and reference voltages having different potentials are applied to the first control terminals from the first differential pair to the nth differential pair, respectively. A reference voltage unit to be supplied is provided, and a control unit that changes a control voltage applied to the second control terminal from the first differential pair to the nth differential pair is provided from the first differential pair to the nth differential. Switches current between each first current switching terminal up to the pair and the second current switching terminal of the nth differential pair. The current switching circuit and the n amplifiers having the amplifier control terminals are provided, and the characteristics are changed by changing the current flowing into the amplifier control terminals of the n amplifiers by the current switching circuit. It is characterized by becoming. In the variable amplifier, an attenuator is preferably provided between the input terminal and the amplifier so as to change the characteristics of the amplifier.

  According to the above configuration, when the differential pair is composed of an NPN-type bipolar transistor or an N-type field effect transistor, current is sucked from the current switching terminal on the side of the control terminal having a high voltage with respect to the common terminal. On the other hand, when the differential pair is composed of a PNP-type bipolar transistor or a P-type field effect transistor, current is discharged (flowed out) from the current switching terminal on the side of the control terminal having a low voltage with respect to the common terminal. ).

  In the above configuration, the control voltage when switching the current path basically matches the reference voltage. Therefore, if a simple resistor division is used as the reference voltage section, the current switching voltage can be easily set only by the resistance value. can do. In the above configuration, since the differential pair is used, no current flows into the control unit, so that the control unit does not need to perform a bipolar operation as in the prior art, and the control unit can be simplified.

  In addition, in the above configuration, if the upper limit of each reference voltage is set low, the control voltage can be lowered. Therefore, even if the power supply voltage is as low as 3 V, switching of the current path can be easily controlled. The range that can be used can be expanded.

  Furthermore, in the above configuration, since n amplifiers whose characteristics are changed by changing the current flowing into the amplifier control terminal by the current switching circuit, the operation of each amplifier is switched by the current switching circuit. Thus, the noise is low when the gain is high, and the linearity can be improved as the gain decreases.

  In the current switching circuit, as a first example, the differential pair is composed of an NPN type bipolar transistor or an N type field effect transistor, and the emitter or source (hereinafter referred to as emitter) of the first transistor and the second transistor. Are connected to each other to form a differential pair, and a base or gate (hereinafter referred to as a base) serving as a first control terminal of the first differential pair is connected to the first reference voltage, and the second difference The first base of the moving pair is connected to a second reference voltage that is higher than the first reference voltage, and thereafter, the first control terminal of the nth differential pair is connected to the first base of the (n−1) th differential pair. Is connected to the nth reference voltage, which is higher than the (n-1) th reference voltage, and the second base serving as the second control terminal from the first differential pair to the nth differential pair is connected to the control unit. Output power of the control unit. As the current rises, the first collector from the first collector or drain (hereinafter referred to as the first collector) to which the current flows becomes the first current switching terminal of the first differential pair to the nth differential pair. The configuration may be such that the second collector becomes the second current switching terminal of the nth differential pair.

  In the above-described configuration, the amount of current sucked in a plurality of current paths can be adjusted by using a differential pair configured by an NPN bipolar transistor or an N-type field effect transistor.

  In the above configuration, when a bipolar transistor is used, current flows only from the control unit to the differential pair, and current does not flow into the control unit. Therefore, the control unit does not need to perform a bipolar operation. In addition, when a field effect transistor is used, almost no current flows from the control unit to the differential pair. For this reason, the said structure can simplify the structure of a control part very much.

  In the current switching circuit, as a second example in place of the first example, the differential pair is composed of a PNP-type bipolar transistor or a P-type field effect transistor, and the emitter or source (hereinafter referred to as the emitter) of the first transistor. ) And the emitter of the second transistor are connected to each other to form a differential pair, and a base or gate (hereinafter referred to as a base) serving as a first control terminal of the first differential pair is used as a first reference voltage. And the first base of the second differential pair is connected to a second reference voltage lower than the first reference voltage, and thereafter, the first base of the nth differential pair is the first of the (n−1) th differential pair. The second base connected to the nth reference voltage, lower than the (n-1) th reference voltage to which one base is connected, and serving as a second control terminal from the first differential pair to the nth differential pair Is connected to the control unit. As the output voltage of the unit decreases, the current flows from the first collector or first drain (hereinafter referred to as the first collector), which becomes the first current switching terminal of the first differential pair, to the nth differential pair. The first collector and the second collector serving as the second current switching terminal of the nth differential pair may be changed.

  In the circuit configuration of the second example, unlike the circuit configuration of the first example, the amount of current flowing out of the plurality of current paths can be adjusted. In general, switching the current supply is easier to use, and with this circuit, it is possible to switch such a current supply.

  Contrary to the previous circuit, when the bipolar transistor is used, the control current is one direction in which the current flows from each differential pair to the control unit. When a field effect transistor is used, almost no control current flows.

  Alternatively, a current mirror circuit using a PNP-type bipolar transistor or a P-type field effect transistor with respect to the collector of each differential pair of the current switching circuit of the first example is configured to fold the current flow back to the current flow. May be provided respectively.

  This circuit operates in the same way as the circuit shown in the second example. However, since the current flow-out circuit is composed of a current mirror circuit, only one transistor is placed in the current path, and the current is kept constant. Thus, the upper limit of the voltage that can be supplied can be expanded to near the upper limit of the power supply voltage, and the voltage range in which a constant current can be supplied can be widened. Further, since the current flows from the control unit to the current switching circuit, the control unit can be easily configured.

  The emitter area or gate width of each transistor of the folded current mirror circuit may be different from each other.

  In the variable amplifier, the n amplifiers are variable in gain by changing a current flowing into the amplifier control terminal, and do not operate when a control current to the amplifier control terminal becomes zero. Also good.

  In the variable amplifier, the amplifier control terminals of the n amplifiers are the first current switching terminals from the first differential pair to the nth differential pair, and the second current switching of the nth differential pair. It may be connected to any one of the terminals.

  In the variable amplifier, the characteristic change may be low noise when the gain is high, and linearity may increase as the gain decreases.

  In order to solve the above-described problem, a mobile terminal according to the present invention includes any of the variable amplifiers described above.

  Compared with the conventional circuit, in the current switching circuit of the variable amplifier of the present invention, by using the switching by the differential pair (differential circuit) connected in cascade (cascade), almost no current flows in the control unit, and the current flows. Bipolar operation is not necessary. In the present invention, since the current switching voltage can be uniquely set by the reference voltage, setting is easy and the degree of freedom is high.

  Furthermore, in the present invention, since a silicon bipolar transistor can switch a current at about 100 mV to 200 mV, a large number of switching is possible even at a low power supply voltage of about 3 V as well as a high voltage. A current switching circuit having a path can be easily configured.

  In the variable amplifier according to the present invention, n amplifiers whose characteristics are changed by changing the current flowing into the amplifier control terminal by the current switching circuit are provided by the current switching circuit. Therefore, the characteristics can be changed so as to improve the linearity as the gain decreases.

  As a result, since the variable amplifier of the present invention can change the above characteristics even at a low power supply voltage, it can be suitably used for a variable amplifier such as a portable terminal.

  Each embodiment of the current switching circuit used in the variable amplifier according to the present invention will be described below with reference to FIGS.

(First embodiment)
FIG. 1 is a block diagram showing the contents of the current switching circuit. In the current switching circuit, the three differential pairs D1, D2, and D3 are cascaded so as to be stacked on each other. In this differential pair, the upper two terminals of the block are current switching terminals, the lower terminal is a common terminal connected to a current source, and the left and right terminals are control terminals. When the differential pair is composed of an NPN-type bipolar transistor or an N-type field effect transistor, current is sucked from the current switching terminal on the side of the control terminal having a high voltage with respect to the common terminal. When the differential pair is composed of a PNP-type bipolar transistor or a P-type field effect transistor, current is discharged from the current switching terminal on the side of the control terminal having a low voltage with respect to the common terminal.

  The common terminal of the first differential pair is connected to the constant current circuit CS (constant current unit), and the common terminal of the next differential pair is connected to the second current switching terminal of the previous differential pair. One input of each differential pair is connected to a reference voltage generation circuit (reference voltage unit) RVC.

  The reference voltage generating circuit RVC generates a reference voltage in which voltages are arranged in the order of VR1, VR2, and VR3. The other input of each differential pair D1, D2, D3 is connected to a control voltage generation circuit (control unit) CVC. In general, voltages VS1, VS2, and VS3 output from the control voltage generation circuit CVC are VS = VS1 = VS2 = VS3.

  When the current switching circuit is used, the control voltage VS when switching the current path basically matches the reference voltage. If a simple resistance division is used as the reference voltage generation circuit RVC, each reference voltage for current switching can be easily set only by the resistance value.

  Next, a circuit diagram of a first example of the first embodiment is shown in FIG. In this circuit, an NPN bipolar transistor is used as a differential pair element. The current amplification factor of the bipolar transistor is about 100, and the total current value to be switched is 100 μA. The reference voltage generation circuit RVC is formed by simple resistance division using the resistors indicated by R1 to R5, and the resistance and the base current limiting resistor are all set to 100 kΩ. The power supply voltage is 5V.

  A switching current of 100 μA is converted into a sink current of Q2 by a current mirror circuit composed of Q1 and Q2, and used as a current source. The emitters of both Q3 and Q4 are connected to the collector of Q2, and Q3 and Q4 form a differential pair (differential circuit). The collector (first current switching terminal) of Q3 becomes the first current suction terminal I1-1.

  The collector of Q4 (second current switching terminal) is further connected to each emitter of the differential pair formed by Q5 and Q6, and the collector of Q5 (first current switching terminal) becomes the second current sink terminal I2-1. , Q6's collector (second current switching terminal) is connected to the emitter of the next differential pair.

  In this way, the number of switching circuits can be set freely by cascading the differential pairs so as to be stacked on each other. In this embodiment, five current sink terminals I1-1, I2-1, I3-1, I4-1, and I4-2 are formed by stacking four differential pairs. A current source of 100 μA can be simply formed using an appropriate resistor between VDD and Q1.

  Further, in order to reduce element variation, power supply voltage dependency, and the like, it is desirable to use a circuit using a power supply circuit such as a band gap power supply circuit whose characteristics do not change depending on element characteristics and temperature. Alternatively, in some cases, it is conceivable to use a power supply circuit having such a characteristic that the temperature characteristic of the current switching circuit can be canceled.

  Next, the operation of the above configuration will be described. When the control voltage VS is sufficiently lower than the first reference voltage VR1, the voltage between the base / emitter of Q4 is low, Q4 is turned off, and all current flows from the I1-1 terminal to Q3. As the control voltage VS increases and approaches VR1, which is the first reference voltage, as the voltage between the base and the emitter of Q4 increases, a current also flows into Q4.

  When the control voltage VS coincides with the first reference voltage VR1, the currents flowing through Q3 and Q4 are the same. In this state, since the control voltage VS is sufficiently lower than the second reference voltage VR2, the voltage between the base / emitter of Q6 is low and Q6 is off, so that all the current flowing through Q4 is the I2-1 terminal. To Q5.

  Further, when the control voltage VS rises above the first reference voltage VR1, the current flowing through Q3 gradually decreases, and most of the current flows through Q4. A similar operation occurs for Q5 and Q6, Q7 and Q8, Q9 and Q10, and the current is I1-1 / Q3 → I2-1 / Q5 → I3-1 / Q7 → I4-1 / Q9 → I4-2. / Change to Q10 and switch. A graph showing changes in the current flowing through each transistor is shown in FIG.

  In the present embodiment, the number of switching circuits is five, but it can be freely set from three to several tens. In actual characteristics, as the control voltage VS increases, the total current value slightly decreases. This is because the base current of the transistors forming each differential pair flows from the control voltage generation circuit CVC and the reference voltage generation circuit RVC into Q2, which determines the total current, so that the actual collector current is lower than the set current. This is because the base current increases as the control voltage VS increases while the collector current decreases as the control voltage VS increases.

  R6 to R9 are resistors added to limit the base current. Without this resistance, when the control voltage VS exceeds a certain level, the base current of Q4 coincides with the total current value to be switched, and current cannot be sucked from the outside.

  On the other hand, if the resistance value is too high, the voltage that is actually switched over the setting switching voltage is increased by a voltage drop calculated by the product of the resistance value and the base current. Therefore, it is desirable that the amount of deviation from the allowable switching setting voltage is about a value obtained by dividing the deviation by the base current.

  Further, when the resistance of the reference voltage generation circuit RVC is low, the power consumption of the reference voltage generation circuit RVC increases. When a high resistance value is used, there arise problems that a desired collector current cannot be obtained, or that the actual switching voltage is lowered from the switching voltage determined by the resistance division ratio.

  Fortunately, the direction in which the switching voltage due to the resistance of the reference voltage generation circuit RVC changes is opposite to the direction in which the switching voltage due to the base current limiting resistor changes, so if an appropriate resistance value is selected, the switching voltage error Can be reduced.

  In the example of the current switching circuit of the present embodiment, in the case of this bipolar transistor, a voltage difference of about 350 mV is required in order for the current of the differential pair to be completely switched due to the influence of the emitter resistance. In the case of a silicon bipolar transistor having a sufficiently small emitter resistance, the switching of the differential pair is about 120 mV.

  The means for adjusting the amount of overlap is such that a resistor is inserted between the emitter of the transistor forming the differential pair and the collector of the transistor serving as a current source connected to the emitter of the differential pair, and the transistor of the differential pair is in a Darling connection. The amount of overlap can be increased by means used to reduce the gain of a normal differential amplifier. On the contrary, if a means for increasing the gain is used, the amount of overlap can be reduced.

  In this embodiment, an NPN type bipolar transistor is used. However, if a PNP type bipolar transistor is used, a circuit that switches current discharge (flow) instead of current sink can be formed as a second example.

(Second embodiment)
A circuit diagram of a second embodiment according to the current switching circuit is shown in FIG. In this circuit, an N-type MOSFET is used as an element. All elements have an Idss of about 1 mA. The power supply voltage was 5V. R1 is a resistor for setting the total current. Since 100 μA was required as the current value, the MOSFET used this time was 45.5 kΩ.

  The current is converted into a sink current of Q2 by a current mirror circuit composed of Q1 and Q2, and operates as a current source. In order to improve the temperature characteristics and increase the tolerance against device variations, this portion may be changed to a circuit such as a band gap reference. In the present embodiment, current setting by a simple resistor is used as an example of a simple circuit.

  Q3 and Q4, Q5 and Q6, Q7 and Q8, and Q9 and Q10 form a differential pair. R2 to R6 generate reference voltages VR1, VR2, VR3, and VR4 for the switching signal. All resistance values were 50 kΩ.

  When the switching element is a MOSFET, no current flows into the gate. Therefore, if the current flowing through the reference voltage generation circuit RVC is sufficiently large compared to the gate leakage current, it is preferable to use a resistor as high as possible in the reference voltage generation circuit RVC. Current consumption can be reduced.

  However, in order to obtain a high resistance value, a considerably long resistor is required on the IC and the chip area becomes large. Therefore, in this design, the resistance is set to 50 kΩ. The reference voltage generation circuit RVC uses a simple resistance voltage dividing circuit, but a band gap reference or the like may be used in a portion where high accuracy is required.

  FIG. 5 shows current switching characteristics in this circuit. When the MOSFET is used, unlike the bipolar transistor, the gate current does not flow, so the total switching current is constant regardless of the control voltage VS. In addition, since gm (mutual conductance) is lower than that of the bipolar transistor, the amount of switching overlap is large, and the switching characteristic is gentle.

  The same resistance value is used for each of the resistors R2 to R6 of the reference voltage generation circuit RVC, but the switching voltage can be freely set by changing the resistance value. For example, FIG. 6 shows changes in current when R2 and R4 are 100 kΩ, R3 and R5 are 50 kΩ, and R6 is 200 kΩ. In this case, it can be seen that the control voltage VS having the same current is equal to 1V, 1.5V, 2.5V, and 3V obtained by dividing the power supply voltage of 5V by the resistance divided voltage.

  When the MOSFET of this circuit is used, about 0.5 V is required as an input voltage difference in order for the current of the differential pair to be completely switched. Therefore, there is an overlap of currents flowing in a plurality of current paths. It can be seen quite a bit. By increasing the gm (mutual conductance) of the MOSFET, such as increasing the gate width, the amount of overlap can be reduced. Conversely, the overlap amount can be increased by reducing the gate width. Furthermore, as described in the first embodiment, the overlap amount can be adjusted using means used for adjusting the gain and linearity of the differential amplifier.

  In this embodiment, since the circuit is configured by using an N-type MOSFET, the current to be fed is controlled. However, by using a P-type MOSFET as shown in FIG. It is also possible to configure a current-type switching circuit as a second example.

(Third embodiment)
As the current source that can be switched, the flow-out type is often easier to use than the suction type. As described above, this can be realized by configuring a current switching circuit using a P-type MOSFET or a PNP-type bipolar transistor.

  However, a PNP bipolar transistor on an IC is generally manufactured in a horizontal type, and characteristics such as a current amplification factor are often extremely poor. Also, as the number of differential pairs increases, the voltage between the source / drain of the MOSFET and the emitter / collector of the bipolar transistor increases, so that the maximum voltage that operates as a constant current source decreases and the characteristics deteriorate. Will be invited.

  Therefore, as shown in FIG. 8, in the third embodiment, the current switching circuit is made of an N-type MOSFET, and the current suction is turned back by a current mirror circuit composed of a P-type MOSFET. Is used. As a result, the maximum voltage that operates as a constant current source does not change depending on the current path, and desired characteristics can be obtained at a voltage equal to or lower than the voltage obtained by subtracting the minimum source / drain voltage at which the P-type MOSFET can operate as a constant current source from VDD. it can. The current switching characteristics are shown in FIG.

  Furthermore, in the current mirror circuit of the P-type MOSFET, the current values of the individual paths can be changed by changing the gate widths of the MOSFETs to each other (partially, the gate widths may be the same). In the circuits so far, the total current is a constant value, and differs slightly depending on the overlap, but basically the current flowing through each circuit path is constant. However, depending on the circuit, it may be better to set a different current depending on the path. In that case, the path current can be freely set by setting the gate width of the P-type MOSFET used in this mirror circuit to an appropriate value. Further, when a PNP bipolar transistor is used for the folded current mirror circuit, the same effect can be obtained by changing the emitter areas of the bipolar transistors.

  In this example, MOSFETs are used for the current switching circuit and the folding circuit, but it is also possible to make a circuit using bipolar transistors for either or both of them. However, in that case, unlike the MOSFET, it is necessary to consider an error due to the base current. However, compared to the lowest source / drain voltage that generally operates as a constant current source of a MOSFET, the bipolar transistor has a lower emitter / collector voltage, and the early voltage, which is the collector voltage dependency of the collector current, is also higher in the bipolar transistor. Since it is high, the bipolar transistor is superior in the constant current characteristic. Therefore, an optimal combination may be considered according to the application.

(Fourth embodiment)
By combining the current switching circuit with a plurality of variable gain amplifiers and passive attenuators, a variable gain variable amplifier according to the present invention capable of realizing high linearity at low gain can be configured. A block diagram showing the configuration is shown in FIG.

  A1 to A4 are amplifiers capable of changing the gain by changing the current flowing into the control terminal. The amplifier does not operate at all when the control current becomes zero. The amplifiers A1 to A4 have a certain linearity during operation and do not generate any distortion when not operating completely.

  As the current switching circuit indicated by C1, a circuit as shown in FIG. 8 capable of discharging current can be used. In addition, other circuits such as the circuit shown in FIG. 7 or a circuit using a PNP bipolar transistor can be used.

  Next, the operation will be described. When the control current is controlled and the amplifier A1 is operating, the other amplifiers are off. When an input conversion distortion (IIP3, 3rd Order Input Intermodulation Intercept Point) is considered as a linearity parameter, IIP3 in this state is equal to that of the amplifier A1.

  When the control current is changed and the amplification operation completely shifts from the amplifier A1 to the amplifier A2, the gain of the entire amplifier is obtained by subtracting the attenuation factor of the attenuator B1 from the gain of the amplifier A2. IIP3 in this operation state is obtained by adding the attenuation factor of the attenuator B1 to the IIP3 of the amplifier A2, and the IIP3 of the entire amplifier increases.

  In this way, the gain becomes the smallest when the amplifier A4 is operating. The gain at that time is obtained by subtracting the sum of the attenuation factors of the attenuators B1, B2, and B3 from the gain of the amplifier A4. It becomes. The IIP3 at that time is obtained by adding the sum of the attenuation factors of the attenuators B1, B2, and B3 to the IIP3 of the amplifier A4, and can greatly improve the linearity compared to the single amplifier A1.

  The variable gain variable amplifier made in this way has low noise when gain is high, and can improve linearity as gain decreases. Further, the circuit format related to the variable amplifier can sufficiently operate even with a power supply voltage of about 3 V that can be generated by a lithium battery or the like.

  A wireless portable terminal requires a wide dynamic range in reception characteristics because reception power changes greatly in the surrounding environment.

  In such a portable terminal, when the reception power is strong, the C / N ratio, which is the ratio of signal to noise, can be obtained sufficiently, so the noise characteristics and high gain of the reception unit of the portable terminal are not very important, The linearity that no distortion occurs in the receiving unit due to a strong received signal is important. Further, when the received power is small, distortion does not occur in the receiving unit, but since the C / N ratio is not sufficient, the amplifier in the receiving unit must have high gain and low noise.

  By applying the variable amplifier of the present invention to such a portable terminal, excellent reception characteristics can be obtained in the portable terminal.

  Since the current switching circuit of the variable amplifier of the present invention can easily configure a switching circuit having a large number of switching paths even with a low power supply voltage of about 3 V, the variable amplifier can be used as an amplifier for a battery-driven portable terminal or the like. It can be suitably used.

1 is a circuit block diagram of a first embodiment according to a current switching circuit used in a variable amplifier of the present invention. It is a circuit diagram of the first embodiment using a bipolar transistor. It is a graph which shows the current switching characteristic of the current switching circuit shown in FIG. FIG. 5 is a circuit diagram of a second embodiment according to the current switching circuit using an N-type MOSFET. It is a graph which shows the current switching characteristic of the current switching circuit shown in FIG. 5 is a graph showing current switching characteristics when the resistance value of a reference voltage setting resistor is changed in the current switching circuit shown in FIG. 4. It is a circuit diagram of a modification in the second embodiment according to the present invention using a P-type MOSFET. FIG. 6 is a circuit diagram of a third embodiment of the current switching circuit in which a current switching circuit using an N-type MOSFET is used to return a current using a P-type MOSFET. It is a graph which shows the current switching characteristic of the circuit shown by FIG. FIG. 6 is a circuit block diagram showing, as a fourth embodiment, an example of a variable gain variable gain amplifier according to the present invention, which is obtained by connecting the current switching circuit and an amplifier having different gains to each other. It is a circuit diagram of an example of the conventional current switching circuit. 12 is a graph showing a control voltage characteristic of a base voltage of the circuit shown in FIG. 12 is a graph showing current switching characteristics of the circuit shown in FIG. 12 is a graph showing current switching characteristics when the number of switching current paths is eight in the circuit shown in FIG.

Explanation of symbols

RVC: reference voltage generation circuit CVC: control voltage generation circuits D1, D2, D3, D4: differential pair CS: constant current source (constant current unit)
VR1, VR2, VR3: Reference voltages VS, VS1, VS2, VS3, VS4: Control voltages I1-1, I2-1, I3-1, I3-2, I4-1, I4-2, I1, I2, I3, I4, I5, I6, I7, I8: Switched current terminals VCTRL, VCTRL1, VCTRL2: Control voltage generation circuits Vb1, Vb2, Vb3, Vb4: Base terminals A1, A2, A3, A4 of the respective transistors: Amplifiers B1, B2, B3: Attenuator C1: Current switching circuit

Claims (1)

A differential pair having two control terminals, two current switching terminals and one common terminal, and a constant current section;
The common terminal of the first differential pair is connected to the constant current portion, the common terminal of the second differential pair is connected to the second current switching terminal of the first differential pair, and thereafter the common of the nth differential pair The terminal is connected to the second current switching terminal of the (n-1) th differential pair (n is an integer of 2 or more);
A reference voltage unit that supplies reference voltages having different potentials to the first control terminals from the first differential pair to the nth differential pair is provided,
The control unit that changes the control voltage applied to the second control terminal from the first differential pair to the nth differential pair is configured to switch each first current from the first differential pair to the nth differential pair. A current switching circuit provided to switch a current between the terminal and the second current switching terminal of the nth differential pair;
N amplifiers having amplifier control terminals,
A variable amplifier, wherein characteristics are changed by changing currents flowing into amplifier control terminals of the n amplifiers by current switching circuits, respectively.

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