JP2005312773A - Ultrasonic diagnostic apparatus - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide an ultrasonic diagnostic apparatus having a wall filter estimating an instantaneous phase change between data constituting clutter components and performing a processing equivalent or approximate to a processing correcting the phase change of Doppler signals according to the instantaneous phase change amount by a simple circuit method without a feedback system. <P>SOLUTION: This ultrasonic diagnostic apparatus is provided with a transmitting/receiving means transmitting ultrasonic signals in the interior of a subject in respective scanning line directions a plurality of times respectively and receiving ultrasonic echo signals reflected from the subject, a means of obtaining IQ signals from the receiving signals of the transmitting/receiving means, a means 52 of obtaining amplification signals from the IQ signals, and means 50A, 50B and 51 removing low frequency components from the plurality of IQ signals and the amplification signals reflected from the same positions respectively in the respective scanning line directions according to different characteristics, respectively. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to an ultrasonic diagnostic apparatus that transmits and receives ultrasonic waves to a subject and visualizes blood flow in the subject two-dimensionally using Doppler signals, and in particular, effectively eliminates motion artifacts due to tissue movement. The present invention relates to an ultrasonic diagnostic apparatus including a wall filter for removal.

  Ultrasound color Doppler method is a technique that uses the Doppler effect of ultrasound signals to obtain information on blood flow dynamics within a subject non-invasively. Has made remarkable progress.

  As one type of this ultrasonic diagnostic apparatus, an apparatus that performs color Doppler tomography (also referred to as color flow mapping: CFM) is known. The color Doppler tomography is based on MTI (moving target indicating device) technology used in the radar field. According to the color Doppler tomography, a two-dimensional distribution image of the blood flow velocity on the tomographic plane in the subject is obtained. Can be obtained.

  In order to obtain this blood flow velocity distribution image, the ultrasonic echo obtained from the subject accompanying the transmission of the ultrasonic pulse is converted into a corresponding electrical signal, and then the real part (I signal), imaginary part (Q signal) echo signals. Both the real part and imaginary part echo signals are phase-detected with respect to the reference signal by a quadrature detector and extracted as a Doppler signal representing a phase change from the reference signal. The real part and imaginary part Doppler signals are each converted into a digital signal by an A / D converter and then temporarily stored in a buffer memory.

  In the CFM mode in which CFM is performed, transmission and reception of ultrasonic pulses are repeated a plurality of N times (for example, 16 times) in the same scanning line direction. For this reason, the digital amount of Doppler data necessary for reconstructing one image is a three-dimensional structure consisting of a first dimension, a second dimension, and a third dimension for each of the real part and imaginary part signals. This becomes data and is stored in the buffer memory. The first dimension represents the number (number) of each scan line, the second dimension represents the number of samples (number) in the depth direction along each scan line, and the third dimension represents transmission / reception for each sample point. This represents the number of Doppler data obtained by repetition (referred to as an ensemble number) (number).

  For this reason, when focusing on the same pixel position in the scanning section, received echoes are obtained in time series by transmitting and receiving N ultrasonic pulses, and the digital data phase-detected based on the received echoes is in the third dimension. They are arranged sequentially in the direction. The speed of change of the signal when viewed in the direction of the third dimension corresponds to the magnitude of the Doppler shift frequency, that is, the magnitude of the moving speed of the object.

  The three-dimensional digital data (Doppler signal) formed in the buffer memory in this way has a clutter component (an echo signal from a tissue such as a real organ) for each data string in the third dimension direction of each sample point. Removed by wall filter. This filtering principle is as follows.

  In the received echo, an echo signal from a moving body that moves at a certain speed or more like a blood cell and a clutter component that is an echo signal from a tissue such as a real organ are mixed. The clutter component is larger than the echo signal from the bloodstream with respect to the signal intensity, but the echo signal from the bloodstream is larger than the clutter component with respect to the moving speed. For this reason, the filter circuit of the wall filter is configured as a high pass filter (HPF: High Pass Filter), and the cut-off frequency is set to a value capable of removing the clutter component. Thereby, the clutter component is removed from the detected Doppler signal, and an echo signal from the blood flow is extracted.

  The echo signal from the blood flow is then subjected to an estimation process of the motion state (blood flow velocity, power, variance, etc.) of the blood flow, and a two-dimensional image is created based on the estimated information.

  Although the wall filter is used to remove the clutter component in this way, the actual organ actually moves slightly or may move due to various causes. The echo signal from the bloodstream cannot be clearly distinguished, and the conventional wall filter has a problem that the clutter component cannot be accurately and sufficiently removed sufficiently.

  In order to solve this problem, a method of changing the characteristics of the wall filter based on an instantaneous phase change between data forming a clutter component has been proposed (see, for example, Patent Document 1). That is, first, an instantaneous phase change between data constituting a clutter component is estimated, and the phase change of the Doppler signal is corrected according to the instantaneous phase change amount. Then, a signal obtained by subtracting a fixed value from the phase-corrected Doppler signal is obtained, and the characteristics of the wall filter are changed according to the obtained signal. For this reason, in the method of changing the characteristics of the wall filter, the wall filter is provided with two paths for the data system and the feedback system.

  Furthermore, a method for changing the characteristics of the wall filter by applying a low pass filter (LPF) to the ensemble sequence before estimating the instantaneous phase change of the clutter component is proposed (for example, Patent Document 2). reference). In this method, an independent path is provided in the wall filter for a signal for changing the characteristics of the wall filter, and a total of three paths including a data system and two feedback systems are provided.

  FIG. 10 is a configuration diagram of a wall filter having two paths for a conventional data system and a feedback system, and applying LPF to an ensemble string.

  In the conventional wall filter 1 shown in FIG. 10, the I signal and the Q signal are respectively input to the data complex multiplier 1a and the feedback LPF 1b provided for removing the clutter component. The I and Q signals input to the feedback system are supplied to the clutter phase change estimator 1c after the signal component having a large fluctuation is removed in the LPF 1b.

  The clutter phase change amount estimator 1c performs an operation on the I signal and the Q signal, estimates an instantaneous phase change amount of the clutter component, generates an estimated signal, and supplies the estimated signal to the multiplication signal generator 1d. The multiplication signal generator 1d generates a complex multiplication signal using the phase change amount estimation signal and supplies the complex multiplication signal to the complex multiplier 1a.

  The complex multiplier 1a performs complex multiplication with the complex multiplication signal generated by the multiplication signal generator 1d and the I signal and Q signal directly input to the data system, and the obtained Doppler signals of the real part and the imaginary part are obtained. Is given to the constant value subtractor 1e. The constant value subtractor 1e performs constant value subtraction on the real part and imaginary part Doppler signals to remove the clutter component and give it to the HPF 1f, and the clutter component remaining after high-pass processing by the HPF 1f is more reliably removed. Is done. Further, the Doppler signal from which the clutter component has been removed is supplied to the complex multiplier 1g.

  That is, the amplitude value at the first phase is regarded as the amplitude value of the clutter component and is uniformly subtracted from the Doppler signal from which the instantaneous phase change of the clutter component is canceled.

  On the other hand, the complex multiplication signal generated by the multiplication signal generator 1d is also supplied to the phase inverter 1h. The phase inverter 1h performs a phase inversion operation on the complex multiplication signal to generate a complex signal having a phase correction amount in an opposite phase, and supplies the complex signal to the complex multiplier 1g. Then, in the complex multiplier 1g, the Doppler signal from which the clutter component has been removed by the HPF 1f is multiplied by the complex signal having the phase correction amount received from the phase inverter 1h in the opposite phase.

  Thereby, in the complex multiplier 1g, the influence of the phase correction on the clutter component is canceled, and the complex multiplier 1g can convert the signal into a signal that can obtain a value such as a blood flow velocity for the ultrasonic probe. The I signal and the Q signal obtained in the complex multiplier 1g are given to a speed / dispersion / power estimation circuit (not shown) and used for calculation of various blood flow information.

The output signal of the constant value subtractor 1e is also given to the clutter information detector 1i, and the clutter information detector 1i calculates a feature amount indicating the degree of clutter components that may remain in the Doppler signal. The The feature quantity which is the output of the clutter information detector 1i is given to the filter characteristic setter 1j, and the cutoff characteristic of the HPF 1f is controlled by the filter characteristic setter 1j according to the feature quantity.
JP-A-10-99333 JP 11-267125 A

  In the conventional ultrasonic diagnostic apparatus, the signal system in the wall filter is multiplexed, and further, since a complex multiplier for phase correction is provided, there is a problem that the circuit scale becomes large. Further, in the wall filter, there is a problem that the control of the data system (data synchronization) becomes complicated due to the presence of the feedback system.

  Furthermore, while the phase correction process is uniquely determined, how to specifically change the filter coefficient of the wall filter (specifically HPF) depends on the target organ, depth, flow velocity range, It is necessary to change according to conditions such as the number of ensembles. However, it is difficult to adjust the filter coefficient of the wall filter so as to be optimal.

  For example, if the cutoff frequency of the wall filter is increased too much, the blood flow component signal is removed. Conversely, if the cutoff frequency is decreased too much, the clutter component signal remains. If the signal of the blood flow component is removed, the diagnosis is hindered, so that the cut-off frequency of the wall filter cannot necessarily be changed greatly. Therefore, in short, it can be said that it is better not to change the cutoff frequency of the wall filter adaptively.

  The present invention has been made in order to cope with such a conventional situation, and estimates an instantaneous phase change between data forming a clutter component, and changes the phase change of a Doppler signal according to the instantaneous phase change amount. It is an object of the present invention to provide an ultrasonic diagnostic apparatus including a wall filter capable of performing a process equivalent to or approximate to the process to be corrected by a simple circuit system without a feedback system.

  In order to achieve the above object, an ultrasonic diagnostic apparatus according to the present invention transmits an ultrasonic signal into a subject a plurality of times in the direction of each scanning line, and the subject as described in claim 1. Transmitting / receiving means for receiving an ultrasonic echo signal reflected from the receiving means; means for obtaining an IQ signal from the received signal of the transmitting / receiving means; means for obtaining an amplitude signal from the IQ signal; and the same position in each scanning line direction And a means for removing a low-frequency component from each of the plurality of IQ signals and amplitude signals reflected from each other by different characteristics.

  Further, in order to achieve the above-described object, the ultrasonic diagnostic apparatus according to the present invention transmits an ultrasonic signal into a subject a plurality of times in each scanning line direction and includes Transmitting / receiving means for receiving an ultrasonic echo signal reflected from a subject; means for obtaining an IQ signal from the received signal of the transmitting / receiving means; means for obtaining an amplitude signal from the IQ signal; and each scanning line direction And a means for removing a low-frequency component from a plurality of amplitude signals reflected from the same position.

  In addition, in order to achieve the above-described object, the ultrasonic diagnostic apparatus according to the present invention transmits an ultrasonic signal into a subject a plurality of times in each scanning line direction as described in claim 7 and A transmission / reception unit that receives an ultrasonic echo signal reflected from a subject, a unit that obtains an IQ signal from the reception signal of the transmission / reception unit, an amplitude acquirer that obtains an amplitude signal from the IQ signal, and the IQ signal While reducing the signal strength of the low frequency component and the medium frequency component, the first high pass filter that passes the high frequency component of the IQ signal, and the low frequency component and frequency of the amplitude signal are A second high-pass filter that reduces the signal strength of the high component and passes the medium-frequency component of the amplitude signal is provided.

  In the ultrasonic diagnostic apparatus according to the present invention, an instantaneous phase change between data forming a clutter component is estimated with respect to a Doppler signal obtained by transmission and reception of ultrasonic waves, and the instantaneous phase change amount is determined. A process equivalent or approximate to the process of correcting the phase change of the Doppler signal can be performed by a simple circuit system without a feedback system.

  Embodiments of an ultrasonic diagnostic apparatus according to the present invention will be described with reference to the accompanying drawings.

  FIG. 1 is a configuration diagram showing a first embodiment of an ultrasonic diagnostic apparatus according to the present invention.

  The ultrasonic diagnostic apparatus shown in FIG. 1 includes an ultrasonic probe 1 capable of bidirectional signal conversion between an ultrasonic signal and an electric signal, a transmission system circuit 2 connected to the ultrasonic probe 1, and a reception / processing system. Circuit 3.

  The ultrasonic probe 1 includes an array-type piezoelectric vibrator disposed at the tip thereof. The array-type vibrator has a plurality of piezoelectric elements arranged in parallel, and the direction of the arrangement is a scanning direction. Each of the plurality of piezoelectric elements forms a transmission / reception channel.

  The transmission system circuit 2 includes a pulse generator 11 that generates a reference rate pulse, and a transmission circuit 12 that generates a drive pulse by delaying the reference rate pulse output from the pulse generator 11 for each channel. The drive pulse for each channel output from the transmission circuit 12 is supplied to each of the plurality of transducers of the ultrasonic probe 1. The transmission delay time of the drive pulse is controlled for each channel and is repeatedly supplied for each rate frequency. In response to the supply of the drive pulse, an ultrasonic pulse is emitted from each transducer. This ultrasonic pulse forms a transmission beam with a controlled transmission delay time while propagating in the subject, and a part of the ultrasonic pulse is reflected at an interface having different acoustic impedances to become an echo signal. Part or all of the returned echo signal is received by the transducer and converted into a corresponding electrical signal.

  On the other hand, the reception / processing system circuit 3 includes a reception circuit 21 connected to the ultrasonic probe 1, a B mode processing circuit 22, a CFM mode processing circuit 23, and a display circuit placed on the output side of the reception circuit 21. 24. The reception circuit 21 includes a preamplifier for each channel connected to the transducer of the probe 1, a delay circuit connected to each of the preamplifiers, and an adder that adds the delay outputs of the delay circuits. For this reason, the echo signal received by the probe 1 is subjected to delay control for reception focus and added after the analog signal of the corresponding electric quantity is taken into the receiving circuit 21 and amplified for each channel. As a result, a reception beam having a focus point determined according to the control of the reception delay time is formed in calculation, and desired directivity is obtained.

  The output terminal of the receiving circuit 21 is branched and connected to a B mode processing circuit 22 and a CFM mode processing circuit 23. The B-mode processing circuit 22 is responsible for creating B-mode black and white tomographic image data, and includes a logarithmic amplifier, an envelope detector, and an A / D converter (not shown). Therefore, the echo signal phased and added by the receiving circuit 21 is logarithmically amplified by the logarithmic amplifier, the envelope of the amplified signal is detected by the envelope detector, and further converted to a digital signal by the A / D converter. Is sent to the display system circuit 24 as a B-mode image signal.

  The display circuit 24 includes a digital scan converter (DSC) 31 having a frame memory for B mode and a CFM and a write / read control circuit, a color processor 32 for performing pixel color application processing, a D / A converter 33, And a TV monitor 34 for display. The digital envelope detection signal output from the B-mode processing circuit 22 is written in the B-mode frame memory of the DSC 31.

  Further, the CFM processing circuit 23 is a circuit group responsible for creating CFM mode image data for observing blood flow dynamics. The input side of the CFM processing circuit 23 converts the echo signal output from the receiving circuit 21 into a real part Q and an imaginary part I. Correspondingly, it is branched to input in two systems. For each signal system of the real part Q and the imaginary part I, a mixer 41A (41B), an LPF 42A (42B), and an A / D converter 43A (43B) are provided in this order. The CFM processing circuit 23 further includes buffer memories 44A and 44B that temporarily store processing signals of the real part and imaginary part from the A / D converters 43A and 43B, a wall filter 45 that performs filtering processing based on the stored signal, and A speed / dispersion / power estimation circuit 46 for performing various calculations related to blood flow dynamics based on the output of the filter unit is provided. The CFM processing circuit 23 further includes a reference oscillator 47 that oscillates a reference signal for reference, and a phase shifter 48 that gives a phase difference of exactly 90 degrees to the reference signal and supplies the phase difference to the mixers 41A and 41B. The reference oscillator 47 and the pulse generator 21 of the transmission system circuit 2 are driven in synchronization with each other. The reference signal has substantially the same frequency as the ultrasonic signal.

  For this reason, the echo signal output from the receiving circuit 21 is multiplied with the reference signal by the mixer 41A (41B) in each of the signal system of the real part and the imaginary part, and then the high-frequency component is output by the LPF 42A (42B). Are removed to provide a baseband signal. That is, the echo signal is subjected to phase detection (quadrature phase detection) by the mixer 41A (41B) and the LPF 42A (42B) for each real part and imaginary part, and as a baseband Doppler signal reflecting the phase difference from the reference signal Extracted. The Doppler signal is converted into digital data by the A / D converter 43A (43B) for each real part and imaginary part, and is temporarily stored in the buffer memory 44A (44B).

  The wall filter 45 is inserted in order to remove unnecessary echo signals reflected from the heart wall or the like using Doppler data groups individually stored in the buffer memories 44A and 44B. The processing of the wall filter 45 achieves the filtering technique according to the present invention. The specific configuration is as shown in FIG. 3, and the processing will be described later. By the wall filter 45, the Doppler component (referred to as clutter component) from the real organ is reliably and accurately removed from the entire Doppler signal, and only the Doppler component from the bloodstream is extracted.

  The real part and imaginary part Doppler data filtered by the wall filter 45 are sent to the speed / dispersion / power estimation circuit 46, respectively. The velocity / dispersion / power estimation circuit 46 estimates blood flow dynamics information using real part and imaginary part Doppler data, for example, an autocorrelator and an average speed calculator, dispersion calculator, power using the correlation result. An arithmetic unit is included to estimate and calculate information such as the average velocity of blood flow, the distribution of velocity distribution, and the power of reflected signals from the blood flow. The calculation result is temporarily stored in the CFM frame memory of the DSC 31 as CFM mode image data.

  In the DSC 31, the image data stored in the B-mode frame memory and the CFM-mode frame memory are read out separately by the standard TV system. Further, in parallel with this reading, one of the common pixels of both frame memories is alternatively selected to form one frame of image data in which the CFM mode image is superimposed on the B mode image (background image). . The image data is subjected to color application processing by the color processor 32, and then converted to an analog signal at a predetermined timing by the D / A converter and displayed on the TV monitor 34. As a result, a two-dimensional color image of blood flow velocity is displayed against a black and white B-mode image.

  Subsequently, the configuration and operation of the wall filter 45 will be described together with the description of the filtering principle of the present invention.

  FIG. 2 is a diagram showing base band digital Doppler data stored in each of the buffer memories 44A and 44B shown in FIG.

  In order to create an image in the CFM mode, transmission / reception of ultrasonic pulses in the same scanning line direction is repeated N times (for example, 16 times). Based on the echo signal obtained every transmission and reception, the Doppler data subjected to quadrature detection is stored in the buffer memories 44A and 44B, respectively. Therefore, as shown in FIG. 2, the baseband digital Doppler data DD stored in each of the buffer memories 44A and 44B is three-dimensional. The first dimension represents the number of scan lines (numbers) 1-L, the second dimension represents the number of samples (numbers) 1-M in the depth direction along each scan line, and the third dimension The numbers (numbers) 1 to N of Doppler data DD obtained by repetition of transmission and reception for each sample point are represented. The number of Doppler data DD is called the ensemble number as described above. In the CFM mode, as shown by the hatched portion in FIG. 2, N pieces of Doppler data DD obtained in time series at each sample point are independently processed to obtain blood flow dynamic information for each sample point.

  For this reason, the wall filter 45 receives a digital Doppler signal (IQ signal) f (k) (k = 1, 2,..., N) of the real part Q and the imaginary part I from the buffer memories 44A and 44B. Supplied for each position on the scanning plane.

  The operational characteristics of the wall filter 45 are summarized in the following three. That is, (1) a signal having a low frequency component among the IQ signal and the amplitude signal reduces the signal strength. (2) For medium frequency components, the signal strength of the IQ signal is reduced, while the amplitude signal is mainly output. (3) For high frequency components, the signal strength of the amplitude signal is reduced, while the IQ signal is mainly output. Here, reducing the signal strength includes making the signal strength zero, that is, cutting the signal.

  FIG. 3 is a block diagram showing detailed configurations of the wall filter 45 and the speed / dispersion / power estimation circuit 46 shown in FIG.

  The wall filter 45 obtains amplitude as a means for obtaining an amplitude signal from the first HPF 50A, 50B for the two I signals and the second HPF 51, IQB having different characteristics from the first HPF 50A, 50B. A circuit 52 and an adder circuit 53 are included. The I signal guided to the wall filter 45 is branched and supplied to the first HPF 50A for I signal and the amplitude acquisition circuit 52. Similarly, the Q signal guided to the wall filter 45 is also branched and supplied to the first HPF 50B for Q signal and the amplitude acquisition circuit 52.

The amplitude acquisition circuit 52 calculates the amplitude from each of the I signal and the Q signal and supplies the amplitude to the second HPF 51 as an amplitude signal. That is, if the data sequence in the ensemble direction of the IQ signal is f (k) (k = 1, 2,..., N) as shown in Expression (1), A (k) is acquired as an amplitude signal, It has a function given to the second HPF 51.
[Equation 1]
f (k) = I (k) + jQ (k) = A (k) exp {jφ (k)} (1)
However, k = 1, 2,..., N.

  Each of the first and second HPFs 50A, 50B and 51 obtains a fitting signal by polynomial fitting a data string in the ensemble direction by the least square method, and subtracts the fitting signal from the original signal. A system that performs the operation is used. This method is described in A.I. P. G. Hoeks et al. In 1991 in the US Ultrasonic Imaging magazine No. 13 “An Effective Algorithm to Remove Low Frequency Doppler Designed by the Sonic Device” It is a method.

  Polynomial fitting by the least square method is equivalent to the result of applying a filter on the orthogonal polynomial space and inversely transforming it, so that HPF operation and LPF operation by the least square method can be performed by matrix operation. That is, an orthogonal matrix such as an orthogonal polynomial transformation is used to define an orthonormal matrix U and a diagonal matrix D, and an N-row N-column filter matrix W is defined by equation (2-1). An HPF operation or an LPF operation can be performed by performing a matrix operation as shown in Expression (2-2), where y is a column vector.

[Equation 2]
W = U ⁻¹ DU (2-1)
y = Wx (2-2)
However, in Equation (2-1), a matrix that performs orthogonal transformation other than orthogonal polynomial transformation can be used as the matrix U.

  For this reason, the first and second HPFs 50A, 50B, 51 perform a matrix operation on the input signal using a linear filter of a real coefficient shown in Expression (2-2) by the filter matrix W and output the result. . That is, the first HPF 50A for the I signal performs the matrix operation shown in the equation (2-2) on the data string vector x in the ensemble direction of the I signal to obtain the LPF or HPF output signal y by the least square method. On the other hand, the first HPF 50B for the Q signal performs the matrix operation represented by the equation (2-2) on the data string vector x in the ensemble direction of the Q signal, while having the function to be obtained and given to the adding circuit 53. It has a function of obtaining an LPF or HPF output signal y by the square method and outputting it to the speed / dispersion / power estimation circuit 46.

  In addition, the second HPF 51 performs a matrix operation shown in Expression (2-2) on the data string vector x in the ensemble direction of the amplitude signal A (k) received from the amplitude acquisition circuit 52 and performs the least square method. It has a function of obtaining the LPF or HPF output signal y and giving it to the adding circuit 53.

  Here, the characteristic of the linear filter is determined by the diagonal elements of the diagonal matrix D as shown in Expression (2-1). That is, when m is set to 1 from the smaller diagonal element of the diagonal matrix D and the rest is set to 0, the linear filter has a characteristic as an LPF, and m is set to 0 from the smaller diagonal element of the diagonal matrix D. When the remaining is set to 1, the linear filter has a characteristic as an HPF. Hereinafter, a configuration example of the first HPFs 50A and 50B and the second HPF 51 when the number of ensembles N = 16 is shown. The diagonal element hpf1 of the diagonal matrix D that determines the characteristics of the first HPFs 50A and 50B is expressed by Equation (3-1), and the diagonal element hpf2 of the diagonal matrix D that determines the characteristics of the second HPF 51 is The value shown in Formula (3-2) is set.

[Equation 3]
hpf1 = [0,0,0,0,0,0,0,0,1,1,1,1,1,1,1,1]
... (3-1)
hpf2 = [0,0,0,0,1,1,1,1,0,0,0,0,0,0,0,0]
... (3-2)

  That is, in the first and second HPFs 50A, 50B, 51, the coefficient of the fourth or lower basis is set to 0 in the orthogonal polynomial space, and the low frequency component IQ signal and amplitude signal are respectively cut. The coefficients of the fifth to eighth bases are 0 in the first HPFs 50A and 50B, and 1 in the second HPF 51. For this reason, among the IQ signals output from the first HPFs 50A and 50B, the medium frequency IQ signal is cut, while the IQ amplitude signal output from the second HPF 51 has a medium frequency. However, it is output without being cut. Further, the ninth and subsequent basis coefficients are set to 1 in the first HPFs 50A and 50B, and set to 0 in the second HPF 51. For this reason, among the IQ signals that are the outputs of the first HPFs 50A and 50B, the high-frequency IQ signal is output without being cut, whereas the frequency of the IQ amplitude signal that is the output of the second HPF 51 is high. The component amplitude signal is cut.

  That is, when defining the filter matrices W of the first and second HPFs 50A, 50B, and 51, respectively, by setting the diagonal elements hpf1 and hpf2 of the diagonal matrix D, the above-mentioned “(1) lower frequency among Doppler signals (2) The amplitude signal of the IQ signal is output for the intermediate frequency component of the Doppler signal, and (3) the IQ signal is output for the high frequency component of the Doppler signal. Features can be imparted to the wall filter 45. The first and second HPFs 50A, 50B, and 51 function as means for removing low frequency components from the IQ signal and the amplitude signal by different characteristics.

  FIG. 4 is a diagram showing an example of the circuit configuration and calculation method of the first and second HPFs 50A, 50B, 51 shown in FIG.

The first and second HPFs 50 </ b> A, 50 </ b> B, 51 include a coefficient generator 70, N multipliers 71, and an adder circuit 72. The coefficient generator 70 generates N elements a _k1 , a _k2 ,..., A _kN in the k-th row of the N-by-N filter matrix W, and individually supplies them to each multiplier 71. Each element x (1), x (2),..., X (N) of the data string vector x in the ensemble direction is individually input to each multiplier 71, and the elements a _k1 , a _k2 ,. • The product with a _kN is calculated. The multiplication results obtained by the multipliers 71 are given to the adding circuit 72 and added. As a result, the element y (k) in the k-th row of the HPF output signal y is obtained.

Further, a similar operation is performed for each row (k = 1, 2,..., N), and an N × N matrix operation shown in Expression (4) is performed.

  Further, the adder circuit 53 of the wall filter 45 adds the I signal received from the first HPF 50A for I signal and the amplitude signal received from the second HPF 51 to the speed / dispersion / power estimation circuit 46 to add the I signal. It has a function to give as a signal.

  On the other hand, the speed / dispersion / power estimation circuit 46 includes a power arithmetic circuit 54, a power integrator 55, a pulse pair arithmetic circuit 56, a PP (pulse pair) integrator 57, a LOG compression circuit 58, a phase detector 59, and a dispersion estimator 60. Prepare.

  The power calculation circuit 54 uses the I signal received from the adder circuit 53 and the Q signal received from the first HPF 50B for the Q signal to perform the calculation shown in the equation (5), and P0 which is each power signal in the ensemble direction. (K) is obtained and provided to the power integrator 55.

[Equation 5]
P0 (k) = {I (k)} ² + {Q (k)} ² (5)

  The power integrator 55 obtains the power signal P1 by adding the power signals P0 (k) in the ensemble direction received from the power calculation circuit 54 as shown in the equation (6), and the obtained power signal P1. It has a function to be given to the LOG compression circuit 58.

[Equation 6]
P1 = P0 (1) + P (2) + ... + P (N) (6)

  The LOG compression circuit 58 performs a calculation shown in Expression (7) on the power signal P1 received from the power integrator 55 to estimate the blood flow power signal P, and variance estimation of the estimated blood flow power signal P It has the function to output to the device 60 and the subsequent DSC 31.

[Equation 7]
P = 10 × log (P1) (7)

  The pulse pair calculation circuit 56 performs the calculation shown in Expression (8) using the I signal received from the adder circuit 53 and the Q signal received from the first HPF 50B for the Q signal, and is each pulse pair signal in the ensemble direction. It has a function of obtaining pp (k) and giving it to the PP integrator 57.

[Equation 8]
pp (k) = conj {IQ (k−1)} IQ (k) (8)
Here, conj (x) is a function for calculating a conjugate complex number of x.

  The PP integrator 57 adds the pulse pair signals pp (k) in the ensemble direction received from the pulse pair calculation circuit 56 as shown in Expression (9), and obtains the output signal ac and the obtained output signal ac. The phase detector 59 and the variance estimator 60 have a function to be given.

[Equation 9]
ac = pp (2) + pp (3) +... + pp (N) (9)

  The phase detector 59 obtains the blood flow velocity signal V by performing the calculation shown in Expression (10) using the output signal ac of the PP integrator 57, and outputs the obtained blood flow velocity signal V to the subsequent DSC 31. It has the function to do.

[Equation 10]
V = atan {image (ac) / real (ac)} (10)

  The variance estimator 60 has a function of obtaining the blood flow variance signal T by the equation (11) using the power signal P received from the LOG compression circuit 58 and the output signal ac of the PP integrator 57 and outputting the blood flow variance signal T to the subsequent DSC 31. .

[Equation 11]
T = 1− | ac | / P (11)

  Next, the operation of the wall filter 45 and the velocity / dispersion / power estimation circuit 46 thus configured will be described.

  When the IQ signal stored in each of the buffer memories 44A and 44B shown in FIG. 1 is input to the wall filter 45, the I signal branches to the first HPF 50A for I signal and the amplitude acquisition circuit 52. The Q signal is branched and supplied to the first HPF 50B for the Q signal and the amplitude acquisition circuit 52. In the amplitude acquisition circuit 52, the amplitude of the IQ signal is obtained and supplied to the second HPF 51 as an amplitude signal.

  Then, in the first HPF 50A, 50B and the second HPF 51, for example, the IQ signal component or the amplitude signal component having a low frequency is cut by the characteristics determined by the equations (3-1) and (3-2). . For this reason, an IQ signal component or an amplitude signal component that has a low frequency and a low speed and is highly likely to be a clutter component from tissue is cut.

  For the IQ signal input to the wall filter 45, the IQ signal input to the wall filter 45 is output from the wall filter 45 due to the characteristics of the first HPFs 50A and 50B. That is, a high-speed IQ signal having a high frequency component is unlikely to be a clutter component due to tissue movement and is highly likely to be a blood flow signal. Provided to the power estimation circuit 46.

  Further, when the frequency of the IQ signal input to the wall filter 45 is medium, the IQ signal is cut by the characteristics of the first HPFs 50A and 50B, and the amplitude signal obtained by the amplitude acquisition circuit 52 is added to the addition circuit 53. The output of the wall filter 45 as an I signal through When the speed of the IQ signal input to the wall filter 45 is a medium component, it is determined whether the IQ signal is due to tissue movement or blood flow even when focusing on the phase change. Difficult to do.

  However, paying attention to the change in the amplitude of the IQ signal, the change in the amplitude of the IQ signal due to the movement of the tissue is small, but in the case of the IQ signal due to blood flow, the amplitude changes with speckle. Therefore, if an IQ signal having a medium frequency component does not pass through the wall filter 45 and a medium frequency signal of the IQ signal having a medium frequency is passed through the wall filter 45, the IQ signal due to tissue movement passes through. Without this, it is possible to pass only IQ signals due to blood flow.

  Here, the amplitude signal that passes through the second HPF 51 from the amplitude acquisition circuit 52 and becomes the I signal output of the wall filter 45 is the output of the constant value subtractor 1e in the conventional wall filter 1 shown in FIG. This is equivalent to a signal after instantaneous phase correction in which an instantaneous phase change between data constituting components is estimated and the phase change of the Doppler signal is corrected according to the instantaneous phase change amount. That is, the wall filter 45 has a function of performing processing substantially equivalent to that of the conventional wall filter 1 shown in FIG.

  Therefore, when the frequency of the IQ signal input to the wall filter 45 is medium, if the amplitude signal, which is the output of the amplitude acquisition circuit 52, is used as the I signal output of the wall filter 45, the clutter component due to the movement of the tissue is greatly increased. Can be reduced.

  Here, the reason why the wall filter 45 in the ultrasonic diagnostic apparatus according to the present invention has the same function as the conventional wall filter 1 shown in FIG. 10 will be described.

  FIG. 5 is a detailed circuit block diagram of main components in the conventional wall filter 1 shown in FIG.

  FIG. 5 shows the details of the remaining circuit blocks in the conventional wall filter 1 with the clutter information detector 1i, filter characteristic setting unit 1j, phase inverter 1h, and complex multiplier 1g for adaptively controlling parameters removed. It is a figure. In FIG. 5, the same reference numerals are given to the components corresponding to the components in FIG. 10.

  That is, if the cut-off frequency of the wall filter is increased too much, the blood flow component signal will be removed. Conversely, if the cut-off frequency is decreased too much, the clutter component signal will remain, but the blood flow component signal will remain. If removed, it will interfere with the diagnosis. Therefore, in the wall filter 45 shown in FIGS. 1 and 3, the filter coefficient is set to a fixed value without changing the cutoff frequency adaptively.

  Therefore, the conventional wall filter 1 shown in FIG. 10 will be described except for a circuit configuration for adaptively controlling parameters. The phase inverter 1h and the complex multiplier 1g are circuits for correcting the shift due to the phase correction. However, the phase inverter 1h and the complex multiplier 1g are not necessarily required to display the true blood flow velocity. The complex multiplier 1g is also omitted from the description.

  Further, in the conventional wall filter 1 shown in FIG. 10, the HPF 1f and the LPF 1b have a circuit configuration called a matrix filter like the HPFs 50A, 50B, and 51 shown in FIG. 3, and the constant value subtractor 1e is included in the HPF 1f. be able to.

  That is, the wall filter 1 has a system for processing the I signal and a system for processing the Q signal, and includes a delay circuit 60 for the I signal, a complex multiplier 1a, HPF1f and LPF1b, a delay circuit 60 for the Q signal, It has a complex multiplier 1a, HPF1f and LPF1b, a clutter phase change estimator 1c and a multiplication coefficient generator 1d which are common to the I and Q signals. Further, the clutter phase change amount estimator 1c includes a pulse pair calculation circuit 61, a phase detector 62, and a phase integrator 63.

  The I signal and Q signal input to the wall filter 1 are applied to the data delay circuit 60 and the feedback LPF 1b, respectively. Here, considering the case where there is no LPF 1b in the feedback system, the I signal and the Q signal are supplied to the pulse pair arithmetic circuit 61 of the feedback system. In the pulse pair calculation circuit 61, the calculation shown in the equation (12) is performed on the data sequence f (k) in the ensemble direction of the IQ signal as shown in the equation (1) to generate the pulse pair signal pp (k). .

[Equation 12]
pp (1) = 1
pp (k) = conj {f (k−1)} f (k) (k = 2,..., N)
(12)

  Next, each pulse pair signal pp (k) in the ensemble direction generated by the pulse pair calculation circuit 61 is given to the phase detector 62. In the phase detector 62, the calculation of Expression (13) is performed on each pulse pair signal pp (k) to generate the signal ph (k).

[Equation 13]
ph (k) = atan {imag (pp (k)) / real (pp (k))}
(K = 1, ..., N)
... (13)
In equation (13), imag (x) is a function that outputs the imaginary part of x, and real (x) is a function that outputs the real part of x.

  Next, each signal ph (k) in the ensemble direction generated by the phase detector 62 is supplied to the phase integrator 63. In the phase integrator 63, the calculation of Expression (14) is performed on each signal ph (k), that is, the signals ph (k) are added to generate the signal c (k).

[Formula 14]
c (k) = ph (1) + ph (2) +... + ph (k) (k = 1,..., N)
(14)

  Next, each signal c (k) generated by the phase accumulator 63 is given to the multiplication coefficient generator 1d. In the multiplication coefficient generator 1d, the calculation of Expression (15) is performed on each signal c (k) to generate the signal mix (k).

[Equation 15]
mix (k) = exp {−jc (k)} (k = 1,..., N) (15)

  Next, the real part of each signal mix (k) generated by the multiplication coefficient generator 1d is a complex multiplier 1a for I signal, the imaginary part of each signal mix (k) is a complex multiplier for Q signal, and Is given to 1a.

  On the other hand, the required delay time is given to the I signal I (k) and the Q signal Q (k) given to the delay circuit 60 of the data system, and the complex multiplier 1a for I signal and Q signal is given. Given each. The complex multiplier 1a for I signal multiplies the real part of mix (k) and the I signal I (k), and the complex multiplier 1a for Q signal uses the imaginary part of mix (k) and Q The signal Q (k) is multiplied. At this time, the delay time is set in the delay circuit 60 so that the signal mix (k) for the same k is multiplied by the I signal I (k) or the Q signal Q (k), and the I signal I (k ) And Q signal Q (k) are delayed.

  That is, the calculation of Expression (16) is performed by the complex multiplier 1a for I signal and Q signal to generate the output signal g (k).

[Equation 16]
g (k) = f (k) · mix (k) (16)

  Here, equation (17) is established from the nature of the pulse pair calculation.

[Equation 17]
ph (k) = φ (k) −φ (k−1) (k = 2,..., N)
ph (1) = 0
... (17)
Therefore, Expression (18) is derived from Expression (17) and Expression (14).

[Equation 18]
c (k) = ph (1) + ph (2) +... + ph (k)
= {Φ (2) −φ (1)} + {φ (3) −φ (2)} +... + {Φ (k) −φ (k−1)}
= -Φ (1) + φ (k)
... (18)

  Furthermore, Expression (19) is derived from Expression (18), Expression (16), and Expression (15).

[Equation 19]
g (k) = f (k) · mix (k)
= A (k) exp {jφ (k)} exp {−jφ (k) + jφ (1)}
= A (k) exp {jφ (1)}
... (19)
Here, in expression (19), exp {jφ (1)} is an initial phase common to all signals, and thus does not affect the subsequent processing. Therefore, exp {jφ (1)} can be ignored in equation (19). If exp {jφ (1)} is ignored in equation (19), equation (20) is derived.

[Equation 20]
g (k) = A (k) = abs {f (k)} (20)
In equation (20), abs (x) is a function that outputs the absolute value of x.

  From the equation (20), the output signal g (k) of the complex multiplier 1a, that is, the wall filter 1 shown in FIG. 5, the data stream f (k) in the ensemble direction of the IQ signal is fed back by the LPF 1b in the feedback system. It can be seen that the signal g (k) obtained by performing the instantaneous phase correction of the IQ signal without processing is equal to the amplitude signal A (k) of the IQ signal f (k) which is the original signal.

  From the above, in FIG. 5, when there is no LPF 1b, the output signal g (k) of the complex multiplier 1a is equivalent to the amplitude signal A (k) that is the output signal of the amplitude acquisition circuit 52 shown in FIG. I understand that.

  Next, in the wall filter 1 shown in FIG. 5, the data sequence f (k) in the ensemble direction of the IQ signal is subjected to filter processing as preprocessing by the LPF 1b in the feedback system, and then instantaneous phase correction is performed. Further, the HPF 1f Consider the case of performing filter processing.

  LPF1b and HPF1f shown in FIG. 5 are similar to the HPFs 50A, 50B, and 51 shown in FIG. 3 with respect to the input signal by the filter matrix W defined by Equation (2-1). A matrix operation using a linear filter is performed and output. Therefore, the characteristics of LPF 1b and HPF 1f are determined by the diagonal elements of diagonal matrix D.

  Here, when m is set to 1 from the smallest of the diagonal elements of the diagonal matrix D and the rest is set to 0, the linear filter has a characteristic as an LPF, and m from the smallest of the diagonal elements of the diagonal matrix D is changed. Since the linear filter has a characteristic as HPF when 0 is set to 0 and the remaining is set to 1, the diagonal elements lpf and hpf of the diagonal matrix D that determine the characteristics of the LPF 1b and HPF 1f are, for example, when the ensemble number is 16, They are set as shown in equations (21-1) and (21-2), respectively.

[Equation 21]
lpf = [1,1,1,1,1,1,1,1,0,0,0,0,0,0,0,0]
... (21-1)
hpf = [0,0,0,0,1,1,1,1,1,1,1,1,1,1,1,1]
... (21-2)

  Then, in the feedback system of the wall filter 1 shown in FIG. 5, the 8 signals on the low frequency side in the orthogonal polynomial space among the IQ signals are not cut in the LPF 1 b, so that the output of the complex multiplier 1 a The signal g (k) is equal to the amplitude signal A (k) of the IQ signal f (k). For the eight base signals on the high frequency side in the orthogonal polynomial space of the IQ signal, the signal is cut in the LPF 1b. Therefore, the output signal g (k) of the complex multiplier 1a is a signal different from the amplitude signal A (k) of the IQ signal f (k).

  That is, the eight base signals on the low frequency side in the orthogonal polynomial space in the IQ signal are likely to be clutter components, and therefore the amplitude signal A (k) whose instantaneous phase is corrected in the feedback system is the complex multiplier 1a. The output signal g (k) is an output signal g (k). Of the IQ signals, the eight base signals on the high frequency side in the orthogonal polynomial space are unlikely to be clutter components, so the IQ signal f (k) input to the data system is corrected. The output signal g (k) of the complex multiplier 1a is not used.

  Further, the output signal g (k) of the complex multiplier 1a is given to the HPF 1f, and the amplitude signals A (k) corresponding to the four base signals on the low frequency side in the orthogonal polynomial space are cut. In other words, the clutter component is more reliably removed from the IQ signal by cutting the low frequency side signal. On the other hand, among the output signal g (k) of the complex multiplier 1a, the four amplitude signals A (k) between the fifth and the eighth from the low frequency side are output without being affected by the HPF 1f.

  As a result, the IQ signal that is the output of the HPF 1 f and the output of the wall filter 1 is four amplitude signals A (k) between the fifth and eighth from the low frequency side, and eight original signals on the high frequency side. It becomes an IQ signal.

  Therefore, in the wall filter 45 shown in FIG. 3, the diagonal element hpf1 of the diagonal matrix D that determines the characteristics of the first HPFs 50A and 50B is set to the value shown in Expression (3-1), and the characteristics of the second HPF 51 are Assuming that the diagonal element hpf2 of the diagonal matrix D to be determined is a value shown in the equation (3-2), the wall filter 45 has the characteristic that the wall filter 1 shown in FIG. It is equivalent to the characteristic of

  That is, since the four elements from the smaller of the diagonal elements hpf1 and hpf2 of the diagonal matrix D of each of the first HPFs 50A and 50B and the second HPF 51 are 0, the four components on the low frequency side of the IQ signal Are cut in any of the first HPFs 50A and 50B and the second HPF 51, and components that are highly likely to be clutter signals are removed.

  Further, among the diagonal elements hpf1 of the diagonal matrix D that determines the characteristics of the first HPFs 50A and 50B, the elements between the fifth and eighth elements from the smallest are 0, so the frequency of the IQ signal is The four components having a medium frequency between the fifth and eighth from the smallest are cut by the first HPFs 50A and 50B. On the other hand, among the diagonal elements hpf2 of the diagonal matrix D that determines the characteristics of the second HPF 51, the element between the fifth and eighth elements from the smallest is 1, so the amplitude obtained by the amplitude acquirer 45c. The signal is output from the wall filter 45 as an I signal via the second HPF 51 and the adder circuit 53.

  Furthermore, among the diagonal elements hpf2 of the diagonal matrix D that determines the characteristics of the second HPF 51, the ninth and subsequent elements from the smallest are zero, so the amplitude signal obtained by the amplitude acquirer 45c is the second The HPF 51 is cut. On the other hand, among the diagonal elements hpf1 of the diagonal matrix D that determines the characteristics of the first HPFs 50A and 50B, the ninth and subsequent elements from the smallest are 1, so the 9th from the smaller frequency of the IQ signals. The eight high frequency components after the first are output from the wall filter 45 via the first HPFs 50A and 50B and the adder circuit 53.

  As a result, similarly to the output of the wall filter 1 shown in FIG. 5, the output of the wall filter 45 includes four amplitude signals A (k) between the fifth and eighth from the low frequency side, and eight high frequency side signals. The IQ signal is the original signal. That is, the first to eighth outputs on the low frequency side of the wall filter 45 coincide with the instantaneous phase correction output in the wall filter 1 shown in FIG. However, since the first HPFs 50A and 50B perform the filtering process on the IQ signal not subjected to the phase correction, the output of the ninth and subsequent IQ signals from the low frequency side of the wall filter 45 is the phase correction. Is strictly different from the output of the wall filter 1 shown in FIG.

  By the way, the characteristics of the first HPFs 50A and 50B are eight diagonal elements hpf1 on the higher frequency side and zero in the other. Accordingly, the first HPFs 50A and 50B respond only to the IQ signal obtained from the fast moving one. In this respect, since the influence of the relatively slow movement of the tissue is small on the IQ signal in the high frequency region, it is not a big problem.

  FIG. 6 is a diagram showing the frequency characteristics of the orthogonal polynomial space at a point where the tissue movement of the actual ultrasonic signal is large.

  In FIG. 6, the vertical axis indicates the signal strength relative value [dB] of the frequency in the orthogonal polynomial space, and the horizontal axis indicates the base number of the diagonal matrix D of the wall filter 45 and 1 corresponding to the column in the ensemble direction. . Here, the number of ensembles and the number of bases are 16. In FIG. 6, square marks and alternate long and short dash lines indicate data D1 indicating the frequency characteristic of the orthogonal polynomial space of the IQ signal of the ensemble sequence input to the wall filter 45 shown in FIG. 3 to the conventional wall filter 1 shown in FIG. Black circles and solid lines indicate data D2 indicating the frequency characteristics of the orthogonal polynomial space of the IQ signal output from the complex multiplier 1a of the conventional wall filter 1 shown in FIG. 5, and white circles and dotted lines indicate the wall filter 45 shown in FIG. This is data D3 indicating the frequency characteristic of the orthogonal polynomial space of the amplitude signal of the IQ signal output from the amplitude acquirer 45c.

  That is, the data D2 is a characteristic of the IQ signal obtained by instantaneous phase correction in the complex multiplier 1a after filtering the data D1 indicating the input IQ signal by the LPF 1b of the conventional wall filter 1 shown in FIG. The data D3 indicates the characteristic of the amplitude signal acquired by the amplitude acquirer 45c of the wall filter 45 shown in FIG. 3 with respect to the data D1 indicating the IQ signal.

  According to FIG. 6, the characteristics of the bases on the lower side of the first to eighth frequencies, that is, the values of the coefficients are the values in the case of the signal subjected to instantaneous phase correction since the data D2 and the data D3 overlap. It can be seen that the values in the case of the amplitude signal completely match. Furthermore, although the data D1 is output from the wall filter 45 shown in FIG. 3 after the ninth frequency on the higher frequency side, the base characteristics are not significantly different between the data D2 and the data D1 subjected to instantaneous phase correction. I understand that there is no.

  The IQ signal, which is the output of the wall filter 45 obtained in this way, is applied to the power calculation circuit 54 and the pulse pair calculation circuit 56 of the subsequent speed / dispersion / power estimation circuit 46. In the power calculation circuit 54, the calculation shown in the equation (5) is performed, and the obtained power signal is supplied to the power integrator 55. Further, in the power integrator 55, the calculation shown in Expression (6) is performed, and the obtained power signal is given to the LOG compression circuit 58. Then, in the LOG compression circuit 58, the blood flow power signal P is estimated by performing the calculation shown in the equation (7). The estimated blood flow power signal P is output to the variance estimator 60 and the subsequent DSC 31 for image generation.

  On the other hand, in the pulse pair calculation circuit 56, the calculation shown in the equation (8) is performed, and the obtained pulse pair signal is given to the PP integrator 57. In the PP integrator 57, the calculation shown in Expression (9) is performed, and the obtained output signal is given to the phase detector 59 and the variance estimator 60. Then, in the phase detector 59, the calculation shown in Expression (10) is performed, and the blood flow velocity signal V is obtained. The obtained blood flow velocity signal V is output to the subsequent DSC 31 and used for image generation.

  In addition, the variance estimator 60 performs the calculation shown in the equation (11) from the power signal received from the LOG compression circuit 58 and the output signal of the PP integrator 57 to obtain the blood flow variance signal T. The obtained blood flow dispersion signal T is output to the subsequent DSC 31 for image generation.

  According to the ultrasonic diagnostic apparatus as described above, with a simpler circuit configuration without a feedback system, the conventional clutter component of the tissue that is moving delicately due to instantaneous phase correction, that is, heartbeat or respiration, is conventionally performed. It is possible to provide a function equivalent to the function of instantaneously canceling the instantaneous phase change. That is, in the conventional method of the wall filter 1 in the ultrasonic diagnostic apparatus, it is necessary to have two systems of data paths for the phase correction feedback mechanism, and the circuit scale is large and the control is complicated. Since the ultrasonic diagnostic apparatus does not have a feedback mechanism and only one data path is required, the circuit scale can be reduced and the control is simple.

  When the IQ signal input to the wall filter 45 is only the base signal between the fifth and eighth frequencies from the lower frequency side, the output of the wall filter 45 is the IQ signal that has passed through the second HPF 51. Only amplitude signal. For this reason, the signal given from the wall filter 45 to the speed / dispersion / power estimation circuit 46 is only the I signal, and the Q signal is zero. Therefore, the blood flow velocity signal V is +0 (a positive number close to 0) or −0 (a negative number close to 0). In this case, the blood flow velocity signal V is not a correct blood flow velocity value but is a signal representing the direction in which the blood flow flows. Therefore, in such a case, only the blood flow power signal P and the direction in which the blood flow flows are displayed.

  On the other hand, the diagonal elements hpf1 and hpf2 are set to 1 or 0 so that the diagonal elements hpf1 and hpf2 of the diagonal matrix D that determine the characteristics of the first HPFs 50A and 50B and the second HPF 51 do not overlap each other. However, for example, the diagonal elements hpf1 and hpf2 may be overlapped with each other as shown in Expression (22-1) and Expression (22-2).

[Equation 22]
hpf1 = [0,0,0,0,0.1,0.1,0.1,0.1,0.5,1,1,1,1,1,1,1,1]
... (22-1)
hpf2 = [0,0,0,0,0.9,0.9,0.9,0.9,0.5,0,0,0,0,0,0,0]
... (22-2)

  If the diagonal elements hpf1 and hpf2 are set so as to overlap each other as in the examples of the expressions (22-1) and (22-2), the IQ signal input to the wall filter 45 becomes 5 from the low frequency side. The Q signal applied to the velocity / dispersion / power estimation circuit 46 does not become 0 even when only the base signal between the 8th and the 8th is present, so that the blood flow velocity signal V is not 0 as the blood flow velocity value. Can be requested.

  FIG. 7 is a block diagram showing a second embodiment of the ultrasonic diagnostic apparatus according to the present invention.

  In the ultrasonic diagnostic apparatus shown in FIG. 7, the circuit configuration of the wall filter 45A and the velocity / dispersion / power estimation circuit 46A is different from that of the ultrasonic diagnostic apparatus shown in FIG. Since other configurations and operations are not substantially different from those of the ultrasonic diagnostic apparatus shown in FIG. 1, only the circuit configurations of the wall filter 45A and the velocity / dispersion / power estimation circuit 46A are shown, and the same components are denoted by the same reference numerals. Therefore, the description is omitted.

  The wall filter 45A includes two first HPFs 50A and 50B for I and Q signals, a second HPF 51 having different characteristics from the first HPFs 50A and 50B, an amplitude acquisition circuit 52, and a first power calculation circuit 54A. , A second power calculation circuit 54B, a pulse pair calculation circuit 56, and an addition circuit 53. The speed / dispersion / power estimation circuit 46A includes a power integrator 55, a PP integrator 57, a LOG compression circuit 58, a phase detector 59, and a dispersion estimator 60.

  The I signal input to the wall filter 45A is supplied to the first HPF 50A for I signal and the amplitude acquisition circuit 52. Similarly, the Q signal input to the wall filter 45 is supplied to the first HPF 50B for Q signal and the amplitude acquisition circuit 52. The output of the first HPF 50A for the I signal is supplied to the first power calculation circuit 54A and the pulse pair calculation circuit 56, and the output of the first HPF 50B for the Q signal is also calculated with the first power calculation circuit 54A. Is provided to circuit 56.

  A second HPF 51 is provided on the output side of the amplitude acquisition circuit 52, and a second power calculation circuit 54B is provided on the output side of the second HPF 51. Further, the output of the first power calculation circuit 54 </ b> A and the output of the second power calculation circuit 54 </ b> B are supplied to the addition circuit 53.

  On the other hand, the power integrator 55 of the speed / dispersion / power estimation circuit 46A receives the output of the adder circuit 53, and the PP integrator 57 receives the output of the pulse pair arithmetic circuit 56. The subsequent configuration of the power integrator 55 and the PP integrator 57 is the same as that of the speed / dispersion / power estimation circuit 46 shown in FIG.

  Further, the first power calculation circuit 54A performs the calculation shown in Expression (5) using the I signal and the Q signal received from the first HPFs 50A and 50B, respectively, and P0 (k The second power calculation circuit 54B uses the square value of the amplitude signal of the IQ signal received from the second HPF 51 as each power signal in the ensemble direction. It has a function to give to.

  The pulse pair calculation circuit 56 performs the calculation shown in the equation (8) using the I signal and the Q signal received from the first HPFs 50A and 50B, respectively, to obtain pp (k) that is each pulse pair signal in the ensemble direction, It has a function to be given to the PP integrator 57. The adder circuit 53 has a function of adding the power signals received from the first and second power calculation circuits 54A and 54B to the power integrator 55. The power integrator 55 uses the power signals in the ensemble direction received from the adder circuit 53 to obtain the power signal P1.

  The functions of the other components are the same as the functions of the components in the wall filter 45 and the velocity / dispersion / power estimation circuit 46 shown in FIG.

  In the wall filter 45A and the speed / dispersion / power estimation circuit 46A, the power calculation is performed by the first power calculation circuit 54A on the I signal that is the output of the first HPF 50A, while the output of the second HPF 51 is output. The power calculation is performed on the amplitude signal by the second power calculation circuit 54B. Then, the power calculation output of the amplitude signal is added to the power calculation output of the I signal in the adding circuit 53.

  That is, the wall filter 45A and the speed / dispersion / power estimation circuit 46A shown in FIG. 7 are configured so that the wall filter 45 and the speed / dispersion / power estimation circuit 46 shown in FIG. While the configuration is such that the amplitude signal that is the output of the second HPF 51 is directly added, the power of the I signal that is the output of the first HPF 50A and the power of the amplitude signal that is the output of the second HPF 51 are obtained. Are added to each other.

  For this reason, in the wall filter 45A and the speed / dispersion / power estimation circuit 46A, the data after the filter processing of the first HPF 50A and the second HPF 51 are added in an incoherent manner without considering the phase. Therefore, the filters of the first HPF 50A and the second HPF 51 do not have to be the same orthogonal transform filter, and the first HPF 50A and the second HPF 51 can be set arbitrarily to some extent.

  In addition to the circuit configuration shown in FIG. 7, for example, each power calculation output of the IQ signal and the amplitude signal obtained by the power calculation circuits 54 </ b> A and 54 </ b> B is given to the individual power integrator 55. A similar effect can be obtained by obtaining the power integrated output of the IQ signal and the amplitude signal based on the power calculation outputs of the IQ signal and the amplitude signal and then adding the obtained power integrated output.

  FIG. 8 is a block diagram showing a third embodiment of the ultrasonic diagnostic apparatus according to the present invention.

  In the ultrasonic diagnostic apparatus shown in FIG. 8, the wall filter 45B and the velocity / dispersion / power estimation circuit 46B have the same circuit configuration as the wall filter 45A and velocity / dispersion / power estimation circuit 46A of the ultrasonic diagnostic apparatus shown in FIG. Is different. Since other configurations and operations are not substantially different from those of the ultrasonic diagnostic apparatus shown in FIG. 1, only the circuit configurations of the wall filter 45B and the velocity / dispersion / power estimation circuit 46B are shown. Therefore, the description is omitted.

  The wall filter 45B includes two first HPFs 50A and 50B for I and Q signals, a second HPF 51 having different characteristics from the first HPFs 50A and 50B, an amplitude acquisition circuit 52, and a first power calculation circuit 54A. The second power calculation circuit 54B and the pulse pair calculation circuit 56 are provided. The speed / dispersion / power estimation circuit 46A includes a first power integrator 55A, a second power integrator 55B, a PP integrator 57, a first LOG compression circuit 58A, a second LOG compression circuit 58B, a phase A detector 59, a coefficient generator 80, a first multiplier 81A, a second multiplier 81B, and an adder circuit 53 are provided.

  The connection configurations of the first HPFs 50A and 50B, the second HPF 51, the amplitude acquisition circuit 52, and the pulse pair calculation circuit 56 are the same as those of the wall filter 45A and the velocity / dispersion / power estimation circuit 46A of the ultrasonic diagnostic apparatus shown in FIG. .

  A first power integrator 55A is provided on the output side of the first power arithmetic circuit 54A, and a first LOG compression circuit 58A is provided on the output side of the first power integrator 55A. A second power integrator 55B is provided on the output side of the second power calculation circuit 54B, and a second LOG compression circuit 58B is provided on the output side of the second power integrator 55B. The output of the first LOG compression circuit 58A is supplied to the first multiplier 81A and the coefficient generator 80, and the output of the second LOG compression circuit 58B is the second multiplier 81B and the first LOG compression circuit. This is supplied to the coefficient generator 80 common to the output destination of 58A.

  Further, the output of the coefficient generator 80 is given to the first multiplier 81A and the second multiplier 81B, and the outputs of the first multiplier 81A and the second multiplier 81B are given to the common adder circuit 53. It is done.

  On the other hand, unlike the velocity / dispersion / power estimation circuit 46 A of the ultrasonic diagnostic apparatus shown in FIG. 7, the variance estimator 60 is not provided and the output of the PP integrator 57 is given to the phase detector 59. Then, the blood flow velocity signal V is obtained by the phase detector 59, and the blood flow power signal P is obtained by the adder circuit 53, which are output from the velocity / dispersion / power estimation circuit 46A.

  Further, the first power calculation circuit 54A performs the calculation shown in Expression (5) using the I signal and the Q signal received from the first HPFs 50A and 50B, respectively, and P0 (k The second power calculation circuit 54B obtains the square value of the amplitude signal of the IQ signal received from the second HPF 51 for each power signal in the ensemble direction. As a second power integrator 55B.

  The first and second power integrators 55A and 55B add the power signals P0 (k) in the ensemble direction received from the first and second power calculation circuits 54A and 54B, respectively, as shown in Expression (6). Thus, the power signal P1 is obtained and the obtained power signal P1 is provided to the first and second LOG compression circuits 58A and 58B.

  The first and second LOG compression circuits 58A and 58B perform the calculation shown in Expression (7) on the power signal P1 received from the first and second power integrators 55A and 55B, respectively, to thereby obtain the blood flow power signal. A function of estimating P, and a function of supplying the estimated blood flow power signal P to the first and second multipliers 81A and 81B and the coefficient generator 80.

  The coefficient generator 80 obtains a difference between the blood flow power signals P received from the first and second LOG compression circuits 58A and 58B, and based on the obtained difference, the first and second LOG compression circuits 58A and 58B. The blood flow power signal P obtained in (1) is weighted and added, and a function for obtaining the coefficient is given to the first and second multipliers 81A and 81B.

  The first and second multipliers 81A and 81B multiply the coefficients received from the coefficient generator 80 by the blood flow power signals P received from the first and second LOG compression circuits 58A and 58B, respectively, and adder 53 The adder circuit 53 has a function of adding the blood flow power signals P received from the first and second multipliers 81A and 81B and outputting the blood flow power signal P as a blood flow power signal P.

  That is, the coefficient generator 80, the first and second multipliers 81A and 81B, and the adder circuit 53 function as means for weighted addition of signals obtained from the IQ signal and the amplitude signal, and the first and second LOGs. The blood flow power signal P, which is the output of the compression circuits 58A and 58B, is configured to be weighted and added.

  In the wall filter 45B and the velocity / dispersion / power estimation circuit 46B configured as described above, the I signal that is the output of the first HPF 50A and the amplitude signal that is the output of the second HPF 51 are individually used. The blood flow power signal P is obtained by the first and second power calculation circuits 54A and 54B, the first and second power integrators 55A and 55B, and the first and second LOG compression circuits 58A and 58B. Further, each obtained blood flow power signal P is weighted and added by the coefficient generator 80, the first and second multipliers 81A and 81B, and the adder circuit 53, and is output as the output of the velocity / dispersion / power estimation circuit 46B. A blood flow power signal P is obtained.

  The weighted addition coefficient at this time is set based on the difference between the blood flow power signals P obtained by the first and second LOG compression circuits 58A and 58B by the coefficient generator 80, and the weight at the time of addition is set. Is adjusted.

  For this reason, according to the wall filter 45B and the velocity / dispersion / power estimation circuit 46B, the same effects as those of the wall filter 45A and velocity / dispersion / power estimation circuit 46A shown in FIG. By weighting and adding the data and the data obtained from the amplitude signal, the data can be arbitrarily adjusted and added according to the value of each data.

  The target of the weighted addition is not limited to the blood flow power signal P that is the output of the first and second LOG compression circuits 58A and 58B, but may be the first, You may perform weight addition with respect to the power calculation output which is the output of 2nd power calculation circuit 54A, 54B, and the power integration output which is the output of 1st, 2nd power integrator 55A, 55B.

  FIG. 9 is a block diagram showing a fourth embodiment of the ultrasonic diagnostic apparatus according to the present invention.

  In the ultrasonic diagnostic apparatus shown in FIG. 9, the circuit configuration of the wall filter 45C and the velocity / dispersion / power estimation circuit 46C is different from that of the ultrasonic diagnostic apparatus shown in FIG. Since other configurations and operations are not substantially different from those of the ultrasonic diagnostic apparatus shown in FIG. 1, only the circuit configurations of the wall filter 45C and the velocity / dispersion / power estimation circuit 46C are shown, and the same components are denoted by the same reference numerals. Therefore, the description is omitted.

  The wall filter 45C includes an amplitude acquisition circuit 52 and a third HPF 90 for amplitude signals. Further, the speed / dispersion / power estimation circuit 46C includes only the power estimation circuit of the speed / dispersion / power estimation circuit 46 shown in FIG. 3, and there is no circuit for estimating the speed / dispersion. Therefore, only the power signal P is output.

  The amplitude acquisition circuit 52 has a function of generating an amplitude signal from the IQ signal input to the wall filter 45C and supplying the amplitude signal to the third HPF 90. The characteristics of the third HPF 90 are set by a method different from that of the first and second HPFs 50A, 50B, and 51 shown in FIG. The third HPF 90 has the same characteristics as the HPF 1 f in the conventional wall filter 1 shown in FIG. 5 or FIG. 10, and is represented by Expression (2-2) by the filter matrix W defined by Expression (2-1). A matrix operation is performed by a linear filter of real coefficients and output.

  Therefore, for example, when the number of ensembles is 16, the diagonal element hpf3 of the diagonal matrix D that determines the characteristics of the third HPF 90 is set as shown in Expression (23).

[Equation 23]
hpf3 = [0,0,0,0,1,1,1,1,1,1,1,1,1,1,1,1,1]
... (23)

  That is, the third HPF 90 cuts the four base signals on the low frequency side in the orthogonal polynomial space out of the amplitude signal in the same manner as the HPF 1 f in the conventional wall filter 1 and passes the others. The output of the third HPF 90 is given to the power arithmetic circuit 54 as an IQ signal that is the output of the wall filter 45C.

  In the wall filter 45C, the amplitude signal that is the output of the amplitude acquisition circuit 52 is equivalent to the output of the complex multiplier 1a in the conventional wall filter 1 shown in FIG. Therefore, in the case where the conventional wall filter 1 shown in FIG. 5 is not provided with the LPF 1b, if the third HPF 90 having the same characteristics as the HPF 1f is provided in the subsequent stage of the amplitude acquisition circuit 52, the wall filter 45C A function equivalent to that of the conventional wall filter 1 in which the LPF 1b is omitted can be provided.

  For this reason, if configured like the wall filter 45C, only the blood flow power information can be displayed, but the function equivalent to that of the conventional wall filter 1 in which the LPF 1b is omitted can be provided with a very simple circuit configuration.

1 is a configuration diagram showing a first embodiment of an ultrasonic diagnostic apparatus according to the present invention. FIG. The figure which shows the digital Doppler data of the base band stored in each of the buffer memory shown in FIG. The block diagram which shows the detailed structure of the wall filter and speed / dispersion / power estimation circuit shown in FIG. The figure which shows an example of the circuit structure of 1st, 2nd HPF shown in FIG. 3, and a calculation method. FIG. 2 is a detailed circuit block diagram of main components in the conventional wall filter 1 shown in FIG. 1. The figure which showed the frequency characteristic of the orthogonal polynomial space in the point where the motion of the structure | tissue of an actual ultrasonic signal is large. The block diagram which shows 2nd Embodiment of the ultrasonic diagnosing device which concerns on this invention. The block diagram which shows 3rd Embodiment of the ultrasonic diagnosing device which concerns on this invention. The block diagram which shows 4th Embodiment of the ultrasonic diagnosing device which concerns on this invention. The block diagram of the wall filter which provided two paths for the conventional data system and a feedback system, and applied LPF to the ensemble string.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Ultrasonic probe 2 Transmission system circuit 3 Reception / processing system circuit 21 Reception circuit 24 Display circuit 41A, 41B Mixer 42A, 42B LPF
43A, 43B A / D converters 44A, 44B Buffer memory 45 Wall filter 46 Speed / dispersion / power estimation circuit 47 Reference transmitter 48 Phase shifters 50A, 50B First HPF
51 Second HPF
52 Amplitude acquisition circuit 53 Addition circuit 54, 54A, 54B Power operation circuit 55, 55A, 55B Power integrator 56 Pulse pair operation circuit 57 PP (pulse pair) integrator 58, 58A, 58B LOG compression circuit 59 Phase detector 60 Variance estimator 70 Coefficient Generator 71 Multiplier 72 Adder Circuit 80 Coefficient Generator 81A, 81B Multiplier 90 Third HPF

Claims (7)

Transmission / reception means for transmitting an ultrasonic signal into a subject a plurality of times in each scanning line direction and receiving an ultrasonic echo signal reflected from the subject, and means for obtaining an IQ signal from the reception signal of the transmission / reception means And means for obtaining an amplitude signal from the IQ signal, and means for removing low frequency components from the plurality of IQ signals and amplitude signals reflected from the same position in the respective scanning line directions by different characteristics. An ultrasonic diagnostic apparatus comprising: The means for removing the low frequency component reduces the signal strength of both the IQ signal and the amplitude signal for the low frequency component, and reduces the signal strength of the IQ signal for the medium frequency component. 2. The ultrasonic diagnostic apparatus according to claim 1, wherein a signal is mainly passed and an IQ signal is mainly passed for a component having a high frequency, while a signal strength of an amplitude signal is reduced. The means for removing the low frequency component operates the same orthogonal transform base coefficient on the IQ signal and the amplitude signal, so that the signal strength of both the IQ signal and the amplitude signal is obtained for the low frequency component. The signal intensity of the IQ signal is reduced for a medium frequency component, while the amplitude signal is mainly passed, and the IQ signal is mainly passed for a high frequency component, while the signal strength of the amplitude signal is reduced. The ultrasonic diagnostic apparatus according to claim 1, wherein the ultrasonic diagnostic apparatus is reduced. The means for removing the low frequency component reduces the signal strength of both the IQ signal and the amplitude signal for the low frequency component, and reduces the signal strength of the IQ signal for the medium frequency component. The signal is mainly passed, and the IQ signal is mainly passed for a component having a high frequency, while the signal strength of the amplitude signal is reduced, and the weighted addition of the passed IQ signal and the amplitude signal is provided. The ultrasonic diagnostic apparatus according to claim 1. The means for removing the low frequency component reduces the signal strength of both the IQ signal and the amplitude signal for the low frequency component, and reduces the signal strength of the IQ signal for the medium frequency component. The signal is mainly passed, and the IQ signal is mainly passed for the component having a high frequency, while the signal strength of the amplitude signal is reduced, and each blood flow obtained from the passed IQ signal and amplitude signal is further reduced. 2. The ultrasonic diagnostic apparatus according to claim 1, further comprising means for weighted addition of power signals. Transmission / reception means for transmitting an ultrasonic signal into the subject a plurality of times in each scanning line direction and receiving an ultrasonic echo signal reflected from the subject, and means for obtaining an IQ signal from the reception signal of the transmission / reception means And means for obtaining an amplitude signal from the IQ signal and means for removing a low-frequency component from a plurality of amplitude signals reflected from the same position in each scanning line direction. Ultrasonic diagnostic equipment. Transmission / reception means for transmitting an ultrasonic signal into the subject a plurality of times in each scanning line direction and receiving an ultrasonic echo signal reflected from the subject, and means for obtaining an IQ signal from the reception signal of the transmission / reception means And an amplitude acquirer for obtaining an amplitude signal from the IQ signal, and reducing the signal strength of the low frequency component and the medium frequency component of the IQ signal, while passing the high frequency component of the IQ signal. And a second high-pass filter that reduces a signal intensity of a low-frequency component and a high-frequency component of the amplitude signal while passing a medium-frequency component of the amplitude signal. An ultrasonic diagnostic apparatus comprising:

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