JP2005259390A - Modular rosette for communication - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a means to keep a near-end crosstalk attenuation and a far-end crosstalk attenuation between pairs at specified values or more, where the pairs are of multi-pair/multi-core modular jack for communication, which is composed of jack and clamp terminals, and a circuit board for electrically connecting both the terminals. <P>SOLUTION: A mutual inductance M<SB>k</SB>and a coupling capacitance C<SB>k</SB>between line pairs are set so that formulas ¾M<SB>k</SB>/C<SB>k</SB>¾=(Z<SB>0</SB>/ 2)<SP>2</SP>and 2πf ¾M<SB>k</SB>¾/Z<SB>0</SB>≤10<SP>-FEXT/20</SP>are satisfied, where Z<SB>0</SB>, f , NEXT, and FEXT are line impedance, frequency, near-end crosstalk attenuation, and far-end crosstalk attenuation of the modular jack, respectively. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

  The present invention relates to a multi-channel transmission system for voice or data, and more specifically, a UTP cable, a modular jack for connection, a clamp terminal, or both, which constitute a wiring system in which cross-talk is a problem. It is about the rosette.

  In an information wiring system represented by a LAN, a multi-wire multi-pair round-trip transmission path is used. As a transmission medium, a metallic UTP (Unshielded Twisted Pair) cable is generally used. As a connector for connection, a modular rosette including a modular jack, a clamp terminal, and a line electrically connecting both of them or a printed wiring board (hereinafter abbreviated as PCB (Printed Circuit Board)) is used.

  FIG. 1 shows the appearance of a unit portion of an exposed modular rosette installed in an arbitrary place in a room. FIG. 2 shows the configuration of the rosette. The rosette has a modular jack 1 that terminates a cable with a plug, a clamp case 2 that houses a terminal for clamping the cable core, a cable core release button 3 that releases the clamped cable, a board 4 on which the above components are mounted, and the board mounted thereon. The base 5 and the cover 6 are configured.

  3 and 4 show the appearance of the embedded modular rosette installed on the wall surface. 3 is a perspective view of the rosette as seen from the modular jack side, and FIG. 4 is a perspective view of the rosette as seen from the clamp terminal side. FIG. 5 shows the structure of an embedded modular rosette. From the modular jack 7 for terminating the cable with plug, the clamp terminal 8 for clamping the cable core, the cable core release button 9 for releasing the clamped cable, the board 10 for mounting the above components, the base 11 for mounting the board and the cover 12 Become.

  The clamp terminals of these exposed and embedded modular rosettes are S spring type clamp terminals that have a non-return action (Nippon Telegraph and Telephone Corporation model numbers NMJ-8 and WUK-8 described in Patent Document 1, etc.) Terminal).

  In multi-wire multi-pair transmission elements (hereinafter referred to as multi-wire elements) such as cables, modular rosettes, modular jacks and clamp terminals that make up the wiring system, the crosstalk between the pairs is not suppressed to a predetermined amount or more. must not. This is because crosstalk between pairs becomes a major factor causing a code error in a transmission signal.

  Crosstalk control in multi-strip elements is generally not easy. In particular, when the transmission rate is increased, the demand for crosstalk attenuation is drastically severe. For example, the near-end and far-end crosstalk attenuation of the modular rosette specified by the international standard TIA / EIA-568-B.2 (non-patent document 1) for LAN wiring systems is 100 MHz in enhanced category 5 (denoted as Cat.5e). In Category 6 (referred to as Cat.6) that uses up to 250 MHz, this is a significant enhancement of 54 dB and 43.1 dB.

  In Cat.5e up to the maximum signal frequency of 100 MHz, the modular rosette (Fig. 6) that uses the U-slit called RJ-45 as the modular jack 14 and the 110 type as the clamp terminal 15 simply minimizes the coupling capacity between pairs. It was possible to realize a predetermined amount of crosstalk attenuation. In the S spring type clamp terminal with non-return action, it is only necessary to pay attention to the coupling capacity in the type with the terminal spacing wide as in the exposed type in Fig. 2, but in the S spring type as in Fig. 5 embedded type In the type where metal spring holders with large opposing areas are arranged at narrow intervals, it is difficult to obtain a sufficient characteristic margin even with Cat.5e without considering not only the coupling capacitance between pairs but also the influence of mutual inductance.

  Conventionally, there is no unified quantitative method as a means of realizing a modular rosette with crosstalk control exceeding a predetermined level, and each manufacturer has been designed with unique know-how. In general, a method of forming a line connecting a modular jack pin and a clamp terminal or a pattern of a PCB by cut-and-try while measuring a crosstalk amount is employed. It is not uncommon for the number of prototype boards to exceed 100.

Furthermore, although there are several types of Cat.6 compatible modular jacks currently on the market, the most important near-end crosstalk attenuation (hereinafter abbreviated as NEXT) or far-end crosstalk attenuation (hereinafter abbreviated as FEXT) in the wiring system. In the current situation, there are many cases where even the standard is broken. It is a big problem in this field.
No. 7-43963 TIA / EIA-568-B.2 “Commercial Building Telecommunications Cabling Standard, Part 2: Balanced Twisted-Pair Cabling Components” May 2001 Takeyama Shuzo, "Electromagnetic Phenomenon Theory", Maruzen 1980 IEICE September 2003 Society Conference Proceedings C-5-1 "A Study on Crosstalk Compensation for Multi-Wire Transmission Lines" IEICE September 2003 Society Conference Proceedings C-5-2 “Cat.5e for telephone embedded rosette” IEICE Technical Report EMD2003-98 “Study on Crosstalk Control of LAN9 Nectar”

  In view of the above situation, it is an object of the present invention to establish a means for realizing reliable crosstalk characteristics and reducing development man-hours in the development and design of modular rosettes for high-speed LANs based on cut-and-try. To that end, first, an equivalent model of the target multi-pair line must be set, and NEXT and FEXT must be analyzed and expressed by controllable parameters. Then, it is an object of the present invention to clarify the specific numerical conditions for making NEXT and FEXT equal to or greater than a predetermined value, thereby realizing a predetermined crosstalk characteristic.

In order to solve the above problems, the present invention assumes a crosstalk model as shown in FIGS. The former corresponds to electromagnetic coupling, and the latter corresponds to electrostatic coupling. In both figures, line 1/2 is an inductive pair and line 3/4 is an induced pair. There are stray capacitance and mutual inductance between the two pairs of round-trip lines. _Let each be M _ij and C _ij . Subscripts i and j correspond to tracks 1 to 4. When a signal current flows through the induction pair in the direction of the solid line, a crosstalk current flows through the induced pair in the direction of the dotted line due to imbalance between M _ij and C _ij .

The bridge equivalent circuit of FIGS. 7 and 8 is shown in FIGS. Assuming that the balance between the two bridges is substantially maintained, the unbalance between the two line pairs, that is, the mutual inductance M _k and the coupling capacitance C _k can be given by a good approximation as follows.
(Formula 1)
M _k = M ₁₃ + M ₂₄ −M ₁₄ −M ₂₃ → 0
(Formula 2)
C _k = C ₁₃ + C ₂₄ −C ₁₄ −C ₂₃ → 0

These regulate the crosstalk current due to electromagnetic coupling and electrostatic coupling between two pairs of round-trip lines, and these can be expressed as I _Mk and I _Ck .
(Formula 3)
I _Mk = j ω (M _k / 2Z ₀ ) I ₀

(Formula 4)
_{I Ck = j ω (C k} Z 0/8) I 0
Is given.

When the equivalent circuit is further simplified, it can be expressed as shown in FIGS. Here, attention must be paid to the direction of crosstalk current. That is, in the case of electromagnetic coupling in FIG. 11, the direction of crosstalk current is reversed at the near end and the far end, whereas in the case of electrostatic coupling in FIG. 12, the direction is the same. Therefore, the crosstalk currents I _NE and I _{NE at} both ends are given by the following equations from equations 3 and 4.
(Formula 5)
_{I NE = I Ck + I Mk} = j ω (C k Z 0/8 + M k / 2Z 0)
(Formula 6)
I _FE = I _Ck −I _Mk = j ω (C _k Z ₀ / 8−M _k / 2Z ₀ )

  It will be understood from equations 5 and 6 that the crosstalk current at the near end and the far end is in a trade-off relationship. This is a routine experience at the connector adjustment stage, and if the near-end crosstalk attenuation (NEXT) is increased, the far-end crosstalk attenuation (FEXT) decreases, but the reverse phenomenon also occurs.

In the TIA standard, NEXT is more important than FEXT as shown in Table 1.
(Table 1)
┌────────┬───┬──┬─────────┬────────┐
│ Item │ Symbol │ Unit │ Cat.5e Standard │ Cat.6 Standard │
│ │ │ │ (frequency 100MHz) │ (frequency 100MHz) │
├────────┼───┼──┼─────────┼────────┤
│Near-end crosstalk attenuation │NEXT │dB │ 43.0min │ 54.0min │
├────────┼───┼──┼─────────┼────────┤
│Near-end crosstalk attenuation │FEXT │dB │ 35.1min │ 43.1min │
├────────┼───┼──┼─────────┼────────┤
│ Return loss │ RL │dB │ 20.0min │ 25.0min │
├────────┼───┼──┼─────────┼────────┤
│ Insertion loss │ IL │dB │ 0.4max │ 0.2max │
└────────┴───┴──┴─────────┴────────┘

For the time being, I _NE = 0, that is, NEXT → ∞, it is easier than Equation 5 (Equation 7)
| M _k / C _k | = (Z _0/2 ) ²
The following conditional expression is obtained.

  That is, it can be seen that when the square root of the ratio between the coupling capacitance and the coupling mutual inductance becomes half of the line impedance, the near-end crosstalk becomes zero (NEXT → ∞). This is the specific condition for NEXT maximization found in the present invention.

When the FEXT value at this time is obtained, FEXT = 20log | I ₀ / I _FE |. Therefore, by reflecting Equation 7 in Equation 6, the following equation can be obtained.
(Formula 8)
FEXT = −20log ₁₀ {ω | M _k | / Z ₀ }

When obtaining the condition of M _k to secure the FEXT required value at the predetermined frequency f from the above equation,
(Formula 9)
2 πf ｜ M _k ｜ / Z ₀ ≦ 10 ^{-FEXT / 20}
It becomes. That is, if Equation 7 and Equation 9 are satisfied, it will be understood that NEXT is infinite and the FEXT value is secured to a predetermined value or more. However, although NEXT will be infinite, the NEXT value will be slightly reduced due to the influence of the series inductance in the line, but the conditions that should be aimed for in the design are as described above.

Here, when the near-end crosstalk attenuation NEXT and the far-end crosstalk attenuation FEXT in the general case are obtained,
(Formula 10)
_{NEXT = -20log 10 {ω (C} k Z 0/8 + M k / 2Z 0)}
(Formula 11)
FEXT = −20log ₁₀ {ω (C _k Z ₀ / 8− M _k / 2Z ₀ )}
The following formula is obtained. FIG. 13 is a graph with Z ₀ = 100 Ω and f = 100 MHz. The horizontal axis is the coupling capacitance C _k , the vertical axis is the crosstalk attenuation, and the coupling mutual inductance M _k is a parameter. According to the figure, when C _k is small, both NEXT and FEXT take a constant value. As C _k increases, NEXT rises, reaching a C _k value that satisfies Equation 7, and becomes infinite. After that, it decreases with an inclination of 6 dB octave. FEXT gradually decreases as C _k increases, and after NEXT infinity, decreases with an octave slope of 6 dB.

Looking at the parameter M _k , it can be seen that the smaller M _k is, the larger both NEXT and FEXT are. Conversely, when M _k is determined, NEXT can be made infinite by C _k , but FEXT cannot be extended beyond a certain value. This is a trade-off.

FIG. 14 is a graph showing the expressions 11 and 12 in another expression. The horizontal axis represents NEXT and FEXT on the vertical axis, and M _k and C _k were scaled in the graph. Thus, given NEXT and FEXT, predetermined M _k and C _k can be obtained. Conversely, if M _k and C _k are known, NEXT and FEXT at that time can be read. This is named a crosstalk chart.

Mutual inductance M _k and coupling capacitance to achieve the specified NEXT and FEXT values
The numerical specific requirements to be satisfied by C _k became clear. As will be described later, it is easier to control the coupling capacitance than the mutual inductance between lines on the substrate. Therefore, in designing the connector, the signal path between the jack and the clamp terminal is first drawn on the board so that _Mk is minimized, and then the coupling capacity between the pairs is controlled to a predetermined value.

  In order to keep the mutual inductance Mk between pairs subject to crosstalk compensation small, it is also effective to create a new Mk on the board, but before that, the Mk of each jack and terminal, which is another device, is effective. It is a more effective means to evaluate and select a terminal arrangement in which both cancel each other with opposite characteristics.

  Prior to a specific embodiment, a method for controlling mutual inductance and coupling capacitance between lines on a substrate will be described.

The mutual inductance between the reciprocating lines can be obtained from the interlinkage ratio of the magnetic flux generated by the closed circuit (Shinzo Takeyama, “Electromagnetism Phenomenon”, Chapter 15, Maruzen 18). Assuming that there are two pairs of reciprocating lines such as (1) (2) and (3) (4) as shown in FIG. 15, the mutual inductance _Mk between the pairs is given by the following equation.
(Formula 12)
Mk = (μ ₀ l / 2π) log _e {(1 + α) (1−α)} (H)
Here (Formula 13)
α = a / b
a is a line interval, b is a pair interval, l is a line length, and μ ₀ is a vacuum permeability. As can be seen from Equations 12 and 13, the polarity of M _k can be reversed by exchanging the forward path and the return path of one of the round-trip lines. Fig. 16 shows the results of experimental verification using test pieces. The calculated value and the measured value are in good agreement. From the above, it will be understood that M _k on the substrate can be controlled by the line spacing, the distance between line pairs, the line length, and the like.

Next described control of the coupling capacitor C _k. C _k control can be performed relatively easily by adding a capacitance between predetermined lines. The type of additional capacitance can be a plate type with the substrate sandwiched, but the interdigital type that can be configured on the same plane is easier to adjust. FIG. 17 shows the results of experimental determination of the number of digital stages and the capacitance value using a test piece. Capacitance control on the order of several pF is possible.

FIG. 18 shows the characteristics of the crosstalk model that was created and verified. The solid line is the calculated value of the crosstalk attenuation obtained from the measured values of M _k and C _k , and the plot is the actual measurement value. Since both show a relatively good agreement, it is understood that the above-described crosstalk analysis and design requirements are valid.

As an actual example, the result of converting the conventional embedded telephone rosette as shown in FIGS. 3 and 4 into Cat.5e only by changing the circuit board will be described. Unlike the normal LAN connector (Fig. 6), this connector is unique in that it uses the aforementioned S-type spring with a release mechanism at the clamp terminal and can be retried after termination. Since the metal spring holders face each other with an area of 10 mm ² or more at 1 mm intervals as shown in FIG. 5, it was expected that the electromagnetic boundary conditions were complicated. Actually, with the use of the IDC terminal, it was easy to obtain the characteristics only by the capacitance balance, but this connector requires a circuit design that also takes into account the mutual inductance as in the present invention.

  In the LAN connector, terminals 12, 36, 54, and 78 of 8-pole 8-core are paired. In the plug and jack sections, the terminal numbers are arranged in order from 1 to 8, so that the pair 54 straddles the pair 54. Therefore, it is well known that the admittance balance between 36/54 pairs is the worst and the biggest bottleneck in crosstalk compensation. Therefore, for the sake of simplicity of explanation, the crosstalk compensation between 36/54 is described below.

19 and 20 show substrate circuit patterns in the embodiment. The coupling mutual inductance Mk between 36/54 was 0.57nH. The optimum value of the coupling capacitance according to the above design requirements is −0.23 pF as calculated from Equation 7 with the characteristic impedance Z ₀ = 100 Ω. By adding an interdigital capacitor between each line, the coupling capacitance was set close to the above value, and the additional capacitance was finely adjusted so that NEXT was maximized. As a result, the NEXT value at 100 MHz was 63 dB. The C _k value at that time was −0.60 pF. NEXT calculated from M _k and C _k, the calculated value of the FEXT (51dB, 44dB) and measured values (63dB, 54dB) are relatively good agreement.

  The optimal value of Ck, NEXT and FEXT are 0. There is a deviation of about several pF or 10 dB, but this is due to the measurement accuracy of Mk and Ck, mismatch of characteristic impedance, etc., and can be regarded as a good response in the high frequency range of the order of 100 MHz.

  For terminals other than 36/54, the Mk of the modular jack and the Mk of the clamp terminal are selected so that the opposite polarities cancel each other, so Mk can be kept as low as 1nH at the maximum. It was. As a result, the frequency characteristics of NEXT and FEXT between all pairs are as shown in Figs. 21 and 22, respectively. The worst-case margin in the entire frequency range of 1MHz to 100MHz is about 5 dB for both NEXT and FEXT compared to the Cat.5e standard. It has been secured.

Perspective view of conventional exposed LAN connector unit Same as above, exploded perspective view Perspective view of a conventional embedded LAN connector observed from the modular jack side Same as above, perspective view observed from clamp terminal side Same as above, exploded perspective view Perspective view of conventional 110-type LAN connector Crosstalk model by electromagnetic coupling regarding means for solving the problems of the present invention Same as above, crosstalk model with stray capacitance Same as above, equivalent circuit representation of crosstalk model by electromagnetic coupling Same as above, bridge equivalent circuit representation of crosstalk model by stray capacitance Same as above, equivalent circuit with simplified crosstalk model by electromagnetic coupling Same as above, equivalent circuit with simplified crosstalk model due to stray capacitance Same as above, a graph showing the calculated value of coupling capacitance-crosstalk characteristics when the coupling mutual inductance is used as a parameter Same as above, crosstalk chart The figure which shows the test piece example same as above Same as above, line length vs. coupled inductance characteristics Same as above, graph showing interdigital capacitor stage-capacitance characteristics Same as above, coupling capacitance-crosstalk characteristics measurement (coupled mutual inductance constant) Part surface pattern diagram of substrate of embedded telephone rosette in the embodiment of the present invention Same as above, solder surface pattern diagram of embedded telephone rosette board Same as above, graph showing NEXT characteristics of embedded telephone rosette Same as above, graph showing FEXT characteristics of embedded telephone rosette

Explanation of symbols

1 Exposed type modular jack 2 Same as above, Clamp case 3 Same as above, Cable core release button,
4 Same as above, Board 5 Same as above, Base 6 Same as above, Cover 7 Modular jack of embedded type rosette 8 Same as above, Clamp terminal 9 Same as above, Cable core release button 10 Same as above, Board 11 Same as above, Base 12 Same as above, Cover 13 110 type LAN connector 14 110 type LAN connector modular jack 15 110 type clamp terminal

Claims (3)

In a multi-channel communication transmission system in which the characteristic impedance of each round-trip line pair is Z ₀ , when the minimum values of the near-end crosstalk attenuation between channels and the far-end crosstalk attenuation at the signal angular frequency ω are NEXT and FEXT, respectively, Pair coupling capacitance C _k and coupling mutual inductance M _k
C _k M _k <0
And
_{| C k | ≦ 4 (10} -NEXT / 20 -10 -FEXT / 20) / 2 ω
_{| M k | ≧ Z 0 (} 10 -NEXT / 20 +10 -FEXT / 20) / ω

A communication cable or a communication modular rosette including a communication modular jack, a clamp terminal, or both. Multi-channel communication transmission system consisting of two or more components such as modular jacks, clamp terminals, or circuit boards that electrically couple them, and requiring near-end crosstalk attenuation greater than far-end crosstalk attenuation The coupling capacitance C _k and the coupling mutual inductance M _k between the channel pair

C _k = -4M _k / Z ₀ ²

For the required value FEXT of the far-end crosstalk attenuation at a predetermined frequency f.

2 πf ｜ M _k ｜ / Z ₀ ≦ 10 ^{-FEXT / 20}

A communication modular rosette including a communication cable, a communication modular jack, a clamp terminal, or both, characterized in that the following inequality is established. A modular rosette for multi-channel communication, wherein a terminal arrangement is selected so that the mutual inductance between the pair of channels in the modular jack and the mutual inductance between the clamp terminals corresponding to each other cancel each other with opposite polarity.