JP2005243992A - Tuning circuit employing tunnel junction element - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a tuning circuit for employing two 1-wavelength tunnel junction elements and a microstrip line with a half wavelength interconnecting the elements. <P>SOLUTION: The tuning circuit employing the tunnel junction elements comprises the two 1-wavelength tunnel junction elements each of which comprises upper and lower electrodes made of a superconducting material and a tunnel junction formed between them and wherein no overhang exists in the upper electrode at least in the widthwise direction of the junction; and the half wavelength microstrip line interconnecting the elements. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

  The present invention relates to a tuning circuit using tunnel junction elements, and more particularly to a tuning circuit using two one-wavelength tunnel junction elements.

A tunnel junction (SIS junction) having a superconductor-insulator-superconductor sandwich structure (SIS structure) has a large capacitance in structure and short-circuits a high-frequency signal by itself. Therefore, in order to efficiently couple the input signal to the SIS junction, it is necessary to reduce the junction size and further integrate a tuning circuit for removing the junction capacitance. In such a tuning circuit, in principle, the tuning can be fractional bandwidth Delta] f / f ₀ is limited by 1 / ωC _J R _N of SIS junction. Here, omega is the angular frequency, the capacitance of C _J is SIS junction, R _N is normal resistance. Therefore, in order to ensure 20% as a ratio band, the value of .omega.C _J R _N product at the center frequency f ₀ is needed to be about 5. .omega.C _J R _N product is strongly dependent on the critical current density J _C of SIS junction, are related by _{J C = ωC S I C R} N / (ωC J R N). Here, C _S is a capacitance per unit area of the SIS junction, and I _C is a critical current. For example, assuming that the junction capacitance per unit area is 100 fF / μm ² , Band10 (789 to 950 GHz), which is the highest frequency band of ALMA (Atacama Large Millimeter-Wave Submillimeter Wave Interferometer), uses approximately 20 kA when using an Nb junction. / cm ^2, a high critical current density of NbN using bonding when about 40 kA / cm ² is required.

In the current SIS junction fabrication technology, the higher the critical current density of the junction, the more the electrical characteristics of the junction tend to deteriorate, which causes an increase in noise temperature due to subgap leakage current and the like. Furthermore, in the ultra-high frequency region where the wavelength is extremely short, such as Band 10, the tuning circuit is also shortened, it becomes difficult to scale down by the conventional design method, and the junction size needs to be about sub μm ^2. It has been difficult to realize a mixer using a SIS junction element having noise and broadband characteristics.

Therefore, the present inventor treats the SIS junction as a distributed constant circuit as a tuning circuit using a relatively large SIS junction in the terahertz band, and configures a resonator with an elongated junction, so that the junction capacitance can be easily increased by the junction itself. A SIS mixer that can be tuned is proposed. However, according to studies of the present inventors, in such a SIS mixer, operating band is limited by conventional manner approximately 1 / ωC _J R _N, is still necessary SIS junction of the high critical current density in a broadband operation Met. The present inventors have further focused on the nature of the distributed constant type tunnel junction, by using a plurality of resonant circuits, it proposes a SIS mixer that can obtain a wide band ratio than 1 / ωC _J R _N (For example, refer to Patent Document 1).
JP 2003-218415 A

As described above, the present inventors focused on the nature of the distributed constant type tunnel junction, and proposed SIS mixer capable of obtaining a wide band ratio than 1 / ωC _J R _N by using a plurality of resonant circuits ing. However, according to the study of the present inventor, the design of such a SIS mixer depends on the experience of the designer, and therefore the design result is not always the optimum design value (optimum solution). It was. Therefore, the present inventor has studied a design apparatus that can design a tuning circuit optimized for low noise and wideband characteristics. In this design apparatus, the configuration of the tuning circuit to be processed is limited to two half-wavelength tunnel junction elements and a half-wavelength microstrip line connecting between the two half-wavelength tunnel junction elements. An optimum circuit parameter that can realize the input performance is obtained only by inputting the loss and the specific band.

  However, in the course of studying this design apparatus, the present inventor has found the following problems.

That is, for the broadband operation of the tuning circuit composed of the distributed constant type SIS junction and the microstrip line connecting between them, the characteristic impedance Z _J of the SIS junction and the characteristic impedance Z _{m of the} microstrip line are The ratio Z _J : Z _m needs to be increased. Usually, the upper limit of the value of Z _m is about 30Ω, and therefore, wide band operation cannot be realized unless Z _J is made extremely small (for example, about 1.3 to 1.4Ω). By the way, as the superconductors (S) above and below the SIS junction, NbN (niobium nitride) is preferable because it has a higher critical temperature than Nb (niobium). However, since NbN has a large inductance component and a large characteristic impedance, the value of Z _J cannot be realized. Therefore, when NbN is used as the upper and lower electrodes 102 and 105 ′, as shown in FIG. 14A, the upper electrode 105 ′ is formed to overhang in the width direction of the SIS junction. That is, the upper electrode width W _s ′ is formed wider than the junction width W _J. Thereby, the characteristic impedance of the SIS junction is set to a small value as described above (for example, about 1Ω) by the action of the parasitic capacitance in the overhang portion (see the dotted line a in FIG. 14B). In FIG. 14A, the insulating substrate is omitted, 103 is an interlayer insulating film, and 104 is a tunnel insulating film (see FIG. 1 and the like).

However, it has been found that when such an overhang exists, the phase velocity changes depending on the width of the SIS junction as indicated by the solid line a in FIG. In addition, the arrow in a figure shows the vertical axis | shaft of the said curve. Therefore, it has been found that even if two half-wavelength SIS junction elements are designed, the half-wavelength may not be maintained at the time of manufacture due to the intentionally formed overhang for the above-described reason, and the optimum design may not be achieved. . On the other hand, it was found that when no overhang exists, the phase velocity does not change depending on the width of the SIS junction, as indicated by the solid line b in FIG. However, if there is no overhang, when NbN is used as the upper and lower electrodes, the characteristic impedance is large (dotted line b in FIG. 14 (b)), so that the necessary Z _J value cannot be realized. It is extremely difficult to use high NbN. Therefore, the present inventor makes the characteristic impedance of the SIS junction half that of the SIS junction having the same junction width by setting the junction length of the SIS junction to one wavelength that is twice (or an integer multiple) of the half wavelength. We thought that it would be possible to achieve characteristic impedance and maintain low noise and broadband characteristics with the same circuit parameters.

  An object of the present invention is to provide a tuning circuit using two one-wavelength tunnel junction elements and a half-wavelength microstrip line connecting between them.

  The tuning circuit using the tunnel junction element of the present invention is two one-wavelength tunnel junction elements, comprising an upper electrode and a lower electrode made of a superconductor, and a tunnel junction formed between them. The tunnel junction element has no overhang of the upper electrode in the width direction of the junction, and a half-wavelength microstrip line connecting between them.

According to the tuning circuit using the tunnel junction element of the present invention, a configuration including two one-wavelength tunnel junction elements and a half-wavelength microstrip line connecting between them is adopted. As a result, the upper electrode has a shape that does not overhang in the width direction of the SIS junction (the junction width W _J and the upper electrode width W _s are substantially equal, see FIG. 2), and the phase velocity depends on the junction width. The characteristic impedance Z _J of the SIS junction is made extremely small (for example, about 1.3 to 1.4Ω) even when there is no overhang. Thus, Z _J and microstrip line characteristic impedance Z _m and the ratio Z _J: Increase the Z _m, it is possible to obtain an optimized tuning circuit for low-noise and broadband characteristics. In such a tuning circuit, NbN having a high critical temperature can be used as superconductors above and below the SIS junction because the inductance component is large and the characteristic impedance is large. Further, for such a tuning circuit, an optimum circuit parameter capable of realizing the input performance can be obtained based on the reflection loss and the specific band in the signal source.

(Configuration of tuning circuit)
The tuning circuit of the present invention is composed of two one-wavelength (d _J ) SIS junction elements J1 and J2 shown in FIG. 1 and a half-wavelength (d _m ) microstrip line M connecting between them. FIG. 1A is a plan view of the tuning circuit, FIG. 1B is a cross-sectional view of the tuning circuit, and shows a cross section taken along the 3b-3b cutting line in FIG. FIG. 1A corresponds to FIG. 5A described later. 2A is a perspective view showing a cut surface perpendicular to the 3b-3b cutting line of the SIS junction element J1 (and J2), and FIG. 2B is a 3b-3b of the microstrip line M. It is a perspective view which shows collectively the cut surface orthogonal to a cutting line.

The reason for using the two SIS junctions J1 and J2 is as follows. That can be obtained as described above, a wide enough bandwidth ratio than 1 / ωC _J R _N by using multiple SIS junction. However, on the other hand, when three or more SIS junction elements are used, when the Josephson current unnecessary for tuning is erased by applying a magnetic field, the SIS junction elements cannot cancel each other. If two are used, the Josephson currents can be canceled with each other by using SIS junctions having the same shape. Therefore, the two SIS junction elements J1 and J2 of the present invention have the same shape.

The reason why the SIS junctions J1 and J2 are set to one wavelength is that the junction length is doubled (or an integer multiple) in the vicinity of a half-wavelength frequency, in order to use the formula (4) described later. This is because the characteristic impedance is reduced (halved) after using the approximation of moving the pure resistance portion of the SIS junction J1 close to the signal source to the signal source side (Z _s side). The reason why the microstrip line M has a half-wavelength is that the pure resistance portion of the SIS junction J1 close to the signal source is placed on the signal source side (Z _s side) using the equation (4) described below in the vicinity of the half-wavelength frequency. This is because the approximation moving to () is used.

Two SIS junction elements J1, J2, to treat as a transmission line having a loss due to quasiparticle tunneling current as described above, and, in order to obtain a wide enough bandwidth ratio than 1 / ωC _J R _N, The shape is elongated. Here, elongate means that the shape can maintain a single propagation mode. Since the propagation mode in this example is the TEM mode, the length (junction length) d _J of the SIS junction elements J1 and J2 is one wavelength so that other propagation modes do not occur. The bonding width (dimension in the direction orthogonal to d _J ) is, for example, 1/6 wavelength or less.

  As shown in FIG. 1B, the SIS junction elements (tunnel junction elements) J1 and J2 include an upper electrode (S) and a lower electrode (S) made of a superconductor, and an insulating film ( I) and a tunnel junction. The microstrip line M includes an upper line and a lower line (and its interlayer insulating film). The upper electrode and the lower electrode of the SIS junction element and the upper line and the lower line of the microstrip line are made of the same layer.

  In this example, the lower electrodes of the SIS junction elements J1 and J2 and the lower line of the microstrip line M are composed of a first NbN (niobium nitride) layer 102 which is a superconductor. The first NbN layer 102 is a ground electrode (ground plane) formed to a thickness of 200 nm by epitaxial growth on the single crystal MgO substrate 101 which is an insulator. As shown in FIG. 1A, the width of the ground electrode 102 is sufficiently wide (about 5 to 10 times) wider than the widths of the SIS junction elements J1 and J2 and the microstrip line M. Therefore, the microstrip line M becomes a coplanar waveguide. On the first NbN layer 102, an MgO film 103, which is an interlayer insulating film formed to a thickness of 200 nm by epitaxial growth, is formed. After the opening of the tunnel portion is formed in the MgO film 103, the tunnel insulating film 104 made of the MgO film is formed extremely thin (for example, 1 nm) on the exposed first NbN layer 102 by epitaxial growth. The upper electrodes of the SIS junction elements J1 and J2 and the upper line of the microstrip line M are composed of the second NbN layer 105 which is a superconductor. The second NbN layer 105 is a wiring and an electrode formed to a thickness of 400 nm by epitaxial growth on the MgO interlayer film 103. The SIS junctions (tunnel junctions) J1 and J2 are made of NbN / MgO / NbN. Therefore, it is an all NbN tuning circuit.

In this example, as shown in FIG. 2A, the upper electrodes 105 of the SIS junction elements J1 and J2 have a shape with no overhang in the direction of the junction width. That is, the junction width W _J and the upper electrode width W _s are formed to be substantially equal. This is to prevent the phase velocity from changing depending on the junction width. In the junction length direction, the upper electrodes 105 of the SIS junction elements J1 and J2 have a shape without an overhang, but an overhang may be formed. On the other hand, the upper line 105 of the microstrip line M is formed with a width W _M (≠ W _s ) as shown in FIG. In FIG. 2, the substrate 101 which is an insulator is omitted.

(Design principle)
Next, the principle used in the design of a tuning circuit using the tunnel junction element of the present invention will be described with reference to FIGS.

In order to follow this design principle, first, the configuration of the tuning circuit is once replaced as shown in FIG. That is, the tuning circuit of the present invention is composed of the two one-wavelength SIS junction elements J1 and J2 and the half-wavelength microstrip line M connecting between them as described above. This can be illustrated as in FIG. On the other hand, a tuning circuit to which a design principle for obtaining an optimum circuit parameter can be applied includes two half-wavelength SIS junction elements J1 and J2 and a half-wavelength microstrip line M connecting between them. This can be illustrated as in FIG. Therefore, the tuning circuit in FIG. 3A is temporarily replaced with the tuning circuit in FIG. 3B to obtain circuit parameters, and then replaced (returned) with the tuning circuit in FIG. The reason why it can be replaced will be described later. 3 'a characteristic impedance of Z _J (J1' SIS junction element J1 of (b) When), 3 SIS characteristic impedance Z _J (J1) of the junction element J1 (and J2) of (a) is a wavelength Since it is doubled from one half wavelength to one wavelength, Z _J (J1 ′) / 2. The same applies to the SIS junction elements J2 ′ and J2. The characteristic impedance of the microstrip line M of the half-wave is Z _m, there is no change de 3 (a) and FIG. 3 (b).

As described above, after the configuration of the tuning circuit is temporarily replaced, the tuning circuit is designed as follows. That is, a SIS junction having a particularly elongated structure among tunnel junction elements can be expressed using transmission line parameters as shown in FIG. This is because the SIS junction having an elongated structure constitutes a microstrip line and can be handled as a transmission line having a loss due to a quasiparticle tunnel current, as is well known. In FIG. 4A, the characteristic impedance of the SIS junction is Z _J , the length (junction length) is d _J , and the propagation constant is γ _J = α _J + jβ _J. α _J is an attenuation constant due to the loss, and β _J is a phase constant.

When the transmission line is assumed to be an open end, the input impedance Z _in is
Z _in = Z _J coth (γ _J d _J ) (1) If this transmission line is
sin (α _J d _J ) ≈α _J d _J ... If the line satisfies the equation (2), the equation (1) is
_{Z in = Z J × (Z} J / α J d J · cos (β J d J) + jZ J · sin (β J d J)) / (Z J · cos (β J d J) + jZ J / α J d _J · sin (β _J d _J )) ··· (3) can be rewritten.

This equation (3) represents the input impedance of the circuit shown in FIG. Therefore, under the conditions described above, the open-ended distributed constant type SIS junction shown in FIG. 4A is at the end of the lossless transmission line shown in FIG. 4B (that is, the line with α _J = 0). It can be said that it is equivalent to a circuit having a pure resistance load Z _J / α _J d _J. From the above, it can be considered that the reactance frequency fluctuation component of the distributed constant SIS junction is simply due to the lossless transmission line connected to the pure resistance.

  As a method for efficiently tuning such reactance frequency fluctuation components, use of a half-wave bandpass filter structure can be considered. In general, the half-wave bandpass filter is composed of a low impedance line and a high impedance line. Therefore, a SIS junction is assigned to the low impedance line, and a microstrip line is assigned to the high impedance line. In designing such a filter, a well-known filter design theory such as Chebyshev theory can be employed. Since Chebyshev theory is similar to a broadband impedance matching circuit, as is well known, the impedance characteristics can be controlled in a wide band by the filter circuit structure, which is suitable for the filter structure of the present invention (RE Collin: Foundations for Microwave Engineering (McGrow-Hill, New York, 1992) 2nd ed., Chap. 5, p.303.).

(Design method)
In the design of the tuning circuit, as a filter circuit structure, two SIS junction elements (tunnel junction elements) having a half wavelength (length d _J ) are connected as shown in FIG. A tuning circuit having a simplest three-stage band-pass filter structure including a microstrip line having a half wavelength (length d _m , d _J ≠ d _m ) is set. In this tuning circuit design, what is different from the design of a normal bandpass filter is that there is a loss in a part of the transmission line constituting the filter structure, and the input / output impedance is not made the same accordingly. . Therefore, in the design apparatus and method of the present invention, a signal source impedance Z _s different from the termination load resistance is assumed, and the maximum reflection loss and its ratio band between this and the tuning circuit are input as design conditions, and the SIS at that time is input. Find the optimal solution that minimizes the critical current density at the junction. This optimum solution is the characteristic impedance of each filter in the three-stage bandpass filter structure, that is, the SIS junction and the microstrip line.

At this time, the signal source impedance can be freely controlled by adding a ¼ wavelength impedance matching circuit (T) to such a tuning circuit. Therefore, the value of the signal source impedance Z _s is not limited. Thereby, the optimal solution can be obtained by freely selecting the signal source impedance Z _s .

As described above, a part of such a tuning circuit includes a loss. Therefore, when the circuit is simplified by using an equivalent circuit, the distributed constant SIS junction in the tuning circuit is expressed as a pure resistance by the equation (2). And a lossless transmission line. Furthermore, the phase β _J d _J of the SIS junction is in the vicinity of π, that is, in the vicinity of the frequency (central frequency) at which the distributed constant SIS junction has a half wavelength.
α _J d _J tan (β _J d _J ) << 1 (1) The pure resistance part of the SIS junction close to the signal source may be moved to the signal source side (Z _s side) under the condition satisfying the equation (4). it can. That is, when the SIS junction in the element structure of FIG. 5A is replaced with the simplified equivalent circuit of FIG. 4B, the pure resistance portion of the SIS junction close to the signal source that should be connected between aa ′ is obtained. It can be moved to the signal source side (Z _s side). Therefore, the circuit shown in FIG. 5A can be rewritten to the circuit shown in FIG.

Next, the circuit constant is normalized by the termination load resistance R _L , and the circuit is divided into three as shown in FIG. The circuit of FIG. 5C is referred to as a simple circuit model. In the simple circuit model, the load side from aa ′ has a normal three-stage bandpass filter structure including a lossless transmission line. This filter part is designed by applying Chebyshev theory, but it is necessary to follow conditions that minimize the critical current density of the SIS junction.

That is, the standardized characteristic impedance of the half-wave line connected to the terminating load resistors R _L and aa ′ is α _J d _J. Since α _J is an attenuation constant, the lower the critical current density of the SIS junction, the lower the normalized characteristic impedance. When the line satisfies the equation (2), α _J d _J << 1, so that the transmission line having a lower critical current density of the SIS junction has a higher ratio with the normalized termination load resistance of 1Ω. According to the filter theory, as is well known, the larger the ratio, the larger the maximum reflection coefficient of the filter characteristics (the above-mentioned RE Collin paper). Accordingly, when the reflection coefficient of the circuit when the signal source impedance for the filter portion (a-a ′ to the load side (right side) circuit) is 1Ω is set to ρ, the maximum value is obtained under the condition where ρ is given. If this is taken, the condition for obtaining the minimum critical current density is obtained.

To find this solution, the impedance of the tuning circuit is first described using ρ. Assuming that the input impedance on the load side from aa ′ is Z _f , the relational expression with ρ is well known,
ρ = | (1−Z _f ) / (1 + Z _f ) | (5) From now on, Z _f can be expressed using ρ.
Z _f = (1 + ρ) / (1-ρ) or Z _f = (1-ρ) / (1 + ρ) (6) Input impedance Z _in of the tuning circuit, as shown in FIG. 5, since 1Ω to Z _f are connected in parallel,
Z _in = (1 ± ρ) / 2 (7) Now, considering that the reflection coefficient between the signal source and the tuning circuit is less than a certain value of ρ _s , the relational expression is described using the signal source impedance Z _s ′ normalized by R _L :
ρ _s ≦ | (Z _s ′ −Z _in ) / (Z _s ′ + Z _in ) | (8) is obtained. Thus, [rho is 'a function of, Z _s that maximizes the [rho' Z _s so that is obtained.

(Tuning circuit design)
A tuning circuit was designed that satisfies the condition that the reflection loss with respect to the signal source impedance with respect to the tuning circuit is -10 dB or less and the ratio band is 20%. That is, “reflection loss−10 dB” and “specific bandwidth 20%” are design conditions.

Since ρ _s = 0.314 is obtained from the reflection loss of −10 dB, the equations (7) and (8) are graphed as shown in FIG. The shaded area satisfies the condition of the reflection coefficient ρ. The maximum value of ρ is when Z _s ′ = 0.409, and the maximum value is 0.575. The maximum value “0.575” of this reflection coefficient is obtained.

Therefore, if the signal source impedance Z _s ′ = 0.409 of the tuning circuit with the circuit constant normalized by the termination load resistance R _L , while achieving a reflection coefficient ρ _s = 0.314 or less with respect to the tuning circuit, FIG. 5 (c), the reflection coefficient ρ in the filter circuit structure on the load side can be set to a maximum value of 0.575. From this result, in applying the Chebyshev theory in the design of the filter portion, the optimum solution of the tuning circuit can be obtained if the filter design is performed under the conditions of the reflection coefficient ρ = 0.575 and the relative bandwidth Δf / f ₀ = 20%. That is, the design conditions of “reflection coefficient ρ = 0.575” and “specific band Δf / f ₀ = 20%” are obtained.

  Therefore, the circuit parameters are calculated by applying the design principle of the half-wave filter having the well-known broadband Chebyshev characteristic (the aforementioned R. E. Collin paper). Usually, the filter characteristic is described by the relationship between the phase θ and the reflection coefficient ρ. The basic three-stage half-wave Chebyshev filter characteristics are designed so that the reflection coefficient becomes zero at a total of three points of the center frequency (π in phase) and the frequencies before and after that. FIG. 7A shows ρ-θ characteristics representing typical characteristics of a filter having a maximum reflection coefficient of 0.575. FIG. 7B shows circuit parameters of the filter. The filter design is based on the design of a broadband 1/4 wavelength impedance matching circuit, and the center phase is described as π / 2, so the specific band is 40%. Since the phase at the filter is 2θ, the specific band is 20%.

Therefore, as circuit parameters of the tuning circuit shown in FIG. 5A, as shown in FIG. 7B, R _L : Z _J : Z _m : Z _s = 1: 0.12: 2.69: 0. 41 is obtained. This is a circuit parameter (impedance ratio of the filter portion) of the tuning circuit of FIG.

When the frequency characteristic of the tuning circuit is calculated, the impedance characteristic shown in FIG. 8A and the reflection loss shown in FIG. 8B are obtained. The frequency is standardized at the center frequency. From FIG. 8, it can be seen that the specific band of 20% is achieved with the signal source impedance and the reflection loss of the tuning circuit of -10 dB or less as designed. As can be seen from the Smith chart of FIG. 8A, by selecting a low signal source impedance, the normal Chebyshev characteristic (the intersection of the impedance trajectory passes through the center of the chart, that is, the reflection coefficient is 0 at the frequency of three points). As a result, it can be seen that the maximum value of ρ in FIG. 5C is obtained. From this, the minimum critical current density is obtained and α _J d _J = 0.12. However, in practice, the in-line wavelength differs depending on the magnetic field penetration length depending on the SIS junction material and the capacitance per unit area. Therefore, the minimum critical current density was obtained as follows in the actual device design. .

(Mixer design)
As a mixer having a tuning circuit composed of two distributed constant SIS junctions, a quasi-optical mixer composed of a non-reflective layered MGO super hemisphere lens and a twin slot antenna is designed as an input optical system. Therefore, FIG. 1 shows the tuning circuit of the mixer chip. A feeding point P of a twin slot antenna (not shown) is arranged at the center using a coplanar waveguide (see FIG. 1). The center frequency is designed to be 870 GHz, and the antenna impedance in the vicinity is about 65Ω (for example, J. Zmuidzinas, NG Ugras, D. Miller, M. Gaidis, HG LeDuc, “Low-noise slot antenna SIS mixers,” IEEE Trans. Appl. Supercond., Vol. 5, pp. 3053-3056, 1995. / or M. Gaidis, HG LeDuc, M. Bin, D. Miller, JA Stern and J. Zmuidzinas, “Characterization of low- noise quasi-optical SIS mixers for the submillimeter band, "IEEE Trans. Microwave Theory Tech., vol. 44, pp. 1130-1139, 1996.). The tuning circuit is integrated with the central conductor of one coplanar waveguide as a ground plane. A ¼ wavelength impedance transformer T for matching with the antenna impedance is added to the tuning circuit. As described above, the mixer is manufactured by the epitaxial NbN / MgO / NbN technology using the single crystal MgO substrate, and the tuning circuit is composed of the NbN / MgO / NbN tunnel junction and the NbN / MgO / NbN microstrip line. (For example, A. Kawakami, Z. Wang, and S. Miki, “Low-loss epitaxial NbN / MgO / NbN trilayers for THz applications,” IEEE. Trans. Appl. Supercond., Vol. 11, pp. 80- 83, 2001. / or A. Kawakami, Z. Wang, and S. Miki, “Fabrication and characterization of epitaxial NbN / MgO / NbN Josephson tunnel junctions,” J. Appl. Phys., Vol. 90, pp. 4796 -4799, 2001.).

  The tuning circuit was designed using the analysis result of the simple circuit model, with a relative bandwidth of 20% (174 GHz) and a reflection loss of −10 dB or less with respect to the center frequency. FIG. 9 shows design parameters used in the design. Although the values of these design parameters are mainly based on actually measured values, the capacitance per unit area of the epitaxial NbN / MgO / NbN junction is assumed to be the same as that of the epitaxially grown NbN / AlN / NbN junction ( Z. Wang, Y. Uzawa, and A. Kawakami, “High current density NbN / AlN / NbN tunnel junctions for submillimeter wave SIS mixers,” IEEE Trans. Appl. Supercond., Vol. 7, pp. 2797-2800, 1997 .) The calculation of the characteristic impedance and propagation constant of the superconducting microstrip line and SIS junction transmission line necessary for the design of the tuning circuit is described by the present inventors in the literature Y. Uzawa and Z. Wang, Studies of High Temperature Superconductor, ed. AV. Narliker (Nova Science, Hauppauge, NY, 2002) Vol. 43, Chap. 9, p. 255.

First, the value of the minimum critical current density of the SIS junction necessary for satisfying the design condition is obtained as described above. As a calculation example, as shown in FIG. 10, the value of α _J d _J = 0.12 of a SIS transmission line having a width of 1 μm at a center frequency of 870 GHz was plotted against the critical current density. From this figure, a critical current density satisfying α _J d _J = 0.12 is obtained as about 16 kA / cm ² (C _S = 110 fF / μm ² ). Compared with the value of 40 kA / cm ² according to the conventional design method, it can be reduced to half or less.

Thereafter, the replacement from FIG. 3B to FIG. 3A is performed. That is, a low characteristic impedance is realized using the circuit parameters obtained as described above. For a half-wave SIS junction element, as shown in FIG. 4A, the characteristic impedance is Z _J , the length (junction length) is d _J , and the propagation constant is γ _J = α _J + jβ _J. Consider a SIS junction element that is different from the half-wave SIS junction element only in the junction length (one wavelength). The characteristic impedance of this one-wavelength SIS junction element is Z _J , the junction length is 2d _J , and the propagation constant is γ _J = α _J + jβ _J. Then, α _J d _J << 1, the case of β _J d _J ≒ π, the input impedance Z _in of the SIS junction element of the one wavelength _{Z in = Z J coth (2γ} J d J) ≒ Z J / 2 · coth (γ _J d _J ).

FIG. 11 shows the characteristic impedance of the SIS junction element under this condition. From FIG. 11, it can be seen that the real part a and the imaginary part b of the input impedance of such a one-wavelength SIS junction element are equal to the real part c and the imaginary part d of the input impedance of the half-wave SIS junction element, respectively. . Therefore, when α _J d _J << 1, β _J d _J ≈π holds, it can be said that a half-wave SIS junction element is equivalent to a one-wavelength SIS junction element that differs only in the junction length. it can. And, since the input impedance of such a one-wavelength SIS junction element is Z _J / 2 · coth (γ _J d _J ), it is shown in comparison with the formula (1) that of the half-wave SIS junction element. It turns out that it is 1/2. Therefore, the half-wavelength SIS junction element shown in FIG. 3B used for calculating the circuit parameters can be replaced with the one-wavelength SIS junction element shown in FIG.

  FIG. 12 shows an outline of a tuning circuit designed to be close to the circuit parameter (characteristic impedance ratio) obtained as described above. FIG. 12A shows a tuning circuit composed of two one-wavelength SIS junction elements J1 and J2 and a half-wavelength microstrip line M without overhang. That is, it is an example of the tuning circuit of the present invention (FIG. 3A). FIG. 12B shows a tuning circuit composed of two half-wavelength SIS junction elements J1 'and J2' having an overhang and a half-wavelength microstrip line M for comparison. That is, it is an example of the tuning circuit of FIG.

12A, the junction length of the SIS junction elements J1 and J2 is 4.2 μm, the junction width is 0.9 μm, and the upper electrode width W _S is also 0.90 μm (FIG. 2A). )reference). In this example, no overhang is formed even in the direction of the junction length. The microstrip line M has a line length of 35.2 μm and a line width of 1.0 μm. 12B, the junction length of the SIS junction elements J1 and J2 is 2.9 μm, the junction width W _J is 1.3 μm, and the upper electrode width W _S ′ is 3.0 μm (FIG. 14 ( a)). The microstrip line M has a line length of 35.2 μm and a line width of 1.0 μm. That is, there is no change and the microstrip line has a half wavelength.

  In FIG. 13A, curves a and b indicate impedance loci when the load side is viewed from the input side of the tuning circuit of the tuning circuit of FIGS. 12A and 12B, respectively. In FIG. 13 (b), curves a and b show the reflection loss characteristics when the antenna impedance of the tuning circuit of FIGS. 12 (a) and 12 (b) is a constant value of 65Ω. These are standardized at 65Ω. From these, it can be seen that the reactance component having the frequency dependence of the half-wavelength junction placed at the terminal is well compensated by the tuning circuit as described in the above design principle. In addition, by adding a ¼ wavelength impedance transformer T (same as in FIG. 1) to the tuning circuit, a ratio band of 20% or more was obtained. Further, even when the half-wavelength SIS junction is replaced with a one-wavelength SIS junction without overhang, the reflection loss characteristic of the junction can be kept wide as shown in FIG. .

As described above, according to the present invention, according to the tuning circuit using a tunnel junction element, a configuration comprising two one-wavelength tunnel junction elements and a half-wavelength microstrip line connecting between them is adopted. As a result, the phase velocity changes depending on the junction width so that the upper electrode does not overhang in the width direction of the SIS junction (with the junction width W _J and the upper electrode width W _s substantially equal). preventing, and even without overhang by very small characteristic impedance Z _J of SIS junction, Z _J and microstrip line characteristic impedance Z _m and the ratio of Z _J: increase the Z _m, low It is possible to obtain a tuning circuit optimized for noise and broadband characteristics, and since the inductance component is large as superconductors above and below the SIS junction, Despite impedance is large, it is possible to use the critical temperature is high NbN, based on and the reflection loss fractional bandwidth of the signal source, it is possible to obtain an optimum circuit parameters that can realize the input performance.

It is the top view and sectional drawing which show the tuning circuit using a tunnel junction element. It is a perspective view which shows the tuning circuit using a tunnel junction element. It is a circuit block diagram which shows the tuning circuit using a tunnel junction element. It is a design principle explanatory view of a tuning circuit using a tunnel junction element. It is a design principle explanatory view of a tuning circuit using a tunnel junction element. It is a figure which shows the example of a design of the tuning circuit using a tunnel junction element. It is a figure which shows the example of a design of the tuning circuit using a tunnel junction element. It is a figure which shows the example of a design of the tuning circuit using a tunnel junction element. It is a figure which shows the design parameter of the tuning circuit using a tunnel junction element. It is a figure which shows the example of a design of the tuning circuit using a tunnel junction element. It is a substitution explanatory view of a tuning circuit using a tunnel junction element. It is a substitution explanatory view of a tuning circuit using a tunnel junction element. It is a figure which shows the example of a design of the tuning circuit using a tunnel junction element. It is a figure which shows the background of this invention.

Explanation of symbols

DESCRIPTION OF SYMBOLS 101 Single crystal MgO substrate 102 1st NbN layer 103 MgO film 104 Tunnel insulating film 105 2nd NbN layer J1, J2 SIS junction element M Microstrip line T Impedance transformer (impedance matching circuit)

Claims (4)

Two one-wavelength tunnel junction elements comprising an upper electrode and a lower electrode made of a superconductor and a tunnel junction formed therebetween, and at least an overhang of the upper electrode in the width direction of the junction No tunnel junction element,
A tuning circuit using a tunnel junction element, characterized by comprising a half-wavelength microstrip line connecting between them. The microstrip line consists of an upper line and a lower line,
The tuning circuit using a tunnel junction element according to claim 1, wherein the upper electrode and the lower electrode of the tunnel junction element and the upper line and the lower line of the microstrip line are each made of the same layer. . The tuning circuit using a tunnel junction element according to claim 2, wherein the upper and lower electrodes of the tunnel junction element and the upper and lower lines of the microstrip line are formed of an NbN layer. The impedance of the signal source is adjusted by connecting the signal source and a tunnel junction element close to the signal source with a ¼ wavelength microstrip line serving as an impedance matching circuit. A tuning circuit using the tunnel junction element according to Item 1.

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