JP2005094835A

JP2005094835A - Switching power supply

Info

Publication number: JP2005094835A
Application number: JP2003321210A
Authority: JP
Inventors: Masaru Nakamura; 勝中村
Original assignee: Sanken Electric Co Ltd
Current assignee: Sanken Electric Co Ltd
Priority date: 2003-09-12
Filing date: 2003-09-12
Publication date: 2005-04-07

Abstract

<P>PROBLEM TO BE SOLVED: To solve the problem of difficulty to accurately detect, on a primary side, an overcurrent state on the secondary side of a transformer in a switching power supply that comprises a transformer. <P>SOLUTION: The switching power supply comprises a transformer 3 containing a primary coil N1, a secondary coil N2, and a third coil N3. A switch 17 is provided to intermittently apply the primary coil N1 with a DC voltage. A feedback control signal forming circuit 7 is connected to the output terminal of an output stage rectifying/smoothing circuit 5. A current detection resistor 8 is connected in series to the switch 17. A switch-on/off control signal forming circuit 23 generates an on/off-control pulse for the switch 17 based on a feedback control signal and a current detection signal. A reference voltage generating circuit 27 generates a reference voltage Vr that changes inversely proportional to the output of an input voltage detecting circuit 26. An overcurrent protection/detection comparator 28 compares a current detection signal Vi with the reference voltage Vr to output a signal representing overcurrent protection. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

本発明は、変動の可能性を有する直流入力電圧を供給する直流電源に接続されたスイッチング電源装置に関する。
The present invention relates to a switching power supply device connected to a DC power supply that supplies a DC input voltage having a possibility of fluctuation.

従来の典型的なスイッチング電源装置は、交流電源に接続された入力段整流平滑回路と、この入力段整流平滑回路の対の出力端子間に接続されたトランスの１次巻線とオン・オフ用のスイッチとの直列回路と、トランスの２次巻線に接続された出力段整流平滑回路と、この出力段整流平滑回路の出力電圧を検出して出力電圧を一定に制御するための帰還制御信号を形成する手段と、１次巻線とスイッチとの直列回路に流れる電流を検出する電流検出器と、電流検出器から得られる鋸波波形が帰還信号に達した時にスイッチオン終了時点を示す信号を得るためのオン終了時点検出用比較器と、スイッチのオン開始を示すクロック信号を繰返して発生するクロック発生回路と、クロック信号の発生時点からスイッチオン終了時点までの幅を有するパルスを形成してスイッチの制御端子に送るスイッチ制御パルス形成回路とから成る。スイッチング電源装置は、更に、負荷、スイッチ等を過負荷又は過電流から保護するための過電流保護回路を有する。典型的な過電流保護回路は、トランスの１次側に設けられた電流検出器と、過電流レベルを示す基準電圧を発生する基準電圧源と、電流検出器の出力と基準電圧とを比較する過電流検出用比較器とから成り、過電流検出用比較器の過電流を示す信号はスイッチ制御パルス形成回路に供給され、スイッチ制御パルスの発生を禁止する。これによりスイッチがオフになり、スイッチ及び負荷等が過電流から保護される。 The conventional typical switching power supply device includes an input stage rectifying / smoothing circuit connected to an AC power supply, a primary winding of a transformer connected between the output terminals of the pair of input stage rectifying / smoothing circuits, and on / off switching. A series circuit with a switch of the output, an output stage rectifying and smoothing circuit connected to the secondary winding of the transformer, and a feedback control signal for detecting the output voltage of the output stage rectifying and smoothing circuit and controlling the output voltage to be constant , A current detector for detecting the current flowing in the series circuit of the primary winding and the switch, and a signal indicating the switch-on end time when the sawtooth waveform obtained from the current detector reaches the feedback signal Comparator for detecting ON end time, a clock generation circuit that repeatedly generates a clock signal indicating ON start of the switch, and a width from the generation time of the clock signal to the switch ON end time Forming a pulse consisting of a switch control pulse forming circuit for sending to the control terminal of the switch. The switching power supply device further includes an overcurrent protection circuit for protecting a load, a switch, and the like from overload or overcurrent. A typical overcurrent protection circuit compares a current detector provided on the primary side of a transformer, a reference voltage source that generates a reference voltage indicating an overcurrent level, and an output of the current detector and a reference voltage. An overcurrent detection comparator is provided, and a signal indicating the overcurrent of the overcurrent detection comparator is supplied to the switch control pulse forming circuit to inhibit the generation of the switch control pulse. As a result, the switch is turned off, and the switch and the load are protected from overcurrent.

もし、電流検出器がトランスの２次側に設けられているか、又はスイッチング電源装置の入力電圧が一定であれば、上述の過電流保護回路であってもほとんど問題なく過電流保護を達成することができる。しかし、電流検出抵抗をトランスの２次側に設けると、トランスの１次側と２次側との絶縁を図るために構成が複雑になり、また、電流検出抵抗における電力損失が大きくなる。
また、スイッチング電源装置の入力電圧を一に保つことは困難であり、更に入力電圧のレンジ切替が要求されることがある。もし、出力電圧帰還制御方式を採用しているスイッチング電源装置において、入力電圧が変化すると、帰還電圧は入力電圧に反比例的に変化し、スイッチの制御パルスを狭める動作が生じる。このため、入力電圧が所定値よりも高くなると、トランスの２次側の過電流状態を正確に検出することができない。即ち、２次側が過電流状態であっても、過電流検出用比較器から過電流を示す信号を得ることができない。
この問題を解決するための技術として、後記特許文献１に電流検出信号に入力補正電圧を加算することが開示されている。また、後記特許文献２に、入力電圧の変動による過電流保護機能の低下を防ぐために、制御回路の電源電圧を得るためのトランスの３次巻線と電流制御端子との間に過電流補正回路を接続することが開示されている。 If the current detector is provided on the secondary side of the transformer or the input voltage of the switching power supply device is constant, the above overcurrent protection circuit can achieve overcurrent protection with almost no problems. Can do. However, when the current detection resistor is provided on the secondary side of the transformer, the configuration becomes complicated in order to achieve insulation between the primary side and the secondary side of the transformer, and power loss in the current detection resistor increases.
Further, it is difficult to keep the input voltage of the switching power supply unit at one, and there is a case where range switching of the input voltage is required. If the input voltage changes in a switching power supply apparatus that employs the output voltage feedback control method, the feedback voltage changes inversely with the input voltage, and an operation of narrowing the control pulse of the switch occurs. For this reason, when the input voltage becomes higher than a predetermined value, the overcurrent state on the secondary side of the transformer cannot be accurately detected. That is, even if the secondary side is in an overcurrent state, a signal indicating overcurrent cannot be obtained from the overcurrent detection comparator.
As a technique for solving this problem, Japanese Patent Application Laid-Open No. H10-228667 discloses adding an input correction voltage to a current detection signal. Patent Document 2 described later discloses an overcurrent correction circuit between a tertiary winding of a transformer and a current control terminal for obtaining a power supply voltage of the control circuit in order to prevent a deterioration of an overcurrent protection function due to a change in input voltage. Is disclosed.

ところで、入力電圧の変化に拘らず過電流保護を正確に行うことができるスイッチング電源装置の小型化及び低コスト化が要求されている。 Incidentally, there is a demand for downsizing and cost reduction of a switching power supply device that can accurately perform overcurrent protection regardless of changes in input voltage.

発明が解決しようとする課題は、スイッチング電源装置の過電流保護を高精度化、小型化及び低コスト化を容易に達成することができないことである。
特開平６−２１７５３３号公報特開平９− ９３９１６号公報 The problem to be solved by the invention is that the overcurrent protection of the switching power supply cannot be easily achieved with high accuracy, miniaturization and cost reduction.
JP-A-6-217533 Japanese Patent Laid-Open No. 9-93916

上記課題を解決するための本発明は、
変動する可能性を有する直流入力電圧を供給する第１及び第２の直流端子と、
前記第１及び第２の直流端子間に単一又は複数のスイッチを介して接続されたトランスの１次巻線と、
前記１次巻線に電磁結合された２次巻線と、
前記２次巻線に接続された整流平滑回路と、
前記整流平滑回路の直流出力電圧を所望値に制御するための帰還制御信号を形成する帰還制御信号形成回路と、
前記帰還制御信号形成回路と前記スイッチの制御端子との間に接続されたスイッチオン・オフ制御信号形成回路と、
前記スイッチを流れる電流を検出する電流検出手段と、
前記第１及び第２の直流端子間の直流入力電圧を検出する入力電圧検出手段と、
前記入力電圧検出手段の出力電圧レベルの変化に応じて連続的又は段階的に変化する参照電圧を発生する参照電圧発生回路と、
前記電流検出手段と前記参照電圧発生回路とに接続されており且つ前記電流検出手段の出力が前記参照電圧に達した時に前記スイッチのオン制御を禁止する出力を発生する過電流検出用比較器と
を備えていることを特徴とするスイッチング電源装置に係わるものである。
本発明における入力電圧検出手段は、後述する実施例の入力電圧検出回路２６に限ることなく、入力電圧供給端子１９とグランド端子２２のみ、又はこれ等に接続された入力電圧検出ラインのみでよい。また、分圧用の第１及び第２の抵抗R1、R2を省いて入力電圧を非分圧で参照電圧発生回路２７に送ることができる。また、第１及び第２の抵抗R1、R2の分圧回路の代わりに半導体等を使用した非直線的分圧回路を設けること、又は参照電圧発生回路２７に分圧機能を含めることができる。 The present invention for solving the above problems is as follows.
First and second DC terminals for supplying a DC input voltage having a possibility of fluctuating;
A primary winding of a transformer connected between the first and second DC terminals via a single or a plurality of switches;
A secondary winding electromagnetically coupled to the primary winding;
A rectifying / smoothing circuit connected to the secondary winding;
A feedback control signal forming circuit for forming a feedback control signal for controlling the DC output voltage of the rectifying and smoothing circuit to a desired value;
A switch on / off control signal forming circuit connected between the feedback control signal forming circuit and a control terminal of the switch;
Current detecting means for detecting a current flowing through the switch;
Input voltage detecting means for detecting a DC input voltage between the first and second DC terminals;
A reference voltage generating circuit for generating a reference voltage that changes continuously or stepwise according to a change in the output voltage level of the input voltage detecting means;
An overcurrent detection comparator that is connected to the current detection means and the reference voltage generation circuit and generates an output that inhibits on-control of the switch when the output of the current detection means reaches the reference voltage; The present invention relates to a switching power supply device characterized by comprising:
The input voltage detection means in the present invention is not limited to the input voltage detection circuit 26 of the embodiment described later, but only the input voltage supply terminal 19 and the ground terminal 22 or only the input voltage detection line connected thereto. Further, the first and second resistors R1 and R2 for voltage division can be omitted, and the input voltage can be sent to the reference voltage generation circuit 27 without voltage division. In addition, a non-linear voltage dividing circuit using a semiconductor or the like can be provided instead of the voltage dividing circuit of the first and second resistors R1 and R2, or the voltage dividing function can be included in the reference voltage generating circuit 27.

なお、請求項２に示すように、更に、前記１次巻線及び前記２次巻線に電磁結合された３次巻線と、前記３次巻線に接続されている制御電源用整流平滑回路と、前記直流電源から前記制御電源用整流平滑回路の平滑用コンデンサに起動電流を供給する起動回路とを有し、少なくとも前記スイッチオン・オフ制御信号形成回路と前記入力電圧検出手段と前記過電流検出用比較器と前記起動回路とが共通の集積回路で構成され、前記集積回路は、前記第１の直流端子に接続される入力電圧供給端子を有し、前記入力電圧供給端子に前記起動回路及び前記入力電圧検出手段が接続されていることが望ましい。
また、請求項３に示すように、前記スイッチが前記集積回路の中に配置され、前記集積回路は、前記入力電圧供給端子を有する他に、前記制御電源用整流平滑回路を接続するための制御電源端子と、前記帰還制御信号形成回路を接続するための帰還端子と、前記第２の直流端子に接続するための共通端子と、前記スイッチの対の主端子の一方を前記１次巻線に接続するための第１の主端子と、前記スイッチの対の主端子の他方に接続された第２の主端子とを有し、前記第２の主端子は前記電流検出手段を介して前記第２の直流端子に接続されていることが望ましい。
また、請求項４に示すように、前記参照電圧発生回路は、前記入力電圧検出手段で検出された入力電圧検出値が増大した時に逆に低下した値を有する参照電圧を発生するものであことが望ましい。換言すれば、前記参照電圧発生回路は、前記入力電圧検出手段の出力電圧に反比例的に変化する参照電圧を発生するものであることが望ましい。前記入力電圧検出手段の出力電圧と参照電圧との相関を示す関数は任意に設定できる。
また、請求項５に示すように、前記参照電圧発生回路は、前記入力電圧検出手段の出力電圧に反比例的に変化する参照電圧を発生するものであって、前記入力電圧検出手段から得られた検出電圧（Ｖa ）に反比例した値（１／Ｖa ）に対して所定の係数（Ｋ）を乗算した値（Ｋ／Ｖa ）を求める第１の演算手段と、前記第１の演算手段の出力値（Ｋ／Ｖa ）に所定のオフセット電圧（Ｖb ）を加算する第２の演算手段とから成ることが望ましい。
また、請求項６に示すように、前記参照電圧発生回路は互いに電圧レベルの異なる複数の参照電圧を前記直流入力電圧の変化に応じて階段状に発生するものであって、少なくとも１つの基準電圧を発生する基準電圧源と、前記入力電圧検出手段の出力電圧が基準電圧源の前記基準電圧よりも低いか否かを判定する少なくとも１つの入力電圧判定用比較手段と、少なくとも第１の参照電圧とこの第１の参照電圧よりも低い第２の参照電圧とを発生する第１及び第２の参照電圧源と、前記入力電圧検出手段の出力電圧が前記基準電圧よりも低いことを示す前記入力電圧判定用比較手段の出力に応答して前記第１の参照電圧を前記過電流検出用比較器に送り、前記入力電圧検出手段の出力電圧が前記基準電圧よりも高いことを示す前記入力電圧判定用比較手段の出力に応答して前記第２の参照電圧を前記過電流検出用比較器に送るスイッチ手段とを有していることが望ましい。
また、請求項７に示すように、前記スイッチオン・オフ制御信号形成回路は、前記帰還制御信号形成回路に接続された第１の入力端子と前記電流検出手段に接続された第２の入力端子と前記電流検出手段の出力が前記帰還制御信号を横切った時に前記スイッチのオン期間の終了を示す信号を出力する出力端子とを有するオン終了時点検出用比較器と、前記スイッチのオン開始を示すオン開始信号を所定の周期で繰返して発生するオン開始信号発生回路と、前記オン終了時点検出用比較器と前記オン開始信号発生回路とに接続され、前記オン開始信号の発生時点から前記オン終了時点までの幅を有するスイッチ制御パルスを発生するパルス形成回路とから成ることが望ましい。 According to a second aspect of the present invention, there is further provided a tertiary winding electromagnetically coupled to the primary winding and the secondary winding, and a control power supply rectifying and smoothing circuit connected to the tertiary winding. And a starting circuit for supplying a starting current from the DC power source to the smoothing capacitor of the control power supply rectifying and smoothing circuit, at least the switch on / off control signal forming circuit, the input voltage detecting means, and the overcurrent The detection comparator and the activation circuit are configured by a common integrated circuit, and the integrated circuit has an input voltage supply terminal connected to the first DC terminal, and the activation circuit is connected to the input voltage supply terminal. The input voltage detecting means is preferably connected.
According to a third aspect of the present invention, the switch is disposed in the integrated circuit, and the integrated circuit has a control for connecting the control power source rectifying and smoothing circuit in addition to the input voltage supply terminal. One of a power supply terminal, a feedback terminal for connecting the feedback control signal forming circuit, a common terminal for connecting to the second DC terminal, and a main terminal of the switch pair is used as the primary winding. A first main terminal for connection; and a second main terminal connected to the other of the main terminals of the pair of switches, wherein the second main terminal is connected to the first main terminal via the current detection means. It is desirable to be connected to two DC terminals.
According to a fourth aspect of the present invention, the reference voltage generating circuit generates a reference voltage having a decreased value when the input voltage detection value detected by the input voltage detecting means increases. Is desirable. In other words, it is desirable that the reference voltage generation circuit generates a reference voltage that varies inversely proportional to the output voltage of the input voltage detection means. A function indicating the correlation between the output voltage of the input voltage detecting means and the reference voltage can be arbitrarily set.
According to a fifth aspect of the present invention, the reference voltage generation circuit generates a reference voltage that varies inversely with the output voltage of the input voltage detection means, and is obtained from the input voltage detection means. First calculation means for obtaining a value (K / Va) obtained by multiplying a value (1 / Va) inversely proportional to the detection voltage (Va) by a predetermined coefficient (K), and an output value of the first calculation means It is desirable to comprise a second calculation means for adding a predetermined offset voltage (Vb) to (K / Va).
According to a sixth aspect of the present invention, the reference voltage generation circuit generates a plurality of reference voltages having different voltage levels in a stepped manner in accordance with a change in the DC input voltage, and includes at least one reference voltage A reference voltage source for generating the input voltage, at least one input voltage determination comparing means for determining whether the output voltage of the input voltage detection means is lower than the reference voltage of the reference voltage source, and at least a first reference voltage And a first reference voltage source for generating a second reference voltage lower than the first reference voltage, and the input indicating that the output voltage of the input voltage detecting means is lower than the reference voltage. The input voltage determination indicating that the output voltage of the input voltage detection means is higher than the reference voltage by sending the first reference voltage to the overcurrent detection comparator in response to the output of the voltage determination comparison means. That in response to the output of the comparison means and a switching means for sending the second reference voltage to the overcurrent detection comparator it is desirable.
According to a seventh aspect of the present invention, the switch on / off control signal forming circuit includes a first input terminal connected to the feedback control signal forming circuit and a second input terminal connected to the current detection means. And an ON end point detection comparator having an output terminal for outputting a signal indicating the end of the ON period of the switch when the output of the current detection means crosses the feedback control signal, and indicates ON start of the switch An on start signal generation circuit that repeatedly generates an on start signal at a predetermined cycle, and is connected to the on end time detection comparator and the on start signal generation circuit, and the on end is started from the generation time of the on start signal. It is desirable to comprise a pulse forming circuit that generates a switch control pulse having a width up to a point in time.

各請求項の発明によれば、過電流検出用比較器に入力させる参照信号を入力電圧検出回路の出力電圧の変化に応じて変化させるという比較的簡単な構成によって入力電圧変動時又は入力電圧レンジ切換時の過電流検出精度の向上を図ることができ、スイッチング電源装置の小型化及び低コスト化を図ることができる。
請求項２の発明によれば、集積回路の入力電圧供給端子に過電流保護のための入力電圧検出回路と起動回路との両方を接続するので、集積回路の端子数の増大を伴なわないで正確な過電流保護を達成することができる。このため、スイッチング電源装置の小型化及び低コスト化を図ることができる。
請求項３の発明によれば、スイッチも集積回路の中に配置されているので、スイッチング電源装置の小型化及び低コスト化を図ることができる。
請求項４の発明によれば、理想又は理想に近い参照電圧を提供できる。
請求項５の発明によれば、演算手段によって目的とする参照電圧を容易且つ正確に得ることができる。
請求項６の発明によれば、参照電圧を階段状に容易且つ正確に発生することができる。
請求項７の発明によれば、スイッチの制御パルス及び過電流保護を容易に達成することができる。 According to the invention of each claim, when the input voltage fluctuates or the input voltage range, the reference signal to be input to the overcurrent detection comparator is changed according to the change of the output voltage of the input voltage detection circuit. The overcurrent detection accuracy at the time of switching can be improved, and the switching power supply device can be reduced in size and cost.
According to the second aspect of the present invention, since both the input voltage detection circuit for overcurrent protection and the start-up circuit are connected to the input voltage supply terminal of the integrated circuit, the number of terminals of the integrated circuit is not increased. Accurate overcurrent protection can be achieved. For this reason, size reduction and cost reduction of a switching power supply device can be achieved.
According to the invention of claim 3, since the switch is also arranged in the integrated circuit, the switching power supply device can be reduced in size and cost.
According to the invention of claim 4, it is possible to provide an ideal or near-ideal reference voltage.
According to the invention of claim 5, the target reference voltage can be obtained easily and accurately by the arithmetic means.
According to the sixth aspect of the present invention, the reference voltage can be easily and accurately generated stepwise.
According to the invention of claim 7, the control pulse of the switch and the overcurrent protection can be easily achieved.

次に、図１〜図１２を参照して本発明の実施形態を説明する。 Next, an embodiment of the present invention will be described with reference to FIGS.

図１に示す実施例１に従うスイッチング電源装置は、大別して商用交流電源１に接続された入力段整流平滑回路２と、トランス３と、集積回路４と、出力段整流平滑回路５と、制御電源用整流平滑回路６と、帰還制御信号形成回路７と、電流検出抵抗８とから成る。 The switching power supply device according to the first embodiment shown in FIG. 1 is roughly divided into an input stage rectifying / smoothing circuit 2, a transformer 3, an integrated circuit 4, an output stage rectifying / smoothing circuit 5, and a control power source connected to a commercial AC power source 1. It comprises a rectifying / smoothing circuit 6, a feedback control signal forming circuit 7, and a current detection resistor 8.

入力段整流平滑回路２は、交流電源１に接続された第１及び第２の交流端子１ａ、１ｂとスイッチング電源装置の直流入力端子として機能する第１及び第２の直流端子２ａ、２ｂとを有し、周知のダイオード整流回路と平滑回路とを含んでいる。第２の直流端子２ｂは共通端子であって、グランドに接続されている。なお、交流電源１を３相交流電源とし、入力段整流平滑回路２を３相の整流平滑回路とすることもできる。また、入力段整流平滑回路２の代わりに力率改善機能を有する交流―直流変換回路を接続することができる。また、交流電源１と直流端子２ａ、２ｂとの間に力率改善回路を接続することができる。また、入力段整流平滑回路２を電池等の直流電源とすることもできる。第１及び第２の直流端子２ａ、２ｂ間には交流電源１の電圧変化に応じて変化する直流入力電圧Ｖｉｎが得られる。 The input stage rectifying / smoothing circuit 2 includes first and second AC terminals 1a and 1b connected to the AC power source 1 and first and second DC terminals 2a and 2b that function as DC input terminals of the switching power supply device. A known diode rectifier circuit and a smoothing circuit. The second DC terminal 2b is a common terminal and is connected to the ground. The AC power source 1 can be a three-phase AC power source, and the input stage rectifying and smoothing circuit 2 can be a three-phase rectifying and smoothing circuit. Further, instead of the input stage rectifying / smoothing circuit 2, an AC-DC conversion circuit having a power factor improving function can be connected. A power factor correction circuit can be connected between the AC power supply 1 and the DC terminals 2a and 2b. Further, the input stage rectifying / smoothing circuit 2 may be a DC power source such as a battery. A DC input voltage Vin that changes in accordance with the voltage change of the AC power supply 1 is obtained between the first and second DC terminals 2a and 2b.

トランス３は、磁気コア９に巻回され且つ相互に電磁結合された１次巻線Ｎ1 、２次巻線Ｎ2 、及び３次巻線Ｎ3 を有する。 The transformer 3 has a primary winding N1, a secondary winding N2, and a tertiary winding N3 wound around a magnetic core 9 and electromagnetically coupled to each other.

出力段整流平滑回路５はダイオード１０と平滑コンデンサ１１とから成る。平滑コンデンサ１１はダイオード１０を介して２次巻線Ｎ2 に並列に接続されている。ダイオード１０は１次巻線Ｎ1 に電流が流れている時の２次巻線Ｎ2 の電圧で逆バイアスされる方向性を有する。
直流出力端子１２、１３は平滑コンデンサ１１の両端に接続され、負荷１４は直流出力端子１２、１３間に接続されている。 The output stage rectifying / smoothing circuit 5 includes a diode 10 and a smoothing capacitor 11. The smoothing capacitor 11 is connected in parallel to the secondary winding N2 via the diode 10. The diode 10 has a direction to be reverse-biased by the voltage of the secondary winding N2 when a current flows through the primary winding N1.
The DC output terminals 12 and 13 are connected to both ends of the smoothing capacitor 11, and the load 14 is connected between the DC output terminals 12 and 13.

制御電源電圧Ｖccを供給するための制御電源用平滑整流回路６は、ダイオード１５と平滑コンデンサ１６とから成る。平滑コンデンサ１６はダイオード１５を介して３次巻線Ｎ3 に並列に接続されている。ダイオード１５は１次巻線Ｎ1 に電流が流れている時に３次巻線Ｎ3 に誘起する電圧で逆バイアスされる方向性を有する。 The control power supply smoothing rectifier circuit 6 for supplying the control power supply voltage Vcc comprises a diode 15 and a smoothing capacitor 16. The smoothing capacitor 16 is connected in parallel to the tertiary winding N3 via a diode 15. The diode 15 has a direction that is reverse-biased by a voltage induced in the tertiary winding N3 when a current flows through the primary winding N1.

集積回路４は混成集積回路であって、第１及び第２の主端子１７’、１８と、入力電圧供給端子１９と、制御電源端子２０と、帰還端子２１と、グランド端子２２とを有する。第１の主端子１７’は１次巻線Ｎ1 を介して第１の直流端子２ａに接続されている。第２の主端子１８は電流検出手段としての電流検出抵抗８を介して第２の直流端子２ｂに接続されている。 The integrated circuit 4 is a hybrid integrated circuit, and includes first and second main terminals 17 ′ and 18, an input voltage supply terminal 19, a control power supply terminal 20, a feedback terminal 21, and a ground terminal 22. The first main terminal 17 'is connected to the first DC terminal 2a via the primary winding N1. The second main terminal 18 is connected to the second DC terminal 2b via a current detection resistor 8 as current detection means.

集積回路４に含まれている絶縁ゲート型電界効果トランジスタから成るスイッチ１７は第１及び第２の主端子１７’、１８間に接続されている。即ち、スイッチ１７のドレインが第１の主端子１７’に接続され、ソースが第２の主端子１８に接続されている。従って、１次巻線Ｎ1 とスイッチ１７と電流検出抵抗８との直列回路が第１及び第２の直流端子２ａ、２ｂ間に接続されている。スイッチ１７は電界効果トランジスタに限るものではなく、ＩＧＢＴ、トランジスタ等の別の制御可能な半導体スイッチとすることができる。 A switch 17 comprising an insulated gate field effect transistor included in the integrated circuit 4 is connected between the first and second main terminals 17 ′ and 18. That is, the drain of the switch 17 is connected to the first main terminal 17 ′, and the source is connected to the second main terminal 18. Accordingly, a series circuit of the primary winding N1, the switch 17, and the current detection resistor 8 is connected between the first and second DC terminals 2a and 2b. The switch 17 is not limited to a field effect transistor, and may be another controllable semiconductor switch such as an IGBT or a transistor.

集積回路４は、スイッチ１７の他に、スイッチオン・オフ制御信号形成回路２３、起動回路２４、低電圧誤動作防止回路２５、入力電圧検出回路２６、参照電圧発生回路２７、及び過電流検出用比較器２８を有する。 In addition to the switch 17, the integrated circuit 4 includes a switch on / off control signal generation circuit 23, a start circuit 24, a low voltage malfunction prevention circuit 25, an input voltage detection circuit 26, a reference voltage generation circuit 27, and an overcurrent detection comparison. A container 28.

起動回路２４は第１及び第２の直流端子２ａ、２ｂ間の電圧によってスイッチング電源装置の動作を開始させるために平滑コンデンサ１６に充電電流を供給するものであり、入力電圧供給端子１９と制御電源端子２０との間に接続された定電流源２９と起動スイッチ３０とから成る。起動スイッチ３０は、低電圧誤動作防止回路２５から制御電圧Ｖccがパルス形成回路３３を正常動作させることが可能な値よりも低いことを示す出力が発生していることに応答してオンになり、定電流源２９を入力電圧供給端子１９と制御電源端子２０との間に接続する。なお、定電流源２９を起動抵抗に置き換えることができる。起動スイッチ３０のオンの期間には、入力電圧供給端子１９が起動スイッチ３０及び定電流源２９及び制御電源端子２０を介して平滑コンデンサ１６に接続されるので、３次巻線Ｎ3 に電圧が発生していない起動時であっても、平滑コンデンサ１６が充電される。スイッチ１７のオン・オフ動作が開始し、３次巻線Ｎ3 に電圧が発生し、平滑コンデンサ１６が所定電圧以上に充電されたことが低電圧誤動作防止回路２５で検出されると、起動スイッチ３０がオフに制御され、起動回路２４は入力電圧供給端子１９から切り離される。これにより、平滑コンデンサ１６の過充電防止及び起動回路２４の電力損失の低減を図ることができる。なお、起動回路２４の損失及び過充電の問題がない時には、起動スイッチ３０を省き、定電流源２９又は起動抵抗を常に接続状態に保つこともできる。なお、低電圧誤動作防止回路２５は起動スイッチ３０の制御回路と呼ぶこともできる。 The starting circuit 24 supplies a charging current to the smoothing capacitor 16 in order to start the operation of the switching power supply device by the voltage between the first and second DC terminals 2a and 2b. It comprises a constant current source 29 and a start switch 30 connected between the terminals 20. The start switch 30 is turned on in response to an output from the low voltage malfunction prevention circuit 25 indicating that the control voltage Vcc is lower than a value that allows the pulse forming circuit 33 to operate normally, A constant current source 29 is connected between the input voltage supply terminal 19 and the control power supply terminal 20. The constant current source 29 can be replaced with a starting resistor. When the start switch 30 is on, the input voltage supply terminal 19 is connected to the smoothing capacitor 16 via the start switch 30, the constant current source 29 and the control power supply terminal 20, so that a voltage is generated in the tertiary winding N3. Even during startup, the smoothing capacitor 16 is charged. When the on / off operation of the switch 17 is started, a voltage is generated in the tertiary winding N3, and the low voltage malfunction prevention circuit 25 detects that the smoothing capacitor 16 is charged to a predetermined voltage or higher, the start switch 30 Is controlled to be off, and the starting circuit 24 is disconnected from the input voltage supply terminal 19. Thereby, the overcharge prevention of the smoothing capacitor 16 and the reduction of the power loss of the starting circuit 24 can be aimed at. When there is no problem of the loss of the starting circuit 24 and overcharge, the starting switch 30 can be omitted, and the constant current source 29 or the starting resistor can always be kept connected. The low-voltage malfunction prevention circuit 25 can also be called a control circuit for the start switch 30.

制御電源端子２０は制御電源電圧Ｖccを供給するためにスイッチオン・オフ制御信号形成回路２３に含まれている種々の回路、参照電圧発生回路２７、及び過電流検出用比較器２８に図示が省略されている配線によって接続されている。 The control power supply terminal 20 is not shown in the various circuits included in the switch on / off control signal forming circuit 23 for supplying the control power supply voltage Vcc, the reference voltage generation circuit 27, and the overcurrent detection comparator 28. Connected by the wiring that is.

スイッチオン・オフ制御信号形成回路２３は、分圧回路３１とオン終了時点検出用比較器３２とパルス形成回路３３とから成る。
分圧回路３１は帰還端子２１とオン終了時点検出用比較器３２との間に接続され、帰還端子２１の電圧信号から成る帰還制御信号Ｖf′を減衰した信号Ｖfを比較器３２に送る。即ち、分圧回路３１は、帰還端子２１とグランド端子（共通端子）２２との間に２つの抵抗３１ａ、３１ｂの直列回路を接続し、２つの抵抗３１ａ、３１ｂの相互接続点を比較器３２の負入力端子に接続することによって構成されている。
なお、帰還端子２１の信号Ｖf′の減衰が不要な時には分圧回路３１を省くことができる。 The switch on / off control signal forming circuit 23 includes a voltage dividing circuit 31, an ON end point detection comparator 32, and a pulse forming circuit 33.
The voltage dividing circuit 31 is connected between the feedback terminal 21 and the ON end point detection comparator 32, and sends the signal Vf obtained by attenuating the feedback control signal Vf ′ composed of the voltage signal of the feedback terminal 21 to the comparator 32. In other words, the voltage dividing circuit 31 connects a series circuit of two resistors 31 a and 31 b between the feedback terminal 21 and the ground terminal (common terminal) 22, and a connection point between the two resistors 31 a and 31 b is a comparator 32. Is connected to the negative input terminal.
The voltage dividing circuit 31 can be omitted when the signal Vf ′ at the feedback terminal 21 does not need to be attenuated.

直流出力端子１２、１３と帰還端子２１との間に接続された帰還制御信号形成回路７は、直流出力端子１２、１３間の直流出力電圧Ｖo の変化に対応して変化する帰還制御信号Ｖf′を形成する。この実施形態の帰還制御信号形成回路７は、図２に示すように図１の直流出力端子１２、１３間に接続される出力電圧検出抵抗７ａ，７ｂの直列回路と、誤差増幅器７ｃと、基準電圧源７ｄと、発光素子７ｅと、抵抗７ｆと、ホトトランジスタ７ｇと、抵抗７ｈと、電源端子７ｉとから成る。誤差増幅器７ｃは出力電圧検出抵抗７ａ，７ｂの分圧点の検出電圧と基準電圧源７ｄの基準電圧との差の信号を出力する。発光素子７ｅは誤差増幅器７ｃの出力に反比例した光出力を発生する。発光素子７ｅに光結合され且つ絶縁されたホトトランジスタ７ｇは発光素子７ｅの光出力に応答して抵抗値が光出力に反比例的に変化する。ホトトランジスタ７ｇの一端は電源端子７ｉ接続され、その他端は抵抗７ｈを介してグランドに接続され、ホトトランジスタ７ｇと抵抗７ｈとの間に帰還制御信号Ｖf′の出力ラインが接続さているので、帰還制御信号Ｖf′は直流出力電圧Ｖo に対して反比例的に変化する。また、帰還制御信号Ｖf′は、交流電源１の電圧Ｖacに対して図６に示すように反比例的に変化する。また、帰還制御信号Ｖf′は、入力直流電圧Ｖinに対しても反比例的に変化する。 The feedback control signal forming circuit 7 connected between the DC output terminals 12 and 13 and the feedback terminal 21 has a feedback control signal Vf ′ that changes in response to a change in the DC output voltage Vo between the DC output terminals 12 and 13. Form. As shown in FIG. 2, the feedback control signal forming circuit 7 of this embodiment includes a series circuit of output voltage detection resistors 7a and 7b connected between the DC output terminals 12 and 13 of FIG. 1, an error amplifier 7c, and a reference It comprises a voltage source 7d, a light emitting element 7e, a resistor 7f, a phototransistor 7g, a resistor 7h, and a power supply terminal 7i. The error amplifier 7c outputs a signal indicating the difference between the detection voltage at the voltage dividing point of the output voltage detection resistors 7a and 7b and the reference voltage of the reference voltage source 7d. The light emitting element 7e generates a light output that is inversely proportional to the output of the error amplifier 7c. The phototransistor 7g optically coupled to and insulated from the light emitting element 7e changes its resistance value in inverse proportion to the light output in response to the light output of the light emitting element 7e. One end of the phototransistor 7g is connected to the power supply terminal 7i, the other end is connected to the ground through the resistor 7h, and the output line of the feedback control signal Vf 'is connected between the phototransistor 7g and the resistor 7h. The control signal Vf 'varies inversely with the DC output voltage Vo. Further, the feedback control signal Vf ′ changes in inverse proportion to the voltage Vac of the AC power supply 1 as shown in FIG. Further, the feedback control signal Vf ′ varies inversely with the input DC voltage Vin.

電流検出手段としての電流検出抵抗８は、１次巻線Ｎ1 及びスイッチ１７に直列に接続されているので、１次巻線Ｎ1 及びスイッチ１７の電流を検出することができる。即ち、電流検出抵抗８の一端は集積回路４の第２の主端子１８に接続され、他端はグランドに接続されているので、第２の主端子１８とグランドとの間に電流検出抵抗８を流れる電流に比例した電圧レベルを有する電流検出信号Ｖi が得られる。なお、電流検出抵抗８の代りに、変流器、磁気センサ型電流検出器等の電流検出手段を使用することができる。 Since the current detection resistor 8 as current detection means is connected in series with the primary winding N1 and the switch 17, the current of the primary winding N1 and the switch 17 can be detected. That is, since one end of the current detection resistor 8 is connected to the second main terminal 18 of the integrated circuit 4 and the other end is connected to the ground, the current detection resistor 8 is interposed between the second main terminal 18 and the ground. A current detection signal Vi having a voltage level proportional to the current flowing through is obtained. In place of the current detection resistor 8, current detection means such as a current transformer or a magnetic sensor type current detector can be used.

１次巻線Ｎ1 はインダクタンスを有するので、スイッチ１７のオン開始後に電流検出信号Ｖiは図５（Ｂ）に示すように傾斜を有して徐々に増大し、鋸波状の信号が得られる。 Since the primary winding N1 has an inductance, after the switch 17 is turned on, the current detection signal Vi gradually increases with a slope as shown in FIG. 5B, and a sawtooth signal is obtained.

オン終了時点検出用比較器３２の正入力端子は第２の主端子１８を介して電流検出抵抗８に接続されているので、オン終了時点検出用比較器３２は図５（Ｂ）に示すように電流検出信号Ｖiと帰還制御信号Ｖf とを比較し、電流検出信号Ｖi が帰還制御信号Ｖf を横切る時に図５（Ｃ）に示すオン終了時点検出信号Ｖ32を発生する。スイッチ１７のオン終了時点はオフ開始時点であるので、オン終了時点検出用比較器３２をスイッチオフ開始時点決定用比較器と呼ぶこと、及びオン終了時点検出信号Ｖ32をオフ開始時点検出信号と呼ぶこともできる。 Since the positive input terminal of the on-end time point detection comparator 32 is connected to the current detection resistor 8 via the second main terminal 18, the on-end time point detection comparator 32 is as shown in FIG. The current detection signal Vi and the feedback control signal Vf are compared with each other, and when the current detection signal Vi crosses the feedback control signal Vf, the ON end point detection signal V32 shown in FIG. 5C is generated. Since the ON end point of the switch 17 is an OFF start point, the ON end point detection comparator 32 is called a switch OFF start point determination comparator, and the ON end point detection signal V32 is called an OFF start point detection signal. You can also.

パルス形成回路３３は、オン終了時点検出用比較器３２とスイッチ１７の制御端子（ゲート）との間に接続され、出力電圧Ｖo を一定に保つように制御された制御パルスを形成し、これをスイッチオン・オフ制御信号としてスイッチ１７の制御端子とソースとの間に送る。パルス形成回路３３には、過電流防止制御ライン３４及び低電圧誤動作防止制御ライン３５が接続されている。このため、過電流防止制御ライン３４が過電流を示している時には、パルス形成回路３３からスイッチ１７への制御パルスの供給が禁止される。また、低電圧誤動作防止回路２５が低電圧を検出している期間には、パルス形成回路３３はスイッチ１７のための制御パルスを発生しない。 The pulse forming circuit 33 is connected between the ON end point detection comparator 32 and the control terminal (gate) of the switch 17, and forms a control pulse controlled so as to keep the output voltage Vo constant. A switch on / off control signal is sent between the control terminal of the switch 17 and the source. An overcurrent prevention control line 34 and a low voltage malfunction prevention control line 35 are connected to the pulse forming circuit 33. For this reason, when the overcurrent prevention control line 34 indicates an overcurrent, the supply of control pulses from the pulse forming circuit 33 to the switch 17 is prohibited. Further, the pulse forming circuit 33 does not generate a control pulse for the switch 17 while the low voltage malfunction prevention circuit 25 is detecting a low voltage.

図３は図１のパルス形成回路３３の１例を概略的に示す。パルス形成回路３３は、発振回路４０とＲＳフリップフロップ４１と論理和回路４２と駆動回路４３とから成る。発振回路４０は図５（Ａ）に示すクロックパルスＶ40を所定の周期Ｔs で繰返して発生する。この実施形態ではクロックパルスＶ40の繰返し周波数は２０〜１００kHz から選択された値とされている。このクロックパルス周波数は負荷の大きさの変化に応じて変化させること、又は段階的に切換えることができる。 FIG. 3 schematically shows an example of the pulse forming circuit 33 of FIG. The pulse forming circuit 33 includes an oscillation circuit 40, an RS flip-flop 41, an OR circuit 42, and a drive circuit 43. The oscillation circuit 40 repeatedly generates the clock pulse V40 shown in FIG. 5A with a predetermined period Ts. In this embodiment, the repetition frequency of the clock pulse V40 is a value selected from 20 to 100 kHz. The clock pulse frequency can be changed according to the change in the load size, or can be switched stepwise.

フリップフロップ４１のセット入力端子Ｓはクロック発振回路４０に接続され、リセット入力端子Ｒは論理和回路４２を介して図１のオン終了時点検出用比較器３２及び過電流検出用比較器２８に接続されている。従って、フリップフロップ４１は、図５のｔ1 、ｔ3 、ｔ5 、ｔ7 、ｔ9 等に示す図５（Ａ）のクロックパルスＶ40に応答してセット状態（第１の状態）となり、ｔ2 、ｔ4 、ｔ6 、ｔ8 、ｔ10等に示す図５（Ｃ）のオフ終了時点検出信号Ｖ32に応答してリセット状態（第２の状態）となり、出力端子Ｑに図５（Ｄ）に示す出力Ｖ41を送出する。図５（Ｄ）のフリップフロップ４１の出力端子Ｑは駆動回路４３を介して図１のスイッチ１７の制御端子に接続されている。従って、パルス形成回路３３の出力ライン３３ａには図５（Ｄ）と同一の幅を有する制御パルスが得られ、これがスイッチオン・オフ制御信号として機能する。なお、スイッチ１７は図５（Ｄ）の出力Ｖ41の高レベル期間にオンになり、低レベル期間にオフになる。 The set input terminal S of the flip-flop 41 is connected to the clock oscillation circuit 40, and the reset input terminal R is connected to the ON end point detection comparator 32 and the overcurrent detection comparator 28 of FIG. Has been. Accordingly, the flip-flop 41 enters the set state (first state) in response to the clock pulse V40 in FIG. 5A shown by t1, t3, t5, t7, t9, etc. in FIG. 5, and t2, t4, t6. In response to the OFF end point detection signal V32 in FIG. 5C shown at t8, t10, etc., the reset state (second state) is entered, and the output V41 shown in FIG. The output terminal Q of the flip-flop 41 in FIG. 5D is connected to the control terminal of the switch 17 in FIG. Therefore, a control pulse having the same width as that in FIG. 5D is obtained on the output line 33a of the pulse forming circuit 33, and this functions as a switch on / off control signal. Note that the switch 17 is turned on during the high level period of the output V41 in FIG. 5D and turned off during the low level period.

フリップフロップ４１のリセット端子Ｒには過電流検出用比較器２８の出力も入力する。従って、過電流検出用比較器２８の出力が過電流を示している時にはフリップフロップ４１が強制的にリセットされ、スイッチ１７がオフ制御される。 The output of the overcurrent detection comparator 28 is also input to the reset terminal R of the flip-flop 41. Accordingly, when the output of the overcurrent detection comparator 28 indicates an overcurrent, the flip-flop 41 is forcibly reset and the switch 17 is controlled to be turned off.

過電流検出時にスイッチ１７を所定時間以上オフ状態に保持するために図３において点線で示すようにライン３２に直列にタイマ回路４４を接続することができる。このタイマ回路４４は、過電流検出用比較器２８が過電流状態を一度検出すると、過電流状態の継続の有無に関係なく、過電流状態を示す信号を所定時間継続して出力する。
また、過電流時のスイッチ１７のオフ制御をフリップフロップ４１で行う代りに図３において点線で示すようにタイマ回路４４の出力を発振回路４０に供給して発振を停止することによって行うことができる。また、過電流時のスイッチ１７のオフ制御をフリップフロップ４１の出力段で行うこともできる。出力段におけるオフ制御は図１０を参照して追って説明する。 In order to hold the switch 17 in an OFF state for a predetermined time or more when an overcurrent is detected, a timer circuit 44 can be connected in series with the line 32 as shown by a dotted line in FIG. When the overcurrent detection comparator 28 detects an overcurrent state once, the timer circuit 44 continuously outputs a signal indicating the overcurrent state for a predetermined time regardless of whether the overcurrent state is continued.
Further, instead of the flip-flop 41 performing the off control of the switch 17 at the time of overcurrent, it can be performed by supplying the output of the timer circuit 44 to the oscillation circuit 40 and stopping the oscillation as shown by a dotted line in FIG. . Further, the switch 17 can be controlled to be turned off at the output stage of the flip-flop 41 in the event of an overcurrent. The off control in the output stage will be described later with reference to FIG.

パルス形成回路３３のクロック発振回路４０はライン３５によって低電圧誤動作防止回路２５に接続されている。従って、３次巻線Ｎ3 から電圧が得られ、制御電源電圧Ｖccが所定電圧になる前には制御パルスが形成されず、所定電圧になった時に制御パルスが形成され、低電圧時の不安定動作による誤動作が防止される。 The clock oscillation circuit 40 of the pulse forming circuit 33 is connected to the low voltage malfunction prevention circuit 25 by a line 35. Therefore, a voltage is obtained from the tertiary winding N3, and a control pulse is not formed before the control power supply voltage Vcc reaches the predetermined voltage, but a control pulse is formed when the predetermined voltage is reached, and unstable at a low voltage. Malfunction due to operation is prevented.

過電流検出用比較器２８の正入力端子は第２の主端子１８を介して電流検出抵抗８に接続され、負入力端子は参照電圧発生回路２７に接続されている。従って、過電流検出用比較器２８は電流検出信号Ｖi が参照電圧Ｖr に達した時に過電流状態を示す信号を出力する。 The positive input terminal of the overcurrent detection comparator 28 is connected to the current detection resistor 8 via the second main terminal 18, and the negative input terminal is connected to the reference voltage generation circuit 27. Accordingly, the overcurrent detection comparator 28 outputs a signal indicating an overcurrent state when the current detection signal Vi reaches the reference voltage Vr.

参照電圧発生回路２７は第１及び第２の直流端子２ａ、２ｂ間の入力直流電圧Ｖinに反比例的に変化する参照電圧Ｖr を図５（Ｂ）で鎖線で示すように発生する。この参照電圧Ｖr の発生を可能にするために入力電圧検出回路２６の出力ライン２６ａが参照電圧発生回路２７に接続されている。入力直流電圧Ｖinと参照電圧Ｖrとの関係は、入力直流電圧Ｖinが増大した時に参照電圧Ｖrの値が減少する関係であればどのような関係でもよい。例えば、参照電圧Ｖrをｙ、入力直流電圧Ｖinをｘ、任意の定数をｋ，ａ，ｂとした時、ｙ＝ｂ−ｘ、又はｙ＝｛ｋ/（ｘ−ａ）｝＋ｂ、又はｙ＝ｋ/ｘ、又はｙ＝ｌｏｇ_aｘ、又はｙ＝（１／ａ）^x等の任意の関数に従う値を参照電圧Ｖrとすることができる。 The reference voltage generating circuit 27 generates a reference voltage Vr that changes inversely proportional to the input DC voltage Vin between the first and second DC terminals 2a and 2b as shown by a chain line in FIG. The output line 26 a of the input voltage detection circuit 26 is connected to the reference voltage generation circuit 27 in order to enable the generation of the reference voltage Vr. The relationship between the input DC voltage Vin and the reference voltage Vr may be any relationship as long as the value of the reference voltage Vr decreases when the input DC voltage Vin increases. For example, when the reference voltage Vr is y, the input DC voltage Vin is x, and arbitrary constants are k, a, b, y = b−x, or y = {k / (x−a)} + b, or y = K / x, or y = log _a x, or y = (1 / a) A value according to an arbitrary function such as ^x can be used as the reference voltage Vr.

入力電圧検出手段としての入力電圧検出回路２６は、第１及び第２の抵抗Ｒ1 、Ｒ2 の直列回路を入力電圧供給端子１９とグランド端子２２との間に接続し、第１及び第２の抵抗Ｒ1 、Ｒ2 の相互接続点に出力ライン２６ａを接続することによって構成されており、入力電圧を分圧した値を出力する。集積回路４には入力電圧検出回路２６を接続するための独立した端子が形成されず、起動回路２４を接続するために必要な入力電圧供給端子１９に入力電圧検出回路２６が接続されている。従って、集積回路４の端子数の増大を抑え、集積回路４の小型化、低コスト化を図ることができる。なお、既に説明したように、本発明における入力電圧検出手段を、入力電圧供給端子１９とグランド端子２２のみ、又はこれ等に接続された入力電圧検出ラインのみと定義し、分圧用の第１及び第２の抵抗R1、R2からなる入力電圧検出回路２６を参照電圧発生回路２７の一部と考えることができる。また、第１及び第２の抵抗R1、R2の分圧回路の代わりに半導体等を使用した非直線的分圧回路を設けることができる。 An input voltage detection circuit 26 as an input voltage detection means connects a series circuit of first and second resistors R1 and R2 between an input voltage supply terminal 19 and a ground terminal 22, and first and second resistors. The output line 26a is connected to the interconnection point of R1 and R2, and a value obtained by dividing the input voltage is output. In the integrated circuit 4, an independent terminal for connecting the input voltage detection circuit 26 is not formed, and the input voltage detection circuit 26 is connected to the input voltage supply terminal 19 necessary for connecting the starting circuit 24. Therefore, an increase in the number of terminals of the integrated circuit 4 can be suppressed, and the integrated circuit 4 can be reduced in size and cost. As already described, the input voltage detection means in the present invention is defined as only the input voltage supply terminal 19 and the ground terminal 22, or only the input voltage detection line connected thereto, and the first and The input voltage detection circuit 26 including the second resistors R1 and R2 can be considered as a part of the reference voltage generation circuit 27. Further, a non-linear voltage dividing circuit using a semiconductor or the like can be provided instead of the voltage dividing circuit of the first and second resistors R1 and R2.

図４は図１の参照電圧発生回路２７の１例を示す。この参照電圧発生回路２７は、所定の係数Ｋを発生する係数発生回路５０と第１及び第２の演算回路５１、５２と、オフセット電圧発生回路５３とから成る。第１の演算回路５１は乗除算回路から成り、定数回路５０及びライン２６ａによって図１の入力電圧検出回路２６に接続され、入力電圧検出回路２６から得られた検出電圧Ｖa に反比例した値１／Ｖa を求め、且つこの値１／Ｖa に所定の係数Ｋを乗算する。第２の演算回路５２は第１の演算回路５１及びオフセット電圧発生回路５３に接続され、第１の演算回路５１の出力Ｋ／Ｖa にオフセット電圧Ｖb を加算した値からなる参照電圧Ｖr ＝（Ｋ／Ｖa ）＋Ｖb を出力する。なお、係数Ｋは回路内部で決まる値である。また、オフセット電圧Ｖb は過電流検出レベルの最低値即ち過電流検出用参照電圧Ｖr の最低値である。 FIG. 4 shows an example of the reference voltage generation circuit 27 of FIG. The reference voltage generation circuit 27 includes a coefficient generation circuit 50 that generates a predetermined coefficient K, first and second arithmetic circuits 51 and 52, and an offset voltage generation circuit 53. The first arithmetic circuit 51 comprises a multiplication / division circuit, is connected to the input voltage detection circuit 26 of FIG. 1 by a constant circuit 50 and a line 26a, and has a value 1 / inversely proportional to the detection voltage Va obtained from the input voltage detection circuit 26. Va is obtained, and this value 1 / Va is multiplied by a predetermined coefficient K. The second arithmetic circuit 52 is connected to the first arithmetic circuit 51 and the offset voltage generation circuit 53, and a reference voltage Vr = (K), which is a value obtained by adding the offset voltage Vb to the output K / Va of the first arithmetic circuit 51. / Va) + Vb is output. The coefficient K is a value determined inside the circuit. The offset voltage Vb is the minimum value of the overcurrent detection level, that is, the minimum value of the overcurrent detection reference voltage Vr.

入力段整流平滑回路２を交流電源１に接続するか、又は入力段整流平滑回路２の入力又は出力ラインに設けられている図示が省略されている周知の電源スイッチをオンにすると、入力段整流平滑回路２から入力直流電圧Ｖinが得られる。起動時においては３次巻線Ｎ3 に電圧が発生していないので、低電圧誤動作防止回路２５は起動スイッチ３０をオンに保ち、且つパルス形成回路３３を非動作状態に保つ。これにより、入力段整流平滑回路２、入力電圧供給端子１９、起動回路２４、制御電源端子２０及び平滑コンデンサ１６の経路に平滑コンデンサ１６の充電電流が流れる。
平滑コンデンサ１６が集積回路４の駆動が可能な電圧まで充電されると、低電圧誤動作防止回路２５の出力は誤動作が生じないことを示す高レベルとなり、パルス形成回路３３が図５（Ｄ）に示す制御パルスＶ41の発生を開始し、同時又は少し遅れて起動スイッチ３０がオフになる。これにより、スイッチ１７がオン・オフ動作し、トランス３の１次巻線Ｎ1 に電圧が断続的に印加される。トランス３には、スイッチ１７のオン期間にエネルギが蓄積され、スイッチ１７のオフ期間にトランス３からエネルギが放出され、２次巻線Ｎ2 及び３次巻線Ｎ3 にダイオード１０及び１５を導通させる向きの電圧が発生し、出力段整流平滑回路５の平滑コンデンサ１１及び制御電源用整流平滑回路６の平滑コンデンサ１６が充電され、所望出力電圧Ｖo 及び所望の制御電源電圧Ｖccが得られる。 When the input stage rectifying / smoothing circuit 2 is connected to the AC power source 1 or a known power switch (not shown) provided in the input or output line of the input stage rectifying / smoothing circuit 2 is turned on, the input stage rectifying / smoothing circuit 2 is turned on. An input DC voltage Vin is obtained from the smoothing circuit 2. Since no voltage is generated in the tertiary winding N3 at the time of start-up, the low voltage malfunction prevention circuit 25 keeps the start switch 30 on and keeps the pulse forming circuit 33 in a non-operating state. As a result, the charging current of the smoothing capacitor 16 flows through the path of the input stage rectifying and smoothing circuit 2, the input voltage supply terminal 19, the starting circuit 24, the control power supply terminal 20 and the smoothing capacitor 16.
When the smoothing capacitor 16 is charged to a voltage at which the integrated circuit 4 can be driven, the output of the low-voltage malfunction prevention circuit 25 becomes a high level indicating that no malfunction occurs, and the pulse forming circuit 33 is shown in FIG. Generation of the control pulse V41 shown is started, and the start switch 30 is turned off simultaneously or with a slight delay. As a result, the switch 17 is turned on and off, and a voltage is intermittently applied to the primary winding N1 of the transformer 3. Energy is stored in the transformer 3 during the ON period of the switch 17, and energy is released from the transformer 3 during the OFF period of the switch 17, and the diodes 10 and 15 are conducted to the secondary winding N2 and the tertiary winding N3. The smoothing capacitor 11 of the output stage rectifying and smoothing circuit 5 and the smoothing capacitor 16 of the control power supply rectifying and smoothing circuit 6 are charged, and the desired output voltage Vo and the desired control power supply voltage Vcc are obtained.

入力直流電圧Ｖinが一定の状態で負荷変動に基づいて出力電圧Ｖo が所望値よりも高くなると、図５（Ｂ）のｔ7 時点以後に示すように帰還制御信号Ｖf の電圧レベルがｔ7 時点前のＶf よりも低下し、図５（Ｄ）に示すようにｔ7 以後の制御パルスＶ41の幅がｔ7 以前の制御パルスの幅よりも狭くなり、出力電圧Ｖo が所望値に戻される。出力電圧Ｖo が所望値よりも低くなった時には上述の高くなった時と逆の動作になる。 When the input DC voltage Vin is constant and the output voltage Vo becomes higher than the desired value based on the load fluctuation, the voltage level of the feedback control signal Vf is the time before the time t7 as shown after the time t7 in FIG. As shown in FIG. 5D, the width of the control pulse V41 after t7 becomes narrower than the width of the control pulse before t7, and the output voltage Vo is returned to the desired value. When the output voltage Vo becomes lower than the desired value, the operation is the reverse of that when the output voltage Vo becomes higher.

負荷１４の大きさが一定で、入力直流電圧Ｖinが比較的低い第１の値からこれよりも高い第２の値になった時には、前述の出力電圧Ｖo の上昇時と同様に図５のｔ7 時点以後の動作が生じる。入力直流電圧Ｖinの変動時には参照電圧Ｖr が入力直流電圧Ｖinに反比例的に変化し、図５（Ｂ）のｔ7 時点以後の参照電圧Ｖr は、ｔ7 以前の参照電圧Ｖr よりも低くなる。このため、参照電圧Ｖr を固定して図５のｔ7 時点以前の参照電圧Ｖr とｔ7 以後に参照電圧Ｖr とを同一値に保つ方式に比べて図５のｔ7 以後では電流検出信号Ｖi が参照信号Ｖr に早く達し、過負荷状態を正確に検出することができる。即ち、入力直流電圧Ｖinが比較的高い第２の値の状態で負荷１４の短絡等の異常が生じると、出力電圧Ｖo は低下し、帰還制御信号Ｖf のレベルは逆に高くなり、制御パルスＶ41の幅が増大する動作が生じる。しかし、帰還制御信号Ｖf が参照信号Ｖr よりも高くなろうとすると、この前に電流検出信号Ｖi が参照信号Ｖr に達し、制御パルスＶ41の送出が禁止される。これにより、負荷１４に過大電流が流れ続けることが禁止され、負荷１４が過電流から保護される。 When the load 14 is constant and the input DC voltage Vin changes from a relatively low first value to a second value higher than this, t7 in FIG. 5 is the same as when the output voltage Vo rises. Operation after the point in time occurs. When the input DC voltage Vin varies, the reference voltage Vr changes in inverse proportion to the input DC voltage Vin, and the reference voltage Vr after time t7 in FIG. 5B is lower than the reference voltage Vr before t7. For this reason, the reference voltage Vr is fixed and the reference voltage Vr before the time t7 in FIG. 5 and the reference voltage Vr after the time t7 are kept at the same value, and the current detection signal Vi is the reference signal after the time t7 in FIG. Vr can be reached early and an overload condition can be accurately detected. That is, when an abnormality such as a short circuit of the load 14 occurs when the input DC voltage Vin is at a relatively high second value, the output voltage Vo decreases, the level of the feedback control signal Vf increases, and the control pulse V41. An operation of increasing the width of the signal occurs. However, if the feedback control signal Vf is going to be higher than the reference signal Vr, the current detection signal Vi reaches the reference signal Vr before this, and transmission of the control pulse V41 is prohibited. Thereby, it is prohibited that an excessive current continues to flow through the load 14, and the load 14 is protected from the overcurrent.

本実施形態は次の効果を有する。
（１）過電流検出用比較器２８に与える参照電圧Ｖr を直流入力電圧Ｖinに対して反比例的に変えるので、直流入力電圧Ｖinの変動及び直流入力電圧Ｖinの例えば１００〜２４０Ｖの範囲での切換えがあっても負荷１４の過電流状態をトランス３の１次巻線Ｎ1 側で正確に検出できる。このため、過電流保護の精度が向上し、スイッチング電源装置の安全性の向上を容易に達成することができる。
（２）過電流検出用入力電圧検出回路２６、参照電圧発生回路２７及び過電流検出用比較器２７が集積回路４に含まれているので、外付け回路を設けることなしに過電流保護を正確に達成することができ、スイッチング電源装置のコストの低減及び小型化を図ることができる。
（３）起動回路２４が接続されている入力電圧供給端子１９に過電流検出用の入力電圧検出回路２６が接続され、且つ第２の主端子１８に過電流検出用比較器２８が接続されているので、過電流検出専用の端子を集積回路４に追加することが不要になり、集積回路４の低コスト化及び小型化を図ることができる。
（４）第１及び第２の演算回路５１、５２によって参照電圧Ｖr を容易且つ正確に得ることができる。
（５）起動スイッチ３０を有するので、起動後に起動回路２４の電流を遮断することができ、電力損失の低減及び過充電防止を達成することができる。
（６）起動期間には低電圧誤動作防止回路２５によって制御パルスの発生が禁止されるので、スイッチ１７のオン・オフを安定的に開始させることができる。 This embodiment has the following effects.
(1) Since the reference voltage Vr applied to the overcurrent detection comparator 28 is changed in inverse proportion to the DC input voltage Vin, fluctuations in the DC input voltage Vin and switching of the DC input voltage Vin within a range of, for example, 100 to 240V Even if there is, the overcurrent state of the load 14 can be accurately detected on the primary winding N1 side of the transformer 3. For this reason, the accuracy of overcurrent protection is improved, and the safety of the switching power supply device can be easily improved.
(2) Since the integrated circuit 4 includes the overcurrent detection input voltage detection circuit 26, the reference voltage generation circuit 27, and the overcurrent detection comparator 27, the overcurrent protection can be accurately performed without providing an external circuit. The switching power supply device can be reduced in cost and size.
(3) An input voltage detection circuit 26 for detecting overcurrent is connected to the input voltage supply terminal 19 to which the starting circuit 24 is connected, and an overcurrent detection comparator 28 is connected to the second main terminal 18. Therefore, it is not necessary to add a dedicated terminal for overcurrent detection to the integrated circuit 4, and the cost and size of the integrated circuit 4 can be reduced.
(4) The reference voltage Vr can be obtained easily and accurately by the first and second arithmetic circuits 51 and 52.
(5) Since the start switch 30 is provided, the current of the start circuit 24 can be interrupted after the start, and reduction of power loss and prevention of overcharge can be achieved.
(6) Since the generation of the control pulse is prohibited by the low voltage malfunction prevention circuit 25 during the start-up period, the on / off of the switch 17 can be started stably.

次に、図８及び図９を参照して実施例２のスイッチング電源装置を説明する。但し、実施例２に従うスイッチング電源装置は、実施例１に従う図４の参照電圧発生回路２７を図８に変形し、この他は図１と同一に形成したものであるので、必要に応じて図１も参照する。 Next, the switching power supply device according to the second embodiment will be described with reference to FIGS. However, in the switching power supply device according to the second embodiment, the reference voltage generating circuit 27 of FIG. 4 according to the first embodiment is modified to FIG. 8 and the other portions are formed in the same manner as FIG. See also 1.

図８に示す変形された参照電圧発生回路２７´は、比較器６０と、基準電圧源６１と、第１及び第２の参照電圧源６２，６３と、切換スイッチ６４とから成る。比較器６０の一方の入力端子は図１の入力電圧検出回路２６の出力ライン２６ａに接続され、他方の入力端子は基準電圧源６１に接続されている。基準電圧源６１の電圧Ｖ１０は、入力電圧検出回路２６の出力電圧Ｖａの変化範囲の任意の中間値に設定されている。なお、この実施形態では、このＶ１０が図１の交流電源１の交流電圧Ｖａｃが１７０Ｖの時の入力電圧検出回路２６の出力電圧Ｖａに相当する値に設定されている。入力電圧検出回路２６の出力電圧Ｖａが基準電圧Ｖ１０よりも低い時には比較器６０の出力が第１の値（低レベル）となり、逆に出力電圧Ｖａが基準電圧Ｖ１０よりも高い時には比較器６０出力が第２の値（高レベル）となる。 The modified reference voltage generation circuit 27 ′ shown in FIG. 8 includes a comparator 60, a reference voltage source 61, first and second reference voltage sources 62 and 63, and a changeover switch 64. One input terminal of the comparator 60 is connected to the output line 26 a of the input voltage detection circuit 26 of FIG. 1, and the other input terminal is connected to the reference voltage source 61. The voltage V10 of the reference voltage source 61 is set to an arbitrary intermediate value in the change range of the output voltage Va of the input voltage detection circuit 26. In this embodiment, this V10 is set to a value corresponding to the output voltage Va of the input voltage detection circuit 26 when the AC voltage Vac of the AC power supply 1 in FIG. When the output voltage Va of the input voltage detection circuit 26 is lower than the reference voltage V10, the output of the comparator 60 becomes the first value (low level). Conversely, when the output voltage Va is higher than the reference voltage V10, the output of the comparator 60 is output. Becomes the second value (high level).

第１及び第２の参照電圧源６２，６３は切換スイッチ６４の接点ａ，ｂを介して参照電圧発生回路２７´の出力ライン６５に接続されている。切換スイッチ６４は比較器６０の第１の値の出力に応答して第１の接点ａがオンになり、比較器６０の第２の値の出力に応答して第２の接点ｂがオンになるように形成されている。図８には切換スイッチ６４が簡略化されて機械的スイッチで示されているが、実際には第１及び第２の参照電圧源６２，６３を出力ライン６５に選択的に接続する２つの電子スイッチから成る。第１及び第２の参照電圧源６２，６３は図９に第１及び第２の参照電圧Ｖ１１（例えば０,７５Ｖ）、Ｖ１２（例えば０,６０Ｖ）を発生する。 The first and second reference voltage sources 62 and 63 are connected to the output line 65 of the reference voltage generation circuit 27 ′ via the contacts a and b of the changeover switch 64. The changeover switch 64 turns on the first contact a in response to the output of the first value of the comparator 60, and turns on the second contact b in response to the output of the second value of the comparator 60. It is formed to become. In FIG. 8, the changeover switch 64 is simplified and shown as a mechanical switch. However, in practice, two electronic devices for selectively connecting the first and second reference voltage sources 62 and 63 to the output line 65 are shown. Consists of switches. The first and second reference voltage sources 62 and 63 generate first and second reference voltages V11 (for example, 0,75V) and V12 (for example, 0,60V) in FIG.

実施例２の参照電圧発生回路２７´においては、商用交流電源１が１００Ｖ系の時に比較器６０の出力が第１の値となって切換スイッチ６４の第１の接点ａがオンになって第１の参照電圧Ｖ１１が出力され、商用交流電源１が２００Ｖ系の時に比較器６０の出力が第２の値となって切換スイッチ６４の第２の接点ｂがオンになって第２の参照電圧Ｖ１２が出力される。図７から明らかなように交流電圧Ｖａｃが高くなるに従って参照電圧Ｖｒを低くすることが望ましい。従って、交流電圧Ｖａｃが１００Ｖ系の値の時と２００Ｖ系の値の時とで参照電圧Ｖｒを図９のように階段状に切換えると、負荷１４の過電流保護を適切に達成することができる。 In the reference voltage generation circuit 27 'of the second embodiment, when the commercial AC power supply 1 is a 100V system, the output of the comparator 60 becomes the first value and the first contact a of the changeover switch 64 is turned on. 1 is output, and when the commercial AC power supply 1 is a 200V system, the output of the comparator 60 becomes the second value and the second contact b of the changeover switch 64 is turned on, and the second reference voltage V12 is output. As is apparent from FIG. 7, it is desirable to lower the reference voltage Vr as the AC voltage Vac increases. Therefore, the overcurrent protection of the load 14 can be appropriately achieved by switching the reference voltage Vr in a stepwise manner as shown in FIG. 9 when the AC voltage Vac is a value of 100V system and a value of 200V system. .

図８の参照電圧発生回路２７´における１個の比較器６０と１個の基準電圧源６１との代りに複数個の比較器とレベルの異なる複数の基準電圧源とを設けることができる。この場合には各比較器に入力電圧検出回路２６の出力電圧Ｖａを入力させ、各比較器によって互いに異なる基準電圧と入力電圧検出回路２６の出力電圧Ｖａとを比較し、入力直流電圧Ｖｉｎの変化を多段階に判定する。また、複数個の比較器に対比させて複数の参照電圧源を設け、複数個の比較器の出力の変化に応じて参照電圧を変化させる。これにより、参照電圧Ｖｒを入力直流電圧Ｖｉｎの変化に応じて多段階に変化させることができ、図７の連続的に変化する参照電圧Ｖｒと実質的に同一に変化する段階状の参照電圧を得ることができる。 In place of one comparator 60 and one reference voltage source 61 in the reference voltage generation circuit 27 ′ of FIG. 8, a plurality of comparators and a plurality of reference voltage sources having different levels can be provided. In this case, the output voltage Va of the input voltage detection circuit 26 is inputted to each comparator, the reference voltage different from each other is compared with the output voltage Va of the input voltage detection circuit 26 by each comparator, and the change of the input DC voltage Vin Is determined in multiple stages. In addition, a plurality of reference voltage sources are provided in contrast to the plurality of comparators, and the reference voltage is changed in accordance with changes in the outputs of the plurality of comparators. As a result, the reference voltage Vr can be changed in multiple stages according to the change in the input DC voltage Vin, and the step-like reference voltage that changes substantially the same as the continuously changing reference voltage Vr in FIG. Can be obtained.

図１０は実施例３のスイッチング電源装置の変形されたパルス形成回路３３´を示す。この図１０のパルス形成回路３３´は、図３のパルス形成回路３３の一部を変え、この他は図３と同一に構成したものであるので、図３の実質的に同一の部分には同一の符号を付してその説明を省略する。 FIG. 10 shows a modified pulse forming circuit 33 ′ of the switching power supply device according to the third embodiment. The pulse forming circuit 33 'shown in FIG. 10 is a part of the pulse forming circuit 33 shown in FIG. 3 and the other parts are the same as those shown in FIG. The same reference numerals are given and description thereof is omitted.

図１０では図３の論理和回路４２及びタイマ回路４４の代りに、論理積回路４４、オン禁止用ＲＳフリップフロップ４５、及びリセット回路４６が設けられている。オン・オフ制御パルス形成用のフリップフロップ４１のリセット端子Ｒには図１のオン終了時点検出用比較器３２の出力ライン３２ａのみが接続されている。論理積回路４４の一方の入力端子は制御パルス形成用ＲＳフリップフロップ４１の正相出力端子に接続され、その他方の入力端子はオン禁止用ＲＳフリップフロップ４５の逆相出力端子に接続され、その出力端子は駆動回路４３を介して図１のスイッチ１７の制御端子に接続されている。従って、オン禁止用ＲＳフリップフロップ４５がセット状態の時には制御パルス形成用ＲＳフリップフロップ４１の出力パルスの伝送が阻止され、論理積回路４４の出力が低レベルを維持し、スイッチ１７がオフに保たれる。 In FIG. 10, in place of the OR circuit 42 and the timer circuit 44 shown in FIG. 3, an AND circuit 44, an on-prohibition RS flip-flop 45, and a reset circuit 46 are provided. Only the output line 32a of the on-end time point detection comparator 32 of FIG. 1 is connected to the reset terminal R of the flip-flop 41 for on / off control pulse formation. One input terminal of the AND circuit 44 is connected to the positive phase output terminal of the control pulse forming RS flip-flop 41, and the other input terminal is connected to the negative phase output terminal of the on-prohibition RS flip-flop 45. The output terminal is connected to the control terminal of the switch 17 in FIG. Therefore, when the on-prohibition RS flip-flop 45 is in the set state, transmission of the output pulse of the control pulse forming RS flip-flop 41 is blocked, the output of the AND circuit 44 is maintained at a low level, and the switch 17 is kept off. Be drunk.

オン禁止用ＲＳフリップフロップ４５のセット入力端子Ｓは図１の過電流検出用比較器２８の出力ライン３４に接続され、リセット入力端子Ｒはリセット回路４６に接続されている。従って、図１の過電流検出用比較器２８が過電流状態を検出すると、オン禁止用ＲＳフリップフロップ４５がオンになり、スイッチ１７がオフ制御され、このオフ制御状態はリセット回路４６からリセット信号が発生するまで保持される。これにより、図１の負荷１４の過電流保護を確実に達成することができる。 The set input terminal S of the on-prohibition RS flip-flop 45 is connected to the output line 34 of the overcurrent detection comparator 28 in FIG. 1, and the reset input terminal R is connected to the reset circuit 46. Therefore, when the overcurrent detection comparator 28 in FIG. 1 detects an overcurrent state, the on-prohibition RS flip-flop 45 is turned on and the switch 17 is controlled to be off. Until it occurs. Thereby, the overcurrent protection of the load 14 of FIG. 1 can be achieved reliably.

本発明は上述の実施例に限定されるものでなく、例えば次の変形が可能なものである。
（１）トランス３の２次巻線Ｎ₂及び３次巻線Ｎ₃の極性を１次巻線Ｎ₁と同一とし、スイッチ１７がオンの時にダイオード１０，１５が導通するフォワード型スイッチング電源装置とすることができる。
（２）スイッチ１７を集積回路４に含めない構成とすることができる。また、電流検出抵抗８を集積回路４に含めることができる。
（３）帰還制御信号形成回路７の一部又は全部を集積回路４に含めることができる。
（４）スイッチ１７に流れる電流の検出信号として電流に反比例的に変化する電流検出信号を形成することができる。即ち、スイッチ１７の電流が徐々に増大している時に徐々に低下する電圧を電流検出信号とすることができる。この場合には、参照電圧発生回路２７，２７´の出力電圧Ｖｒを直流入力電圧Ｖｉｎに比例的に変化させる。また、帰還信号形成回路７を図１１の帰還信号形成回路７´に変形する。図１１の帰還信号形成回路７´では、ホトトランジスタ７ｈが抵抗７ｇとグランドとの間に接続されている。従って、出力電圧Ｖo に対して比例的に変化する帰還制御信号Ｖf′がえられる。
（５）図１のオン終了時点検出用比較器３２に電流検出信号Ｖｉからなる鋸波を入力させるに代りに図１２に示す鋸波発生回路７０から鋸波電圧を入力させることができる。鋸波発生回路７０はスイッチ１７のオン開始を示す信号、例えば図３のクロック発振回路４０のクロックパルスで鋸波の発生を開始し、スイッチ１７のオン終了を示す信号、例えばオン終了時点検出用比較器３２の出力で鋸波の発生を終了させる。
（６）１つのスイッチ１７の代わりに複数のスイッチを１次巻線N1に直列に接続して１つのスイッチの電圧分担を低減することができる。 The present invention is not limited to the above-described embodiments, and for example, the following modifications are possible.
(1) A forward-type switching power supply device in which the secondary winding N ₂ and the tertiary winding N ₃ of the transformer 3 have the same polarity as the primary winding N ₁ and the diodes 10 and 15 are conductive when the switch 17 is on. It can be.
(2) The switch 17 may not be included in the integrated circuit 4. In addition, the current detection resistor 8 can be included in the integrated circuit 4.
(3) A part or all of the feedback control signal forming circuit 7 can be included in the integrated circuit 4.
(4) As a detection signal for the current flowing through the switch 17, a current detection signal that varies inversely with the current can be formed. That is, a voltage that gradually decreases when the current of the switch 17 gradually increases can be used as the current detection signal. In this case, the output voltage Vr of the reference voltage generation circuits 27 and 27 ′ is changed in proportion to the DC input voltage Vin. Further, the feedback signal forming circuit 7 is transformed into the feedback signal forming circuit 7 ′ of FIG. In the feedback signal forming circuit 7 ′ of FIG. 11, the phototransistor 7h is connected between the resistor 7g and the ground. Therefore, a feedback control signal Vf 'that changes in proportion to the output voltage Vo is obtained.
(5) Instead of inputting the sawtooth wave composed of the current detection signal Vi to the ON end point detection comparator 32 of FIG. 1, a sawtooth voltage can be input from the sawtooth wave generation circuit 70 shown in FIG. The sawtooth wave generation circuit 70 starts generation of a sawtooth wave with a signal indicating the start of turning on of the switch 17, for example, the clock pulse of the clock oscillation circuit 40 of FIG. The generation of the sawtooth wave is terminated at the output of the comparator 32.
(6) It is possible to reduce the voltage sharing of one switch by connecting a plurality of switches in series with the primary winding N1 instead of one switch 17.

本発明は直流電源装置等に利用可能である。 The present invention is applicable to a DC power supply device and the like.

本発明の実施例１に従うスイッチング電源装置を示す回路図である。It is a circuit diagram which shows the switching power supply device according to Example 1 of this invention. 図１の帰還信号形成回路を詳しく示す回路図である。It is a circuit diagram which shows the feedback signal formation circuit of FIG. 1 in detail. 図１のパルス形成回路を詳しく示すブロック図である。FIG. 2 is a block diagram showing in detail the pulse forming circuit of FIG. 1. 図１の参照電圧発生回路を詳しく示すブロック図である。FIG. 2 is a block diagram illustrating in detail the reference voltage generation circuit of FIG. 1. 図３の各部の状態を示す波形図である。It is a wave form diagram which shows the state of each part of FIG. 図１の交流電源の電圧Ｖａｃの実効値と帰還制御信号Ｖｆ´との関係を示す図である。It is a figure which shows the relationship between the effective value of the voltage Vac of the alternating current power supply of FIG. 1, and the feedback control signal Vf '. 図１の交流電源の電圧Ｖａｃの実効値と参照電圧Ｖｒとの関係を示す図である。It is a figure which shows the relationship between the effective value of the voltage Vac of the alternating current power supply of FIG. 1, and the reference voltage Vr. 実施例２の参照電圧発生回路を示す回路図である。FIG. 6 is a circuit diagram illustrating a reference voltage generation circuit according to a second embodiment. 図１の交流電源の電圧Ｖａｃの実効値と図８の参照電圧Ｖｒとの関係を示す図である。It is a figure which shows the relationship between the effective value of the voltage Vac of the alternating current power supply of FIG. 1, and the reference voltage Vr of FIG. 実施例３のパルス形成回路を示す回路図であるFIG. 6 is a circuit diagram illustrating a pulse forming circuit according to a third embodiment. 変形例の帰還制御信号形成回路を示す回路図である。It is a circuit diagram which shows the feedback control signal formation circuit of a modification. 変形例のスイッチ制御信号形成回路の一部を示す回路図である。It is a circuit diagram which shows a part of switch control signal formation circuit of a modification.

Explanation of symbols

２ａ，２ｂ第１及び第２の直流端子
３トランス
４集積回路
５出力段整流平滑回路
６制御電源用整流平滑回路
７帰還制御信号形成回路
８電流検出抵抗
１７スイッチ
２３スイッチオン・オフ制御信号形成回路
２４起動回路
２６入力電圧検出回路
２７参照電圧発生回路
２８過電流検出用比較器
３２オン終了時点検出用比較器 2a, 2b First and second DC terminals 3 Transformer 4 Integrated circuit 5 Output stage rectifying and smoothing circuit 6 Control power source rectifying and smoothing circuit 7 Feedback control signal forming circuit 8 Current detection resistor 17 Switch 23 Switch on / off control signal forming circuit 24 Start-up circuit 26 Input voltage detection circuit 27 Reference voltage generation circuit 28 Overcurrent detection comparator 32 On-end time detection comparator

Claims

First and second DC terminals for supplying a DC input voltage having a possibility of fluctuating;
A primary winding of a transformer connected between the first and second DC terminals via a single or a plurality of switches;
A secondary winding electromagnetically coupled to the primary winding;
A rectifying / smoothing circuit connected to the secondary winding;
A feedback control signal forming circuit for forming a feedback control signal for controlling the DC output voltage of the rectifying and smoothing circuit to a desired value;
A switch on / off control signal forming circuit connected between the feedback control signal forming circuit and a control terminal of the switch;
Current detecting means for detecting a current flowing through the switch;
Input voltage detecting means for detecting a DC input voltage between the first and second DC terminals;
A reference voltage generating circuit for generating a reference voltage that changes continuously or stepwise according to a change in the output voltage level of the input voltage detecting means;
An overcurrent detection comparator that is connected to the current detection means and the reference voltage generation circuit and generates an output that inhibits on-control of the switch when the output of the current detection means reaches the reference voltage; A switching power supply device comprising:

Further, a tertiary winding electromagnetically coupled to the primary winding and the secondary winding, a control power rectifying and smoothing circuit connected to the tertiary winding, and the DC power supply to the control power supply A starting circuit for supplying a starting current to the smoothing capacitor of the rectifying and smoothing circuit,
At least the switch on / off control signal formation circuit, the input voltage detection means, the overcurrent detection comparator, and the start circuit are configured by a common integrated circuit,
The integrated circuit has an input voltage supply terminal connected to the first DC terminal, and the start-up circuit and the input voltage detection means are connected to the input voltage supply terminal. The switching power supply device according to 1.

The switch is disposed in the integrated circuit, and the integrated circuit has the input voltage supply terminal, a control power supply terminal for connecting the control power supply rectifying and smoothing circuit, and the feedback control signal forming circuit. A feedback terminal for connecting the first DC terminal, a common terminal for connecting to the second DC terminal, and a first main terminal for connecting one of the main terminals of the switch pair to the primary winding; And a second main terminal connected to the other main terminal of the pair of switches, and the second main terminal is connected to the second DC terminal via the current detecting means. The switching power supply device according to claim 2.

4. The reference voltage generation circuit according to claim 1, wherein the reference voltage generation circuit generates a reference voltage having a decreased value when an input voltage detection value detected by the input voltage detection means increases. The switching power supply device described in any one.

The reference voltage generating circuit generates a reference voltage that varies inversely with the output voltage of the input voltage detecting means,
First calculating means for obtaining a value (K / Va) obtained by multiplying a value (1 / Va) inversely proportional to the detected voltage (Va) obtained from the input voltage detecting means by a predetermined coefficient (K);
The switching according to any one of claims 1 to 3, further comprising: second arithmetic means for adding a predetermined offset voltage (Vb) to the output value (K / Va) of the first arithmetic means. Power supply.

The reference voltage generating circuit generates a plurality of reference voltages having different voltage levels from each other in a stepped manner according to a change in the DC input voltage,
A reference voltage source for generating at least one reference voltage;
At least one input voltage determination comparison means for determining whether an output voltage of the input voltage detection means is lower than the reference voltage of a reference voltage source;
First and second reference voltage sources for generating at least a first reference voltage and a second reference voltage lower than the first reference voltage;
The first reference voltage is sent to the overcurrent detection comparator in response to the output of the input voltage determination comparison means indicating that the output voltage of the input voltage detection means is lower than the reference voltage, and the input Switch means for sending the second reference voltage to the overcurrent detection comparator in response to the output of the input voltage determination comparison means indicating that the output voltage of the voltage detection means is higher than the reference voltage. The switching power supply device according to claim 1, wherein the switching power supply device is provided.

The switch on / off control signal forming circuit includes:
A first input terminal connected to the feedback control signal forming circuit, a second input terminal connected to the current detection means, and an ON period of the switch when the output of the current detection means crosses the feedback control signal An on-end time point detection comparator having an output terminal for outputting a signal indicating the end of
An on start signal generating circuit for repeatedly generating an on start signal indicating on start of the switch at a predetermined period;
And a pulse forming circuit that is connected to the on-end time point detection comparator and the on-start signal generation circuit and generates a switch control pulse having a width from the on-start signal generation time to the on-end time point. The switching power supply device according to any one of claims 1 to 6.