JP2005071874A - Discharge lamp lighting device - Google Patents

Discharge lamp lighting device Download PDF

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JP2005071874A
JP2005071874A JP2003301878A JP2003301878A JP2005071874A JP 2005071874 A JP2005071874 A JP 2005071874A JP 2003301878 A JP2003301878 A JP 2003301878A JP 2003301878 A JP2003301878 A JP 2003301878A JP 2005071874 A JP2005071874 A JP 2005071874A
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resistor
discharge lamp
capacitor
frequency
value
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JP4676685B2 (en
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Masahiro Naruo
誠浩 鳴尾
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Panasonic Electric Works Co Ltd
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Matsushita Electric Works Ltd
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    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B20/00—Energy efficient lighting technologies, e.g. halogen lamps or gas discharge lamps

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Abstract

<P>PROBLEM TO BE SOLVED: To provide a discharge lamp lighting device which reduces a crest factor and prevents a lamp from flickering and going out by suppressing a lamp current oscillation in dimming light for a low luminous flux, and can perform stable feedback control. <P>SOLUTION: The discharge lamp lighting device comprises: a power converting section 1 which converts a DC power of a DC power source Vdc into a high frequency power by performing switching control of switching elements Q1, Q2 and supplies the high frequency power to a discharge lamp FL; and a feedback section 2 which compares a high frequency power output from the power converting section 1 with a reference value corresponding to a dimming ratio for the discharge lamp FL and adjusts a controlled variable of the switching control so as for the high frequency power output to coincide with the reference value, where the feedback section 2 has a error amplifier 3. A series circuit of a resistor R1 and a capacitor C1 is connected in parallel to an input resistor R3 of an operational amplifier OP forming the error amplifier 3. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

本発明は、高周波電力を供給して放電灯を点灯する放電灯点灯装置に関するものである。   The present invention relates to a discharge lamp lighting device for lighting a discharge lamp by supplying high frequency power.

（従来例１）
図１４に示すように、スイッチング素子Ｑ１，Ｑ２をスイッチング制御することで直流電源Ｖdcの直流電力を高周波電力に変換して放電灯ＦＬに供給する電力変換部１と、放電灯ＦＬの調光比に応じた基準値を電力変換部１の高周波出力と比較し該高周波出力を基準値に一致させる方向へスイッチング制御の制御量を調整するフィードバック部２とを備えた放電灯点灯装置が提供されている。電力変換部１は、例えば直流電源Ｖdcの両極間に接続される一対のスイッチング素子Ｑ１，Ｑ２と、何れか一方のスイッチング素子Ｑ１又はＱ２の両端に接続されるインダクタ及びコンデンサ（何れも図示せず）の共振回路とを有するインバータ回路からなり、一対のスイッチング素子Ｑ１，Ｑ２を交互にオン／オフ（スイッチング）することで直流電力を高周波電力に変換するものであって、スイッチング素子Ｑ１，Ｑ２のスイッチング周波数を変化させて高周波電力を増減することにより放電灯ＦＬの調光が可能である。尚、スイッチング素子Ｑ１，Ｑ２がバイポーラトランジスタからなる場合には、それぞれ逆方向に電流を流すためのダイオードＤ１，Ｄ２が逆並列に接続される。 (Conventional example 1)
As shown in FIG. 14, the power conversion unit 1 that converts the DC power of the DC power source Vdc into high-frequency power by switching control of the switching elements Q1 and Q2 and supplies it to the discharge lamp FL, and the dimming ratio of the discharge lamp FL A discharge lamp lighting device is provided that includes a feedback unit 2 that compares the reference value according to the frequency with the high-frequency output of the power converter 1 and adjusts the control amount of the switching control in a direction that matches the high-frequency output with the reference value. Yes. The power conversion unit 1 includes, for example, a pair of switching elements Q1 and Q2 connected between both poles of the DC power supply Vdc, and an inductor and a capacitor (both not shown) connected to both ends of either one of the switching elements Q1 or Q2. ) Of the inverter circuit having the resonance circuit of FIG. 5 and converting the DC power into the high frequency power by alternately turning on / off (switching) the pair of switching elements Q1 and Q2, The discharge lamp FL can be dimmed by changing the switching frequency to increase or decrease the high-frequency power. When the switching elements Q1 and Q2 are bipolar transistors, diodes D1 and D2 for flowing current in the opposite directions are connected in antiparallel.

フィードバック部２は高周波出力に対応する検出電圧Ｖｓと基準値に対応する基準電圧Ｖｄとの差分を増幅する誤差増幅器３と、Ｖ−ｆ変換器４と、ドライバ回路５とを有する。誤差増幅器３は、反転入力端子に検出電圧Ｖｓが入力されるとともに非反転入力端子に基準電圧Ｖｄが入力されるオペアンプＯＰと、反転入力端子に直列接続される入力抵抗Ｒ３と、オペアンプＯＰの出力端子と反転入力端子との間に接続される抵抗Ｒ２及びコンデンサＣ２の並列回路とを具備する。Ｖ−ｆ変換器４は誤差増幅器３の出力電圧を周波数に変換して出力し、ドライバ回路５はＶ−ｆ変換器４から出力された周波数に応じて検出電圧Ｖｓと基準電圧Ｖｄとの差分をゼロとする方向へスイッチング素子Ｑ１，Ｑ２のスイッチング周波数を調整する。すなわち、電力変換部１はフィードバック部２により基準値（基準電圧）に応じた高周波電力を出力するようにフィードバック制御されるのである。   The feedback unit 2 includes an error amplifier 3 that amplifies the difference between the detection voltage Vs corresponding to the high frequency output and the reference voltage Vd corresponding to the reference value, a Vf converter 4, and a driver circuit 5. The error amplifier 3 includes an operational amplifier OP in which the detection voltage Vs is input to the inverting input terminal and the reference voltage Vd is input to the non-inverting input terminal, an input resistor R3 connected in series to the inverting input terminal, and an output of the operational amplifier OP. And a parallel circuit of a resistor R2 and a capacitor C2 connected between the terminal and the inverting input terminal. The Vf converter 4 converts the output voltage of the error amplifier 3 into a frequency and outputs it, and the driver circuit 5 outputs a difference between the detection voltage Vs and the reference voltage Vd according to the frequency output from the Vf converter 4. The switching frequency of switching elements Q1 and Q2 is adjusted in a direction in which is set to zero. That is, the power conversion unit 1 is feedback-controlled by the feedback unit 2 so as to output high-frequency power corresponding to the reference value (reference voltage).

図１５は放電灯ＦＬを抵抗負荷とみなしたときの電力変換部１の電圧−電流特性を示しており、実線イが放電灯ＦＬの静特性、破線ロが放電灯ＦＬの定格出力付近の電圧−電流特性、一点破線ハが低光束調光時の電圧−電流特性をそれぞれ表している。尚、ここでいう低光束調光は定格点灯時の光束を１００％としたときに２５％以下に光束を減少させることを意味している。電力変換部１のスイッチング周波数を一定とすると、定格点灯においては実線イと破線ロが一点Ａでのみ交わるから、基本的にはフィードバック制御を行わなくても安定点灯が可能である。ところが、低光束調光時の電圧−電流特性を表す一点破線ハと放電灯ＦＬの静特性を表す実線イとは２つの交点Ｂ１，Ｂ２を持つので、フィードバック部２によるフィードバック制御を行わなければ放電灯ＦＬのちらつきや所謂ジャンプ現象といった不安定な動作をしてしまう。これを回避するには、電力変換部１の電圧−電流特性（一点破線ハ）を定電流特性に近付けて放電灯ＦＬの静特性（実線イ）との交点を一つにすることが考えられるが、このためには電力変換部１が有する共振回路のインダクタンス値を大きくしなければならない。しかしながら、インダクタンス値を大きくすれば、定格出力を得るために直流電源Ｖdcの直流電圧を上げる必要があり、また、ストレスに耐え得るだけの直流重畳特性を得るためには共振回路のインダクタを大型化する必要があり、放電灯点灯装置のコストアップは免れないことになる。従って、上述のように電力変換部１における共振電流やランプ電流などの電気量を検出してフィードバック制御する方法が一般的に採用されている。   FIG. 15 shows the voltage-current characteristics of the power converter 1 when the discharge lamp FL is regarded as a resistive load. The solid line A is the static characteristic of the discharge lamp FL, and the broken line B is the voltage near the rated output of the discharge lamp FL. -Current characteristics and one-dot broken lines C represent voltage-current characteristics during low beam dimming. The low beam dimming here means that the light flux is reduced to 25% or less when the light flux at the rated lighting is 100%. If the switching frequency of the power conversion unit 1 is constant, the solid line A and the broken line B intersect only at one point A in rated lighting, so that stable lighting is basically possible without performing feedback control. However, since the one-dot broken line C representing the voltage-current characteristic during low beam dimming and the solid line a representing the static characteristic of the discharge lamp FL have two intersections B1 and B2, feedback control by the feedback unit 2 must be performed. An unstable operation such as flickering of the discharge lamp FL or a so-called jump phenomenon is caused. In order to avoid this, it is conceivable that the voltage-current characteristic (one-dot broken line c) of the power converter 1 is brought close to the constant current characteristic so that the intersection with the static characteristic (solid line a) of the discharge lamp FL becomes one. However, for this purpose, the inductance value of the resonance circuit included in the power converter 1 must be increased. However, if the inductance value is increased, it is necessary to increase the DC voltage of the DC power supply Vdc in order to obtain the rated output, and in order to obtain a DC superposition characteristic that can withstand stress, the inductor of the resonance circuit is enlarged. Therefore, the cost increase of the discharge lamp lighting device is unavoidable. Therefore, as described above, a method is generally employed in which the amount of electricity such as a resonance current and a lamp current in the power conversion unit 1 is detected and feedback controlled.

通常、上述のようなフィードバック部２におけるフィードバック制御はスイッチング周波数によって決まる一周期の電気量（電流、電力など）の安定化を図るものであるので、スイッチング周波数以上の周波数の影響をほとんど受けないように設計される。これはスイッチング周波数に比べてかなり低い周波数にフィードバックの回路網関数が極を持つようなものである。ところが、蛍光灯などの放電灯ＦＬにおける熱力学特性は動作点及び周囲温度にも依存する変化を持って数百Ｈｚオーダーの極を生じる。そのため、低光束調光時（特に低温時）において、フィードバック部２の応答性が放電灯ＦＬの挙動に追随できないと、フィードバックループ内に発振が生じてしまう。この発振はフィードバック部２の出力に現れ、その出力に応じて電力変換部１のスイッチング周波数が変調されるため、図１６に示すように放電灯ＦＬに流れるランプ電流の包絡線が発振し、結果的にランプ電流のクレストファクタが悪化し、ちらつきや、ランプ電流の低下したところでアークの消滅を引き起こす虞がある。ここで、図１６（ａ）はランプ電流の波形図、（ｂ）はドライブ回路５からスイッチング素子Ｑ１，Ｑ２に与えられる駆動信号の波形図をそれぞれ示している。   Normally, the feedback control in the feedback unit 2 as described above is intended to stabilize the amount of electricity (current, power, etc.) in one cycle determined by the switching frequency, so that it is hardly affected by the frequency above the switching frequency. Designed to. This is such that the feedback network function has a pole at a much lower frequency than the switching frequency. However, the thermodynamic characteristics of a discharge lamp FL, such as a fluorescent lamp, generate poles on the order of several hundred Hz with changes depending on the operating point and the ambient temperature. Therefore, at the time of low luminous flux dimming (particularly at low temperatures), if the response of the feedback unit 2 cannot follow the behavior of the discharge lamp FL, oscillation will occur in the feedback loop. This oscillation appears at the output of the feedback unit 2, and the switching frequency of the power conversion unit 1 is modulated according to the output, so that the envelope of the lamp current flowing through the discharge lamp FL oscillates as shown in FIG. In particular, the crest factor of the lamp current deteriorates, and there is a risk of causing flickering or arc extinction when the lamp current decreases. Here, FIG. 16A shows a waveform diagram of the lamp current, and FIG. 16B shows a waveform diagram of the drive signal given from the drive circuit 5 to the switching elements Q1 and Q2.

（従来例２）
上記従来例１の不具合を改善した放電灯点灯装置の一例を図１７に示す（特許文献１参照）。電力変換部１は電界効果トランジスタからなるスイッチング素子Ｑ１，Ｑ２と、ローサイドのスイッチング素子Ｑ２の両端に直流カット用のコンデンサＣｄを介して接続されたインダクタＬｘ及びコンデンサＣｘの共振回路とで構成され、コンデンサＣｘの両端に放電灯ＦＬが接続されている。また、フィードバック部２は誤差増幅器３、Ｖ−ｆ変換器４、ドライバ回路５で構成される点は従来例１と共通であるが、誤差増幅器３がオペアンプＯＰの反転入力端子と出力端子の間にコンデンサＣ２のみが接続されるとともに反転入力端子とグランドの間にコンデンサＣ４が接続されて構成される点が従来例１と異なっている。尚、検出電圧Ｖｓはローサイドのスイッチング素子Ｑ２に直列接続された検出抵抗Ｒ１０の両端電圧からとっている。 (Conventional example 2)
FIG. 17 shows an example of a discharge lamp lighting device that improves the problem of the conventional example 1 (see Patent Document 1). The power conversion unit 1 includes switching elements Q1 and Q2 including field effect transistors, and an inductor Lx and a resonance circuit of a capacitor Cx connected to both ends of the low-side switching element Q2 via a DC cut capacitor Cd. A discharge lamp FL is connected to both ends of the capacitor Cx. The feedback unit 2 includes the error amplifier 3, the Vf converter 4, and the driver circuit 5 in common with the conventional example 1, but the error amplifier 3 is provided between the inverting input terminal and the output terminal of the operational amplifier OP. Is different from the conventional example 1 in that only the capacitor C2 is connected to the inverting input terminal and the ground is connected to the capacitor C4. The detection voltage Vs is taken from the voltage across the detection resistor R10 connected in series to the low-side switching element Q2.

図１８に従来例２における誤差増幅器３の周波数−ゲイン特性（破線ニ）と、従来例１における誤差増幅器３の周波数−ゲイン特性（一点破線ホ）とを示す（但し、周波数軸及びゲイン軸は何れも対数）。従来例１の誤差増幅器３はオペアンプＯＰの反転入力端子と出力端子の間に抵抗Ｒ２が接続されており、この抵抗Ｒ２と入力抵抗Ｒ３との比（Ｒ２／Ｒ３）によって誤差増幅器３の直流ゲインが決定される。これに対して従来例２の誤差増幅器３では、理想的なオペアンプＯＰであれば実線ヘで示すように直流ゲインが直線状になって無限大まで取り得るが、実際にはオペアンプＯＰが持つ最大ゲインＡmaxによって制限されるため、従来例１の誤差増幅器３との違いは直流ゲインの違いだけとなる。而して、このフィードバック部２では、抵抗Ｒ１０の両端電圧を入力抵抗Ｒ３とコンデンサＣ４からなる積分回路で積分した検出電圧Ｖｘと基準電圧Ｖｄの差分を誤差増幅器３で増幅し、Ｖ−ｆ変換器４とドライバ回路５を介して検出電圧Ｖｘと基準電圧Ｖｄの差分が小さくなるようにスイッチング素子Ｑ１，Ｑ２のスイッチング周波数を制御しており、入力抵抗Ｒ３とコンデンサＣ２，Ｃ４で決まる時定数をスイッチング周波数ｆの逆数以上且つ１００μｓ以下の範囲に設定することによって従来例１の不具合を改善している。   FIG. 18 shows the frequency-gain characteristic (dashed line d) of the error amplifier 3 in the conventional example 2 and the frequency-gain characteristic (one-dot broken line e) of the error amplifier 3 in the conventional example 1 (however, the frequency axis and the gain axis are All are logarithmic). In the error amplifier 3 of the conventional example 1, a resistor R2 is connected between the inverting input terminal and the output terminal of the operational amplifier OP, and the DC gain of the error amplifier 3 is determined by the ratio (R2 / R3) of the resistor R2 and the input resistor R3. Is determined. On the other hand, in the error amplifier 3 of the conventional example 2, if it is an ideal operational amplifier OP, the DC gain becomes linear as shown by the solid line and can be taken to infinity, but in reality the maximum operational amplifier OP has. Since it is limited by the gain Amax, the difference from the error amplifier 3 of the conventional example 1 is only the difference in DC gain. Thus, in the feedback unit 2, the difference between the detection voltage Vx obtained by integrating the voltage across the resistor R10 by the integrating circuit composed of the input resistor R3 and the capacitor C4 and the reference voltage Vd is amplified by the error amplifier 3, and Vf conversion is performed. The switching frequency of the switching elements Q1 and Q2 is controlled so as to reduce the difference between the detection voltage Vx and the reference voltage Vd via the device 4 and the driver circuit 5, and the time constant determined by the input resistor R3 and the capacitors C2 and C4 is set. The problem of the prior art example 1 is improved by setting it in the range not less than the reciprocal of the switching frequency f and not more than 100 μs.

（従来例３）
上記従来例１の不具合を改善した放電灯点灯装置の他の例を図１９に示す（特許文献２参照）。この従来例３と従来例２との回路構成上の相違点は、コンデンサＣ４がない点と、抵抗Ｒ１１及びコンデンサＣ１１の直列回路がコンデンサＣ２と並列に接続されている点と、放電灯ＦＬに流れるランプ電流をランプ電流検出回路６で検出した検出電圧を誤差増幅器３に入力している点とにあり、その他の構成は共通である。この従来例３では、抵抗Ｒ１１の抵抗値とコンデンサＣ１１の容量値とで決まるカットオフ周波数１／（２π×Ｒ１１×Ｃ１１）の零点を導入することにより、従来例１で説明した放電灯ＦＬの周波数特性に生じる極を打ち消して、ランプ電流の包絡線の発振を抑えるようにしている。尚、従来例３におけるコンデンサＣ２はコンデンサＣ１１に比較して遙かに小さい容量値しか持たないものであり、電力変換部１のスイッチング周波数において十分に低いゲインとするために付加されている。
特開２０００−３０８８７号公報 特開平６−２２３９７９号公報 (Conventional example 3)
FIG. 19 shows another example of a discharge lamp lighting device that improves the problem of the conventional example 1 (see Patent Document 2). The difference in circuit configuration between Conventional Example 3 and Conventional Example 2 is that there is no capacitor C4, a series circuit of a resistor R11 and a capacitor C11 is connected in parallel with the capacitor C2, and the discharge lamp FL. The other configuration is common in that the detected voltage detected by the lamp current detection circuit 6 is input to the error amplifier 3. In the conventional example 3, by introducing a zero point of a cut-off frequency 1 / (2π × R11 × C11) determined by the resistance value of the resistor R11 and the capacitance value of the capacitor C11, the discharge lamp FL described in the conventional example 1 is provided. The pole generated in the frequency characteristic is canceled out to suppress the oscillation of the lamp current envelope. The capacitor C2 in the conventional example 3 has a capacitance value much smaller than that of the capacitor C11, and is added in order to obtain a sufficiently low gain at the switching frequency of the power converter 1.
Japanese Patent Laid-Open No. 2000-30887 Japanese Patent Laid-Open No. 6-2231979

ここで、図２０は従来例１における誤差増幅器３のボード線図を破線トで表し、従来例２における誤差増幅器３のボード線図を実線チで表している。但し、ｆｓは電力変換部１のスイッチング周波数である。実線チ上の点Ａは、入力抵抗Ｒ３とコンデンサＣ２，Ｃ４で決まる時定数がスイッチング周波数ｆの逆数以上且つ１００μｓ以下の範囲に収まるという条件を満たしており、従来例１におけるジャンプ現象等の不具合が改善可能である。しかしながら、スイッチング周波数ｆｓにおけるゲインを比較した場合、従来例１におけるゲイン（点Ｃ）よりも従来例２におけるゲイン（点Ｂ）が増加しており、スイッチング周波数ｆｓあるいはそれ以上の周波数成分の影響が大きくなり、フィードバック部２におけるフィードバック制御が不安定になる虞がある。すなわち、一般的に電力変換部１の出力を増大させる場合にはスイッチング周波数ｆｓを低くし、放電灯ＦＬの定格点灯時にスイッチング周波数ｆｓが最も低くなり、且つスイッチング周波数ｆｓが低くなるに従って誤差増幅器３のゲインが増加するので、定格点灯時にフィードバック制御が不安定になる虞がある。また放電灯ＦＬの光出力を下限まで落とした調光下限時、特に放電灯ＦＬのインピーダンス値が上昇する低温時において、放電灯ＦＬの印加電圧（電力変換部１の出力電圧）の上昇に対して放電灯ＦＬの点灯を維持するために、フィードバック部２におけるフィードバック制御のスイッチング周波数ｆｓの変化幅が定格点灯時等に比べて非常に大きくなり、電力変換部１のスイッチング素子Ｑ１，Ｑ２に過大なストレスが加わったり、インダクタＬｘから騒音発生といった新たな問題が生じる虞があった。ここで、図２１は上述の調光下限時にフィードバック制御を行ったときのランプ電流並びにスイッチング周波数ｆｓの発振の様子を表している。   Here, FIG. 20 shows a Bode diagram of the error amplifier 3 in the conventional example 1 by a broken line and a Bode diagram of the error amplifier 3 in the conventional example 2 by a solid line h. However, fs is the switching frequency of the power converter 1. The point A on the solid line H satisfies the condition that the time constant determined by the input resistor R3 and the capacitors C2 and C4 is within the range of the reciprocal of the switching frequency f and 100 μs or less. Can be improved. However, when the gain at the switching frequency fs is compared, the gain (point B) in the conventional example 2 is larger than the gain (point C) in the conventional example 1, and the influence of the frequency component of the switching frequency fs or higher is increased. There is a possibility that the feedback control in the feedback unit 2 becomes unstable due to the increase. That is, in general, when the output of the power conversion unit 1 is increased, the switching frequency fs is lowered, the switching frequency fs is lowest when the discharge lamp FL is rated-lit, and the error amplifier 3 is decreased as the switching frequency fs is lowered. As the gain increases, feedback control may become unstable when the rated lighting is on. In addition, when the light control lower limit when the light output of the discharge lamp FL is lowered to the lower limit, particularly at low temperatures where the impedance value of the discharge lamp FL increases, the applied voltage of the discharge lamp FL (output voltage of the power conversion unit 1) increases. In order to maintain the lighting of the discharge lamp FL, the change width of the switching frequency fs of the feedback control in the feedback unit 2 becomes very large compared to the rated lighting or the like, and the switching elements Q1 and Q2 of the power conversion unit 1 are excessively large. There is a possibility that a new problem such as the generation of noise from the inductor Lx may occur or the stress may be applied. Here, FIG. 21 shows the state of oscillation of the lamp current and the switching frequency fs when feedback control is performed at the dimming lower limit.

一方、従来例３においては、既に説明したように従来例１の不具合が改善できるとともにコンデンサＣ２によってスイッチング周波数ｆｓにおけるゲインを低くしていることから従来例２のようにフィードバック制御が不安定になることを抑えることが可能である。尚、従来例３における誤差増幅器３では直流ゲインがオペアンプＯＰが持つ最大ゲインで制約され、オペアンプＯＰのばらつきや最大ゲインの変動によりフィードバックループの極にばらつきや変動を生じることになるので、実用上は従来例１のようにコンデンサＣ２と並列に抵抗Ｒ２を接続して安定に動作するようにしている。ここで、抵抗Ｒ２を含めた誤差増幅器３の伝達関数Ｇ２（ｓ）は、下記の式１で表される。   On the other hand, in the conventional example 3, as described above, the problem of the conventional example 1 can be improved, and the gain at the switching frequency fs is lowered by the capacitor C2, so that the feedback control becomes unstable as in the conventional example 2. It is possible to suppress this. In the error amplifier 3 in the conventional example 3, the direct current gain is limited by the maximum gain of the operational amplifier OP, and the variation of the operational amplifier OP and the variation of the maximum gain cause variations and variations in the poles of the feedback loop. As in Conventional Example 1, a resistor R2 is connected in parallel with the capacitor C2 so as to operate stably. Here, the transfer function G2 (s) of the error amplifier 3 including the resistor R2 is expressed by the following Equation 1.

Figure 2005071874
Figure 2005071874

尚、ｓはラプラス演算子を表し、基準電圧Ｖｄはゼロとしている。また比較のために従来例１における誤差増幅器３の伝達関数Ｇ１（ｓ）を求めると下記の式２で表される。 Note that s represents a Laplace operator, and the reference voltage Vd is zero. For comparison, the transfer function G1 (s) of the error amplifier 3 in the conventional example 1 is obtained by the following equation 2.

Figure 2005071874
Figure 2005071874

式１の右辺第２項の分子（ｓＣ１１Ｒ１１＋１）がフィードバック制御の周波数応答の零点であり、Ｃ１１Ｒ１１の値を放電灯ＦＬの動特性の極を打ち消すように選択することでランプ電流包絡線の発振を抑えることができる。しかしながら、式１の右辺第２項の分母で決まるフィードバック制御の極はＲ２，Ｃ２，Ｒ１１，Ｃ１１の４つの変数を適切に設定しなければならず、また従来例１の抵抗Ｒ２及びコンデンサＣ２の値から大きく変更することを余儀なくされる。従来の放電灯点灯装置において、種々の放電灯に対して電力変換部１や検出回路の回路定数、その他の周辺回路の回路定数がそれぞれ最適値に設定されており、さらなる低光束調光の実現に向けて従来例２の構成を採用すると、フィードバック部２（誤差増幅器３）の回路定数を一から設計し直す必要があった。 The numerator (sC11R11 + 1) in the second term on the right side of Equation 1 is the zero point of the frequency response of the feedback control. By selecting the value of C11R11 so as to cancel the pole of the dynamic characteristics of the discharge lamp FL, the oscillation of the lamp current envelope is performed. Can be suppressed. However, the pole of the feedback control determined by the denominator of the second term on the right side of Equation 1 must appropriately set four variables R2, C2, R11, and C11, and the resistance R2 and the capacitor C2 of the conventional example 1 Forced to change significantly from the value. In the conventional discharge lamp lighting device, the circuit constants of the power conversion unit 1 and the detection circuit and the circuit constants of other peripheral circuits are set to optimum values for various discharge lamps, thereby realizing further low luminous flux dimming. When the configuration of the conventional example 2 is adopted toward the above, it is necessary to redesign the circuit constant of the feedback unit 2 (error amplifier 3) from scratch.

本発明は上記事情に鑑みて為されたものであり、その目的は、低光束調光時のランプ電流の発振を抑えてクレストファクタの低減及びちらつきや立ち消えを防止するとともに安定したフィードバック制御が可能な放電灯点灯装置を提供することにある。   The present invention has been made in view of the above circumstances, and its purpose is to suppress lamp current oscillation during low beam dimming, to reduce crest factor, to prevent flicker and extinction, and to enable stable feedback control Is to provide a simple discharge lamp lighting device.

請求項１の発明は、上記目的を達成するために、スイッチング素子をスイッチング制御することで直流電源の直流電力を高周波電力に変換して放電灯に供給する電力変換部と、放電灯の調光比に応じた基準値を電力変換部の高周波出力と比較し該高周波出力を基準値に一致させる方向へスイッチング制御の制御量を調整するフィードバック部とを備え、該フィードバック部は、高周波出力に対応する検出電圧と基準値に対応する基準電圧との差分を増幅する誤差増幅器を有する放電灯点灯装置において、前記誤差増幅器は、第１の入力端子に検出電圧が入力されるとともに第２の入力端子に基準電圧が入力されるオペアンプと、第１の入力端子に直列接続される入力抵抗と、オペアンプの出力端子と第１の入力端子との間に接続されるインピーダンス素子と、前記入力抵抗と並列に接続される第１の抵抗及び第１のコンデンサの直列回路とを具備することを特徴とする。   In order to achieve the above object, a first aspect of the present invention provides a power conversion unit that converts DC power of a DC power source into high-frequency power by switching control of a switching element and supplies the high-frequency power to a discharge lamp, and dimming of the discharge lamp A feedback unit that compares the reference value according to the ratio with the high-frequency output of the power conversion unit and adjusts the control amount of the switching control in a direction to match the high-frequency output with the reference value, the feedback unit corresponds to the high-frequency output In a discharge lamp lighting device having an error amplifier for amplifying a difference between a detection voltage to be detected and a reference voltage corresponding to a reference value, the error amplifier has a detection voltage input to a first input terminal and a second input terminal An operational amplifier to which a reference voltage is input, an input resistor connected in series to the first input terminal, and an impedance connected between the output terminal and the first input terminal of the operational amplifier. Dance elements, characterized by comprising a series circuit of a first resistor and a first capacitor connected in parallel with the input resistor.

この発明によれば、第１の抵抗の抵抗値及び第２のコンデンサの容量値を適当な値に設定することにより、低光束調光時のランプ電流の発振を抑えてクレストファクタの低減及びちらつきや立ち消えが防止できるとともに安定したフィードバック制御が可能となる。   According to the present invention, by setting the resistance value of the first resistor and the capacitance value of the second capacitor to appropriate values, the oscillation of the lamp current during low beam dimming can be suppressed to reduce the crest factor and flicker. In addition, it is possible to prevent disappearance and stable feedback control.

請求項２の発明は、請求項１の発明において、前記第１の抵抗の抵抗値をＲ１、第１のコンデンサの容量値をＣ１、円周率をπとしたとき、０．１ｋＨｚ≦１／（２π×Ｃ１×Ｒ１）≦１０ｋＨｚの関係を満たすように第１の抵抗の抵抗値及び第１のコンデンサの容量値が設定されることを特徴とする。   According to a second aspect of the present invention, in the first aspect of the invention, when the resistance value of the first resistor is R1, the capacitance value of the first capacitor is C1, and the circumferential ratio is π, 0.1 kHz ≦ 1/1 / The resistance value of the first resistor and the capacitance value of the first capacitor are set so as to satisfy the relationship of (2π × C1 × R1) ≦ 10 kHz.

この発明によれば、より効果的に低光束調光時の安定点灯が実現可能となる。   According to the present invention, it is possible to more effectively realize stable lighting at the time of low beam dimming.

請求項３の発明は、請求項１の発明において、前記インピーダンス素子は第２の抵抗と第２のコンデンサの並列回路からなり、前記第１の抵抗の抵抗値をＲ１、第２の抵抗の抵抗値をＲ２、入力抵抗の抵抗値をＲ３、第１のコンデンサの容量値をＣ１、第２のコンデンサの容量値をＣ２、円周率をπとしたとき、１／（２π×Ｃ２×Ｒ２）≦１／｛２π×Ｃ１×（Ｒ１＋Ｒ３）｝の関係を満たすように第１の抵抗、第２の抵抗、入力抵抗の各抵抗値及びに第１及び第２のコンデンサの容量値が設定されることを特徴とする。   According to a third aspect of the present invention, in the first aspect of the present invention, the impedance element comprises a parallel circuit of a second resistor and a second capacitor, the resistance value of the first resistor is R1, and the resistance of the second resistor When the value is R2, the resistance value of the input resistor is R3, the capacitance value of the first capacitor is C1, the capacitance value of the second capacitor is C2, and the circumference is π, 1 / (2π × C2 × R2) The resistance values of the first resistor, the second resistor, and the input resistor and the capacitance values of the first and second capacitors are set so as to satisfy the relationship of ≦ 1 / {2π × C1 × (R1 + R3)}. It is characterized by that.

この発明によれば、より効果的に低光束調光時の安定点灯が実現可能となる。   According to the present invention, it is possible to more effectively realize stable lighting at the time of low beam dimming.

請求項４の発明は、請求項１又は２又は３の発明において、前記第１の抵抗の抵抗値をＲ１、入力抵抗の抵抗値をＲ３としたとき、０．１≦Ｒ１／（Ｒ１＋Ｒ３）≦０．５の関係を満たすように第１の抵抗及び入力抵抗の各抵抗値が設定されることを特徴とする。   According to a fourth aspect of the present invention, in the first, second or third aspect of the invention, when the resistance value of the first resistor is R1 and the resistance value of the input resistance is R3, 0.1 ≦ R1 / (R1 + R3) ≦ Each resistance value of the first resistor and the input resistor is set so as to satisfy the relationship of 0.5.

この発明によれば、より効果的に低光束調光時の安定点灯が実現可能となる。   According to the present invention, it is possible to more effectively realize stable lighting at the time of low beam dimming.

請求項５の発明は、請求項１〜４の何れかの発明において、前記誤差増幅器は、第２の抵抗と第２のコンデンサの並列回路からなる前記インピーダンス素子と、第１の入力端子とグランドの間に接続される第３のコンデンサとを具備し、第２の抵抗の抵抗値をＲ２、第３のコンデンサの容量値をＣ３、円周率をπとしたとき、１／（２π×Ｒ２×Ｃ３）＞１ｋＨｚの関係を満たすように第２の抵抗の抵抗値及び第３のコンデンサの容量値が設定されることを特徴とする。   According to a fifth aspect of the present invention, in any one of the first to fourth aspects, the error amplifier includes the impedance element formed of a parallel circuit of a second resistor and a second capacitor, a first input terminal, and a ground. And a third capacitor connected between the first and second capacitors, wherein the resistance value of the second resistor is R2, the capacitance value of the third capacitor is C3, and the circumference is π, 1 / (2π × R2 The resistance value of the second resistor and the capacitance value of the third capacitor are set so as to satisfy the relationship of × C3)> 1 kHz.

この発明によれば、高周波出力に含まれるノイズ成分による誤動作が防止できる。   According to the present invention, it is possible to prevent malfunction due to noise components included in the high-frequency output.

請求項６の発明は、請求項１〜４の何れかの発明において、入力抵抗の前段にローパスフィルタが設けられ、該ローパスフィルタのカットオフ周波数が１ｋＨｚよりも大きい値に設定されることを特徴とする。   According to a sixth aspect of the present invention, in any one of the first to fourth aspects, a low-pass filter is provided in front of the input resistance, and the cutoff frequency of the low-pass filter is set to a value larger than 1 kHz. And

この発明によれば、高周波出力に含まれるノイズ成分による誤動作が防止できる。   According to the present invention, it is possible to prevent malfunction due to noise components included in the high-frequency output.

請求項７の発明は、請求項１〜６の何れかの発明において、前記フィードバック部は、電力変換部のスイッチング素子に流れるスイッチング電流を前記高周波出力として基準値と比較することを特徴とする。   The invention of claim 7 is the invention according to any one of claims 1 to 6, wherein the feedback unit compares a switching current flowing through a switching element of a power conversion unit with a reference value as the high-frequency output.

請求項８の発明は、請求項１〜６の何れかの発明において、前記フィードバック部は、放電灯に流れるランプ電流を前記高周波出力として基準値と比較することを特徴とする。   The invention of claim 8 is characterized in that, in the invention of any one of claims 1 to 6, the feedback section compares the lamp current flowing in the discharge lamp with a reference value as the high-frequency output.

本発明によれば、第１の抵抗の抵抗値及び第２のコンデンサの容量値を適当な値に設定することにより、低光束調光時のランプ電流の発振を抑えてクレストファクタの低減及びちらつきや立ち消えが防止できるとともに安定したフィードバック制御が可能となる。   According to the present invention, by setting the resistance value of the first resistor and the capacitance value of the second capacitor to appropriate values, the crest factor can be reduced and the flicker can be suppressed by suppressing the oscillation of the lamp current at the time of low beam dimming. In addition, it is possible to prevent disappearance and stable feedback control.

（実施形態１）
本実施形態は、図１に示すように従来例１とほぼ共通の回路構成を有し、誤差増幅器３を構成するオペアンプＯＰの入力抵抗Ｒ３に抵抗Ｒ１とコンデンサＣ１の直列回路を並列接続した点に特徴がある。但し、他の構成は従来例１と共通であるから、共通の構成要素には同一の符号を付して説明を省略する。 (Embodiment 1)
As shown in FIG. 1, the present embodiment has a circuit configuration almost the same as that of the conventional example 1, and a series circuit of a resistor R1 and a capacitor C1 is connected in parallel to an input resistor R3 of an operational amplifier OP constituting the error amplifier 3. There is a feature. However, since the other configuration is common to Conventional Example 1, common components are denoted by the same reference numerals and description thereof is omitted.

フィードバック部２では高周波出力に対応する検出電圧Ｖｓと基準値に対応する基準電圧Ｖｄとの差分を誤差増幅器３で増幅し、検出電圧Ｖｓが基準電圧Ｖｄよりも高ければＶ−ｆ変換器４とドライバ回路４を介してスイッチング素子Ｑ１，Ｑ２のスイッチング周波数を上昇方向へ制御することで電力変換部１の出力を低下させ、反対に検出電圧Ｖｓが基準電圧Ｖｄよりも低ければＶ−ｆ変換器４とドライバ回路４を介してスイッチング素子Ｑ１，Ｑ２のスイッチング周波数を下降方向へ制御することで電力変換部１の出力を増大させている。このようなフィードバック部２のフィードバック制御によって放電灯ＦＬへの供給電力を一定値に維持し、基準電圧Ｖｄを変化させることで放電灯ＦＬへの供給電力を増減して放電灯ＦＬを調光可能としている。   The feedback unit 2 amplifies the difference between the detection voltage Vs corresponding to the high frequency output and the reference voltage Vd corresponding to the reference value by the error amplifier 3, and if the detection voltage Vs is higher than the reference voltage Vd, By controlling the switching frequency of the switching elements Q1, Q2 in the increasing direction via the driver circuit 4, the output of the power conversion unit 1 is decreased. Conversely, if the detection voltage Vs is lower than the reference voltage Vd, a Vf converter. 4 and the driver circuit 4 to control the switching frequency of the switching elements Q1 and Q2 in the downward direction, thereby increasing the output of the power converter 1. By such feedback control of the feedback unit 2, the power supplied to the discharge lamp FL is maintained at a constant value, and by changing the reference voltage Vd, the power supplied to the discharge lamp FL can be increased or decreased to dim the discharge lamp FL. It is said.

ここで、本発明者は従来例１における誤差増幅器３の回路定数（抵抗Ｒ２，Ｒ３の抵抗値やコンデンサＣ２）を変更して周波数特性や位相特性を求める実験を行った結果、図２０に示した誤差増幅器３の場合、図１６に示すように放電灯ＦＬに流れるランプ電流の発振する周波数が、図２０において位相が−９０ｄｅｇとなる付近の周波数、すなわち本実施形態における抵抗Ｒ２とコンデンサＣ２で決まるカットオフ周波数ｆｃ（＝１／２πＣ２Ｒ２）の約１０倍付近にあることを見いだした。但し、放電灯ＦＬの出力レベルや周囲温度により、放電灯ＦＬの特性の違いに起因して上記発振周波数は変化するものと考えられる。   Here, the inventor conducted an experiment to obtain frequency characteristics and phase characteristics by changing the circuit constants (resistance values of the resistors R2 and R3 and the capacitor C2) of the error amplifier 3 in the conventional example 1. As a result, as shown in FIG. In the case of the error amplifier 3, as shown in FIG. 16, the oscillation frequency of the lamp current flowing through the discharge lamp FL is a frequency in the vicinity of the phase of −90 deg in FIG. 20, that is, the resistor R2 and the capacitor C2 in this embodiment. It was found that the cutoff frequency was about 10 times the determined cutoff frequency fc (= 1 / 2πC2R2). However, it is considered that the oscillation frequency changes due to the difference in characteristics of the discharge lamp FL depending on the output level of the discharge lamp FL and the ambient temperature.

ここで２つの例のランプ電流、スイッチング周波数、発振周波数並びにカットオフ周波数ｆｃの関係を図２に示し、各例のボード線図を図３に示す。すなわち、第１例では図２（ａ）に示すようにカットオフ周波数ｆｃが約４８Ｈｚであるときにランプ電流の発振周波数が約５３０Ｈｚとなり、第２例では同図（ｂ）に示すようにカットオフ周波数ｆｃが約１６０Ｈｚであるときにランプ電流の発振周波数が約１ｋＨｚとなっていた。図２（ｂ）から明らかなように、カットオフ周波数ｆｃが１ｋＨｚ付近若しくはそれ以上になると、ランプ電流の発振振幅が減少してクレストファクタが低減されているので、ちらつきが低減できるとともに一時的に電力変換部１の出力が低下して放電灯ＦＬが立ち消えを起こすこともない。つまり、カットオフ周波数ｆｃを１ｋＨｚ付近での位相が−９０ｄｅｇよりも大きくなるようにすれば、ランプ電流の発振振幅を抑制してクレストファクタを低減し、ちらつきや立ち消えを抑えることが可能である。   Here, the relationship between the lamp current, the switching frequency, the oscillation frequency, and the cutoff frequency fc of the two examples is shown in FIG. 2, and the Bode diagram of each example is shown in FIG. That is, in the first example, as shown in FIG. 2A, when the cutoff frequency fc is about 48 Hz, the oscillation frequency of the lamp current is about 530 Hz. In the second example, the cutoff frequency is cut as shown in FIG. When the off-frequency fc is about 160 Hz, the oscillation frequency of the lamp current is about 1 kHz. As apparent from FIG. 2B, when the cut-off frequency fc is around 1 kHz or higher, the oscillation amplitude of the lamp current is reduced and the crest factor is reduced, so that flicker can be reduced and temporarily. The output of the power conversion unit 1 does not decrease and the discharge lamp FL does not turn off. In other words, if the cutoff frequency fc is set so that the phase in the vicinity of 1 kHz is larger than −90 deg, it is possible to suppress the oscillation amplitude of the lamp current to reduce the crest factor, and to suppress flickering and extinction.

ここで、本実施形態における誤差増幅器３の伝達関数Ｇ０（ｓ）を求めると下記の式３で表される。   Here, when the transfer function G0 (s) of the error amplifier 3 in the present embodiment is obtained, it is expressed by the following Expression 3.

Figure 2005071874
Figure 2005071874

上記式３で表される本実施形態における誤差増幅器３の伝達関数Ｇ０（ｓ）、並びに式１で表される従来例２における誤差増幅器３の伝達関数Ｇ２（ｓ）のボード線図を図４に示し、実線リがＧ０（ｓ）のゲイン及び位相、破線チがＧ１（ｓ）のゲイン及び位相、一点破線ルがＧ２（ｓ）のゲイン及び位相をそれぞれ表している。また、伝達関数Ｇ０（ｓ）をＧ１（ｓ）とＧａ（ｓ）に分けたボード線図を図５に示し、破線ヲがＧ１（ｓ）のゲイン及び位相、実線ワがＧａ（ｓ）のゲイン及び位相をそれぞれ表している。但し、放電灯ＦＬや電力変換部１の回路定数などによって誤差増幅器３に必要なゲイン（直流ゲイン）が変わるため、図４及び図５では縦軸のゲイン値を明記していない。 4 is a Bode diagram of the transfer function G0 (s) of the error amplifier 3 in the present embodiment expressed by the above equation 3 and the transfer function G2 (s) of the error amplifier 3 in the conventional example 2 expressed by the equation 1. FIG. The solid line represents the gain and phase of G0 (s), the broken line represents the gain and phase of G1 (s), and the dashed line represents the gain and phase of G2 (s). Further, a Bode diagram in which the transfer function G0 (s) is divided into G1 (s) and Ga (s) is shown in FIG. 5, the broken line is the gain and phase of G1 (s), and the solid line is Ga (s). Each represents a gain and a phase. However, since the gain (DC gain) required for the error amplifier 3 varies depending on the circuit constants of the discharge lamp FL and the power conversion unit 1, the gain value on the vertical axis is not clearly shown in FIGS.

式３から明らかなように、本実施形態における誤差増幅器３の伝達関数Ｇ０（ｓ）は、従来例１における誤差増幅器３の伝達関数Ｇ１（ｓ）とＧａ（ｓ）との積で表される。これは図５のボード線図上において、ゲイン及び位相とも２つの伝達関数Ｇ１（ｓ），Ｇａ（ｓ）の和が伝達関数Ｇ０（ｓ）となることを示している。従って、伝達関数Ｇａ（ｓ）の特性のみに着目して回路定数を設定することにより、従来の放電灯点灯装置から簡単に調光下限を引き下げることが可能となる。この点において、大幅な回路定数の変更が必要となる従来例３に比較して本実施形態が設計上有利となる。また、上述のように１ｋＨｚ付近での位相が−９０ｄｅｇよりも大きくなるようにした場合、図４に示すようにスイッチング周波数ｆｓ（１ｋＨｚ）付近での誤差増幅器３のゲインを従来例２における誤差増幅器３のゲインよりも減少させることができる。   As apparent from Equation 3, the transfer function G0 (s) of the error amplifier 3 in the present embodiment is represented by the product of the transfer function G1 (s) and Ga (s) of the error amplifier 3 in the conventional example 1. . This indicates that the sum of the two transfer functions G1 (s) and Ga (s) in both the gain and phase becomes the transfer function G0 (s) on the Bode diagram of FIG. Therefore, by setting the circuit constant while paying attention only to the characteristics of the transfer function Ga (s), it is possible to easily lower the dimming lower limit from the conventional discharge lamp lighting device. In this respect, the present embodiment is advantageous in design as compared with the conventional example 3 in which a large change in circuit constant is required. When the phase near 1 kHz is larger than −90 deg as described above, the gain of the error amplifier 3 near the switching frequency fs (1 kHz) is set to the error amplifier in the conventional example 2 as shown in FIG. The gain can be reduced to less than 3.

而して、本実施形態における低光束調光時のランプ電流並びにスイッチング周波数を計測したところ、図６に示すようにランプ電流のクレストファクタを１．５以下にまで抑えることができ、目視によるちらつきも確認されなかった。また、スイッチング周波数ｆｓの変動幅も抑えられており、インダクタによる騒音などの不具合も問題はなかった。さらに、定格点灯時の動作確認も行ったが、従来例２のような不安定な動作も生じなかった。   Thus, when the lamp current and the switching frequency at the time of low luminous flux dimming in this embodiment were measured, the crest factor of the lamp current can be suppressed to 1.5 or less as shown in FIG. Also not confirmed. Further, the fluctuation range of the switching frequency fs is suppressed, and there is no problem with problems such as noise caused by the inductor. Further, although the operation was confirmed at the time of rated lighting, the unstable operation as in the conventional example 2 did not occur.

ところで、コンデンサＣ１と抵抗Ｒ１で決まるカットオフ周波数ｆp2（＝１／２πＣ１Ｒ１）を変えて誤差増幅器３のゲイン特性及び位相特性を調べたところ、図７に示す結果が得られた。ここで、図７における破線ｄはカットオフ周波数ｆp2を低くした場合の伝達関数Ｇ０（ｓ）、一点破線ｅはカットオフ周波数ｆp2を高くした場合の伝達関数Ｇ０（ｓ）、実線ｆはカットオフ周波数ｆp2をその中間とした場合の伝達関数Ｇ０（ｓ）のゲイン特性及び位相特性をそれぞれ示している。上述のように従来例１〜３の課題を解決するには１ｋＨｚ付近における誤差増幅器３の位相を−９０ｄｅｇよりも大きくすることが必要であるが、カットオフ周波数ｆp2を低くすると従来例１における位相特性に近付いてしまうことから、少なくともカットオフ周波数ｆp2を０．１ｋＨｚ以上に設定する必要のあることが判った。一方、カットオフ周波数ｆp2を高くすると１ｋＨｚ付近での位相は大きくなるが、それとともにスイッチング周波数付近のゲインも上昇してしまうことから、カットオフ周波数ｆp2は少なくとも１０ｋＨｚ以下とし、好適には１ｋＨｚ以下に設定することが望ましいことが判った。すなわち、カットオフ周波数ｆp2が０．１ｋＨｚ以上且つ１０ｋＨｚ以下、好適には０．１ｋＨｚ以上且つ１ｋＨｚ以下の関係を満たすようにコンデンサＣ１の容量値及び抵抗Ｒ１の抵抗値を設定すれば、一層効果的に低光束調光時の安定点灯が実現可能となる。   By the way, when the gain characteristic and the phase characteristic of the error amplifier 3 were examined by changing the cutoff frequency fp2 (= 1 / 2πC1R1) determined by the capacitor C1 and the resistor R1, the result shown in FIG. 7 was obtained. Here, the broken line d in FIG. 7 is the transfer function G0 (s) when the cut-off frequency fp2 is lowered, the one-dot broken line e is the transfer function G0 (s) when the cut-off frequency fp2 is raised, and the solid line f is the cut-off. The gain characteristic and phase characteristic of the transfer function G0 (s) when the frequency fp2 is in the middle are shown. As described above, in order to solve the problems of the conventional examples 1 to 3, it is necessary to make the phase of the error amplifier 3 near 1 kHz larger than −90 deg. However, if the cutoff frequency fp2 is lowered, the phase in the conventional example 1 is reached. Since it approaches the characteristics, it has been found that at least the cutoff frequency fp2 needs to be set to 0.1 kHz or more. On the other hand, if the cutoff frequency fp2 is increased, the phase near 1 kHz increases, but the gain near the switching frequency also increases. Therefore, the cutoff frequency fp2 is at least 10 kHz or less, preferably 1 kHz or less. I found it desirable to set. That is, it is more effective if the capacitance value of the capacitor C1 and the resistance value of the resistor R1 are set so that the cutoff frequency fp2 satisfies the relationship of 0.1 kHz to 10 kHz, preferably 0.1 kHz to 1 kHz. In addition, stable lighting at the time of low beam dimming can be realized.

尚、図８に示すように電力変換部１を従来例２と同様のハーフブリッジ型のインバータ回路で構成し、ローサイドのスイッチング素子Ｑ２に直列接続された検出素子Ｚ１の両端電圧（検出電圧Ｖｓ）をフィードバック部２に入力して基準電圧Ｖｄと比較する構成とすれば、放電灯ＦＬへの供給電力を安定化することが可能であり、図９に示すように従来例３と同様のランプ電流検出回路６で放電灯ＦＬに流れるランプ電流を検出した検出電圧Ｖｘをフィードバック部２に入力して基準電圧Ｖｄと比較する構成とすれば、放電灯ＦＬに流れるランプ電流を安定化することが可能である。   As shown in FIG. 8, the power conversion unit 1 is configured by a half-bridge inverter circuit similar to that of the conventional example 2, and the voltage across the detection element Z1 connected in series to the low-side switching element Q2 (detection voltage Vs). Is input to the feedback unit 2 and compared with the reference voltage Vd, the power supplied to the discharge lamp FL can be stabilized, and as shown in FIG. If the detection voltage Vx obtained by detecting the lamp current flowing through the discharge lamp FL by the detection circuit 6 is input to the feedback unit 2 and compared with the reference voltage Vd, the lamp current flowing through the discharge lamp FL can be stabilized. It is.

（実施形態２）
ところで実施形態１の回路構成において、入力抵抗Ｒ３とコンデンサＣ１及び抵抗Ｒ１の並列回路におけるカットオフ周波数ｆz1（＝１／２πＣ１（Ｒ１＋Ｒ３））と、オペアンプＯＰの入出力間に並列接続されているコンデンサＣ２及び抵抗Ｒ２のカットオフ周波数ｆｃとの値が近くなるように抵抗Ｒ１〜Ｒ３の抵抗値及びコンデンサＣ１，Ｃ２の容量値を設定したときの誤差増幅器３のゲイン特性及び位相特性を調べたところ、図１０に示す結果が得られた。ここで、図１０における破線ｇはスイッチング周波数付近のゲインを実施形態１とほぼ等しくした場合の伝達関数Ｇ０（ｓ）、一点破線ｈは１ｋＨｚ付近での位相を実施形態１とほぼ等しくした場合の伝達関数Ｇ０（ｓ）、実線ｉはｆz1とｆｃの値を近付けていない場合の伝達関数Ｇ０（ｓ）のゲイン特性及び位相特性をそれぞれ示している。一点破線ｈではスイッチング周波数付近のゲインが高くなっており、破線ｇでは従来例１の特性に近いものとなっている。すなわち、カットオフ周波数ｆz1を高くすれば一点破線ｈに対してゲインを抑えつつ位相を補償することが可能であり、これらのことから少なくともｆｃ≦ｆz1の関係を満たすように抵抗Ｒ１〜Ｒ３の抵抗値及びコンデンサＣ１，Ｃ２の容量値を設定すれば、より効果的に低光束調光時の安定点灯が実現可能となる。 (Embodiment 2)
By the way, in the circuit configuration of the first embodiment, the cutoff frequency fz1 (= 1 / 2πC1 (R1 + R3)) in the parallel circuit of the input resistor R3, the capacitor C1, and the resistor R1 is connected in parallel between the input and output of the operational amplifier OP. When the resistance values of the resistors R1 to R3 and the capacitance values of the capacitors C1 and C2 are set so that the values of the cutoff frequency fc of C2 and the resistor R2 are close, the gain characteristic and phase characteristic of the error amplifier 3 are examined. The result shown in FIG. 10 was obtained. Here, a broken line g in FIG. 10 is a transfer function G0 (s) when the gain near the switching frequency is substantially equal to that of the first embodiment, and a one-dot broken line h is a case where the phase near 1 kHz is substantially equal to that of the first embodiment. The transfer function G0 (s) and the solid line i indicate the gain characteristic and phase characteristic of the transfer function G0 (s) when the values of fz1 and fc are not close to each other. In the one-dot broken line h, the gain near the switching frequency is high, and in the broken line g, the characteristic is similar to that of the conventional example 1. That is, if the cut-off frequency fz1 is increased, it is possible to compensate the phase while suppressing the gain with respect to the one-dot broken line h. From these facts, the resistances of the resistors R1 to R3 satisfy at least the relationship of fc ≦ fz1. If the value and the capacitance values of the capacitors C1 and C2 are set, stable lighting at the time of low beam dimming can be realized more effectively.

（実施形態３）
実施形態１の回路構成において、カットオフ周波数ｆp2を略一定とする条件下で入力抵抗Ｒ３と抵抗Ｒ１の和と、抵抗Ｒ１との比ｋ（＝Ｒ１／（Ｒ１＋Ｒ３））を変えて誤差増幅器３のゲイン特性及び位相特性を調べたところ、図１１に示す結果が得られた。ここで、図１１における破線ｊは上記比ｋを大きくした場合の伝達関数Ｇ０（ｓ）、一点破線ｍは上記比ｋを小さくした場合の伝達関数Ｇ０（ｓ）、実線ｎは上記比ｋをその中間とした場合の伝達関数Ｇ０（ｓ）のゲイン特性及び位相特性をそれぞれ示している。破線ｊからは比ｋを大きく、すなわち１に近付けることは抵抗Ｒ１の抵抗値を大きくすることであり、これにより従来例１の特性に近付くことが判る。そのため、１ｋＨｚ付近で望ましい位相とするために、比ｋの値に上限を設定する必要があり、実験結果などからｋ≦０．５とすることが望ましいことが判った。一方、一点破線ｍから比ｋを小さくすると、１ｋＨｚ付近での位相が大きくなるが、それとともにスイッチング周波数付近のゲインが上昇するから、ゲインをある程度制限するために比ｋの値に下限を設定する必要があり、実験結果などからｋ≧０．０５とすることが望ましいことが判った。 (Embodiment 3)
In the circuit configuration of the first embodiment, the error amplifier 3 is changed by changing the ratio k (= R1 / (R1 + R3)) of the sum of the input resistance R3 and the resistance R1 and the resistance R1 under the condition that the cutoff frequency fp2 is substantially constant. When the gain characteristic and the phase characteristic were examined, the results shown in FIG. 11 were obtained. Here, the broken line j in FIG. 11 indicates the transfer function G0 (s) when the ratio k is increased, the one-dot broken line m indicates the transfer function G0 (s) when the ratio k is decreased, and the solid line n indicates the ratio k. The gain characteristic and phase characteristic of the transfer function G0 (s) in the middle are shown. From the broken line j, it can be seen that increasing the ratio k, that is, approaching 1 means increasing the resistance value of the resistor R1, thereby approaching the characteristics of Conventional Example 1. For this reason, in order to obtain a desirable phase in the vicinity of 1 kHz, it is necessary to set an upper limit for the value of the ratio k, and it has been found that k ≦ 0.5 is desirable from experimental results and the like. On the other hand, if the ratio k is decreased from the one-dot broken line m, the phase near 1 kHz increases, but the gain near the switching frequency increases with this, so a lower limit is set to the value of the ratio k to limit the gain to some extent. Therefore, it was found that k ≧ 0.05 is desirable from the experimental results.

而して、上記比ｋが０．０５≦ｋ≦０．５の関係を満たすように抵抗Ｒ１，Ｒ３の抵抗値を設定すれば、より効果的に低光束調光時の安定点灯が実現可能となる。   Thus, if the resistance values of the resistors R1 and R3 are set so that the ratio k satisfies the relationship of 0.05 ≦ k ≦ 0.5, stable lighting at the time of low beam dimming can be realized more effectively. It becomes.

（実施形態４）
本実施形態は、図１２に示すように実施形態１とほぼ共通の回路構成を有し、誤差増幅器３を構成するオペアンプＯＰの反転入力端子とグランド（基準電位）との間にコンデンサＣ３を接続するとともに、抵抗Ｒ２とコンデンサＣ３で決まるカットオフ周波数ｆp3（＝１／２πＣ３Ｒ２）が１ｋＨｚよりも大きくなるようにコンデンサＣ３の容量値を設定した点に特徴がある。但し、他の構成は実施形態１と共通であるから、共通の構成要素には同一の符号を付して説明を省略する。 (Embodiment 4)
As shown in FIG. 12, the present embodiment has a circuit configuration almost the same as that of the first embodiment, and a capacitor C3 is connected between the inverting input terminal of the operational amplifier OP constituting the error amplifier 3 and the ground (reference potential). In addition, the capacitance value of the capacitor C3 is set so that the cut-off frequency fp3 (= 1 / 2πC3R2) determined by the resistor R2 and the capacitor C3 is larger than 1 kHz. However, since other configurations are the same as those in the first embodiment, the same components are denoted by the same reference numerals and description thereof is omitted.

而して、図８に示した回路構成ではスイッチング素子Ｑ２に流れるスイッチング電流の検出電圧Ｖｓを誤差増幅器３に入力しており、この検出電圧Ｖｓに含まれる高周波成分がノイズとなってフィードバック部２が誤動作する虞があるが、本実施形態では検出電圧Ｖｓが入力するオペアンプＯＰの反転入力端とグランドの間にコンデンサＣ３を接続することで高周波成分（ノイズ）を低減し、フィードバック部２の誤動作を防止することができる。   Thus, in the circuit configuration shown in FIG. 8, the detection voltage Vs of the switching current flowing through the switching element Q2 is input to the error amplifier 3, and the high frequency component contained in this detection voltage Vs becomes noise and the feedback unit 2 In this embodiment, a high-frequency component (noise) is reduced by connecting a capacitor C3 between the inverting input terminal of the operational amplifier OP to which the detection voltage Vs is input and the ground, and the feedback unit 2 malfunctions. Can be prevented.

一方、コンデンサＣ３は誤差増幅器３の周波数特性に影響を与え、抵抗Ｒ３，Ｒ１やコンデンサＣ１を取り外したときに上記カットオフ周波数ｆp3に極を持つことになる。位相を補償したい周波数（１ｋＨｚ）付近にこの極があるとすると、実際にはカットオフ周波数ｆp3の１０分の１の周波数から位相が遅れるので、実施形態１で説明した効果が低下し、ひいては従来例１のような不具合が現れる虞がある。このことから、カットオフ周波数ｆp3は少なくとも１ｋＨｚよりも高くする必要があり、好適には１０ｋＨｚ以上となるようにコンデンサＣ３の容量値を設定することが望ましい。そうすれば、高周波ノイズによるフィードバック部２の誤動作を防止しつつ実施形態１と同様の効果を奏し得る。   On the other hand, the capacitor C3 affects the frequency characteristics of the error amplifier 3, and has a pole at the cut-off frequency fp3 when the resistors R3 and R1 and the capacitor C1 are removed. If this pole is in the vicinity of the frequency (1 kHz) at which the phase is to be compensated, the phase is actually delayed from one-tenth of the cut-off frequency fp3. There is a possibility that a problem such as that in Example 1 may appear. Therefore, the cut-off frequency fp3 needs to be higher than at least 1 kHz, and it is desirable to set the capacitance value of the capacitor C3 so that it is preferably 10 kHz or higher. If it does so, the same effect as Embodiment 1 can be produced, preventing malfunction of feedback part 2 by high frequency noise.

（実施形態５）
本実施形態は、図１３に示すように実施形態１とほぼ共通の回路構成を有し、誤差増幅器３の前段に抵抗Ｒ４とコンデンサＣ４からなるローパスフィルタ７を設けるとともに、ローパスフィルタ７のカットオフ周波数ｆｘ（＝１／２πＣ４Ｒ４）が１ｋＨｚよりも大きくなるように抵抗Ｒ４の抵抗値及びコンデンサＣ４の容量値を設定した点に特徴がある。但し、他の構成は実施形態１と共通であるから、共通の構成要素には同一の符号を付して説明を省略する。 (Embodiment 5)
As shown in FIG. 13, the present embodiment has a circuit configuration that is substantially the same as that of the first embodiment. A low-pass filter 7 including a resistor R4 and a capacitor C4 is provided in front of the error amplifier 3, and the low-pass filter 7 is cut off. It is characterized in that the resistance value of the resistor R4 and the capacitance value of the capacitor C4 are set so that the frequency fx (= 1 / 2πC4R4) is greater than 1 kHz. However, since other configurations are the same as those in the first embodiment, the same components are denoted by the same reference numerals and description thereof is omitted.

而して、本実施形態も実施形態４と同様にスイッチング素子Ｑ２に流れるスイッチング電流の検出電圧Ｖｓに含まれる高周波成をローパスフィルタ７により低減し、フィードバック部２の誤動作を防止することができる。しかしながら、抵抗Ｒ４及びコンデンサＣ４が誤差増幅器３の周波数特性に影響を与え、上記カットオフ周波数ｆｘに極を持つことにより、実際にはカットオフ周波数ｆｘの１０分の１の周波数から位相が遅れるので、実施形態１で説明した効果が低下し、ひいては従来例１のような不具合が現れる虞がある。このことから、カットオフ周波数ｆｘを少なくとも１ｋＨｚよりも高くする必要があり、好適には１０ｋＨｚ以上となるように抵抗Ｒ４の抵抗値及びコンデンサＣ４の容量値を設定することが望ましい。但し、抵抗Ｒ４の抵抗値を大きくしすぎると、ローパスフィルタ７を含めたフィードバック部２の直流ゲインが小さくなって必要がゲインが得られなくなるので、入力抵抗Ｒ３に対してローパスフィルタ７の抵抗Ｒ４の抵抗値を十分に小さくしておくことが望ましい。そうすれば、高周波ノイズによるフィードバック部２の誤動作を防止しつつ実施形態１と同様の効果を奏し得る。   Thus, in the present embodiment, similarly to the fourth embodiment, the high-frequency component included in the detection voltage Vs of the switching current flowing through the switching element Q2 can be reduced by the low-pass filter 7, and malfunction of the feedback unit 2 can be prevented. However, since the resistor R4 and the capacitor C4 affect the frequency characteristics of the error amplifier 3 and have a pole at the cut-off frequency fx, the phase is actually delayed from the 1 / 10th frequency of the cut-off frequency fx. In addition, the effect described in the first embodiment is deteriorated, and as a result, there is a possibility that a problem such as that of the conventional example 1 appears. For this reason, it is necessary to set the cutoff frequency fx to be higher than at least 1 kHz, and it is desirable to set the resistance value of the resistor R4 and the capacitance value of the capacitor C4 so as to be preferably 10 kHz or more. However, if the resistance value of the resistor R4 is excessively increased, the DC gain of the feedback unit 2 including the low-pass filter 7 is reduced and the necessary gain cannot be obtained. Therefore, the resistance R4 of the low-pass filter 7 with respect to the input resistor R3. It is desirable to make the resistance value of this sufficiently small. If it does so, the effect similar to Embodiment 1 can be show | played, preventing malfunctioning of the feedback part 2 by a high frequency noise.

実施形態１の概略回路構成図である。1 is a schematic circuit configuration diagram of Embodiment 1. FIG. 同上の動作説明図である。It is operation | movement explanatory drawing same as the above. 同上における誤差増幅器のボード線図である。It is a Bode diagram of an error amplifier in the same as the above. 同上における誤差増幅器のボード線図である。It is a Bode diagram of an error amplifier in the same as the above. 同上における誤差増幅器のボード線図である。It is a Bode diagram of an error amplifier in the same as the above. 同上におけるランプ電流及びスイッチング周波数の波形図である。It is a wave form diagram of a lamp current and switching frequency in the same as the above. 同上における誤差増幅器のボード線図である。It is a Bode diagram of an error amplifier in the same as the above. 同上における他の概略回路構成図である。It is another schematic circuit block diagram in the same as the above. 同上におけるさらに他の概略回路構成図である。It is another schematic circuit block diagram in the same as the above. 実施形態２における誤差増幅器のボード線図である。6 is a Bode diagram of an error amplifier in Embodiment 2. FIG. 実施形態３における誤差増幅器のボード線図である。FIG. 10 is a Bode diagram of an error amplifier according to a third embodiment. 実施形態４の概略回路構成図である。FIG. 6 is a schematic circuit configuration diagram of a fourth embodiment. 実施形態５の概略回路構成図である。10 is a schematic circuit configuration diagram of Embodiment 5. FIG. 従来例１の概略回路構成図である。It is a schematic circuit block diagram of the prior art example 1. 同上の動作説明図である。It is operation | movement explanatory drawing same as the above. 同上におけるランプ電流及びスイッチング周波数の波形図である。It is a wave form diagram of a lamp current and switching frequency in the same as the above. 従来例２の概略回路構成図である。It is a schematic circuit block diagram of the prior art example 2. 同上における誤差増幅器のボード線図である。It is a Bode diagram of an error amplifier in the same as the above. 従来例３の概略回路構成図である。It is a schematic circuit block diagram of the prior art example 3. 従来例１，２における誤差増幅器のボード線図である。It is a Bode diagram of an error amplifier in conventional examples 1 and 2. 従来例２におけるランプ電流及びスイッチング周波数の波形図である。It is a wave form diagram of a lamp current and switching frequency in example 2 of the past.

符号の説明Explanation of symbols

１ 電力変換部
２ フィードバック部
３ 誤差増幅器
Ｒ１，Ｒ２，Ｒ３ 抵抗
Ｃ１，Ｃ２ コンデンサ DESCRIPTION OF SYMBOLS 1 Power converter 2 Feedback part 3 Error amplifier R1, R2, R3 Resistor C1, C2 Capacitor

Claims (8)

スイッチング素子をスイッチング制御することで直流電源の直流電力を高周波電力に変換して放電灯に供給する電力変換部と、放電灯の調光比に応じた基準値を電力変換部の高周波出力と比較し該高周波出力を基準値に一致させる方向へスイッチング制御の制御量を調整するフィードバック部とを備え、該フィードバック部は、高周波出力に対応する検出電圧と基準値に対応する基準電圧との差分を増幅する誤差増幅器を有する放電灯点灯装置において、前記誤差増幅器は、第１の入力端子に検出電圧が入力されるとともに第２の入力端子に基準電圧が入力されるオペアンプと、第１の入力端子に直列接続される入力抵抗と、オペアンプの出力端子と第１の入力端子との間に接続されるインピーダンス素子と、前記入力抵抗と並列に接続される第１の抵抗及び第１のコンデンサの直列回路とを具備することを特徴とする放電灯点灯装置。   A power converter that converts the DC power of the DC power source into high-frequency power by switching control of the switching element and supplies it to the discharge lamp, and a reference value corresponding to the dimming ratio of the discharge lamp is compared with the high-frequency output of the power converter And a feedback unit that adjusts a control amount of the switching control in a direction to match the high-frequency output with a reference value, and the feedback unit calculates a difference between a detection voltage corresponding to the high-frequency output and a reference voltage corresponding to the reference value. In the discharge lamp lighting device having an error amplifier to be amplified, the error amplifier includes an operational amplifier in which a detection voltage is input to a first input terminal and a reference voltage is input to a second input terminal, and a first input terminal An input resistor connected in series, an impedance element connected between the output terminal of the operational amplifier and the first input terminal, and connected in parallel with the input resistor. The discharge lamp lighting apparatus characterized by comprising a series circuit of a first resistor and a first capacitor. 前記第１の抵抗の抵抗値をＲ１、第１のコンデンサの容量値をＣ１、円周率をπとしたとき、０．１ｋＨｚ≦１／（２π×Ｃ１×Ｒ１）≦１０ｋＨｚの関係を満たすように第１の抵抗の抵抗値及び第１のコンデンサの容量値が設定されることを特徴とする請求項１記載の放電灯点灯装置。   When the resistance value of the first resistor is R1, the capacitance value of the first capacitor is C1, and the circumference is π, the relationship of 0.1 kHz ≦ 1 / (2π × C1 × R1) ≦ 10 kHz is satisfied. The discharge lamp lighting device according to claim 1, wherein a resistance value of the first resistor and a capacitance value of the first capacitor are set. 前記インピーダンス素子は第２の抵抗と第２のコンデンサの並列回路からなり、前記第１の抵抗の抵抗値をＲ１、第２の抵抗の抵抗値をＲ２、入力抵抗の抵抗値をＲ３、第１のコンデンサの容量値をＣ１、第２のコンデンサの容量値をＣ２、円周率をπとしたとき、１／（２π×Ｃ２×Ｒ２）≦１／｛２π×Ｃ１×（Ｒ１＋Ｒ３）｝の関係を満たすように第１の抵抗、第２の抵抗、入力抵抗の各抵抗値及びに第１及び第２のコンデンサの容量値が設定されることを特徴とする請求項１記載の放電灯点灯装置。   The impedance element comprises a parallel circuit of a second resistor and a second capacitor, wherein the resistance value of the first resistor is R1, the resistance value of the second resistor is R2, the resistance value of the input resistor is R3, The relationship of 1 / (2π × C2 × R2) ≦ 1 / {2π × C1 × (R1 + R3)} where C1 is the capacitance value of the capacitor, C2 is the capacitance value of the second capacitor, and π is the circumference ratio. 2. The discharge lamp lighting device according to claim 1, wherein capacitance values of the first and second capacitors are set in the first resistance, the second resistance, and the input resistance so as to satisfy . 前記第１の抵抗の抵抗値をＲ１、入力抵抗の抵抗値をＲ３としたとき、０．１≦Ｒ１／（Ｒ１＋Ｒ３）≦０．５の関係を満たすように第１の抵抗及び入力抵抗の各抵抗値が設定されることを特徴とする請求項１又は２又は３記載の放電灯点灯装置。   When the resistance value of the first resistor is R1 and the resistance value of the input resistor is R3, each of the first resistor and the input resistor is set so as to satisfy the relationship of 0.1 ≦ R1 / (R1 + R3) ≦ 0.5. The discharge lamp lighting device according to claim 1, wherein a resistance value is set. 前記誤差増幅器は、第２の抵抗と第２のコンデンサの並列回路からなる前記インピーダンス素子と、第１の入力端子とグランドの間に接続される第３のコンデンサとを具備し、第２の抵抗の抵抗値をＲ２、第３のコンデンサの容量値をＣ３、円周率をπとしたとき、１／（２π×Ｒ２×Ｃ３）＞１ｋＨｚの関係を満たすように第２の抵抗の抵抗値及び第３のコンデンサの容量値が設定されることを特徴とする請求項１〜４の何れかに記載の放電灯点灯装置。   The error amplifier includes the impedance element including a parallel circuit of a second resistor and a second capacitor, and a third capacitor connected between the first input terminal and the ground, and a second resistor The resistance value of the second resistor and the resistance value of the second resistor so as to satisfy the relationship 1 / (2π × R2 × C3)> 1 kHz The discharge lamp lighting device according to any one of claims 1 to 4, wherein a capacitance value of the third capacitor is set. 入力抵抗の前段にローパスフィルタが設けられ、該ローパスフィルタのカットオフ周波数が１ｋＨｚよりも大きい値に設定されることを特徴とする請求項１〜４の何れかに記載の放電灯点灯装置。   The discharge lamp lighting device according to any one of claims 1 to 4, wherein a low-pass filter is provided in front of the input resistor, and a cutoff frequency of the low-pass filter is set to a value larger than 1 kHz. 前記フィードバック部は、電力変換部のスイッチング素子に流れるスイッチング電流を前記高周波出力として基準値と比較することを特徴とする請求項１〜６の何れかに記載の放電灯点灯装置。   The discharge lamp lighting device according to any one of claims 1 to 6, wherein the feedback unit compares a switching current flowing through a switching element of a power conversion unit with a reference value as the high-frequency output. 前記フィードバック部は、放電灯に流れるランプ電流を前記高周波出力として基準値と比較することを特徴とする請求項１〜６の何れかに記載の放電灯点灯装置。   The discharge lamp lighting device according to any one of claims 1 to 6, wherein the feedback unit compares a lamp current flowing through the discharge lamp with a reference value as the high-frequency output.
JP2003301878A 2003-08-26 2003-08-26 Fluorescent lamp lighting device Expired - Fee Related JP4676685B2 (en)

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Cited By (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN100410742C (en) * 2005-08-02 2008-08-13 凹凸科技（中国）有限公司 Converter controller having feedforward compensation, converting system and method for controlling same
JP2009224244A (en) * 2008-03-18 2009-10-01 Mitsubishi Electric Corp Discharge lamp lighting device, lighting device
WO2010079539A1 (en) * 2009-01-08 2010-07-15 パナソニック株式会社 Integrator circuit and δς modulator equipped with same
JP2014130061A (en) * 2012-12-28 2014-07-10 Fuji Electric Fa Components & Systems Co Ltd Direct current detecting device

Cited By (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN100410742C (en) * 2005-08-02 2008-08-13 凹凸科技（中国）有限公司 Converter controller having feedforward compensation, converting system and method for controlling same
JP2009224244A (en) * 2008-03-18 2009-10-01 Mitsubishi Electric Corp Discharge lamp lighting device, lighting device
WO2010079539A1 (en) * 2009-01-08 2010-07-15 パナソニック株式会社 Integrator circuit and δς modulator equipped with same
JP2014130061A (en) * 2012-12-28 2014-07-10 Fuji Electric Fa Components & Systems Co Ltd Direct current detecting device

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