JP2004500760A - Relaxation oscillator for transponder - Google Patents

Abstract

One or more parameters (eg, temperature, pressure) at the object (eg, tire 104) can be measured and the data stream (FIGS. 3C, 4B) can be sent to an external reader / interrogator (106). Relaxation oscillator for transponder. This transponder normally operates in passive mode, extracting its power (Vxx, Vcc, Vdd) from the RF interrogation signal received by the antenna device (210, 410), but also in active mode powered by battery. Can work. The transponder includes memories (238, 438) for storing measurements, calibration data, programmable trim settings (436b), and transponder IDs and the like. The measurement indication (N _T , N _P ) comprises a counting oscillation of the measurement signal (Fosc ′) during a fixed time window (t _W , t _T , t _P ). The measurement signal is generated by a relaxation oscillator (252, 312, 452, 512) driven by alternating charging and discharging of the measurement capacitors (C _FX and C _P ), and the charging rate of the capacitor can indicate current (temperature). It is a function of the measured current I (T) B) and the capacity (which can indicate pressure).

Description

【０１７２】
となり、位相経路φ２信号はインバータ５２０ｂにより反転されて信号
【０１７３】
【外２】

【０１７４】
となる。ここで名称の上に示されている線は論理が反転されていること、すなわち、「ＮＯＴ」、を示すので、たとえば、
【０１７５】
【外３】

【０１７６】
は「位相１ではない」こと（すなわち、ＰＨＡＳＥ１信号が高い、すなわち、論理レベル１であると、
【０１７７】
【外４】

【０１７８】
信号は低い、すなわち、論理レベル０）を意味する。ＰＨＡＳＥ１、ＰＨＡＳＥ２、
【０１７９】
【外５】

【０１８０】
、
【０１８１】
【外６】

【０１８２】
信号は、論理回路５３１への入力としてＣＡＰＴＵＲＥ　ＮＴ信号およびＣＡＰＴＵＲＥ　ＮＰ信号に組合わされる。その論理回路は６つの制御信号−２つのコンデンサ選択信号、すなわち、ＣＦＸ　ＴＯ　ＯＳＣ（＃１）およびＣＰ　ＴＯ　ＯＳＣ（＃２）と、４つの測定制御信号ＰＨＡＳＥ１　ＮＴ（＃３）、ＰＨＡＳＥ２　ＮＴ（＃４）、ＰＨＡＳＥ１　ＮＰ（＃５）、およびＰＨＡＳＥ２　ＮＰ（＃６）とを出力する。
【０１８３】
論理５３１に含まれているデジタル信号の詳細が、図５に示されている論理回路例５３１ａ．．５３１ｆ（それぞれ信号（＃１）．．（＃６）のため）として示されている。各論理回路５３１ａ．．５３１ｆは論理回路５３１から出力すべきそれの制御信号を生ずるためにＮＡＮＤゲートと１つのインバータを利用する。ここで説明しているように挙動する信号（＃１）．．（＃６）を発生するその他の形態の論理回路５３１を含むことは本発明の範囲内である。出力信号（＃１）、（＃２）は、１つのパラメータ（Ｎ_ＴまたはＮ_Ｐ）を測定するのであって、他のパラメータ（Ｎ_ＰまたはＮ_Ｔ）を測定するのではない時であることを一般に意味することがわかる。それらの信号は、「ｃａｐｔｕｒｅ．．．」信号の設定とは無関係に、反対のパラメータを測定するのではない時には常に真（論理１）でもあることに注目されたい。このように、信号ＣＦＸ　ＴＯ　ＯＳＣ（＃１）はＣＡＰＴＵＲＥ　ＮＰが偽である時は常に真であり、信号ＣＰ　ＴＯ　ＯＳＣ（＃２）はＣＡＰＴＵＲＥ　ＮＴが偽である時は常に真である。（＃１）、（＃２）信号はスイッチＳＴ２とＳＰ２をそれぞれ制御するので、この論理は、逆のコンデンサＣＰ／ＣＦＸがセット‐リセット回路５１４に接続されていない時には、スイッチＳＴ２／ＳＰ２にコンデンサＣＦＸ／ＣＰをセット‐リセット回路５１４に常に接続させる。これはセット‐リセット回路５１４を定められた状態に始終留める助けをする。
【０１８４】
各出力信号（＃３）ないし（＃６）は、真である時に、発振器がどの位相（位相１または位相２）にあるか、およびどのコンデンサ（温度のためのＣ_ＦＸ、または圧力のためのＣ_Ｐ）、もしあれば、が測定のために使用中であるかを主として決定する信号の状態を一般に示す。図５Ａに示されているＮＡＮＤゲート論理の例でわかるように、各出力信号（＃３）ないし（＃６）は、反対の測定が選択された時、または測定が選択されないとしても、位相とは無関係に、常にまた真であることがわかる。たとえば、論理回路５３１ｃ中のＮＡＮＤゲートは信号ＰＨＡＳＥ１　ＮＴ（＃３）を出力する。温度が測定されているとすると、ＰＨＡＳＥ１　ＮＴは位相１（Ｃ_ＦＸ充電）中は高く、位相２（Ｃ_ＦＸ放電）中は低い。しかし、温度が測定されていないと（すなわち、ＣＡＰＴＵＲＥ　ＮＴが低、真でない）、発振器の位相とは無関係に、温度や圧力が測定されていないとしても、ＰＨＡＳＥ１　ＮＴは高い。同様な論理が、丸で囲まれている数＃３、＃４、＃５、および＃６により示され、かつ図５Ａに論理回路５３１ｃないし５３１ｆとして示されている４つの制御信号（＃３）ないし（＃６）のおのおのに適用される。
【０１８５】
線５２１（４５３と同じ、３２１と２５３を参照）、における
【０１８６】
【外７】

【０１８７】
信号は、温度カウントと圧力カウントを対応するレジスタ４３２と４３４にそれぞれ累積するためにトランスポンダ４００により使用される、（反転された）弛張発振器出力信号Ｆｏｓｃ′である。弛張発振器出力信号Ｆｏｓｃ′はバッファされてレベルをデジタル論理信号レベルに移行させられることがわかる。弛張発振器５１２は、トランスポンダ４００を（能動モード）試験状態、「ＲＥＡＤ　ＴＥＭＰ」と「ＲＥＡＤ　ＰＲＥＳＳ」、のいずれかに置くことにより試験でき、または直接読取ることができる。それらの試験状態により弛張発振器５１２を温度および圧力のためにそれぞれ可能にし、弛張発振器出力信号Ｆｏｓｃ′をＤＡＴＡパッドへ送る。
【０１８８】
弛張発振器５１２のフロントエンドは２つのほぼ同じである測定スイッチング回路５１５ａと５１５ｂで構成されている。それらの回路は温度測定コンデンサＣ_ＦＸまたは圧力測定コンデンサＣ_Ｐと、適切なフロントエンドスイッチング回路５１５をセット‐リセット回路５１４に接続する温度測定直列スイッチＳＴ２または圧力測定直列スイッチＳＰ２を利用している。スイッチＳＴ１とＳＰ１について上で説明したように、スイッチＳＴ２とＳＰ２はＮチャネルトランジスタを用いて実現できる、またはＣＭＯＳで実現できるスイッチなどの標準（非反転）半導体スイッチである。チップ４０２におけるコンデンサはポリ‐ポリ・コンデンサであって、非常に低い温度係数と、高い分解度を与える面積（容量）を有するが、エッチング誤差や位置合わせ誤差などの処理要因に対する感度が低い。温度測定コンデンサＣ_ＦＸは、たとえば、６ｐｆ（＋／−１０％）のポリ‐ポリ・コンデンサである。圧力測定コンデンサＣ_Ｐ４１８（２１８参照）はオフチップであって、上記したものと同様な頑丈で安定な特性を持つことが好ましい（たとえば、直線容量対絶対圧力応答を持ち、４〜４０ｐｆの間で変化する接触モード容量性圧力センサ）。
【０１８９】
温度測定スイッチング回路５１５ａは、たとえば、ＰチャネルトランジスタＰ９、Ｐ１０、Ｐ１３と、ＮチャネルトランジスタＮ９、Ｎ１０、Ｎ１３、Ｎ１４と、インバータ５１７ａと、半導体スイッチＳＴ２（ＮチャネルＣＭＯＳトランジスタなど）と、温度測定コンデンサＣ_ＦＸとで構成されている。圧力測定スイッチング回路５１５ｂは、たとえば、ＰチャネルトランジスタＰ１１、Ｐ１２、Ｐ１４と、ＮチャネルトランジスタＮ１１、Ｎ１２、Ｎ１５、Ｎ１６と、インバータ５１７ｂと、半導体スイッチＳＰ２と、圧力測定コンデンサＣ_Ｐ（４１８、トランスポンダチップ４０２の外部で、接続パッドＣｐを介して接続されている）とで構成されている。上記した種々の制御入力が丸で囲まれている数字１により示されているように測定スイッチング回路５１５に接続されかつ電流Ｉｂｉａｓも図示のように入力される。
【０１９０】
以下、含まれている技術を示すために、温度測定スイッチング回路５１５ａについて説明する。圧力測定スイッチング回路５１５ｂは適切な並列のようにして機能する。
【０１９１】
トランスポンダ４００が弛張発振器５１２を温度測定モードに置くと、行デコーダ＆Ｎ_Ｔ、Ｎ_Ｐ制御回路４４２は、温度測定ウィンドウＷ_Ｔとしてそれが選択する時間ｔ_Ｔの間ＣＡＰＴＵＲＥ　ＮＴを高くセットし（およびＣＡＰＴＵＲＥ　ＮＰを低くセットする）、それにより論理回路５３１ａと５３１ｂを介してスイッチＳＴ２を閉じ（およびスイッチＳＰ２を開放状態に保つ）。セット‐リセット回路５１４は動作を位相１（すなわち、φ１、５１４の出力が高く、φ２、位相２、５１４ｂの出力が低い）でスタートする。その理由は弛張発振器が測定のために使用されていない時には測定スイッチング回路５１５が常に低い電圧にされ（下で説明するように）、この低い電圧は、上記のように信号＃１、＃２のための論理によって比較器５１６ａと５１６ｂに供給されるからである。信号ＰＨＡＳＥ１＝真かつＰＨＡＳＥ２＝偽であるので、論理回路５３１は次の信号、丸で囲まれている数字により特定されている、を出力する、すなわち、ＰＨＡＳＥ１　ＮＴ（＃３）、ＰＨＡＳＥ１　ＮＰ（＃５）、ＰＨＡＳＥ１　ＮＰ（＃６）＝真、ＰＨＡＳＥ１　ＮＴ（＃４）＝偽。それらの信号はトランジスタＰ１３をオフにし、Ｐ９をオンにすることにより、スケールされた電流Ｉ（Ｔ）ＢをＰ９を通じてコンデンサＣ_ＦＸを充電する。スイッチＳＴ２は信号（＃１）により閉じられているので、セット‐リセット回路５１４はコンデンサＣＦＸの立ち上がる電圧を検出する。同時に、トランジスタＮ１４がオンに保たれて、トランジスタＮ９のゲートを接地し、充電されているコンデンサＣ_ＦＸの電流漏れ経路が存在しないようにそれをオフに保つ。トランジスタＰ１０とＮ１３もオフに保たれているので、トランジスタＮ９への他の入力はなく、トランジスタＮ１０の状態は問題ではない。
【０１９２】
セット‐リセット回路５１４が位相２（ＰＨＡＳＥ１＝偽、ＰＨＡＳＥ２＝真）へフリップフロップすると、信号ＰＨＡＳＥ１　ＮＴ（＃３）は偽へ変化し、ＰＨＡＳＥ２　ＮＴ（＃４）は真（ただしＰＨＡＳＥ１　ＮＰ（＃５）とＰＨＡＳＥ２　ＮＰ（＃６）は「真」に留まる）へ変化する。それらの信号はトランジスタＰ１３をオンにし、Ｐ９をオフにすることにより、スケールされた電流Ｉ（Ｔ）ＢをトランジスタＮ１０へ送る。トランジスタＮ１３がオンにされ、トランジスタＮ１４がオフにされているので、トランジスタ対Ｎ１０／Ｎ９は今は電流ミラーを形成し、したがって、トランジスタＮ１０を通ってアースへ流れる電流は、トランジスタＮ９を通じてアースへ流れてコンデンサＣ_ＦＸを放電する等しい電流によりミラーされる。いまはターンオンされている制御トランジスタＰ１０を通じて電流Ｉｂｉａｓを接続することの意図されなかった副作用は、トランジスタＮ１０／Ｎ９を流れる電流がスケールされた電流Ｉ（Ｔ）Ｂとバイアス電流Ｉｂｉａｓの和であり、それによりコンデンサＣ_ＦＸの放電速度を充電速度と比較してわずかに高くする。バイアス電流Ｉｂｉａｓは比較的小さいので、これは弛張発振器５１２のデューティサイクルを５０％−５０％から僅かにずれるように変化するが、カウンティングプロセスには影響を及ぼさない。（意図的に対称的なデューティサイクルで動作させられる、上記の、従来のモデルのトランスポンダ４００と比較されたい。）スイッチＳＴ２は信号（＃１）により依然として閉じられているので、セット‐リセット回路５１４はコンデンサＣ_ＦＸの降下する電圧を検出できる。
【０１９３】
温度測定ウィンドウＷ_Ｔが閉じられていると（トランスポンダ４００はもはや温度測定モードにない）、行デコーダ＆Ｎ_Ｔ、Ｎ_Ｐ制御回路４４２に温度レジスタ４３２内のデータ獲得を停止させる。この点では、ＣＡＰＴＵＲＥ　ＮＴ信号が依然として低いので、スイッチＳＴ２は閉じられたままである（信号ＣＦＸ　ＴＯ　ＯＳＣ）。ＣＡＰＴＵＲＥ　ＮＰ信号が高くセットされた（およびＣＡＰＴＵＲＥ　ＮＴ信号が低い）とするとスイッチＳＴ２は他の時に開くだけである、すなわち、圧力が測定され、温度は測定されない。ＣＡＰＴＵＲＥ　ＮＴ信号が低いと、論理回路５３１は信号ＰＨＡＳＥ１　ＮＴ（＃３）とＰＨＡＳＥ２　ＮＴ（＃４）を真にする。それらの信号はトランジスタＰ１３とＰ９をオフにすることにより、温度測定スイッチング回路５１５ａを流れるスケールされた電流Ｉ（Ｔ）Ｂの流れを遮断する。しかし、トランジスタＮ１３がオンにされ、トランジスタＮ１４がオフにされているので、トランジスタ対Ｎ１０／Ｎ９は電流ミラーを依然として形成し、バイアス制御トランジスタＰ１０がオンであるのでトランジスタＮ１０をアースまで流れるバイアス電流Ｉｂｉａｓは、トランジスタＮ９を通じてアースへ流れてコンデンサＣ_ＦＸを放電する等しい電流によりミラーされる。（小さい）バイアス電流Ｉｂｉａｓはミラー回路Ｎ１０／Ｎ９を流れることを許されている唯一の電流であるので、コンデンサはアース（０ボルト）まで徐々に放電させられ、ＣＡＰＴＵＲＥ　ＮＴ制御信号を高く（真）にすることにより次の温度測定ウィンドウＷ_Ｔが開かれるまでそこに保持される。スイッチＳＴ２は依然として閉じられているので、比較器５１６ａと５１６ｂへの接地された入力はセット‐リセット回路５１４に、ＰＨＡＳＥ１＝高い、ＰＨＡＳＥ２＝低いという定められた状態で発振を停止させる。温度カウントＮ_Ｔまたは圧力カウントＮ_Ｐを累積するために測定コンデンサＣ_ＦＸとＣ_Ｐの各使用の前に、測定コンデンサＣ_ＦＸとＣ_Ｐを定められた状態（零電圧）にリセットするために小さいバイアス電流Ｉｂｉａｓを利用するここで開示している手段を設けることが本発明の特徴である。本発明の別の特徴は、セット‐リセット回路５１４の各使用前にセット‐リセット回路５１４を定められている状態に更に置くためにここで説明している手段を使用することである。
【０１９４】
上記の説明例は、トランスポンダ４００回路の種々の部分において、たとえば、電流変換器４５０（電流スケーリング回路５１０）へのベース‐エミッタ電圧において、弛張発振器４５２（５１２）において、およびデータ獲得回路４５４（図３Ａに示されている回路２５４とほぼ同じ）において、種々の目的のために制御信号ＣＡＰＴＵＲＥ　ＮＴおよびＣＡＰＴＵＲＥ　ＮＰを利用する。適切な順序で事象が起きるように、それらの回路４５０、４５２、および４５４のおのおのへの制御信号のタイミングを変えることは本発明の範囲内である。たとえば、回路４５４は、弛張発振器５１２の時間が安定な動作することができるようにして、測定コンデンサＣ_ＦＸとＣ_Ｐにおける電圧Ｖｂｇ／２とＶｂｇの間で発振するように、ターンオンを遅延させることができる（たとえば、２５６μｓｅｃの遅延１ビット幅時間）。
【０１９５】
トランスポンダ４００の典型的な受動モード動作について上で注意したように、行デコーダ＆Ｎ_Ｔ、Ｎ_Ｐ制御回路４４２は制御信号をセンサインタフェース回路４０６へ（線４８７を介して）供給して、データ送信の１つの部分（たとえば、語／行２ないし６を送信している間に）中に温度に関連するカウントを温度レジスタ４３２に累積し、データ送信の別の部分（たとえば、語／行９ないし１３を送信している間に）中に圧力に関連するカウントを圧力レジスタ４３４に累積する。データ獲得回路４５４について遅延されたターンオンを実現するために、行デコーダ＆Ｎ_Ｔ、Ｎ_Ｐ制御回路４４２は、たとえば、行２のビット１へクロッキングした時にＣＡＰＴＵＲＥ　ＮＴ信号をプログラム可能な電流スケーリング回路５１０と、弛張発振器回路５１２へ送ることができ、その後で、行２のビット２へクロッキングした時に異なる「Ｃａｐｔｕｒｅ　Ｔｅｍｐ」信号をデータ獲得回路４５４へ送ることができる。同様にして、ＣＡＰＴＵＲＥ　ＮＴ信号と「Ｃａｐｔｕｒｅ　Ｔｅｍｐ」信号を異なる時刻に、あるいは一緒にも（たとえば、行７のビット１へクロッキングした時に）送ることができる。
【０１９６】
図面および以上の説明で本発明を示し、記述したが、それは例示であって特徴を限定するものではないと考えるべきであり、好適な実施形態のみを示して説明したものであることを理解すべきであり、全ての改変は本発明の要旨内に入り、保護されることが望まれる。疑いもなく、上記「主題」に対する他の多くの「変化」は本発明が最も近く属している技術の当業者には着想されるものであり、それらの変化はここに開示している本発明の範囲内に含まれるべきであることを意図している。
【図面の簡単な説明】
【図１】
外部読取器／質問器および空気入りタイヤ内のＲＦトランスポンダを備えている本発明のＲＦトランスポンダ装置の一般化した図である。
【図２】
本発明の以前のモデルのＲＦトランスポンダの主要部品のブロック図である。
【図３】
本発明の以前のモデルの図２のＲＦトランスポンダの主要部の回路図である。
【図３Ａ】
本発明の以前のモデルの図２のＲＦトランスポンダの一部の回路図である。
【図３Ｂ】
本発明の以前のモデルの図２のＲＦトランスポンダの一部の回路図である。
【図３Ｃ】
本発明の以前のモデルに従って、データをどのように配列し、送信するかを示す図２のＲＦトランスポンダのメモリ空間内の図である。
【図３Ｄ】
本発明の以前のモデルの図２のＲＦトランスポンダのトランスポンダ指示値とトランスポンダ電力のグラフである。
【図４Ａ】
本発明のＲＦトランスポンダの主な部品のブロック図である。
【図４Ｂ】
本発明の以前のモデルに従って、データをどのように配列し、送信するかを示す図４ＡのＲＦトランスポンダのメモリ空間内の図である。
【図５】
本発明の、図４ＡのＲＦトランスポンダの電流スケーリング部と弛張発振器部の回路図である。
【図５Ａ】
本発明の、図５の弛張発振器部の論理部の回路図である。[0001]
Cross reference with related application
This application is a PCT application entitled RELAXATION OSCILLATOR FOR FOR TRANSPONDER, filed on the same date as the filing date of that application. PCT Application No. DN199999116 PCT and a programmable modulation index for the transponder (PROGRAMMABLE \ MODULATION \ INDEX \ FORTRANSPONDER); No. DN199999118 PCT and a PCT application entitled PROGRAMMABLE \ TRIMING \ FOR \ TRANSPONDER with programmable trimming for transponders, Attorney Case No. DN 19999256 PCT.
[0002]
This application is pending and is hereby incorporated by reference and filed on May 17, 1999 by Yones. 60 / 134,398 is a continuation-in-part application.
[0003]
(Technical field)
The present invention relates to pressure and temperature monitoring, and more particularly to tires for sending pressure and temperature measurements to an external (vehicle) receiver (reader or reader / interrogator). For the combined transponder.
[0004]
(Background technology)
The safe, efficient and economical operation of a vehicle relies heavily on maintaining all (each) tires of the vehicle at the correct pressure. Driving a car at low air pressures can result in excessive tire wear, difficult steering, unsteady running, and increased fuel consumption. All of these difficulties are magnified when the tire pressure goes to zero if the tire gets punctured.
[0005]
The need to monitor tire pressure while the tires are in use is emphasized in connection with "run-flat" tires, which are tires that can be used completely deflated. Such run flat tires are disclosed, for example, in U.S. Pat. No. 5,368,082, owned by the assignee of the present invention, and all of which are incorporated herein by reference. Incorporating reinforced sidewalls, a mechanism to secure the tire bead to the rim, and non-pneumatic tires in pneumatic tires that allow the driver to maintain control of the vehicle after a catastrophic pressure drop Yes, and they have progressed to a point where the driver becomes increasingly unaware that the tire has collapsed. The broad objective behind using runflat tires was to puncture a limited distance (eg, 80 kilometers) before repairing the tires, rather than stopping at the shoulder to repair the punctured tires The purpose is to allow a driver of a car to continue running with pneumatic tires. Therefore, it is desirable to provide a low air pressure alarm device in the vehicle that warns the driver that the air pressure of the pneumatic tire is leaking (for example, with an indicator light or a buzzer).
[0006]
To this end, some electronic devices monitor the pressure of the pneumatic tire and provide an indication of the current tire pressure to the driver of the vehicle or warn the driver when the pressure drops below a predetermined threshold level. And electronic systems are known.
[0007]
For example, U.S. Pat. No. 4,578,992 (Galasko et al., April 1986) discloses a coil forming a passive oscillation circuit having a natural resonance frequency that changes with tire pressure, which changes the capacitance value of a pressure-sensitive capacitor; A tire pressure indicating device including the pressure-sensitive capacitor is disclosed. The circuit is excited by pulses supplied by a coil located outside the tire and fixed to the vehicle, and the natural frequency of the passive oscillation circuit is detected. The natural frequency of the coil / capacitor indicates the pressure on the pressure sensitive capacitor.
[0008]
There are also known electronic devices that can transmit radio frequency (RF) signals indicative of tire pressure to remotely located receivers rather than merely passive resonant circuits. Such a "transmitting device" can have its own power supply, or can only be activated when the pressure drops below a predetermined threshold. Alternatively, the transmitting device can be activated ("turned on") from a remotely located receiver. In this case, the receiver is considered an "interrogator". Further, the transmitting device can receive power by the RF signal from the interrogator. In addition, the tire pressure monitoring electronics may be capable of receiving information from the interrogator. In this case, the electronic device is called a "transponder".
[0009]
As used herein, a “transponder” transmits and receives radio frequency signals and a single state (eg, tire pressure) or multiple states (eg, tire pressure, temperature, rpm) at which the transmitted signal is measured. ), Variable information (data) can be sent in an appropriate format and, depending on the selection, a response to information that may be present in the received signal can be made. The most interesting and typical condition for pneumatic tires is tire pressure. A "passive" transponder is a transponder that receives power from the energy of the signal received from the interrogator. An "active" transponder has its own power supply (e.g., a battery) and uses minimal power until it is "woken up" by a signal from an interrogator, or by an internal periodic timer, or by an attached device. And a transponder including an active transponder in a "sleep mode". As used herein, the term "tag" refers to a transponder that has transmit and receive capabilities, or a transponder that has only transmit capabilities. Generally, tags that are transponders are suitable in the system of the present invention. As used herein, a "tire pressure monitoring system" (TPMS) is an entire system that includes a tag in a tire and a receiver, which can be an interrogator located in the vehicle. Is shown.
[0010]
A tag and its associated condition sensor (eg, a pressure sensor) are mounted in each tire of the vehicle and a single interrogator (or receiver) collects information from each of those transponders; It is known to alert a vehicle driver of a low tire pressure condition that requires correction (eg, tire replacement). For example, U.S. Pat. No. 5,540,092 (Hndfield et al., 1996), which is incorporated herein by reference in its entirety, discloses an apparatus and method for monitoring pneumatic tires. FIG. 1 of that patent discloses a pneumatic tire monitoring device (20) comprising a transponder (22) and a receiver (24).
[0011]
Examples of RF transponders suitable for mounting in pneumatic tires are U.S. Pat. No. 5,451,959 (Schuermann, May 1995) and U.S. Pat. No. 5,661,651 (Geschke et al., 1997). August 1996) and U.S. Patent No. 5,581,023 (Handfield et al., February 1996). The entirety of those U.S. patents are hereby incorporated by reference. The transponder systems described in those U.S. patents disclose an interrogation unit associated with the transponder, at least one of a pressure sensor and a temperature sensor, and various types of tire / transponder identification among multiple transponder systems. Including technology. In most cases, such transponders require battery power.
[0012]
In some cases, the transponder can be implemented as an integrated circuit (IC) chip. Typically, IC chips and other components are mounted, connected, mounted and connected to a substrate such as a printed circuit board (PCB).
[0013]
Some proposed devices have relatively complex transponder sensor capabilities, including measurement and reporting of tire rotation and speed, tire ID, tire temperature, and tire pressure. For example, U.S. Pat. No. 5,562,787 (Koch et al., 1996) and U.S. Pat. No. 5,731,754 (Lee, Jr. et al., 1998), which are hereby incorporated by reference in their entirety. Included here.
Environmental considerations for transponders
The environment in which a transponder mounted on a tire must operate reliably presents a number of problems for successful operation of the transducer, including during manufacture and use of the tire. For example, the temperature range of the sensors (eg, for pressure and temperature) used in the transponder is preferably 125 ° C and must be able to withstand a manufacturing temperature of about 177 ° C. For truck tire applications, the operating range of the pressure sensor must be from about 345 kPa to about 827 kPa (from about 50 psi to about 120 psi) and can withstand a pressure of about 2759 kPa (about 400 psi) during tire manufacture. Must be able to do it. Its accuracy, including the sum of all contributors to its inaccuracy, must be on the order of plus or minus 3% of full scale. The repeatability and stability of the pressure signal must be within the specified accuracy range.
[0014]
However, it must be fulfilled, so that the tire transponder (tag) must be able to operate reliably despite a wide range of pressures and temperatures. Also, the tire transponder must be able to withstand the large mechanical shocks encountered when the vehicle is traveling on road speed bumps or depressions.
[0015]
A device that can be used to indicate whether a transponder or tire has been exposed to too high a temperature that could cause damage is a "MTMS" element or high temperature memory switch developed by Professor Mehran Mehregany of Case Western Reserve University. is there. This is a micromachined element that switches to a closed state at some high temperature point. The sensor switches from an "open" high resistance state, for example, greater than 1 megohm, to a "closed" low resistance state, for example, less than 100 ohms. Although it is generally known to use pressure transducers in pneumatic tires in combination with electronics to send pressure data, those pressure data systems for tires have suffered from the difficulties inherent in the tire environment. Was. Such difficulties include the effective and reliable coupling of RF signals to and from the tire, the rigorous use of tires and electronics, and the inclusion of pressure transducers and electronics in the tire / wheel system. This includes adverse effects on the tire. Another problem in the context of a "passive" RF transponder powered by an external reader / interrogator is a predictable and stable voltage level within the transponder so that the circuit within the transponder can execute its design specifications. May occur.
[0016]
Pressure transducers suitable for use on transponders mounted on tires include:
(A) a piezoelectric transducer,
(B) piezoresistive elements such as those disclosed in U.S. Pat. No. 3,893,228 (George et al., 1975) and U.S. Pat. No. 4,317,216 (Gragg, Jr., 1982);
(C) U.S. Patent No. 4,701,826 (Mikor, 1987); U.S. Patent No. 5,528,452 (Ko, 1996); U.S. Patent No. 5,706,565 (Sparks et al., 1998). ) And PCT / US99 / 16140 (Ko et al., Filed July 7, 1999), silicon capacitive pressure transducers,
(D) an element formed of a variable conductive laminate having conductance of ink,
(E) an element formed of a variable conductive elastic composition;
Is included.
Effect of temperature on gas pressure
In a broad sense, the pressure P, temperature T, and volume V can be easily measured for any gas mass in thermal equilibrium. For sufficiently low density values, (1) for a given mass of gas kept at a constant temperature, pressure is inversely proportional to volume (Boil's law), and (2) For a given mass of retained gas, volume is directly proportional to temperature (Charles and Gay-Lussac's law). This leads to the “state equation” of the ideal gas, that is, the “law of the ideal gas”.
[0017]
PV = μRT
here,
μ is the mass expressed in moles of gas,
R is a constant related to gas,
It is.
[0018]
Thus, for a trapped (fixed) volume of gas, such as air trapped in a pneumatic tire, a temperature (T) rise indicates a pressure (P) rise.
[0019]
Due to the ideal gas law relationship, in the context of pneumatic tires, one problem that arises during the operation of any kind of pressure sensor is that the tire heats up over long runs. As the tire heats up, the air trapped in the substantially constant closed volume of the tire expands, increasing the pressure in the tire, but the total amount of air in the tire remains the same. Since the nominal pressure is different, the tire pressure sensor will give a different instruction when the tire temperature is high than when it is low. This is why tire manufacturers and vehicle manufacturers recommend that vehicle owners check tire pressure when the tires are cold. Of course, a remote tire pressure sensor allows the driver to receive a continuous indication of tire pressure in the vehicle, but the indication may be incorrect due to temperature changes. Therefore, there is a need to compensate for temperature changes of the expanding medium ("gas" or air) in the pneumatic tire.
[0020]
U.S. patents that deal in some way with the effects of the gas law in pneumatic tires include:
[0021]
U.S. Patent No. 3,596,509 (Raffelli, 1971); U.S. Patent No. 4,335,283 (Migrin, 1982); U.S. Patent No. 4,126,772 (Pappas et al., 1978); U.S. Pat. No. 4,909,074 (Gerreheim et al., 1990); U.S. Pat. No. 5,050,110 (RotT, 1991); U.S. Pat. No. 5,230,243 (Reinecke, 1993); U.S. Pat. No. 4,966,034 (Bock et al., 1990); U.S. Pat. No. 5,140,851 (HetTrich et al., 1992); U.S. Pat. No. 4,567,459 (Folger et al., 1986). All of those US patents are hereby incorporated by reference in their entirety.
[0022]
U.S. Pat. No. 4,893,110 (Hebert, 1990), which is hereby incorporated by reference in its entirety, discloses a tire monitoring device that uses pressure and temperature measurements to detect abnormalities. I have. As stated therein, the ratio of temperature to pressure gives a first approximation of the number of moles of gas in the tire. It must be kept constant to prevent inflation fluid from leaking from the tire. (Column 1, lines 18-26). More specifically, U.S. Pat. No. 4,703,650 discloses a tire pressure sensor (4) and a tire temperature sensor (6) for each wheel, and the measured values as encoded signals. Elements (8 and 10) for sending to an on-board computer (12), such as an object, are incorporated. The computer processes the pressure and temperature measurements for each tire and estimates the calculated pressure / temperature ratio (P / T estimate) for each wheel. Generally, the ratio for one tire is compared with the ratio for at least one other tire, and an alarm is output when the comparison result (N) deviates from a predetermined value range.
Technology for sending pressure and temperature indications from tires
Given that both pressure and temperature conditions in a pneumatic tire can be measured, techniques have been developed to send signals indicative of the measured pressure and temperature conditions to an external interrogator / receiver. For example, reference is made to the following U.S. Patents, which are incorporated herein in their entirety:
[0023]
Transmit signals individually identified by phase shift: US Patent No. 4,174,515 (Marzolf, 1979);
Multiplexing signals: US Pat. No. 5,285,189 (Nowicki et al., 1994); US Pat. No. 5,297,424 (SacketT, 1994);
Encoding a signal as separate segments of a data word: US Pat. No. 5,231,872 (Bowler et al., 1993); US Pat. No. 4,695,823 (Vernon, 1987); At least one of the remote meter, the pressure sensor, and the temperature sensor is also included on the same integrated circuit chip;
Transmission between wheels and coils mounted on automobiles: U.S. Pat. No. 4,567,459 (Folger et al., 1986);
Using Frequency Displacement Key (FSK) Signals: US Pat. No. 5,228,337 (Sharpe et al., 1993);
U.S. Pat. No. 5,731,754 (Lee, Jr. et al., US Pat. No. 5,731,754) backscatter modulates an RF signal from an interrogator with state parameter data from a sensor and then returns the backscattered modulated signal to the interrogator. 1998)
U.S. Pat. No. 4,703,650 (Dosjoub et al., 1987), which is hereby incorporated by reference in its entirety, but which encodes the values of two variables measured in the tire; An apparatus for monitoring a tire using the circuit is disclosed. The encoding circuit includes an astable multivibrator that converts measurements of variables, eg, pressure and temperature, to time measurements. The astable multivibrator produces a pulse signal having a pulse width that is a function of temperature and a period ratio that is a function of pressure.
[0024]
U.S. Patent No. 5,054,315 (Dosjoub et al., 1991), which is hereby incorporated by reference in its entirety, describes a technique for encoding several quantity values measured in a tire. Has been disclosed. As disclosed there,
"The encoding of the value of any number of quantities measured in the tire, for example its pressure and temperature, is performed using the ratios of the time intervals TP / Tr, TT / Tr. This modulates the device. It should not be affected by the time lapse of the device, which simultaneously affects the numerator and denominator of the ratio. "
(Summary of the Invention)
In accordance with the present invention, a radio frequency (RF) transponder has circuitry that can transmit information unique to what ties the transponder to an external reader / interrogator. Also, one or more transponder sensors (transducers) perform real-time parameter measurements at the location of the transducer. The measurements are in the form of data in the data stream, superimposed on the signal output by the transponder, such as adding (modulating) the data stream to the RF signal sent by the transponder to an external reader / interrogator. Sent to reader / interrogator.
[0025]
According to the invention, usually two real-time parameters, pressure and temperature, are measured. The pressure is preferably measured by an independent ("off-chip") pressure sensor. The pressure sensor is of a type that changes its capacitance value in a known manner, such as a polynomial, or preferably as a substantially linear function of the ambient pressure. Preferably, the temperature sensor is embedded within the IC chip of the transponder ("on-chip") and is exposed to about the same ambient temperature as the pressure sensor receives so that the true temperature compensated pressure can be easily calculated. Be placed.
[0026]
According to an aspect of the invention, a transponder includes a measurement switching circuit, a set-reset circuit having a first phase path and a second phase path, and a measurement current source that supplies a measurement current to a relaxation oscillator having an output. A logic circuit, a measurement capacitor disposed in the measurement switching circuit, a first transistor controlled by the logic circuit to supply a measurement current to the measurement capacitor for charging the measurement capacitor, and a transistor controlled by the logic circuit And a current mirror for copying the measurement current in the second transistor and charging the measurement capacitor, wherein the logic circuit combines a signal indicative of the phase of the set-reset circuit with a signal from the timing generator. Control the alternating charging and discharging of the measuring capacitor according to the phase of the set-reset circuit. It is an.
[0027]
In accordance with a feature of this aspect of the invention, the measured current indicates the measured parameter, the measurement capacitor has a fixed capacitance value, and thus the relaxation oscillator output is a signal having a frequency indicative of the measured parameter. Alternatively, the measurement capacitor has a varying capacitance value indicating the measured parameter (eg, ambient pressure), and thus the relaxation oscillator output is a signal having a frequency indicating the measured parameter. Alternatively, the measured current indicates a first measured parameter, and the measuring capacitor has a varying capacitance value indicating a second measured parameter, so that the relaxation oscillator output is different from the first measured parameter and the second measured parameter. Is a signal having a frequency that is indicative of the measured parameter. A programmable current scaling circuit may be inserted between the measurement current source and the relaxation oscillator such that the measurement current is a scaled current, the current scaling circuit being controlled by setting at least one scaling trim bit. A switch is provided, each switch being connected in series with at least one scaling transistor to vary the scaled measured current according to a programmable trim setting. The relaxation oscillator compares the voltage on the measurement capacitor with a first fixed reference voltage and a second fixed reference voltage that is lower than the first fixed reference voltage. The relaxation oscillator comprises two comparators and is therefore characterized in that it oscillates between two fixed reference voltages. A bias control transistor can be added to be controlled by the logic circuit to connect a small bias current to the current mirror and discharge the measurement capacitor.
[0028]
According to an aspect of the invention, the relaxation oscillator may comprise a first measurement switching circuit and a second measurement switching circuit, each measurement switching circuit supplying a measurement capacitor and which measurement switching circuit supplies a voltage to the set-reset circuit. And a switch for controlling the operation. When used for measurement, each measurement switching circuit functions similarly to the individual measurement switching circuits described above. Typically, the measured current indicates the measured parameter (e.g., temperature) and the first measuring capacitor has a fixed capacitance value, so that the relaxation oscillator output is switched on when the first switch is closed and the second A signal having a frequency indicative of a first measured parameter when the switch is open, wherein a second measuring capacitor indicates a varying capacitance value indicative of a second measured parameter (eg, pressure); The relaxation oscillator output has a signal having a frequency indicative of the first measured parameter and the second measured parameter when the first switch is open and the second switch is closed. It is. As described above, a programmable current scaling circuit is provided for the measurement current source and the relaxation oscillator such that the measurement current used by the first and second measurement circuits in the relaxation oscillator is a scaled current. Can be inserted in between. Further, the programmable current scaling circuit switch can be controlled to provide different scaling factors for the scaled measurement current used by the two measurement switching circuits.
[0029]
According to an aspect of the invention, a method for generating an oscillation measurement signal on an RF transponder, such as the RF transponder described above, wherein the frequency of the oscillation measurement signal indicates a measured parameter is described. The method includes supplying a measurement current to charge a measurement capacitor in each measurement switching circuit of the relaxation oscillator, and then supplying a measurement current copied to the measurement capacitor to discharge the measurement capacitor. And alternately performing. A programmable scaled measurement current can be provided to the measurement switching circuit to optimize the measurement signal frequency to produce an optimized indication for each measured parameter. By grounding each measurement capacitor when each measurement capacitor is not used for measurement, the measurement capacitors can be placed in a defined state, with the input terminal grounded when the set-reset circuit is not used for measurement. The set-reset circuit can likewise be placed in a defined state. Finally, by adding a delay between when the relaxation oscillator is started and when the measurement signal is gated into the measurement counter / register, the relaxation oscillator can be stabilized before the indication of the measured parameter begins.
[0030]
Other objects, other aspects, and other advantages of the present invention will become apparent from the following description.
[0031]
(Detailed description of the invention)
Reference will now be made in detail to the preferred embodiments of the invention. Examples of these embodiments are shown in the accompanying drawings. The drawings are intended to be illustrative, not limiting. While the invention will be described in the context of those embodiments, it should be understood that it is not intended to limit the spirit and scope of the invention to those particular embodiments.
[0032]
Certain elements in the selected drawings may not be drawn to scale for clarity.
[0033]
Similar elements are often provided with similar reference numbers throughout the drawings. For example, element 199 in one figure (or embodiment) may be similar in many aspects to element 299 in another figure (or embodiment). Such relationships between similar elements in the various figures or various embodiments, if applicable, will be apparent throughout the specification, including the claims and the abstract. There will be.
[0034]
In some cases, similar elements can be referred to by similar numbers in a single drawing. For example, a plurality of elements 199 can be referenced by 199a, 199b, 199c, and the like.
[0035]
For simplicity of illustration, the cross-section shown here is in the form of a "round slice", if any, which is visible in the true cross-section, or a "myopia" cross-section. May be presented.
[0036]
The structure, operation and advantages of this preferred embodiment of the present invention will become more apparent from a consideration of the following description when taken in conjunction with the accompanying drawings.
[0037]
As described above, it is an aspect of the present invention to provide a device that monitors tire pressure of a vehicle and alerts a driver when a low tire inflation pressure condition occurs.
[0038]
FIG. 1 shows a prior art RF transponder device 100 that includes an RF (radio frequency) transponder 102 disposed within (eg, mounted on an inner surface of) a pneumatic tire 104. (An antenna (not shown) is mounted in the tire 104 and connected to the transponder 102.) The transponder 102 has unique identification (ID) information (eg, its own serial number, or a combination thereof). The identification number of the object, eg, tire 104, and data indicative of parameter measurements, such as ambient pressure, detected by a sensor (not shown) associated with the transponder 102, are provided to an external reader / reader. An electronic device that can be transmitted to the interrogator 106. An external reader / interrogator 106 supplies an RF signal interrogating the transponder 102, an antenna 110, a display panel 112 for displaying information sent to / from the transponder 102, and a user operating the reader / interrogator 106. And an operation device (switch, button, knob, etc.) for operating the function of (1). Although shown as a hand-held device, the reader / interrogator can be an electronic device mounted in a vehicle (not shown). The present invention is primarily directed to implementing an RF transponder.
[0039]
As is known, at least one of the ID and the parameter measurement information can be encoded (added) in various ways on the signal sent by the transponder 102 and then read / read for display to the user. Interrogator 106 "decrypts" (restores). The RF transponder 102 can be “passive” and powered by the RF signal generated by the external reader / interrogator 106 and emitted by the antenna 108. Alternatively, the RF transponder 102 can be "active" and powered by a battery. Transponder devices such as the transponder device 100 described herein are well known.
[0040]
The pending PCT Patent Application No. Ser. PCT / US98 / 07338 discloses that a transponder, in particular its operating power is obtained from an external radio frequency (RF) source, and which is at least one of pressure and / or temperature data, hereby incorporated by reference in its entirety Disclose a "passive" transponder associated with a pneumatic tire for use in identification and transmission.
[0041]
The PCT patent application no. PCT / US98 / 07338 discloses a transponder that is an earlier model of the transponder of the present invention (model number "3070C"), which includes commonality with earlier models and improvements. Relevant parts of the previous model are described below with reference to FIGS. 2, 3, 3A, 3B and 3C.
[0042]
FIG. 2 is a block diagram of a model 3070C @ RF transponder 200 (compare 102) showing the main functional components. While this exemplary device is described as a preferred embodiment for measuring pressure and temperature, it is within the scope of the invention to include the measurement of other parameters, including suitable sensors.
[0043]
Transponder 200 is preferably implemented on a single integrated circuit (IC) chip, shown within dashed line 202. Several external components are connected to the integrated circuit. The other dashed lines in this figure include the main functional “blocks” of transponder 200, including transponder “heart” 204 and sensor interface 206. Components external to the IC chip 202 include an antenna system 210 including an antenna 212, a capacitor 214 connected in parallel to the antenna 212 to form an LC resonance tank circuit, and an external precision resistor (Rext) 216. And an external pressure sensing capacitor (Cp) 218 and an external maximum temperature measurement switch (MTMS) 220. Whether or not to use the maximum temperature measurement switch 220 is optional. Antenna 212 may be in the form of a coil antenna, loop antenna, dipole antenna, or the like. Alternatively, the signal output by the transponder can be provided on a transmission line. For some of these antenna embodiments (eg, a loop antenna), the capacitor 214 can be omitted. The reason is that the tuning of such an antenna does not benefit the capacitor. Most of the following description will refer to a transponder having a coil antenna.
[0044]
The pressure sensing capacitor Cp is preferably a rugged, low temperature coefficient sensor having a capacitance-to-pressure response with high sensitivity and good linearity within the pressure range of interest. One example is an all-silicon "touch mode" capacitive pressure sensor, such as the pressure sensor known in the art and described above.
[0045]
The transponder heart 204 processes an RF signal, such as an unmodulated 125 kHz (kilohertz) carrier signal, received by the antenna 212, rectifies the received RF signal, and supplies power to other circuits in the IC chip 202. An interface circuit 222 is provided. For example, the interface circuit provides a regulated power supply voltage (Vdd) of 2.5 volts and a bandgap voltage (Vbg) of 1.32 volts independent of temperature. The supply of various power supply voltages and reference voltages to the transponder circuit is described in detail below with reference to FIG. 3B. The interface circuit 222 receives the RF signal, preferably at the input frequency (Fi) at which it is received, and generates a clock signal in a known manner to control the timing of other circuits in the IC chip. And the output frequency (Fc) of the signal sent by the transponder 200 to an external reader / interrogator (eg, 106).
[0046]
Timing generator / sequencer circuit 226 receives clock pulses from clock generation circuit 224, processes (eg, divides) those clock pulses, and measures parameters (eg, temperature and pressure) for a predetermined time (eg, temperature and pressure). Each t_TAnd t_P) During the timing window (W_TAnd W_P, Described below). Timing window W_TAnd W_PCan be of approximately equal length or of different lengths. The timing generator / sequencer circuit 226 also controls the timing and sequence of various functions performed at the sensor interface 206 (eg, pressure measurement and capture, temperature measurement and capture, described below). The timing generator / sequencer circuit 226 is preferably implemented as an algorithm state machine (ASM).
[0047]
The transponder heart 204 includes a temperature register 232 (eg, 12 bits) that captures and stores the temperature measurement (count) and pressure measurement (count), a pressure register 234 (eg, 12 bit), and an EEPROM array. And a register / counter circuit 230 including a block 236 of addressable memory (e.g., 120 bits). Registers 232, 234 and EEPROM array 236 are shown within dashed lines 238 representing blocks of addressable memory on IC chip 202.
[0048]
Register / counter circuit 230 also includes a multiplexer and column decoder 240 and a row decoder 242 to control the order in which signals (eg, data) are sent over lines 244 to modulation circuit 246. Modulation circuit 246 sends the selected tire operating characteristic measurements in the data stream to a reader / interrogator (eg, 106) via interface circuit 222 and antenna device 210.
[0049]
The transponder heart 204 also includes a baud rate generator 248 that controls the rate at which modulation information (eg, temperature and pressure measurements) is provided to the modulation circuit 246. The baud rate generator 248 includes a data carrier clock that controls the output frequency Fc of the transponder, and a data rate that controls the rate at which the data stream containing measured values, calibration information, identification codes, etc. modulates the output carrier signal of the transponder 200. Also supplies a clock.
[0050]
Sensor interface 206 provides an output current I (T) / N associated with a predictable characteristic voltage of a temperature sensing component (eg, Vbe of transistor Q1, described below) superimposed on external resistance (Rext) 216 on line 251. Includes circuitry 250 generated above. The output current I (T) / N on line 251 is provided to relaxation oscillator 252. In general terms, relaxation oscillator 252 includes an output current I (T) / N on line 251 and an internal capacitance C associated with relaxation oscillator 252._FX1, C_FX2It oscillates at a frequency controlled by the voltage change rate (dV / dT), which is a function of an external capacitance (Cp) 218 that can be connected to the oscillator circuit by a switch. As will be described in greater detail below, an output signal Fosc 'from the relaxation oscillator 252 is provided on line 253, indicating the ambient temperature and pressure. As used herein, the term "ambient" refers to a parameter measured near the transponder 200, or more specifically, near each sensor associated with the transponder 200. When the transponders 200, 102 are mounted in a pneumatic tire (eg, 104), “ambient pressure” and “ambient temperature” refer to the pressure and temperature of the inflation medium (eg, air) in the tire 104.
[0051]
In operation, an RF signal from an external source (ie, a reader / interrogator, not shown, see 106) is received by antenna 212. This RF signal is rectified and used to supply power to the RF transponder 200. The modulation information applied to the modulation circuit 246 is used to change various characteristics (for example, impedance, resonance frequency, and the like) of the antenna device 210. These changes are detected by the external reader / interrogator 106 and decoded to send temperature and pressure information from the RF transponder 200 to the external reader / interrogator 106.
[0052]
The timing generator / sequencer circuit 226 controls when the generation of the frequency Fosc ′ output by the relaxation oscillator 252 includes the external pressure detection capacitance (Cp) 218 and counts the pressure via the data acquisition circuit 254. It also controls the acquisition of the temperature count. For example, to measure the temperature, the temperature sensing current I (T) is applied to the internal oscillator capacitor (C_FX1And C_FX2), But the pressure sensing volumes (Cp) 218 are disconnected (not included) from those volumes. This means that the frequency Fosc 'of the oscillator output signal seen on line 253 is a function of temperature only. When the pressure sensing capacitor (Cp) 218 is "connected", the frequency Fosc 'of the output signal of the oscillator 252 on line 253 is a function of both pressure and temperature, as will be described in greater detail below. As will be described in greater detail below, an algorithm that is a reader / interrogator 106 is employed to derive a "pressure only" indication from the pressure-temperature measurement.
[0053]
References made herein to “pressure indication”, “pressure count”, “pressure response”, “pressure register”, etc., refer to the “pressure indication” actually measured by this transponder technique which produces a hybrid pressure-temperature indication. ] As a whole. Once the hybrid instruction has been processed to remove the temperature component of the hybrid instruction, the instruction is referred to as a "pressure only" instruction.
[0054]
As controlled by the timing generator / sequencer circuit 226, the data acquisition circuit 254 outputs the relaxation oscillator output signal Fosc 'to the temperature register 232 via line 255, depending on whether temperature or pressure is being measured. Or via line 257 to the pressure register 234. The counter converts the oscillator frequency Fosc 'into a count stored in registers 232,234. The timing "window" provided by the timing generator / sequencer circuit 226 has a known controlled duration. As a result, the count remaining (acquired) in the temperature register 232 or the pressure register 234 when the timing window is "closed" is a function of (proportional to) the frequency Fosc 'of the relaxation oscillator 252, and therefore its timing window Whatever temperature or pressure is measured during it is a function of temperature or pressure.
[0055]
An EEPROM array 236 is provided to read / write the temperature and pressure counts (NT, NP, respectively, described in greater detail below) into temperature and pressure indications that can be displayed to a user via display 112, for example. Used to hold calibration constants used by the interrogator (eg, 106). EEPROM array 236 may also store the transponder ID, calibration data for the transponder, and other data specific to a given transponder.
[0056]
FIG. 3 is a more detailed circuit diagram 300 of some of the components of the transponder 200 of FIG. 2, mainly those described above with respect to the sensor interface unit 206 of FIG.
[0057]
In this circuit diagram 300, ordinary circuit symbols are used. For example, lines that intersect each other are not connected to each other unless there is a “black point” at the intersection, and lines are connected to each other if there is a black point. The usual symbols are also used for transistors, diodes, ground connections, resistors, capacitors, switches, comparators, inverters, logic gates (eg, “AND”, “NAND”, “OR”, “NOR”). I have.
[0058]
This circuit is described with respect to a CMOS embodiment. Here, the one with a number after “P” (for example, “P1”) indicates a PMOS (P-channel) transistor, and the one with a number after “N” (for example, “N1”). ) Indicates an NMOS (N-channel) transistor. CMOS transistors are of the FET (field effect transistor) type and have three "nodes" or "terminals"-a "source" (S), a "drain" (D), and a current flow between the source and the drain. Have a “gate” (G) that controls In the following description, some PMOS and CMOS transistors are "diode-connected," meaning that their drains (D) are connected to their gates (G). It will be obvious. The general theory of operation of transistors, especially CMOS, is well known to those skilled in the art to which this invention most closely belongs.
[0059]
As will be apparent from the description below, several CMOS transistors are connected in a "current mirror" configuration. The concept of a current mirror is well known, and its simplest form uses two transistors of the same polarity (eg, two PMOS transistors), the gates of which are connected together and the pair of transistors One is diode-connected. Current mirrors generally involve passing current through a diode-connected transistor. As a result, the gate voltage required to produce that current will be present in the diode-connected transistor. Generally, the gate voltage of a diode-connected transistor is forced to the voltage required to produce a mirror current flowing through the transistor. A diode-connected transistor, by definition, is mirrored because no gate current flows by applying the gate voltage of the diode-connected transistor to any other similarly connected transistor. Current flows through similarly connected transistors. Normally, current mirror transistors have the same physical area. In that case, the mirrored current is essentially the same as the current mirrored current. It is known that making one (of the area) of a transistor physically larger or smaller than the other results in a mirrored current that is larger or smaller than the current mirrored current. If such similarly connected transistors with different areas are connected in a current mirror configuration, the areas of the transistors that differ in scale (larger or smaller) will have correspondingly different magnitudes ( (Large or small) current.
[0060]
Hereinafter, in general, numerous connections between various parts of the circuit will be clearly shown in the figures, and the description will be directed to any connection between various parts of the circuit, all of which are clearly shown in the figures. Rather than listing, the emphasis will be placed on the various functions of the various components of the circuit and the various interactions between the various components.
[0061]
The antenna device 210 includes a coil antenna 212 and a capacitor 214 (connected between terminals of the coil antenna 212 to form an LC resonance tank circuit), and supplies an alternating current (AC) to the full-wave rectifier circuit 302. Whether or not the capacitor 214 is used is optional.
[0062]
The full-wave rectifier circuit 302 (equivalent to 222), as shown, has two PMOS transistors and two diodes connected in a conventional manner to provide a full-wave rectified direct current (DC) voltage. Is output to the line 303. A capacitor 304 is connected between line 303 and ground to "smooth" (filter) the fluctuations ("ripple") in the full-wave rectified direct current (DC) voltage present on line 303. The voltage on line 303 then becomes the voltage available for the remaining components of the transponder, in this case the positive supply voltage Vcc on line 303.
[0063]
A temperature detection circuit 306, substantially corresponding to the base-emitter voltage-to-current converter 250 of FIG. 2, is connected between line 303 (Vcc) and ground, and includes four CMOS transistors P1, P2 and N1, N2, And a bipolar transistor Q1 and is connected to an external resistor 216 (Rext). The transistors P2 and N1 are diode-connected as shown. The two transistors P1 and P2 are connected in a current mirror configuration, and the two transistors N1 and N2 are also connected in a current mirror configuration that can be generally considered. The source (S) of the transistor N1 is grounded via the transistor Q1, and grounded via the source external resistance (Rext) 216 of the transistor N2.
[0064]
As will be apparent, the ability of the temperature sensing circuit 306 to generate a signal (ie, current) that is proportional to the sensed ambient temperature (eg, in the tire to which the transponder is associated) is the performance of the transistor Q1. The base-emitter voltage largely depends on the very predictable and repeatable characteristics of temperature. Resistor (Rext) 216 is an external precision reference resistor whose value is almost independent of temperature (as opposed to the temperature dependence of transistor Q1). A suitable value for the resistor (Rext) 216 is, for example, 20.5 kOhm or 455 kOhm.
[0065]
Transistor N2 is connected in "source-follower" mode between transistor P2 and an external resistor (Rext) 216. When a voltage is applied to the gate (G) of transistor N2, its source voltage "follows" its gate voltage (minus the inherent voltage drop (Vgs) between its gate and its source).
[0066]
When current flows through transistor N1, its gate voltage is canceled by the gate-source voltage drop (Vgs) of transistor N1 which is higher than the emitter voltage of transistor Q1. Since transistors N1 and N2 are substantially identical, the same current flows through each of the two transistors N1 and N2, and their gate-source voltage drop (Vgs) is the same. As a result, the voltage across the external resistor (Rext) 216 at the source of transistor N2 is approximately equal to the voltage at the emitter of transistor Q1. Therefore, applying Ohm's law (E = IR or I = E / R), the current flowing through the external resistor (Rext) 216 is obtained by dividing the emitter voltage of the transistor Q1 by the external resistor (Rext) 216 resistance value. equal.
[0067]
In normal operation, all current flowing through external resistor (Rext) 216 flows through the source of transistor N2, and thus through diode-connected transistor P2. Due to the current mirror connection, the current flowing through transistor P2 is duplicated (mirrored) in transistor P1. This ensures that the currents flowing through the transistors N1 and N2 are always the same, thereby ensuring that the emitter voltage of the transistor Q1 and the terminal voltage of the external resistor (Rext) 216 are the same independently of voltage fluctuations and processing differences. Help them do more. As described above, transistors N1 and N2 are connected in a manner that can be generally considered a current mirror configuration. However, since they are not connected exactly the same, the function of those transistors in circuit 306 is primarily to "match" Q1 and the external resistance (Rext).
[0068]
In short, the temperature detection circuit 306 makes it possible to predict the current I (T) flowing through the external resistance (Rext), and to make it a function of the absolute temperature (T) of the transistor Q1. As will be described in more detail below, the current I (T) flowing through the external resistor (Rext) 216 is mirror-transferred to a relaxation oscillator (316, described below) to provide a signal indicative of the temperature of transistor Q1 to an external source. To the reader (106, FIG. 1). As will be described in more detail below, the output frequency Fosc 'of the relaxation oscillator 312 is a function of the absolute temperature (T) of the transistor Q1.
[0069]
In this regard, it is useful to note that it is essentially the transistor Q1 that is employed as the temperature sensing element of the entire transponder circuit. This transponder circuit takes advantage of the inherent characteristics of such transistors manufactured in CMOS technology that the base-emitter voltage of transistor Q1 varies by a predictable amount -2.2 mv / ° C (millivolts per degree Celsius). This is advantageous because it is employed.
[0070]
It should be noted that the transponder of the present invention is described in terms of a "passive" device, which relies on RF power supplied by an external source (106, FIG. 1) to power its circuitry. . However, it is within the scope of the present invention that the transponder includes its own power source, such as a battery. In either case, it is important that when initially applying power, as described with respect to the temperature detection circuit 306, "gradual" reach from its quiescent state to its normal operating state. To that end, two lines 305 and 307 connected between the temperature detection circuit 306 and the “start” circuit 308 are shown.
[0071]
The start-up circuit 308 (which is also part of the base-emitter voltage-to-current converter 250 of FIG. 2) is connected between the power supply voltage (Vcc) on line 303 and ground to serve two primary purposes. (I) flowing a current to the temperature detection circuit 306 when the transponder (200) is first started from a state in which power is not supplied; and (ii) a current based on a power supply based on a current flowing through the transistor P2. To a mirror-to-ground current.
[0072]
Startup is initiated by transistor P3. Transistor P3 is manufactured to have a high channel resistance to function in "weak pull mode". It is always "on" because its gate is grounded, and behaves essentially like a resistor with a high resistance value (eg,> 1 megohm).
[0073]
At start-up, since no current flows elsewhere in the circuit, transistor P3 operates to turn on transistor N3 by pulling the gate of transistor N3 toward the power supply voltage (Vcc). This effectively connects the grounded source of transistor N3 to its drain (D), thereby grounding the gates of transistors P1, P2, P4, and also grounding the drain of diode-connected transistor P2. Is done. Then, a current flows to the drain of the transistor N3 through the transistor P2 of the temperature detection circuit 306. Since transistors P1, P2, P4 are current mirrored (via "Pbias" line 305), the current now flowing through transistor P is mirrored at transistors P1 and P4. As current flows through transistor P4 into the drain of transistor N5, the current mirror connection between transistors N4 and N5 causes a corresponding current to flow through transistor N4, thereby pulling the gate of transistor N3 down to ground potential, thereby turning transistor N3 off. It effectively "stops" the flowing current.
[0074]
However, now the current is flowing through the current mirror transistors P1, P2 and P4, so the current flowing from the transistor P1 into the transistor Q1 through the diode-connected transistor N1 causes the temperature detection circuit 306 to operate in its stable operating point state ( "Start" in its zero current state). After start-up, transistor N3 is approximately "away" from the circuit and performs its intended function.
[0075]
Transistor N5 is connected to transistor N4 (and to transistor N6, as described below) in a current mirror configuration. Thus, essentially the same current flows through transistor N5 as the current flowing through external resistor (Rext) 216 flows through transistor N5, thereby setting the reference voltage (Nbias) on line 309. . Its reference voltage (Nbias) on line 309 and the supply voltage (Vdd) on line 309 ′ are provided to current scaling circuit 310.
[0076]
The power supply voltage (Vdd) on line 309 'can be provided in any suitable manner, such as a number of bandgap voltages (Vbg) generated in a conventional manner elsewhere on the chip, and its magnitude (e.g., 1.32 volts) must be independent of temperature, as is inherent in the silicon process employed for chip fabrication. The provision of such a stable (eg, bandgap) voltage (eg, Vbg) and a power supply voltage derived therefrom (eg, Vdd) is well within the purview of those skilled in the art to which this invention most closely belongs. Which will be described in greater detail later with reference to FIG.
[0077]
The current scaling circuit 310 (also part of the base-emitter voltage-to-current converter 250 of FIG. 2) is configured in an exemplary manner as follows. The sources of the transistors P5 and P6 are connected to the power supply voltage Vdd. The gate of transistor N6 receives a reference voltage (Nbias) on line 309. Transistor N6 is connected to transistor N5 (and said transistor N4) in a current mirror configuration, and thus mirrors the flow of current I (T) through transistors N4 and N5. Thus, the current flow through the diode-connected transistor P5 mirrors the current flow through the transistors N4, N5, N6.
[0078]
Although the transistors P5 and P6 are connected in a current mirror configuration, the current flowing through the transistor P6 is increased or decreased by the ratio (N) between the physical area of the transistor P5 and the physical area of the transistor P6. For example, if the size of transistor P6 is smaller than the size of transistor P5 (i.e., the face of transistor P5 is "N" times the area of transistor P6), the current flowing through transistor P6 is correspondingly smaller than the current flowing through transistor P5. And small (1 / N). Thus, the "scaled" current flowing through transistor P6 is labeled "I (T) / N" in the figure and is provided via line 311 to relaxation oscillator circuit 312 (see 252). The ratio of the currents flowing through transistors P5 and P6 may be such that one of the transistors is simply made larger than the other, or the other of the two transistors may be two or It can be easily set by a conventional circuit processing technique such as manufacturing as a set of transistors having the same dimensions.
[0079]
Relaxation oscillator circuit 312 (see 252) is of a fairly conventional configuration and includes a measurement switching circuit 315 at the "front end" of set-reset circuit 314, which provides two phase paths 314a and 314b. This circuit 315 includes a measuring capacitor C at the front end of one phase path (φ1) 314a._FX1And a pair of complementary transistors P7 and N7 connected to the side to be charged and another measuring capacitor C_FX2And a pair of complementary transistors P8 and N8 connected to the side to be charged and a switch 350 for adding a positive and another measuring capacitor Cp. The measuring capacitor CFX2, the switch 350 and the measuring capacitor CP are all at the front end of the other phase path (φ2).
[0080]
Because the connections are shown as shown for a given transistor pair (eg, P7 and N7), if their common gate voltage is high (ie, toward a positive power supply), their output (eg, phase path) 314a) is grounded (grounded and isolated from the current I (T) / N on line 311), and when the common gate voltage of those transistors is low, its output is connected through line 311 to the phase of relaxation oscillator 312. It supplies a current I (T) / N that flows through each one of the paths (eg, 314a). As is known for circuits such as relaxation oscillator 312, if the common gate voltage at one of the pair of transistors (eg, P7 and N7) is high, the common gate voltage at the other of the pair of transistors (eg, P7 and N7) Is low, and vice versa. Thus, each phase path 314a and 314b has a certain duty cycle (ie, its "on" time). Its duty cycle can be the same or different from the duty cycle of the other phase paths 314a and 314b, respectively. Then, each transistor pair (eg, P7 and N7) can be considered an “input switch” to its respective phase path (eg, 314a).
[0081]
Each phase path 314a and 314b of relaxation oscillator 312 has a comparator 316a and 316b at its input terminal, respectively, and is fixed between the negative (-) input terminal of comparators 316a and 316b and ground. Capacitors CFX1 and CFX2 having different values. Capacitor C_FX1And C_FX2Have a capacitance of, for example, 2-5 pf (picofarad) and 2-5 pf, respectively, and can be implemented as equal value "on-chip" devices, such as poly-poly capacitors that exhibit a low temperature coefficient (eg, less than 20 ppm). preferable. The positive (+) input terminals of comparators 316a and 316b are connected together and set to a temperature independent reference threshold voltage Vbg, such as 1.32 volts.
[0082]
"NOR" gates 318a and 318b are connected to the input terminals of each phase path 314a and 314b, respectively, and the two NOR gates 318a and 318b are cross-coupled to form a holding circuit that produces an output on lines 319a and 319b. . Thus, cross-coupled NOR gates 318a and 318b can function as flip-flops, ie, RS (reset / set) latches.
[0083]
If the common gate voltage of one of the input switches (eg, P7 and N7) is high, the respective capacitor (eg, CFX1) for its phase path (eg, 314a) is grounded (shortened to lose charge). Is). Conversely, if the common gate voltage of one of the input switches (eg, P7 and N7) is low, the scaled current I (T) / N from line 311 will be due to its phase path (eg, 314a). Of each capacitor (for example, C_FX1) Is added (allowed to flow), and the capacitor begins to charge (the voltage across the capacitor increases). Capacitor C_FX1/ C_FX2When the voltage between the terminals reaches the reference voltage Vbg of the comparator, the output of the comparator 316a / 316b goes low, changing the state of the output of the latch 318a / 318b on the line 319a / 319b. Thus, the relaxation oscillator 312 is connected to the capacitor C_FX1And C_FX2Rise time, and importantly, capacitor C_FX1And C_FX2Oscillates at a frequency Fosc determined by the scaled current I (T) / N supplied to When a larger current I (T) / N is supplied, the capacitor C_FX1And C_FX2Rises faster, crosses the threshold voltage faster, causing the relaxation oscillator 312 to oscillate faster, thereby increasing the frequency Fosc of the signal on line 319a. The signal on line 319a is inverted by inverter 320 as shown to provide frequency Fosc 'on line 321.
[0084]
As will be described in more detail below, the oscillator 312 operates in two mutually exclusive modes of a temperature detection mode (between times t0 and t1) and a pressure detection mode (between times t1 and t2). , A timing generator / sequencer. The frequency Fosc (and Fosc ') of the oscillator output signal is different in each of these modes.
Generation of temperature and pressure readings
In the example of the situation of the transponder 200 associated with a pneumatic tire, it is mainly desirable to determine the pressure in the pneumatic tire. For example, a typical passenger car can properly inflate at about 221 kPa (about 32 psi). Tire inflation pressure is usually specified as the "normal temperature" pressure (measured when the tire is not generating heat from running) and the monitoring device is most likely in use and therefore a "hot" tire. It is secondly desirable to determine the temperature of the inflation medium (e.g., air) inside the pneumatic tire because it signals the pressure measured within. Using the temperature measurements, the monitoring device (eg, 106) can convert the measured pressure to “normal” pressure, for example, with a simple calculation based on the ideal gas law (PV = μRT). This "normal temperature" pressure can be considered a "temperature independent" pressure, which is also indicative of the mass of air contained within the tire. Referring to transponder 200, the hybrid "pressure" measurement it produces must be able to convert only the true pressure into a measurement before it can be used for such gas law calculations (by various calculations described in detail below). ).
[0085]
For example, it has been estimated that a 10% reduction in fuel consumption can be achieved if the pneumatic tire of the vehicle runs at its specified pressure. Professional truck drivers are usually susceptible to this problem and frequently check and adjust tire pressure, while the average driver of a passenger car is aware of tire pressure, for example, until the tire is visibly dented Things are often less. In such a case, an LCD (Liquid Crystal Display) indication on the vehicle dashboard provides the driver with dynamic tire inflation information. A transponder such as the transponder described here is mounted on the pneumatic tire of the vehicle. Significantly, there is the emergence of "run flat" tires marketed by various tire manufacturers. Goodyear's EMT (extended @ mobility @ technology) series tires are an example of a "run flat" tire. Its overall purpose is to maintain normal vehicle control with punctured ("flattened") tires at "reasonable" driving speed (eg, about 144 kilometers per hour (about 60 miles per hour)) While traveling about 120 kilometers (up to about 50 miles). Such run-flat tires are generally well known and as such do not form a part of the present invention. If you are running “flat” on runflat tires, the tires are actually “flat” and must be located at the closest convenient location to the driver (and before the runflat mileage limit). It is particularly important that the driver be alerted to the fact that the driver is driving for "borrowed time" primarily indicated visually or by voice (beep) that repairs are needed It is.
[0086]
By allowing the relaxation oscillator 312 to operate, the frequency Fosc (and Fosc ') of its output signal is a function of the (detected) absolute temperature of transistor Q1. This is the case in temperature sensing mode operation and pressure sensing mode operation.
[0087]
In the temperature detection mode, the capacitor C_FX1And C_FX2In this case, relaxation oscillator 312 has a symmetric (balanced, 50%) duty cycle. In the pressure detection mode, the pressure detection capacitor (Cp) 218_FX2Is switched by a semiconductor switch 350 between the terminals of the first and second terminals. It changes the duty cycle of the relaxation oscillator 312 and the output frequency Fosc (and Fosc ').
[0088]
In the pressure detection mode, the fixed capacitor C_FX1And C_FX2Only are alternately charged (and discharged) resulting in a 50% duty cycle whose period is proportional to ambient temperature. In the pressure detection mode, the pressure detection capacitor (Cp) 218 is switched to the phase path 314b of the oscillator 312. Thus, for a given temperature, the phase path 314a behaves the same as in the temperature sensing mode during the first half of the oscillator period and the phase period 314b during the second half of the oscillator period has a fixed capacitor C_FX2And the capacitance value of the pressure detection capacitor (Cp) 218. This in effect slows down the oscillator and changes its duty cycle. The duty cycle changes are Cp and C_FX2Shows the ratio of Thus, from the ratio of the two periods (with and without Cp in the circuit), it is easy to calculate what the additional capacitance Cp is and thus what the detected pressure is. is there. As will be described in detail later, the temperature dependence of the oscillator output in the pressure detection mode can be completely eliminated in a linear manner.
[0089]
When the operation of the oscillator "slows down" when the pressure sensing capacitor (Cp) 218 is switched to the oscillator circuit, the counting oscillator output pulse during a given pressure measurement window (eg, Wp) will be similar. Temperature measurement window (eg, W_TIt is inevitable that the number of output pulses of the counting oscillator in () becomes relatively smaller (the output frequency becomes lower). In other words, a "slow" oscillator reduces the rate at which counts indicative of parameter measurements are collected. In order to increase the resolution (number) of the count (Np) generated during the pressure measurement window (Wp), the size of the pressure measurement window (Wp) is adjusted so that an appropriate number of pressure counts can be obtained. Can be increased (variable duration). This means that the value at time t2, which defines the end of the pressure measurement window (WP) in the pressure detection mode (between times t1 and t2), is simply increased by the timing generator / sequencer circuit 226 controlling that value (others). Than in the case). For example, the temperature measurement window W_T(Between times t0 and t1) can be on the order of a few milliseconds (eg, 8 milliseconds). Alternatively, increasing the fundamental frequency of the relaxation oscillator 312 by allowing the scaled current (I (T) / N) flowing from the current scaling circuit 310 to the relaxation oscillator 312 to increase during the pressure measurement window (Wp). Is intended to increase the overall resolution of the pressure count. This can be easily performed, for example, in the case of the transistor P6 having a smaller size (area) than the transistor P5, by simply turning on the transistor P6 '(not shown) instead of the transistor P6. The area of transistor P6 'is increased such that the area ratio of transistors P5 and P6 is close to 1 (i.e., the degree of decrease is small), and the current to relaxation oscillator 312 increases, and thus its counting speed increases. Larger than the area. Such conduction of the other transistor P6 'is easily accomplished with a switch (not shown) comparable to the switch 350 that switches the pressure sensing capacitor (Cp) 218. One of ordinary skill in the art to which the present invention most closely pertains will appreciate how to avoid "slowing down" the oscillator when the pressure sensing capacitor (Cp) 218 is connected to the oscillator circuit in accordance with the teachings herein. It will be easily understood.
Optimization of pressure response
In order to obtain (and display) a precise pressure reading that is of utmost importance when monitoring the pressure of a pneumatic tire, certain parameters of the transponder circuit are determined to maximize its pressure responsiveness, and thus external reading The accuracy of the pressure reading displayed by the instrument / interrogator (eg, 106) can be increased.
[0090]
As described above, this transponder responds to the changing capacitance of the pressure sensor or pressure sensing capacitor (Cp) 218 by changing the value of the 12-bit word sent to the external reader / interrogator 106. This binary word is a count of the oscillator frequency during the pressure measurement window (Wp) (between times t1 and t2) set by the timing generator sequencer 226. Accordingly, the pressure response can be described as a count change per unit change in the capacitance of the pressure detection capacitor (Cp) 218.
[0091]
It has been found that the transponder pressure response (and resolution) depends on several factors. Each of these factors can be analyzed. For example,
(A) Temperature measurement window W_TIncreasing the pressure measurement window Wp to make it larger increases the pressure count Np for a given value of the pressure sensing capacitor (Cp) 218 to compare the frequency of the oscillator that occurs during pressure measurement with that during temperature measurement. And relatively low (as detailed above).
[0092]
(B) Increasing the scaled current I (T) / N supplied to the oscillator 312 causes the pressure count Np to increase proportionally for a given value of the pressure sensing capacitor (Cp) 218.
[0093]
(C) Capacitor C_FX1And C_FX2Increases the pressure count Np for a given value of the pressure sensing capacitor (Cp) 218.
[0094]
(D) Increasing the scaled current I (T) / N increases the pressure count Np (for a given value of the pressure sensing capacitor Cp)_FX1And C_FX2It is faster than the decrease in the value and increases in proportion to the increase.
[0095]
(E) If the scaled current I (T) / N is increased and the current cannot be increased only during the pressure measurement window Wp, the pressure count Np and the temperature count N_TIncrease.
[0096]
(F) Capacitor C_FX1And C_FX2Decreases the pressure count Np and the temperature count Np, even if only one of the capacitors is changed._TIncrease.
[0097]
As a general proposal, it is desirable to increase the pressure count Np. However, those skilled in the art to which the invention most closely pertains will recognize that there is a practical upper limit to increasing the pressure count at frequencies that may be unacceptably high for the performance of certain circuits of the IC chip. It will be easily understood.
Measuring parameters
FIG. 3A shows the components involved in the final stage of obtaining the temperature and pressure measurements at the transponder. The signal Fosc 'output by the relaxation oscillator 312 is supplied to the input terminals of the two AND gates 360 and 362 of the data acquisition circuit 254 via the line 321 (see 253). Count ("data" or "indicated value") indicating the measured temperature N_TIs loaded into the temperature register 232 via line 255 so that the temperature measurement window (W_T) Is supplied to the other input of the AND gate 360 by the timing generator / sequencer 226. During the pressure measurement window (Wp), another data signal (“Capture @ Press”) is generated by the timing generator so as to load the pressure register 234 with a count (“data” or “indicative value”) indicating the measured pressure. / Sequencer 226 supplies the other input terminal of AND gate 362. Each register 232, 234 is associated with a counter (not shown) to convert the incoming oscillation signal Fosc 'into a stored count. Then two counts N_T, Np are sent out from the registers 232 and 234 to the modulation circuit 246 via the MUX 240.
[0098]
When the transponder is powered, the temperature and pressure are measured continuously and those measurements are sent to the external reader / interrogator 106 as data words in the data stream. For example, each of the temperature and pressure parameters can be sent to the reader / interrogator 106 as a 12-bit data word in a selected (known) portion of a larger (eg, 144-bit) data stream. One bit in the entire data stream can be dedicated to the state of the MTMS switch 220 (eg, “closed” or “open”). A complete description of an example of a data stream sent by a transponder to an external reader / interrogator is described below with reference to FIG. 3C.
[0099]
Temperature is a fixed time (window W of time from t0 to t1) with period tT_T) Is properly measured by counting the number of cycles output from the oscillator 312. For example, a down counter (not shown, but associated with temperature register 232) may be clocked by an oscillator to provide a window W_TPeriod t_TTemperature count N at the end of_TTo be able to occur. Temperature count N_TThe relationship between temperature and temperature is nearly linear for the circuit 300 of this embodiment.
[0100]
Similarly, the period t_PIs properly measured by counting the number of cycles output from the oscillator 312 during a fixed time (window Wp from t1 to t2) having For example, a down counter (not shown, but associated with pressure register 234) may be clocked by an oscillator to generate a period t of window Wp._PAt the end of the pressure count Np. The relationship between the pressure count Np and the pressure is a predictable function of the actual pressure and temperature for the circuit 300 of this embodiment. As described below, the temperature count (N_T) And the "pressure" count (Np), this hybrid pressure-temperature value can be used to determine a pressure-only value.
Obtain pressure-only readings in reader / interrogator
The fundamental frequency of oscillator 312 is set on the IC chip (eg, 202) and, as described above, is temperature dependent. Thus, the pressure response, or pressure count Np, is a function (hybrid) of temperature and pressure, and the relationship between Np and Cp is non-linear. Therefore, using linear equations to calculate the pressure response will inevitably result in large errors over the range of pressures measured. However, to measure over a limited range of pressures, for example, a range of about 138 kPa (20 psi), using a linear equation may be acceptable. A better approximation may be obtained using multi-dimensional equations, but this complicates the reader / interrogator logic, slows down the response, and requires additional calibration constants.
[0101]
The advantage of using the above transponder circuit is that N_T/ Np and the pressure sensor capacitance Cp are linear, and the reader / interrogator 106 does not require a temperature compensation term in the equation (algorithm) used by it to calculate the pressure, and thus the reader / Make interrogator design extremely simple. (This also assumes the use of a pressure sensor (Cp) 218 that has a nearly linear relationship between pressure and capacity.) This advantageous "ratiometric" relationship is easily illustrated by the following equation: Can be.
[0102]
In general,
Count = window time to be counted (t) * frequency (F)
H = 1 / period
Charging time = V * C / I
For a capacitor whose capacitance to be charged to voltage V with current I is C.
[0103]
Since the period of the relaxation oscillator 312 having the output signal whose frequency is Fosc 'is the sum of the charging times of the capacitors in the two phase paths 314a and 314b, C_FX1And C_FX2Can be handled to obtain a general equation for counting from such relaxation oscillators with For example,
Count = t / (V * C_FX1/ I + C_FX2/ I)
= T * I / (V * (C_FX1+ C_FX2))
Substituting values for temperature count and pressure count
N_T= (T_T* I (T) / n_T) / (Vbg * (C_FX1+ C_FX2))
... [Formula A]
Np = (t_P* I (T) / n_P) / (Vbg * (C_FX1+ C_FX2+ Cp)) where n_TAnd n_PIs the value of the conversion factor N in the scaled current I (T) / N, which can be different between the temperature measurement window and the pressure measurement window.
[0104]
Formula N_TDivided by Np
N_T/ Np = (t_T/ T_P) * (N_T/ N_P) * (C_FX1+ C_FX2+ Cp) / (C_FX1+ C_FX2)
That is,
N_T/ Np = (t_T/ T_P) * (N_T/ N_P) * (1+ (Cp / (C_FX1+ C
_FX2))
... [Equation B]
Is obtained. Since the terms on the right side of the equation B are all constants except for the pressure detection capacity Cp, N_TIt can be seen that there is a linear relationship between / Np and Cp (and therefore pressure). This is N_TThis means that / Np is only a function of pressure and is not affected by temperature changes or fluctuations in capacitor charging current.
[0105]
If none of the response optimization steps described above are used, capacitor C_FX1And C_FX2Have the same value C_FXAnd the measurement window W_TAnd Wp have the same time width t_T= T_P= T_W(For example, 8.192 ms) and the current reduction coefficient n_TAnd n_PHave the same value N, so equation B can be simplified.
[0106]
N_T/ Np = 1 + (Cp / 2 * C_FX)
From equation A, the temperature count N_TIt can be seen that a linear relationship already exists between the current and the current I (T), which is proportional to temperature.
[0107]
It can be seen that there is a linear relationship in both measurement equations A and B, but the slope and intersection of those equations is a complex combination of many parameters specific to a given transponder configuration, and the manufacturing error Even for each transponder in a given configuration. In a simple embodiment of the invention, the transponder has a count N_TAnd Np can be sent to the reader / interrogator, and the reader / interrogator must use the assumed mean values for slope and intersection to determine temperature and pressure. Because this introduces significant inaccuracies, the preferred embodiment described herein stores the calibration constants in the transponder's memory (eg, 236) and the reader / interrogator (eg, 106) In order to be able to accurately calculate the temperature and pressure using the linear equations specifically prepared / optimized for the individual transponders generating the_TAnd with Np. The linear equation used in the reader / interrogator example (eg, 106) is in the well-known “point-slope” form.
[0108]
yy₁= M (xx₁)
here,
(X₁, Y₁) Is the defining point (defining point)
m is a gradient.
The gradient m is a straight line (x₁, Y₁), (X₂, Y₂) Can be determined from any two points above.
[0109]
m = (y₂-Y₁) / (X₂-X₁)
Substituting for x and y, the specific equation for the temperature line is
N_T-N_T1= M_T(TT₁)
become. Temperature T₁If you select a value such as 25 ° C for
N_T-N_T25= M_T(T-25)
Is generated. This equation is_TSolving for yields the following equation for the temperature response line:
[0110]
N_T= M_T(T-25) + N_T25
Where the slope m of this temperature response line_TIs
m_T= (N_T2-N_T1) / (T₂−T₁)
It is. As long as the reader / interrogator “knows” the assumed decision point temperature (eg, 25 ° C.), the calibration constant sent to the reader / interrogator, ie, the decision point temperature count N_T25And the gradient m_TAnd the temperature count N_TThe temperature (T) can be calculated from the value received. A similar set of equations can be applied to determine pressure from sent pressure (and temperature) counts and pressure calibration constants. As noted above, the pressure-only reading is the ratio N instead of the pressure count Np itself._TIt can best be determined from a linear equation using / Np (temperature count divided by pressure count).
[0111]
The calibration constant determines the exposure of each transponder to a controlled, known combination of temperature and pressure conditions, and the temperature count N generated by that transponder._TAnd a corresponding calibration of the pressure count Np. Calculations on those calibration test results determine four calibration constants. The results are then stored in a transponder memory (eg, 236). The four calibration constants are temperature versus temperature count N_TSlope and decision point for pressure linear response, and pressure (only) to count ratio N_T/ Np is a number representing the slope and decision point for a linear response.
Reliable supply and reference voltage generation
As described above, the positive (+) input terminals (terminals) of comparators 316a and 316b are connected together and set to a temperature independent reference "bandgap" voltage Vbg, such as 1.32 volts. . Also, as mentioned above, the power supply voltage (Vdd) on line 309 'is provided as a number of reference bandgap voltages (Vbg) so as to be a stable operating voltage for current scaling circuit 310 and relaxation oscillator 312. it can.
[0112]
FIG. 3B shows a circuit 370 suitable for generating the power supply voltage Vdd. A bandgap voltage Vbg, which can be calculated independently of temperature, can be easily obtained as specific to the selected process (eg, CMOS) based on the processing techniques employed in IC chip fabrication. . This bandgap voltage Vbg is supplied to a positive (+) input terminal of an operational amplifier connected as shown to a feedback loop having a gain, and supplies a power supply voltage Vdd as an integral multiple of the bandgap voltage Vbg. .
Data flow example
As mentioned above, the information (data) from the transponder is sent in the form of a data stream to an external reader / interrogator. Part of that data stream is the temperature count N_TAnd another part is the pressure count Np and another part represents the state of the MTMS switch (220) (eg, "closed" or "open"). The remainder of the data stream may include information specific to that transponder, such as the identity information (eg, serial number), calibration constants, etc., for a given transponder.
[0113]
FIG. 3C shows an example of the architecture of the information stored in memory (eg, 238) within transponder 200 and the data stream sent by transponder 200 to external reader / interrogator 106. The memory 238 of the transponder core 204 has an address space of, for example, 144 bits. Its address space is 119 bits of programmable memory and one address location dedicated to the state of the MTMS switch 220-those 120 bits of programmable memory make up the EEPROM 136. -Plus a temperature register 232 and a pressure register 234, each of 12 bits.
[0114]
Each of the 119 bits of programmable memory is separately written with any combination of data including sync pattern information, general data, error detection codes, and temperature and pressure calibration data. Can be included. The EEPROM is "writable in blocks". This means that in the "write mode", all 120 bits of the EEPROM are programmed to a logical (binary) value of "1". Individual bits can be “erased” (set to a logical value of “0” simply by locking the chip to the physical address of the bit and placing the chip in “erase” mode). Address locations are preserved.
[0115]
In this example, the first twelve data locations (000..011 in row 1) are reserved for synchronization. The next 71 data locations (012.082 in rows 2-7) are for general information and data validation algorithms such as CRC (cyclic redundancy check). The next data location (083) contains the logic level (state) of the MTMS switch 220. A logical value “1” indicates that the MTMS switch is open, and a logical value “0” indicates that the MTMS switch is closed.
[0116]
As mentioned above, each transponder unit is properly calibrated before being mounted in the tire. The next twelve data locations (084.095 in row 8) hold the temperature calibration (eg, decision point) data ("TEMP @ COMP"). The next twelve data locations (096..107 in row 9) hold pressure calibration (eg, decision point) data ("PRESS @ COMP"). The next twelve data locations (108..113 and 114.119 in row 10) each hold temperature and pressure calibration (eg, slope) information.
[0117]
As described above, count N for temperature and pressure_TAnd Np are generated, they are stored in rows 11 and 12 of the overall memory space. The rows correspond to temperature register 232 and pressure register 234, respectively. Various predetermined values can be stored to indicate error conditions such as overflow and short circuit.
Operating frequency and modulation
The transponder of the present invention is not limited to any particular frequency. The choice of operating frequency depends on the location where the transponder is mounted, the location of the reader / interrogator antenna (108) and the selected part of the total RF frequency spectrum, in relation to the object to be monitored by the transponder. Permitting (or limiting) the transmission of data depends heavily on factors such as relevant national laws and regulations.
[0118]
Examples of suitable operating frequencies for transponder operation in the United States are 60 KHz to 490 KHz.
[0119]
The transponder can be polled (and powered) at a first "interrogation" frequency (Fi) by a reader / interrogator 106, and is a data stream divided by a factor of the interrogation frequency or an integer. Can be sent to the reader / interrogator at a convenient second "data carrier" frequency (Fc). For example, Fc = Fi / 2 or Fc = Fi / 4. The frequency (Fc) at which the data stream is sent to the reader / interrogator is independent of the data rate. The data transmission speed is set by a clock generator 224 and a baud rate generator 248. However, those skilled in the art to which the invention most closely pertains will recognize that the range of available baud rates is typically much lower than the interrogation frequency (Fi). The baud rate is preferably derived from the interrogator frequency (Fi), such as the integer divided by the interrogator interrogator frequency (Fi). For example, the baud rate can be set to Fi / 32 (or if Fc = Fi / 2, the baud rate can be set to Fc / 16).
[0120]
For example, the interrogation frequency (Fi) can be set to 125 KHz, the data carrier (Fc) to 62.5 KHz, or half the interrogation frequency.
[0121]
In another example, a query frequency (Fi) of 13.56 MHz has been found to be appropriate.
[0122]
A data stream, such as the example data stream described with reference to FIG. 3C, is provided by modulator circuit 246 to antenna 212 and sent to reader / interrogator 106. It is within the scope of the present invention to employ any suitable modulation method, including amplitude modulation (AM), frequency modulation (FM), frequency displacement keying (FSK), and phase displacement keying (PSK). However, phase displacement keying (PSK) is preferred. AM modulation is not particularly well suited for digital transmission. Frequency modulation methods, such as FM or FSK, have some problems in propagating data-modulated transponder output signals through the media of a pneumatic tire (eg, 104).
Ratio vs. signal strength
Ratio N for pressure indication_TThere is another advantage to using / Np. The reason for this is that the ratioed value is N to the change in coupling between the reader / interrogator and the transponder._TThis is because it has been confirmed that the user does not feel any more than when the measured value of Np is taken alone. This is shown in FIG. 3D, which shows a graph of measurement count (vertical axis 394) and power (horizontal axis 392). For a passive transponder 200, such as the transponder described in the preferred embodiment of the present invention, the transponder power is provided by an RF signal from a reader / interrogator (eg, 106). The power in the transponder 22 circuit may be reduced if the RF coupling strength is weakened by transmission or reception problems, including too long distances or jamming. Some value PWR₁At lower power levels, the relaxation oscillator 312 outputs a lower Fosc 'than the normal frequency signal, and thus the temperature count N_TAnd the pressure count Np are made smaller than would be naturally obtained for a given temperature and pressure. This effect is the minimum power PWR₁Temperature count N extending below_TAnd a downward curve 398 of the pressure count Np. Fortunately, the effect of low power is proportionally the same for both counts, so the ratio N_T/ Np (curve 399) is the minimum power PWR required to operate the transponder 200₀Relatively stable for all power levels up to. Therefore, N at the transponder_TBy determining the ratioed value of / Np (during calibration) and storing the calibration data, a pressure-only reading that is less sensitive to variations in the coupling between the reader / interrogator and the transponder The ability to determine is simplified and made more reliable.
Improved, general
The present invention deals with a new "RFIQ" transponder 400 (see FIG. 4A) that implements an improvement over the conventional model "3070C" transponder 200 described above. Several improvements have been made and new features obtained, including but not limited to the following:
・ Small power consumption,
.Improvement of oscillator stability with respect to power or frequency,
・ Improved resolution of temperature count and pressure count,
・ Electrostatic discharge (ESD) prevention is improved to 2200V or more,
・ Improvement of programmable modulation degree (the magnitude of PSK modulation added to RF signal)
Reduced number of external connection pins for programming and testing,
・ Improved testability in digital and analog,
An increase in the data stream to 192 bits,
An increase in the programmable EEPROM to 156 bits,
・ Programmable by the attached antenna,
Parity 6 bits -N_T, 1 bit for every 4 bits of Np data,
The ability to programmably reduce current to independently optimize the indicated counts for pressure and temperature;
A mode of supplying power from a 3V battery for use in the "active" implementation of the transponder ("active tags",
・ Power on reset
・ Baseband data output to test pins.
Problems to be solved
In particular, the configuration of the above-described conventional transponder 200 has a general problem. Some of these problems are ameliorated by the improvements that are the subject of the present invention.
[0123]
Pressure count Np and temperature count N for a desired pressure range and a desired temperature range_TAlthough the relaxation oscillator of the transponder 200 of the conventional configuration can be adjusted to optimize, the adjustment can only be performed during the manufacture of the integrated circuit. After manufacture of the integrated circuits their counts Np and N_TIt is desirable to have a method that optimizes
[0124]
When using a transponder 200 of conventional configuration (as a passive transponder), incorrect results may be obtained when the power input is too low, such as during start-up, or when far away from the reader / interrogator. May be sent. (See, for example, FIG. 3D.) Also, conventional models start modulation (transmission) immediately upon receiving an interrogation signal of any strength and start transmission at random locations in the data word. Given that the initial signal (from the remote reader / interrogator) was weak, the voltage supply may not be sufficient to produce reasonable temperature and pressure readings. Also, if the modulation is started before a sufficient power level occurs, the power consumption of the signal modulation exacerbates the problem of insufficient power. Another problem arises if the relaxation oscillator does not start every cycle in a defined and consistent state.
[0125]
-The transponder 200 having the conventional configuration has a certain degree of modulation (the magnitude of RF signal modulation) determined during the manufacture of the integrated circuit. It is desirable to have a way to optimize the transponder for different combinations of antennas ("coils") and reader / interrogator 106, and for different operating conditions. Also, some transponder applications use an external zener clamp between the terminals of the antenna 210 to increase stability, which Zener "limits reading" of the transponder 200 in a conventional configuration.
[0126]
• Transponders 200 of conventional configuration can adjust certain operating characteristics during manufacture (eg, adjust transistor size to adjust current), but changes during manufacture are permanent. However, it is not easy to change for each transponder. Further, if adjustment settings are made in a programmable memory (e.g., an EEPROM), those settings may be accessed continuously during power up rather than through the row decoder (242) and column decoder (240). Doing so creates other problems.
[0127]
Other improvements and solutions will be apparent from the description below.
General description
The improved RFIQ ™ transponder 400 (comparable to 200, 102) is a low frequency “passive” (powered by RF signal) transponder or “active” (battery powered) measurement device As a custom CMOS, low voltage integrated circuit ("IC", or "chip") that can measure temperature and pressure. The IC includes (a) whether the component is active or passive, (b) adjusts the temperature and pressure resolution, and (c) the degree of modulation and the clamping strength of the coil (when in passive mode). Perform programmable trimming to determine,
[0128]
In passive mode, the IC generates power from a reader / interrogator RF signal that is coupled to an external LC circuit that is coupled to the transponder's antenna input. The transponder uses the received signal to generate power and an on-chip clock. The transponder modulates the impedance of the antenna, which is known as "backscattered modulation", thereby sending its memory content back to the reader. The reader demodulates the returned data to obtain the sensor identification ("ID") and sensor data and the calibration constants needed to interpret the data.
[0129]
The transponder sends a 196-bit serial phase shift keyed ("PSK") data stream as 16 12-bit words. The first 12 bits are a programmable sync word, and each sync bit is 1.5 bits wide. Next, 144 bits of the EEPROM having the normal bit width are transmitted. The 144 bits contain the transponder's unique ID code, temperature and pressure data calibration constants, and a CRC for error checking. Thereafter, the transponder sends 36 bits of data. The 36 bits are a 12-bit temperature count (N_T), A 12-bit pressure count (Np), a 12-bit word, which contains 5 unused bits as 1, 1 bit indicating the state of the MTMS for the temperature sensor, and even parity. 6 bits. One parity bit is Np data and N is one._TFor every 4 bits of data.
[0130]
In the active mode, the transponder must be controlled by external hardware. The control hardware powers the IC and provides a clock signal via a CLK pad or VB pad. The clock must be provided at the exact clock rate so that temperature and pressure are collected during a fixed period, and therefore, to achieve the highest resolution of temperature and pressure, the hardware must Clocking time can be optimized. The IC can operate down to voltages as low as 2.8V in active mode.
[0131]
As a passive tag or an active tag (transponder), the IC can be tested after assembly by supplying power and communicating with the chip via a 4-pin interface. This interface allows the user to access the EEPROM of the chip and to test the functionality of the chip. The EEPROM data can be read, cleared, or programmed, and the oscillator is tested by reading the frequency directly. Chip sensor oscillators can also be tested, providing a second method of reading temperature and pressure in active mode.
Overview of RFIQ ™ Transponder Circuit
FIG. 4A is a diagram comparable to FIG. 2 and is a block diagram of relevant portions of an improved RFIQ ™ transponder 400 (see 102, 200), with subsequent signals, terminals, and functional blocks (see FIG. 4A). Sections) and connections between them. Although the illustrated device is described as a preferred embodiment for measuring pressure and temperature, it is within the scope of the present invention to include employing appropriate sensors to measure other parameters.
[0132]
Transponder 400 is preferably implemented on a single integrated circuit (IC) chip, shown within dashed line 402 (see 202). Several external components are connected to the chip. The other dashed lines in the figure indicate the large functional “blocks” (“compartments”) of the transponder 400 and include the addressable memory block 438 (see 238) and the sensor interface 406 (see 206).
[0133]
Components external to the IC chip 402 include an antenna (coil) 412, a capacitor 414 connected between terminals of the coil 412 to form an LC resonance tank circuit, an external precision resistor (Rext) 416, an external pressure It includes a sensing capacitor (Cp) 418 (see 218) and an external maximum temperature measurement switch (MTMS) 420 (see 220). The use of the capacitor 414 and switch 420 is optional. Each external component has appropriately labeled connection pads as shown in FIG. 4A: VA and VB for antenna device 410, Rext of precision resistor 416, high side of pressure sensing capacitor 418. And the MTMS for the high side of the maximum temperature measurement switch 420. The grounding of the analog external components Rext, Cp, and MTMS should be through an analog grounding AGND pad. Another ground pad (GND) is for grounding the external digital connection. The remaining connection pads are for use in the active or test operation mode of the transponder 400, as described above.
[0134]
Antenna 412 can be in the form of a coil antenna, loop antenna, dipole antenna, or the like. Antennas are mainly used when the transponder is in passive mode. Alternatively, when the transponder 400 is in the active mode, the antenna device 410 may not be present and the signal output by the transponder 400 can be made via a direct connection to the DATA pad. In the following main part, a transponder having a coil antenna and used in a passive mode will be described.
[0135]
The transponder IC 402 processes an RF signal, such as an unmodulated carrier signal with a frequency of Fi (eg, 125 KHz) received by the antenna 412, and is used to power the transponder 400 operating in a passive mode. To the extent possible, it includes an interface circuit 422 (see 222) that rectifies the received RF signal. Signal processing is performed on the incoming signal in a form suitable for use to generate timing / clock pulses for transponder 400, and modulates the carrier signal for transmission at antenna unit 410. Including adding.
[0136]
The rectified carrier signal is limited to a maximum of about 13.0 volts to prevent breakdown of the IC 402 substrate. The limited rectified signal has a voltage Vpp in the range of 0 to 13 volts that can be read on the VPP pad. Thereafter, the voltage Vpp is shunted up to 6.5 volts. The regulated voltage is named the power supply voltage (or "input voltage") Vxx that can be read on the VXX pad. The Vxx voltage level is adjusted to prevent it from becoming high enough to damage the CMOS in IC 402.
[0137]
The voltage Vxx is controlled by a power supply on reset (POR) circuit 482 (which is new to this model of transponder) and a regulation and bandgap reference circuit 423 (222) for supplying various voltages to the IC chip 402. See).
[0138]
The power supply on-reset circuit 482 may be used to initiate the recording of the sensor readings by the transponder 400 or to provide data until sufficient power is provided by the received carrier signal to allow the transponder 400 to function properly. Is provided to send. POR circuit 482 evaluates the voltage Vxx level and outputs a reset signal that is not output until voltage Vxx is deemed sufficient. If desired, a reset signal can be applied from outside of chip 402 via a RES connection pad, such as for testing.
[0139]
In passive mode, as long as power supply voltage Vxx is sufficient (determined by POR circuit 482, but at least 4 volts), regulation and bandgap reference circuit 423 adjusts voltage Vxx to provide an operating range of 2.8V (stable). (The lowest voltage of the relaxation oscillator 452) provides a regulated analog power supply voltage Vcc of about 3.5 volts, which is about 3.5 volts. The digital power supply voltage Vdd is regulated by a source follower connected to the voltage Vcc and can supply current independently of the voltage Vcc (comparable to the earlier method shown in FIG. 3B described above). Voltage Vdd is typically one threshold below Vcc, about 2.5 volts, and ranges from 1.2V (the lowest voltage for stable logic and memory operation) to about 3.5V. , And are supplied to the digital logic and EEPROM 436 (memory) array 436 during reading. During transponder programming, voltages Vcc and Vdd are not affected by the power supplied to pad VPP. Voltages Vcc and Vdd can be read externally through pads (VCC and VDD) named correspondingly. Also, in the active (battery powered) mode, external power is supplied to pads VCC and VDD, eliminating the need for a regulator. Finally, for circuits requiring a stable reference voltage, the regulation and bandgap reference circuit 423 provides a temperature independent bandgap voltage Vbg. The voltage Vbg is independent of the voltage Vxx as long as the chip power supply voltage Vxx is above the minimum operating level. The conditioning and bandgap reference circuit 423 outputs a temperature-independent voltage Vbg of about 1.20 volts over a transponder operating range of, for example, -40 to 150 ° C. The bandgap voltage Vbg is used as a reference voltage by Vxx, Vcc and Vdd adjusters 423, relaxation oscillator 452 and power supply on / reset circuit 482.
[0140]
The interface and rectifier circuit 422 determines the received RF signal, the input frequency (Fi) it receives, preferably at a timing that generates a clock signal in a known manner to control the timing of other circuits in the IC chip 402 and The clock generator circuit 424 (see 224 and 226). The generated clock signal is a square wave having a duty cycle of 50% and preferably a frequency of Fi, independent of any modulation applied to the antenna device 410 by the transponder 400 for transponder PSK transmission. . Timing and clock generator circuit 424 also divides the system clock to generate timing for addressing data in addressable memory 438 and for modulation. For example, the system clock frequency Fi is reduced by half to determine the frequency of the PSK-modulated return carrier signal. Another division of the frequency Fi is used to determine the baud rate for data transmission. In the active or test mode of operation, the timing and clock generator circuits may not be used, or may be used as buffers for direct input of a clock signal via the CLK pad or VB antenna pad.
[0141]
Various clock signals from timing and clock generator circuit 424 control column decoder 440 (see 240), column-to-data converter 441, and row decoder to control access to data stored in addressable memory 438. & N_T, Np control circuit 442 (see 242). The clock signal is supplied to the row decoder & N to control the timing of the relaxation oscillator 452 and the data acquisition circuit 454._T, Np control circuit 442. The data acquisition circuit 454 calculates the temperature indication value (count) N_TAnd a pressure indication value (count) Np are stored in the temperature register 432 and the pressure register 434, respectively.
[0142]
The addressable memory block 438 includes an EEPROM array 436 (see 236) and several hardware registers 432, 434, and 435 (see 232 and 234). The EEPROM array 436 is programmed with various stored information described below.
[0143]
The last two rows of the EEPROM (eg, rows 14 and 15) comprise a trimming bit portion 436b that is programmed to store trimming information. The trimming information includes (a) controlling the reduction of the current supplied to the relaxation oscillator 452 (in the base-emitter voltage-current converter 450) to control the resolution of the pressure count and the temperature count, and (b) modulating The modulation depth in circuit 446 is set to optimize signal transmission for a given antenna device 410 and a given transponder application, (c) the mode of operation (active or passive) is set, and (d) the terminals of rectifier circuit 422. Optimize the impedance of the clamp at the intermediate voltage Vpp. Using the trimming line 485, the trimming information stored in the trimming bit section 436b can be transferred to a circuit (for example, 450, 446, 484, 422, 422, 422, 422, 422, 422, 422) at which the information operates at any time during the transponder operation (active or passive) 482) can be read directly. Alternatively, as in certain test and programming modes, controlled by test logic 484, the trimming bit portion 436b is accessed for external reading and writing (programming) along with the rest of the EEPROM 436 memory via the DATA connection pad. And can communicate with the transponder data line 444 via the column-data converter 441.
[0144]
As described above for the conventional model transponder 200, each of the temperature register 432 and the pressure register 434 is a count of a down counter (eg, a 12 bit clock) clocked by a sensor data signal of frequency Fosc ′ coming from the sensor interface unit 406. ) Hold.
[0145]
The parity and memory MTMS section 435 is also new to this transponder 400 model. It is implemented, for example, as a hardware register that stores 12-bit data. Five of the twelve bits are set permanently (value = 1) and then one bit that is set / cleared according to the open / closed (I / O) state of the MTMS switch (closed when exposed to too high temperature) Finally, there are six bits that temporarily store the parity of the pressure and temperature counts. Three of those parity bits are for the count Np of the pressure register 434, and the next three parity bits are for the temperature register 432 count Np._TIs for The parity bit indicates the Np down counter and N during acquisition of sensor data._TIt is constantly updated according to the changing count in the down counter (pressure register 434 and temperature register 432, respectively). Each parity bit represents the parity of four sequential bits ("nibbles") in the corresponding pressure register 434 or temperature register 432. The most significant bit represents the most significant nibble of the count, the middle parity bit represents the middle nibble of the count, and the least significant parity bit represents the least significant nibble of the count.
[0146]
The sensor interface unit 406 (see 206) of the transponder chip 402 includes a base-emitter voltage-current converter 450 (see) having a connection pad “Rext”, a relaxation oscillator 452 (see 252) having a connection pad “CP”, and , "MTMS" connection pad and a line 459 connecting it to the parity, the MTMS bit in the MTMS register 435 (comparable to 236 for the previous location of the MTMS bit).
[0147]
Base-emitter voltage-to-current converter 450 functions in a manner similar to converter 250 described in detail above with reference to portions 306-310 of FIG. To improve performance, the circuitry of portions 306, 308, 310 in the new base-emitter voltage-to-current converter 450 replaces a one-stage current mirror (eg, for transistors P1, P2, P4, P6). Utilizing cascode, the external resistor Rext 416 can have a different suitable resistance value, such as 500 kOhm. The reason cascodes are desirable is that they have a high power rejection ratio (PSSR). Also, in a feature described in more detail below with reference to FIG. 5, the final stage 510 (see 310) of the base-emitter voltage-to-current converter 450 is connected differently to the portion 308 to provide a trimming bit register section. Working in concert with the programmed setting at 436b, the reduced proportional-temperature current I (T) B on lines 451 and 455 (equivalent to the current I (T) / N on lines 251 and 311) is obtained. It is supplied to the relaxation oscillator 452 (see 252). In contrast to previous configurations, current scaling circuit 510 (see 310) can scale current I (T) by a variable scaling factor "B" determined by a programmed setting in trimming bit register section 436b. .
[0148]
The relaxation oscillator 452 includes a row decoder & N_T, Np under the control of the Np control circuit 442, operates in conjunction with the external capacitive pressure sensor Cp418 to operate the row decoder & N_T, Depending on the timing window determined by the Np control circuit 442, produces a signal at frequency Fosc 'indicative of the ambient temperature or pressure on line 453 (see 253). The data acquisition circuit 454 includes a row decoder & N_T, Under the timing control of the Np control circuit 442, the Fosc ′ signal to the appropriate hardware register (depending on the timing window), ie, to the temperature register 432 via line 455, and to the pressure register 434 via line 457. Send to
[0149]
The current scaling circuit 510 (part of the base-emitter voltage-to-current converter 450) and the relaxation oscillator 452 have several improvements over the relaxation oscillator 252, and the state of the MTMS 420 is different from that of the previous implementation (EEPROM 236). Are sent to a different memory location (register 435). Otherwise, the sensor interface unit 406 operates essentially the same as the corresponding part 206 in the previous model 3070C transponder 200.
[0150]
As described above with reference to the previous model 200, the term "ambient" refers to the parameter measured near the transponder 400, and more specifically, near each sensor associated with the transponder 400. Point. Also, the term "pressure", referred to herein as "pressure reading", "pressure count", "pressure response", "pressure register", etc., is measured by this transponder technique which actually produces a hybrid pressure-temperature reading. The term "pressure" refers to the overall pressure. When the hybrid indication is processed to remove the temperature component of the hybrid indication, the indication is referred to as a "pressure only" indication.
[0151]
Along with column decoder 440 and row decoder 442, column-to-data converter 441 communicates selected ones of the measured tire operating characteristics in the data stream to an external reader / interrogator (e.g., via antenna device 410). , 106), the order in which the signal (ie, data) is output via line 444 (equivalent to 244) to modulation circuit 446 (equivalent to 246), which is sent via interface and rectifier circuit 422 (see 222). Control. Line 444 also sends the data stream to test logic 484, where it can be directly accessed via the DATA connection pad.
[0152]
Modulation circuit 446 converts the data stream from line 444 into a sequence (modulation) of impedance changes representing it. Those changes are applied to the antenna device 410 via the interface and the rectifier circuit 422. A novel feature of the transponder 400 of the present invention is that the modulation index (modulation magnitude) is adjusted to match the operating power level, the individual transponder 400 in use, the antenna device 410, and the reader / interrogator 106. This is the ability to select a modulation index (via the trimming bit section 436b) that is optimized for.
[0153]
In passive mode operation, an RF carrier signal from an external source (eg, reader / interrogator 106) is received by antenna 412. This RF signal is rectified and used to power the RF transponder 400 and to provide a timing / clock signal. The modulation information added by the modulation circuit 446 is used to change the characteristics (for example, impedance, resonance frequency, etc.) of the LC tank circuit of the antenna device 410. These changes are detected and decoded by the external reader / interrogator 106 as a change in load, and the data is sent back from the RF transponder 400 to the external reader / interrogator 106. Because the passive power of transponder 400 is derived from the received RF signal, and the modulation of that signal is performed from some of that power, POR circuit 482 holds the reset signal during the supply of passive power and the transponder Does not clear its reset signal until it is sufficiently high to ensure stable operation of the transponder 400 during modulation (thus allowing modulation).
[0154]
Test logic 484 may be used to sort the wafer, assemble-level programming of the first board, programming in the pre-calibration stage, calibrating and trimming the transponder, and post-calibration to adjust trimming bits 436b for offset errors caused by encapsulation. Enables tests that can be performed at any stage of transponder manufacture and use, including programming.
[0155]
A more detailed description of the important parts of the RFIQ ™ transponder 400 of the present invention is provided in the following sections.
Memory allocation and data flow
Addressable memory block 438 is configured to provide an improved data flow over previous model 3070C transponder 200. Column decoder 440 and column-to-data converter 441 operate together to control the flow of data between addressable memory blocks 438. When operating in an active or passive mode (ie, not in a test or programming mode), the circuits 440, 441, 442 move the memory locations from the first address to the last (most significant in each data "word"). Accessing one at a time (words are ordered from least numbered word to most significant word, from bit to least significant) produces a serial data stream for transmission. From the following description, it will be seen that the rows to be included in the data stream are selected according to the mode of operation (ie, passive / active, or various test and programming modes). Row decoder & N_T, Np control circuit 442 stores a temperature-related count in temperature register 432 during one portion of the data transmission (e.g., during transmission of word / rows 2 through 6), and other data transmissions. To control the sensor interface circuit 406 (via line 487) to accumulate pressure related counts in the pressure register 434 (eg, during transmission of words / lines 9-13). . Their accumulation should both end by the time for the count to be read from registers 432 and 434 when their count portion of the data stream is valid (eg, words 14 and 15).
[0156]
FIG. 4B (comparable to FIG. 3C) is a "map" of addressable memory blocks, showing its physical organization (by "rows"), and showing the organization of data streams in active or passive mode. (By "word"). In the preferred embodiment of the transponder 400 described individually, each word or row is 12 bits (or column) long, with 16 words in the data, for a total of 192 bits (12 times 16). . The depiction of memory block 438 in FIG. 4A helps to understand the physical organization. It should be noted that rows 1 to 13 of the physical memory (EEPROM 436) correspond to words 1 to 13 of the data stream, except that rows 14 and 15 of the physical memory (EEPROM 436) are not part of the data stream. . Instead, the temperature register 432 is read as word 14 and the pressure register 434 is read as word 15 of the data stream. The last word of the data stream, word 16, is read from the hardware registers of physical memory (parity, MTMS register section 435).
[0157]
The EEPROM 436 portion of the example addressable memory block 438 has 180 cells arranged in an array of 12 columns × 15 rows. Each cell comprises at least one n-channel select gate and one corresponding EEPROM transistor. The first thirteen rows (sync, ID, calibration, CRC 436a) are readable in the normal read mode, whether the transponder 400 is programmed to be active or passive. The EEPROM memory locations in those first 13 rows are selected in the usual manner, one n-channel transistor row select gate for each EEPROM transistor. The EEPROM cells in row 14 and 15 (trimming bit section 436b) hold data for 12 bits of trimming information and are configured differently to facilitate its special role in the transponder. The EEPROM cell in each trimming bit section 436b has two gates instead of one, and is READ as in the test mode with transponder operation.　If the TRIM select gate is not enabled, a special READ is used so that the trimming bit 436b cannot be added to the data stream.　Add TRIM select gate in series with row select gate. Another feature of the special trimming bit register 436b is that each trimming bit 436b has an added detection line to signal the EEPROM bit setting (programmed trimming information) to the appropriate portion of the transponder 400 circuit as needed. It is also a thing.
[0158]
Thus, the logic associated with EEPROM 436 reads trim bit 436b sequentially as "READ" bits 14 and 15 in the data stream instead of temperature and pressure counts.　The TRIM test mode allows external reading. In the "read" test mode, as in normal operation, the trim bits are not seen in the data stream, but are still accessed by the sense amplifier to signal the programmed trim settings to the appropriate portions of transponder 400. In the "write" test mode, all EEPROM 436 cells (including trim bits 436b) are addressed and written simultaneously. This will actually write a "1" to all EEPROM cells, giving them a high threshold. In the "erase" test mode, individual cells can be erased (given a low negative threshold and programmed to "0"). As the clock CLK signal searches through the EEPROM cell array, the voltage at the VPP pad is raised to the programming temperature, allowing erasure by raising the DATA pad, thereby allowing the erased column to be addressed at the intersection of the addressed column and the addressed row. The cell is erased.
[0159]
Referring to FIG. 4B, the map shows an example of an organization in which the first twelve data locations (bits 000..011 in row 1) are reserved for synchronous ("sync") data. The next 48 data locations (bits 012.0.59 in rows 2 through 5) are for general information identifying individual transponders 400. As mentioned above, each transponder unit is properly calibrated before being mounted on the tire. The next 24 data locations (bits 060..083 in rows 6-7) hold the temperature calibration data (eg, decision point and slope). The next 24 data locations (bits 08.107 in rows 8-9) hold pressure calibration data (eg, decision point and slope). The next 36 data locations (bits 108... 143 in rows 10-12) hold additional identification information for IC chip 402. The next twelve data locations (in row 13) are the four bits of identification information for the chip calibration (bits 144 ... 147) and the eight bit value for the data validation algorithm such as CRC (cyclic redundancy check). (Bits 148..155). The next two words in the data stream (word 14..15, bits 156.179) are read from temperature register 432 and pressure register 434, respectively. The last word (word 16, bits 180 ... 191) of this data stream is read from the parity, MTMS register 435. The register contains five "open" bits (bits 180..184) and the logic level (state) of the MTMS switch 420. One subsequent bit (bit 185) and finally six bits (bits 186 ... 191), including three bits, each bit for the parity of the pressure count and the parity of the subsequent temperature count. In. The five open bits are not used and are fixed as logic "1" values.
[0160]
Column decoder 440, column-data converter 441, row decoder & N_T, Np controller 442 circuit coordinates addressing and access to addressable memory block 438.
[0161]
The column decoder 440 is constituted by a 4-bit synchronization counter for addressing 12 1 decoders. The output from column decoder 440 addresses 12 columns of the memory array during programming and reading. To address a column, an address n-channel device routes a low power current source to charge the addressed column. If a bit is to be written, the column is pulled high, but if it is to be erased, the column is pulled low. The output of the current source is buffered and drives a transponder data line 444 that leads through a column to a data converter 441, a modulation circuit 446, and a test logic circuit 484. The columns are sequentially addressed from column 1 to column 12 at the rate of the timing logic clock signal supplied by the timing and clock generator 424 or externally via the CLK pad. After addressing column 12, column decoder 440 provides row decoder & N_T, Np controller 442 to clock it to the next row and cycle itself back to column 1. After a reset signal (turned on and then turned off) is received from the power supply on reset circuit 482 (or via the RES pad), the column decoder 440 and the row decoder & N_T, Np controllers 442 reset their addressing order and start at row 1, column 1, ie, the first bit or cell of addressable memory block 438. Column decoder 440 serially addresses EEPROM array 436 during any of the read operation modes or during erase. In the write test mode, it has no effect since the entire EEPROM memory is simultaneously addressed for a write operation.
[0162]
Row decoder & N_T, Np controller 442 is a 4-bit synchronization counter that addresses 16 one decoders. The decoder addresses thirteen rows of data memory 436a from rows one through thirteen. Write test mode, erase test mode and READ#In TRIM test mode, it also addresses rows 14 and 15 of EEPROM memory 436b for trimming bits. During a normal read (active or passive mode), the decoder addresses the temperature hardware register 432 at row address 14 and the pressure hardware register 434 at row address 15. Regardless of the mode of operation, the row 16 address is sent to a special data row: parity, MTMS hardware register 435. In the read test mode and the erase test mode, each row is addressed sequentially on the falling edge of the last bit of the column decoder 440. In the passive mode or the active mode, the reset signal is applied to the row decoder & N_T, Np controller 442 is set to row 1. It is a synchronization word. In write mode, all rows are addressed simultaneously. It raises all select lines and control gates to voltage Vpp. As mentioned above, row decoder & N_T, Np control circuit 442 stores a count related to temperature in temperature register 432 during a specified time (data collection window) of temperature register 432 and pressure register 434, and stores a count related to pressure in pressure register 432. 434 is to control the sensor interface circuit 406 so as to accumulate it.
Programmable current scaling
As described above for the conventional model 3070C transponder 200, the scaled current I (T) / N} input to the relaxation oscillators 312, 252 via the lines 311, 251 is stored in the current scaling circuit 310 of FIG. Must be scaled by a fixed amount determined by the ratio of the physical areas of the current mirror transistors, such as P5, P6. The physical area is set during manufacture.
[0163]
Part of FIG. 5 shows an example of an improved (programmable) current scaling circuit 510 that is part of the base-emitter voltage-to-current converter 450 of the RFIQ ™ transponder 400 of the present invention. Current scaling circuit 510 (see 310) produces a scaled current I (T) B on line 511 (see 311). Here, the scaling factor "B" can be programmed into the transponder 400 via some of the trimming bits 436b at any time after the transponder 400 is manufactured. Current scaling is now a temperature measurement (eg, temperature scaling factor B_T= 1 or 1.5) and a pressure measurement (eg, pressure scaling factor B_P= 0.5 to 1 to 8.5), and the current supplied to the relaxation oscillator 512 (see 312) is proportional to 1.5 times the proportional-ambient temperature (PTAT) current I (T). To 8.5 can be programmably increased. By programming the current mirror in the current scaling circuit 510, the counts per degree or counts per PSI can be programmably maximized, thereby allowing temperature and pressure counts (N_TAnd N_P) To increase the degree of decomposition and stability. As mentioned above, it is particularly desirable to optimize the pressure count, and thus a wide range of pressure scaling factors B_PAre made available for programming (trimming). Narrower temperature scaling factor B_TIs generally suitable for compensating for process movement at the die level and changes in external precision resistance (Rext) 416. In a similar manner, the temperature scaling factor B_TAnd pressure scaling factor B_PIt is within the scope of the present invention to provide other suitable scaling factor values for
[0164]
Referring to FIG. 5, programmable current scaling circuit 510 is surrounded by a dashed line. Transistor P6 is a current mirror configuration, and transistors P1, P2, P4 in the temperature sensing and pressure mirror circuit (comparable to 306, 308 in FIG. 3, but only P4 and P5 are shown in FIG. 5) , So that their current mirror gates are connected to a line 505 (comparable to 305) that supplies a reference voltage Pbias to the gate of transistor P6. Due to the current mirror connection, the current flowing through transistors P4 and P5 mirrors the PTAT current I (T) flowing through external resistors Rext 416, 316. The voltage source of the programmable current scaling circuit 510 is the voltage Vcc provided on line 503 (comparable to 303). As described above, the voltage Vcc is adjusted and is almost independent of the temperature. The use of the analog power supply voltage Vcc is an improvement over the conventional use of the digital power supply voltage Vdd (circuit 310). The reason is that the voltage Vcc is "cleaner" and does not include the switching noise included in the digital voltage. For simplicity, transistors P4, P6, P6.05, P6.1, P6.2, P6.4 are shown as a single transistor. It is within the scope of the present invention to implement those transistors as cascodes, as described above for the transistors P1, P2, P4, etc. in circuits 306, 308 of FIG. Such cascode transistors used in a current mirror configuration have separate bias lines (eg, Pbias divided into Pbias' and Pbias ") connecting the gates of the transistors in each stage of the cascode current mirror.
[0165]
As in current scaling circuit 310, transistors P4 and P6 are connected in a current mirror configuration. However, instead of the transistors P5 and P6 of the circuit 310 having a fixed ratio of physical area to scale the current to be mirrored, the transistor P6 in the programmable circuit of the present invention is such that P6 is mirrored. It has a programmable ratio of physical area compared to transistor P4 (and transistors P1, P2 of circuits 306, 308). The physical area of transistor P6 is added in parallel with transistor P6, with the trimming bit 436b and the row decoder & N_T, N_PUnder the control of a control signal from the control circuit 442, use is switched by a switch. Each of the transistors P4, P6, P6.05, P6.1, P6.2, and P6.4 shown in FIG. 5 can be determined by the size of the transistors here, or combined together in parallel to form " It should be understood that there is a suitable area that can be the result of adding together the physical areas of multiple transistors, described as "single transistors." For example, a transistor P5 having a physical area of 1 (in arbitrary “units”) may be replaced by a single transistor having a physical area of 1.0 units, or a transistor 2 having a physical area of 0.5 units each. It can be manufactured as individual transistors. In the embodiment shown in FIG. 5, each of the transistors P4, P6, P6.05, P6.1, P6.2, and P6.4 is a relative physical, marked as "A =" n. Has an area. Here, n is a physical area in an arbitrary unit. The physical area of each transistor P4, P6, P6.1 is 1.0 unit, and the transistors P6.05, P6.2, and P6.4 are 0.5, 2.0, and 4.0, respectively. With area. It should be understood that the mirrored transistors P1, P2, P4 of FIG. 3 also have the same physical area as P6, so their physical area is 1.0 unit. By switching between various combinations of their added transistors P6.05, P6.1, P6.2, and P6.4 ("P6.n"), sets from 0.0 to 7.5 units The combined physical area can be added to the area of transistor P6 (1.0 unit) in 0.5 unit increments. As noted above, the ratio of the physical area of the mirrored transistor (eg, the ratio (P6 + P6.n) / 4) is determined by the mirrored current I (T) B emerging from the programmable current scaling circuit 510 on line 511. From the PTAT current I (T), increase by a factor B equal to the ratio of the total physical area.
[0166]
Each added transistor P6. n is one or more control solid state switches (scaling trim switches S5, S6, S7, S8, and S8), shown here as boxes with control inputs indicated by lines on one side. S9, a temperature scaling switch ST1, and a pressure scaling switch SP1). Two types of semiconductor switches, inverted and standard (non-inverted), are shown. Switches ST1 and SP1 are, for example, standard semiconductor switches that can be realized as N-channel transistors, and whose control signals are applied to their gates. Such a switch is conductive (switch "on" or "closed") when the control signal is high (voltage above ground, such as a logic "1", "true", or "set") and low (switch "on"). It does not conduct when ground, logic "0", "false" or "clear". Inverting switches S5, S6, S7, S8, and S9 are programmed "on" (closed) if the corresponding trim bits 5-9 are programmed as "cleared". Note that in the embodiment of the transponder 400 described herein, trim bits 5-9 are the only "low-true" logic bits in the trim bit 436b register. The advantage of using such P-channel transistor switches is that they function faster than N-channel transistor switches as the voltage level increases during transponder power up. Trim bit is the bit address in EEPROM trim bit register 437b that is communicated over line 485. In this example, trim bit 5 controls temperature count precision trimming, trim bit 6 controls pressure count precision trimming, and trim bits 7, 8, 9 control pressure count total trimming. The setting of scaling / trim bit “n” is “TRIMBIT”　N (ie, scaling trim bit signal TRIMBIT for scaling trim bits 5 through 9)　5. . TRIMBIT　9). Non-inverting scaling switches ST1 and SP1 are row decoder & N_T, N_PSignal CAPTURE coming on line 847 from controller 442　NT and CAPTURE　On / off control is performed by the NP. The scaling switches ST1 and SP1 apply the various scaled currents I (T) B during the temperature measurement period or the pressure measurement period (respectively the time window W_TAnd W_P) Is enabled by the relaxation oscillators 452 and 512. Therefore, the value of the current scaling coefficient B is determined by the temperature measurement window W_TMedium (signal CAPTURE　NT is on) is B_T(1 or 1.5) and the pressure measurement window W_PMedium (signal CAPTURE　NP is on) is B_P(1 to 8.5 in 0.5 increments). FIG. 5 shows a series arrangement of switches (eg, S5 and ST1), but using a digital logic to avoid the cumulative switching losses of the I (T) B signal and combining the control signals to form a single switch. Controlling is within the scope of the present invention. For example, an AND gate, whose output controls switch ST1, causes a CAPTURE　TRIMBIT with NT signal inverted　By combining the five signals, the switch S5 can be eliminated.
Relaxation oscillator
As described above for the previous model 3070C of the transponder 200, the relaxation oscillators 312, 252 provide a capacitance C for temperature reading._FX1And C_FX2Or capacity C for pressure reading_FX1And (C_FX2Plus C_P) And a frequency Fosc 'determined by the alternate charging of the two capacitors. The rate of charging these capacitances is determined by the magnitude of the capacitance and the magnitude of the scaled current I (T) / N used to charge them. The capacity discharge rate is not a factor. The reason for this is that each capacitor discharges faster than the other because the other capacitor discharges while charging and the resistance of the discharge path is low. The set-reset portion of the relaxation oscillator 312 flip-flop triggered only when one of the capacitors has charged to a level just above the bandgap voltage Vdg.
[0167]
Part of FIG. 5 shows an example of an improved relaxation oscillator circuit that substantially comprises the relaxation oscillator 452 of the RFIQ ™ transponder 400 of the present invention.
[0168]
Circuit 512 is still a relaxation oscillator and is driven by the scaled current (measured current) provided on line 511 from the now programmable current scaling circuit 510 (see 310). The set-reset unit 514 (see 314a and 314b) of the relaxation oscillator 512, having phase paths φ1 and φ2 (phase 1 path 514a and phase 2 path 514b, respectively), functions in the same general manner, but with the comparator 516 ( 516a, 516b, 316a, 316b) are slightly modified in a known manner to allow for input changes. A major circuit change is for the front end, switching device, which is divided into two measurement switching circuits 515a and 515b, respectively, for temperature measurement and pressure measurement. Capacitor C_FX2Has been lost, the capacitor C is now used during the temperature measurement period._FX(C_FX1) Is used and during the pressure measurement period the capacitor C_P(See 418, 218) only. During one measurement period, the frequency output signal frequency Fosc 'is now_FXOr C_PIs determined by alternate charging and discharging of Those capacitors are row decoder & N_T, N_PSelected by controller 442. Since comparators 516a and 516b have reference voltages of, for example, Vbg and Vbg / 2 (bandgap voltage and half the bandgap voltage), the selected capacitor C_FXOr C_PUses the (scaled) measured current I (T) B to be charged to just above the voltage level Vbg. It triggers the set-reset circuit 514 to flip it from its first state (eg, PHASE1 = true) to a second state (eg, PHASE2 = true). It turns off the charging current I (T) B, and instead reduces the scaled current I (T) B by a mirrored current equal to I (T) B, to the selected capacitor C (T) B._FXOr C_PFrom the voltage level Vbg to the voltage level Vbg / 2 or less. It triggers the set-reset circuit 514 to flip-flop again to its first state, and to select the selected capacitor C_FXOr C_PRestart charging. After the first charge from ground (zero volts) to Vbg / 2, the described operation is performed with the scaled current I (T) B and the selected capacitor C_FX(For temperature measurement) or C_PA uniform 50% duty cycle waveform with a frequency determined by the magnitude (for pressure measurement). Another novel feature of the relaxation oscillator 512, 452 of the present invention is that the capacitor C is not used for measurement use._FXOr C_PTo supply a small bias current Ibias, which is mirrored to be gradually discharged to ground, and to ground those capacitors when not in use, the set-reset circuit 514 It is also used to set the circuit to a defined state before each time used for measurement.
[0169]
An example of the relaxation oscillator circuit 512 has two analog inputs, a (programmably) scaled PTAT current I (T) B provided on line 511 and a bias current Ibias provided on line 513. Bias current Ibias is a small fraction (eg, I (T) / 10) of PTAT current I (T) obtained in a known manner, such as the current scaling procedure described above.
[0170]
An example of the relaxation oscillator circuit 512 is a row decoder & N_T, N_PTwo digital (logic or control) inputs provided on line 487 from controller 442, ie, CAPTURE to determine respectively whether temperature or pressure is being measured.　NT signal and CAPTURE　NP signal. The control signal input is indicated in several places by its corresponding number. For example, CFX　TO　The OSC signal is indicated by the numeral 1 in the circle, from which it exits the logic circuit 531 and from there is input to the switch ST2. The outputs of the phase paths φ1 and φ2 are used as control signals PHASE1 and PHASE2, respectively, and are inverted and used as control signals. That is, the phase path φ1 signal is inverted by the inverter 520a and
[0171]
[Outside 1]

[0172]
And the phase path φ2 signal is inverted by the inverter 520b and
[0173]
[Outside 2]

[0174]
Becomes Here, the line shown above the name indicates that the logic is inverted, that is, "NOT".
[0175]
[Outside 3]

[0176]
Is "not phase 1" (ie, if the PHASE1 signal is high, ie, at logic level 1,
[0177]
[Outside 4]

[0178]
The signal is low, meaning logic level 0). PHASE1, PHASE2,
[0179]
[Outside 5]

[0180]
,
[0181]
[Outside 6]

[0182]
The signal is CAPTURE as an input to logic circuit 531.　NT signal and CAPTURE　Combined with the NP signal. The logic circuit has six control signals-two capacitor selection signals, ie, CFX　TO　OSC (# 1) and CP　TO　OSC (# 2) and four measurement control signals PHASE1　NT (# 3), PHASE2　NT (# 4), PHASE1　NP (# 5) and PHASE2　NP (# 6) is output.
[0183]
The details of the digital signal included in the logic 531 are described in the logic circuit example 531a. . 531f (for signals (# 1)... (# 6) respectively). Each of the logic circuits 531a. . 531f utilizes a NAND gate and one inverter to generate its control signal to be output from logic circuit 531. Signal (# 1) that behaves as described here. . It is within the scope of the present invention to include another form of logic circuit 531 that generates (# 6). The output signals (# 1) and (# 2) have one parameter (N_TOr N_P) Is measured, and other parameters (N_POr N_T) Is generally understood to mean that it is not time to measure. Note that these signals are also true (logic 1) whenever the opposite parameter is not measured, regardless of the setting of the "capture ..." signal. Thus, the signal CFX　TO　OSC (# 1) is CAPTURE　True whenever NP is false, the signal CP　TO　OSC (# 2) is CAPTURE　True whenever NT is false. Since the (# 1) and (# 2) signals control switches ST2 and SP2, respectively, this logic states that when the reverse capacitor CP / CFX is not connected to the set-reset circuit 514, the switch ST2 / SP2 has a capacitor. CFX / CP is always connected to the set-reset circuit 514. This helps to keep the set-reset circuit 514 in a defined state.
[0184]
Each output signal (# 3) to (# 6), when true, indicates which phase (phase 1 or phase 2) the oscillator is in and which capacitor (C for temperature)_FXOr C for pressure_P), Generally indicates the state of the signal, which, if any, primarily determines whether it is in use for measurement. As can be seen in the example of NAND gate logic shown in FIG. 5A, each output signal (# 3) through (# 6) has a phase and phase signal when the opposite measurement is selected or even if no measurement is selected. Is always true again, regardless. For example, the NAND gate in the logic circuit 531c has the signal PHASE1　NT (# 3) is output. Assuming that the temperature is being measured, PHASE1　NT is phase 1 (C_FXHigh during charging, phase 2 (C_FXLow during discharge). However, if the temperature is not measured (ie, CAPTURE　NT is low, not true), PHASE1 even if temperature and pressure are not measured, regardless of the phase of the oscillator.　NT is high. Similar logic is indicated by the circled numbers # 3, # 4, # 5, and # 6 and four control signals (# 3) shown in FIG. 5A as logic circuits 531c through 531f. Or (# 6).
[0185]
At line 521 (same as 453, see 321 and 253)
[0186]
[Outside 7]

[0187]
The signal is the (inverted) relaxation oscillator output signal Fosc 'that is used by the transponder 400 to accumulate the temperature and pressure counts in the corresponding registers 432 and 434, respectively. It can be seen that the relaxation oscillator output signal Fosc 'is buffered and the level can be shifted to a digital logic signal level. The relaxation oscillator 512 puts the transponder 400 in the (active mode) test state, “READ　TEMP ”and“ READ ”　PRESS ", or can be read directly. These test conditions enable the relaxation oscillator 512 for temperature and pressure, respectively, and send the relaxation oscillator output signal Fosc 'to the DATA pad.
[0188]
The front end of the relaxation oscillator 512 is comprised of two substantially identical measurement switching circuits 515a and 515b. The circuits are temperature measuring capacitors C_FXOr pressure measuring capacitor C_PAnd a temperature measuring series switch ST2 or a pressure measuring series switch SP2 connecting an appropriate front end switching circuit 515 to a set-reset circuit 514. As described above for switches ST1 and SP1, switches ST2 and SP2 are standard (non-inverting) semiconductor switches such as switches that can be implemented using N-channel transistors or implemented in CMOS. The capacitor in chip 402 is a poly-poly capacitor, which has a very low temperature coefficient and an area (capacitance) that provides high resolution, but has low sensitivity to processing factors such as etching errors and alignment errors. Temperature measurement capacitor C_FXIs, for example, a 6pf (+/- 10%) poly-poly capacitor. Pressure measuring capacitor C_P418 (see 218) is off-chip and preferably has robust and stable characteristics similar to those described above (e.g., a contact mode with a linear capacity versus absolute pressure response and varying between 4 and 40 pf). Capacitive pressure sensor).
[0189]
The temperature measurement switching circuit 515a includes, for example, P-channel transistors P9, P10, P13, N-channel transistors N9, N10, N13, N14, an inverter 517a, a semiconductor switch ST2 (such as an N-channel CMOS transistor), and a temperature measurement capacitor. C_FXIt is composed of The pressure measurement switching circuit 515b includes, for example, P-channel transistors P11, P12, P14, N-channel transistors N11, N12, N15, N16, an inverter 517b, a semiconductor switch SP2, and a pressure measurement capacitor C_P(418, connected outside of the transponder chip 402 via the connection pad Cp). The various control inputs described above are connected to the measurement switching circuit 515 as indicated by the circled numeral 1 and the current Ibias is also input as shown.
[0190]
In the following, the temperature measurement switching circuit 515a will be described to illustrate the techniques involved. The pressure measurement switching circuit 515b functions in a suitable parallel fashion.
[0191]
When the transponder 400 places the relaxation oscillator 512 in the temperature measurement mode, the row decoder & N_T, N_PThe control circuit 442 controls the temperature measurement window W_TTime it chooses as_TCAPTURE for　Set NT high (and CAPTURE　NP is set low), thereby closing switch ST2 (and keeping switch SP2 open) via logic circuits 531a and 531b. The set-reset circuit 514 starts operation at phase 1 (ie, the output of φ1, 514 is high and the output of φ2, phase 2, 514b is low). The reason is that when the relaxation oscillator is not used for measurement, the measurement switching circuit 515 is always at a low voltage (as described below), and this low voltage is, as described above, the signal # 1, # 2 Is supplied to the comparators 516a and 516b according to the logic for this. Since signal PHASE1 = true and PHASE2 = false, logic circuit 531 outputs the next signal, identified by the number in the circle, ie, PHASE1.　NT (# 3), PHASE1　NP (# 5), PHASE1　NP (# 6) = true, PHASE1　NT (# 4) = false. These signals turn off transistor P13 and turn on P9, thereby causing the scaled current I (T) B to pass through P9 through capacitor C_FXCharge. Since the switch ST2 is closed by the signal (# 1), the set-reset circuit 514 detects the rising voltage of the capacitor CFX. At the same time, transistor N14 is kept on, grounding the gate of transistor N9 and charging capacitor C_FXKeep it off so that there is no current leakage path. Since transistors P10 and N13 are also kept off, there is no other input to transistor N9 and the state of transistor N10 does not matter.
[0192]
When the set-reset circuit 514 flip-flops to phase 2 (PHASE1 = false, PHASE2 = true), the signal PHASE1　NT (# 3) changes to false and PHASE2　NT (# 4) is true (but PHASE1　NP (# 5) and PHASE2　NP (# 6) remains "true"). These signals send the scaled current I (T) B to transistor N10 by turning on transistor P13 and turning off P9. Since transistor N13 is turned on and transistor N14 is turned off, transistor pair N10 / N9 now forms a current mirror, so that current flowing through transistor N10 to ground flows through transistor N9 to ground. Capacitor C_FXIs mirrored by an equal current discharging. An unintended side effect of connecting the current Ibias through the now turned on control transistor P10 is the sum of the scaled current I (T) B and the bias current Ibias, the current flowing through the transistors N10 / N9, Thereby, the capacitor C_FXThe charging rate is slightly higher than the charging rate. Since the bias current Ibias is relatively small, this changes the duty cycle of the relaxation oscillator 512 to slightly deviate from 50% -50%, but does not affect the counting process. (Compare transponder 400 of the previous model, described above, which is intentionally operated with a symmetric duty cycle.) Set-reset circuit 514 because switch ST2 is still closed by signal (# 1). Is the capacitor C_FXCan be detected.
[0193]
Temperature measurement window W_TIs closed (transponder 400 is no longer in temperature measurement mode), row decoder & N_T, N_PThe control circuit 442 stops acquiring data in the temperature register 432. In this regard, CAPTURE　Since the NT signal is still low, switch ST2 remains closed (signal CFX).　TO　OSC). CAPTURE　The NP signal is set high (and CAPTURE　If the NT signal is low), the switch ST2 only opens at other times, i.e. the pressure is measured and the temperature is not measured. CAPTURE　When the NT signal is low, the logic circuit 531 outputs the signal PHASE1　NT (# 3) and PHASE2　Make NT (# 4) true. These signals shut off the flow of the scaled current I (T) B through the temperature measurement switching circuit 515a by turning off the transistors P13 and P9. However, since transistor N13 is turned on and transistor N14 is turned off, transistor pair N10 / N9 still forms a current mirror, and bias current Ibias flowing through transistor N10 to ground since bias control transistor P10 is on. Flows to the ground through the transistor N9 and the capacitor C_FXIs mirrored by an equal current discharging. Since the (small) bias current Ibias is the only current allowed to flow through the mirror circuits N10 / N9, the capacitor is gradually discharged to ground (0 volts) and CAPTURE　By setting the NT control signal high (true), the next temperature measurement window W_TIs held there until it is opened. Since the switch ST2 is still closed, the grounded inputs to the comparators 516a and 516b cause the set-reset circuit 514 to stop oscillating in the prescribed state of PHASE1 = high, PHASE2 = low. Temperature count N_TOr pressure count N_PMeasurement capacitor C to accumulate_FXAnd C_PBefore each use of the measuring capacitor C_FXAnd C_PIt is a feature of the present invention to provide a means as disclosed herein that utilizes a small bias current Ibias to reset to a defined state (zero voltage). Another feature of the present invention is to use the means described herein to further place the set-reset circuit 514 in a defined state before each use of the set-reset circuit 514.
[0194]
The above illustrative example illustrates the various portions of the transponder 400 circuit, for example, at the base-emitter voltage to the current converter 450 (current scaling circuit 510), at the relaxation oscillator 452 (512), and at the data acquisition circuit 454 (FIG. 3A), the control signal CAPTURE for various purposes.　NT and CAPTURE　Use NP. It is within the scope of the invention to change the timing of the control signals to each of those circuits 450, 452, and 454 so that the events occur in the proper order. For example, the circuit 454 enables the time of the relaxation oscillator 512 to operate stably, and the measurement capacitor C_FXAnd C_PCan be delayed (e.g., a delay of 1 bit width of 256 [mu] sec) so that oscillation occurs between the voltages Vbg / 2 and Vbg.
[0195]
As noted above for typical passive mode operation of transponder 400, row decoder & N_T, N_PControl circuit 442 provides a control signal to sensor interface circuit 406 (via line 487) to control the temperature during one portion of the data transmission (eg, while transmitting words / rows 2 through 6). The associated count is accumulated in temperature register 432 and the count associated with pressure is accumulated in pressure register 434 during another portion of the data transmission (eg, while transmitting words / rows 9-13). To achieve delayed turn-on for data acquisition circuit 454, row decoder & N_T, N_PControl circuit 442, for example, may clock CAPTURE when clocking to bit 1 of row 2.　The NT signal can be sent to the programmable current scaling circuit 510 and the relaxation oscillator circuit 512, after which a different “Capture @ Temp” signal can be sent to the data acquisition circuit 454 when clocking to bit 2 of row 2. it can. Similarly, CAPTURE　The NT signal and the "Capture @ Temp" signal can be sent at different times or together (e.g., when clocking to bit 1 of row 7).
[0196]
While the invention has been illustrated and described in the drawings and foregoing description, it should be understood that it is by way of example only and not limitation of the invention, but is shown and described of only the preferred embodiments. It is desirable that all modifications fall within the spirit of the invention and be protected. Undoubtedly, many other "changes" to the above "subject" will occur to those skilled in the art to which the invention pertains most closely, and those changes will not be It is intended to be included within the scope of.
[Brief description of the drawings]
FIG.
1 is a generalized view of an RF transponder device of the present invention with an external reader / interrogator and an RF transponder in a pneumatic tire.
FIG. 2
FIG. 3 is a block diagram of the main components of an RF transponder of an earlier model of the present invention.
FIG. 3
FIG. 3 is a circuit diagram of the main part of the RF transponder of FIG. 2 in a previous model of the invention.
FIG. 3A
FIG. 3 is a circuit diagram of a portion of the RF transponder of FIG. 2 of a previous model of the present invention.
FIG. 3B
FIG. 3 is a circuit diagram of a portion of the RF transponder of FIG. 2 of a previous model of the present invention.
FIG. 3C
FIG. 3 is a diagram in the memory space of the RF transponder of FIG. 2 showing how data is arranged and transmitted according to a previous model of the present invention.
FIG. 3D
FIG. 3 is a graph of transponder indication and transponder power of the RF transponder of FIG. 2 for a previous model of the present invention.
FIG. 4A
It is a block diagram of the main components of the RF transponder of the present invention.
FIG. 4B
FIG. 4B is a diagram in the memory space of the RF transponder of FIG. 4A showing how data is arranged and transmitted according to a previous model of the present invention.
FIG. 5
FIG. 4B is a circuit diagram of a current scaling unit and a relaxation oscillator unit of the RF transponder of FIG. 4A according to the present invention.
FIG. 5A
FIG. 6 is a circuit diagram of a logic unit of the relaxation oscillator unit of FIG. 5 according to the present invention.

Claims (25)

A measurement switching circuit (315, 515a, 515b), a set-reset circuit (314, 514) having a first phase path (314a, 514a) and a second phase path (314b, 514b), and an output (Fosc '). ), Which supply measurement currents (I (T), I (T) / N, I (T) B) to the relaxation oscillators (252, 312, 452, 512). RF transponders (102, 200, 400)
A logic circuit (531);
Measuring capacitors (C _FX , C _P ) arranged in the measuring switching circuit;
A first transistor (P9, P11) controlled by the logic circuit to supply a measurement current for charging the measurement capacitor to the measurement capacitor;
A current mirror (N10, N12) having transistors (P13, N13, P14, N15) controlled by the logic circuit, copying the measurement current in the second transistor (N9, N11) and charging the measurement capacitor; With
The logic circuit includes a measurement control signal (PHASE1 　 NT, PHASE2 　 NT, PHASE1 　 NP, PHASE2 　 NP) and signals (PHASE1, PHASE2) indicating the phase of the set-reset circuit are output from a signal (CAPTURE) from the timing generator (442). 　 NT, CAPTURE 　 An RF transponder (102, 200, 400) characterized in that, in combination with NP), the alternating charging and discharging of the measuring capacitor is controlled according to the phase of the set-reset circuit. The measured current indicates the measured parameter,
A measurement capacitor having a fixed capacitance value, the relaxation oscillator output being a signal having a frequency indicative of the measured parameter;
The RF transponder according to claim 1, wherein: 3. The RF transponder according to claim 2, wherein the measured parameter is temperature. The measurement capacitor is a signal having a varying capacitance value indicating the measured parameter, and the relaxation oscillator output having a frequency indicating the measured parameter;
The RF transponder according to claim 1, wherein: Indicating a first parameter at which the measured current was measured;
The measurement capacitor has a varying capacitance value indicative of a second measured parameter, and the relaxation oscillator output is a signal having a frequency indicative of the first measured parameter and the second measured parameter;
The RF transponder according to claim 1, wherein: The RF transponder according to claim 5, wherein the first measured parameter is temperature and the second measured parameter is pressure. A programmable current scaling circuit (510) inserted between the measurement current source and the relaxation oscillator to scale (I (T) B) the measurement current supplied to the relaxation oscillator. The RF transponder according to claim 1. Programmable current scaling circuit
At least one scaling trim bit (TRIMBIT) 　 5. . TRIMBIT 　 9) At least one scaling trim switch (S5, S6, S7, S8, S9) controlled by the setting of 9), wherein each scaling trim switch has at least one scaling trim switch to control the scaling of the measured current. 8. The RF transponder according to claim 7, wherein the RF transponder is connected in series to the transistor (P6.05, P6.1, P6.2, P6.4). Programmable current scaling circuit
The RF transponder according to claim 7, further comprising a power supply (Vcc) that is an analog voltage source. A first in the first phase path having a negative input terminal connected to a first fixed reference voltage (Vbg) and a positive input terminal connected to the charged side of the measurement capacitor. A comparator (516a),
A positive input terminal connected to a second fixed reference voltage (Vbg / 2) lower than the first fixed reference voltage, and a negative input terminal connected to the charged side of the measuring capacitor A second comparator (516b) in a second phase path having:
The RF transponder according to claim 1, further comprising: 2. The RF transponder according to claim 1, further comprising a bias control transistor (P10, P12) controlled by a logic circuit for connecting a small bias current (Ibias) to a current mirror for discharging the measuring capacitor. . A relaxation oscillator (312, 252, 512) comprising a set-reset circuit (314, 514) having a first phase path (314a, 514a) and a second phase path (314b, 514b) and an output (Fosc '). , 452) to the RF transponder (102, 200, 400) comprising a measurement current source (306, 308) for supplying a measurement current (I (T), I (T) / N, I (T) B). ,
A first measurement switching circuit having an output (515a) and a second measurement switching circuit having an output (515b);
A logic circuit (531);
A first measurement capacitor (C _FX ) located at the output of the first measurement switching circuit (515a);
A first switch (ST2) controlled by a logic circuit and connecting an output of the first measurement switching circuit to a set-reset circuit;
A second measurement capacitor (C _P ) located at the output of the second measurement switching circuit (515b);
A second switch (SP2) controlled by the logic circuit and connecting the output of the second measurement switching circuit to the set-reset circuit;
A first transistor (P9), controlled by a logic circuit, for supplying a measurement current for charging the first measurement capacitor to the first measurement capacitor;
A first current mirror (N10) having a transistor (P13, N13) controlled by a logic circuit, copying a measurement current in the second transistor (N9) and charging a first measurement capacitor;
A third transistor (P11), controlled by a logic circuit, for supplying a measurement current for charging the second measurement capacitor to the second measurement capacitor;
A second current mirror (N12) that has a transistor (P14, N15) controlled by a logic circuit, captures a measurement current in the fourth transistor (N11), and charges a second measurement capacitor. ,
The logic circuit applies a signal (CAPTURE) from the timing generator (442) to signals (PHASE1, PHASE2) indicating the phase of the set-reset circuit. 　 NT, CAPTURE 　 NP) to control the alternating charging and discharging of the first measuring capacitor according to the phase of the set-reset circuit (PHASE1). 　 NT, PHASE2 　 NT),
The logic circuit applies a signal (CAPTURE) from the timing generator (442) to signals (PHASE1, PHASE2) indicating the phase of the set-reset circuit. 　 NT, CAPTURE 　 NP) in combination with a second measurement control signal (PHASE1) for controlling the alternating charging and discharging of the second measuring capacitor according to the phase of the set-reset circuit. 　 NP, PHASE2 　 NP),
The logic circuit receives a signal (CAPTURE) from the timing generator (442). 　 NT, CAPTURE 　 NP) in combination with a capacitor selection signal (CFX) for controlling the opening and closing of the first switch and the second switch. 　 TO 　 OSC, CP 　 TO 　 An RF transponder (102, 200, 400), characterized in that the transponder (OSC) is supplied. The measured current indicates the measured parameter,
The first measurement capacitor has a fixed capacitance value, and the relaxation oscillator output has a frequency indicative of the first measured parameter when the first switch is closed and the second switch is open. Signal
A second measurement capacitor has a varying capacitance value indicative of a second measured parameter, and the relaxation oscillator output has a first capacitance when the first switch is open and the second switch is closed. A signal having a frequency indicative of the measured parameter and the second measured parameter.
The RF transponder according to claim 12, wherein: 14. The RF transponder of claim 13, wherein the first measured parameter is temperature and the second measured parameter is pressure. A programmable current scaling circuit (510) inserted between the measurement current source and the relaxation oscillator to scale (I (T) B) the measurement current supplied to the relaxation oscillator. The RF transponder according to claim 12, wherein Programmable current scaling circuit
At least one temperature scaling trim bit (TRIMBIT) 　 5) At least one temperature scaling trim switch (S5) controlled by the setting of 5), each connected in series with at least one scaling transistor (P6.05, P6.1, P6.2, P6.4). When,
The acquired temperature signal (CAPTURE) from the timing generator 　 NT), each connected in series with at least one of the at least one scaling transistor to form one of the at least one temperature scaling trim switch and a corresponding one of the at least one temperature scaling switch. At least one temperature scaling switch (ST1) for controlling the scaling of the measured current when closed;
At least one pressure scaling trim bit (TRIMBIT) 　 6, TRIMBIT 　 7, TRIMBIT 　 8, TRIMBIT 　 9) at least one pressure scaling trim switch (S6, S7, S8, S9) each controlled by the setting of 9), each connected in series with at least one scaling transistor;
Acquired pressure signal (CAPTURE) from timing generator 　 NP), each connected in series with at least one of the at least one scaling transistor to close at least one pressure scaling trim switch and at least one pressure scaling switch; At least one pressure scaling switch (SP1) for controlling the scaling of the measured current;
16. The RF transponder according to claim 15, comprising: Programmable current scaling circuit
Each one of the at least one temperature scaling trim switch forming a single switch for each one of the at least one temperature scaling trim switch controlled by the at least one temperature scaling trim bit and the acquired temperature signal. One and at least one of the temperature scaling switches;
Each one of the at least one pressure scaling trim switch forming a single switch for each one of the at least one pressure scaling trim switch controlled by the at least one pressure scaling trim bit and the acquired pressure signal. One and at least one of the pressure scaling switches;
The RF transponder according to claim 16, further comprising: Programmable current scaling circuit
Power supply (Vcc) which is an analog power supply,
16. The RF transponder according to claim 15, comprising: A negative input terminal connected to a first fixed reference voltage (Vbg); and charging of the first and second measuring capacitors selected by the first and second switches. A first comparator (516a) in a first phase path having a positive input terminal connected to the side to be connected;
A positive input terminal connected to a second fixed reference voltage (Vbg / 2) that is lower than the first fixed reference voltage, and a positive input terminal selected by the first switch and the second switch; A second comparator (516b) in a second phase path having a first measuring capacitor and a negative input terminal connected to the charged side of the second measuring capacitor;
13. The RF transponder according to claim 12, comprising: A first bias control transistor (P10), controlled by a logic circuit, for connecting a small bias current (Ibias) to the first current mirror to discharge the first measurement capacitor;
A second bias control transistor (P12), controlled by a logic circuit, for connecting a small bias current (Ibias) to the second current mirror to discharge the second measurement capacitor. An RF transponder according to claim 12. The RF transponder (102, 200, 400) generating an oscillation measurement signal (Fosc ') at 400)
Supplying a measurement current (I (T), I (T) / N, I (T) B) to at least one measurement switching circuit;
Providing a measuring capacitor (C _FX , C _P ) in each one of the at least one measuring switching circuit (515a, 515b);
Alternately supplying a measuring current to the measuring capacitor to charge the measuring capacitor and supplying a measuring current copied to the measuring capacitor to discharge the measuring capacitor;
A method for generating an oscillation measurement signal (Fosc ') in an RF transponder (102, 200, 400), comprising: Comprising scaling the measured current (I (T) B), wherein the current scaling factor (B) is set to a programmed setting (TRIMBIT). 　 5. . TRIMBIT 　 9) the type of measured parameter (T, is determined by the P), whereby the indicated value (N T of measured _parameters, to adjust the frequency of the oscillating measurement signal for optimizing the _{N P)} 22. The method of claim 21, wherein: By imaging a small bias current (Ibias) to discharge the measurement capacitor while at least one measurement switching circuit is not being used for measurement, each of the at least one measurement switching circuit is used for measurement. 22. The method according to claim 21, wherein when not in effect, the measuring capacitors of each of the at least one measuring switching circuit are placed in a defined state. Sending the output of at least one measurement switching circuit to a set-reset circuit (514);
A defined state when the set-reset circuit is not used for measurement by grounding the output terminal of at least one measurement switching circuit while the set-reset circuit is not used for measurement 22. The method of claim 21, wherein The reading of the measured parameter is started by adding a delay between the time the relaxation oscillator starts and the time the oscillation measurement signal is sent to the measurement counter / register (230, 232, 234, 432, 434). 22. The method according to claim 21, wherein the relaxation oscillator is allowed to settle before starting.

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