JP2003535552A - Ultra wideband spread spectrum communication method and system - Google Patents

Abstract

(57) Abstract: An ultra-wideband, high-speed, spread-spectrum communication system that transmits information through objects such as walls and ground using short wavelets (103, 104) of electromagnetic energy. The communication system encodes data on the RF signal using a baseband code formed from time shifted and inverted wavelets (103, 104). Typical wavelet pulse widths are on the order of 100 to 1000 picoseconds and bandwidths are about 8 GHz to 1 GHz. Uses a combination of short width wavelets and encoding techniques to ensure that energy is not concentrated in a particular energy band (eg, VHF (30-300 MHz) or UHF (300-1000 MHz)) and is not detected by conventional narrowband receivers. Spreads the signal energy over a very wide frequency band and does not interfere with these communication systems.

Description

Detailed Description of the Invention

(Cross Reference of Related Application) In this specification, "Ultra Wide Bandwidth Spread-spectrum Communications System" dated December 11, 1998 is referred to.
Co-pending U.S. patent application Ser. No. 09 / 209,460, Docket No. 10188-0006-8 dated May 3, 2000, entitled "Planar Ultra Wide Band Antenna with Highly Integrated Electronic Circuits ( Planar Ultra Wide Band Antenna With In
co-pending US patent application Ser. No. 09 /
563, 292 and Docket No. 192408US8PROV 200
"Ultra Wide Band Spread Spectrum Communication System (Ultra Wide Ban
dwidth Spread-spectrum Communications System) ", which includes the subject matter disclosed in co-owned co-pending U.S. Provisional Patent Application No. TBD,
All of these respective contents are hereby incorporated by reference.

FIELD OF THE INVENTION The present invention relates to wireless communication transmitters, receivers, systems, and methods thereof that use wireless digital communication with ultra-wide band (UWB) signaling technology.

BACKGROUND OF THE INVENTION There are many wireless communication technologies for digital data, and recently,
Wireless digital communication has been applied to applications such as mobile phone systems, pagers, remote data collection, and computer wireless networks. The book on wireless digital communications includes the Prentice Hall, Englewood Cliffs, New Jersey, New Jersey.
Kamilo Feher, published by entice-Hall, "Wireless Digital Communications, Modulation, and Spread Spectrum Applications (Wireless Digital
Communications, Modulation & Spread Spectrum Applications ”and Bernard Sklar,“ Digital Communications: Fundamentals and Applications (Digital Com
mnications Fundamentals and Applications) (ISBN 0-13-211939-1), the entire contents of both of which are hereby incorporated by reference. In these books, for example, conventional carrier modulation (ie, FSK), such as phase or frequency modulation.
, MSK, GMSK, BPSK, DBPSK, QPSK, O-QPSK, FQP
It mainly deals with SK, π / 4-DEQPSK, and pulse position modulation (PPM) (for reference, π / 4-DEQPSK is used in the US and Japanese cellular standards).

In these systems, the allocated bandwidth is conventionally shared among a plurality of users by time division multiple access (TMDA) or code division multiple access (CDMA), and FHSS (frequency hopping), which is a spread spectrum technique, is used. Spread spectrum) or CDMA code (direct spread method) is used to spread the spectrum. This "spread spectrum" is a technique that provides a method for sharing bandwidth among multiple users and provides a robust signal that is relatively resistant to background noise.

Spread spectrum techniques improve the robustness of a transmitted signal by a certain amount of repetition within the signal related to the data stored in the signal. This redundancy is often described in terms of "chips" per data bit, but such conventional spread spectrum systems, codes, and techniques are described in 1988 by Computer Science Press. By David L. Nicholson, published by The Company, "Signal Design LPE for Spread Spectrum"
And AJ Systems (Spread Spectrum Signal Design LPE and AJ Systems) "(
ISBN 0-88175-102-2), the entire contents of which are hereby incorporated by reference.

In such a spread spectrum signal, the information division of the chip is realized by a predetermined number of carrier cycles so that the conventional frequency analysis (spectral analysis such as FFT technique) can be used for signal analysis (that is, reception). , Such analysis and iterative techniques assume the continuity of the time-continuous signal for optimal detection.

In these conventional radio frequency communication systems, in each case, a carrier is carried by the data (data) to enable transmission by a relatively small device and efficient propagation through a line-of-sight (LOS) communication channel. It is usually the microwave frequency). However, when the transmitted energy is concentrated at such a high frequency, the energy is easily blocked by the terrain and other inclusions existing between the transmitter and the receiver. In order to understand why such radio frequency energy blocking occurs in a communication system, the interaction between radio frequency energy and objects will briefly be reviewed below.

To obtain the ability to communicate through a physical barrier such as a building, wall, foliage, soil, or tunnel, while communicating at high data rates over a wireless channel, a particular frequency (ie, frequency band) The spectral energy must contain a significant amount of "color" in order to minimize the risk of being blocked. If the transmission signal contains low frequencies, two major advantages can be enjoyed. The first is that low frequencies penetrate high loss media, which is why the US Navy uses very low frequency radio frequency transmissions to communicate with submarines through seawater. This penetration phenomenon can be regarded as a "skin effect" in which the attenuation is in dB and proportional to the transmission frequency. An example of the effectiveness of low frequencies compared to high frequencies in penetrating a structure is shown in FIG. FIG. 1 shows the amount of attenuation (unit: dB) in various materials in relation to the frequency of radio frequency energy. From this figure, it can be seen that the low attenuation at low frequencies minimizes reflection. You can take the view that it is due to being done. That is, in order to reflect waves, the object usually has to be large (greater than 1/4 wavelength) and many small objects reflect high frequency microwaves but low frequency In the case of the wave, the size of the object is too small as compared with the wavelength of the transmission signal, and thus the waves do not interact with each other.

Therefore, to the inventor's knowledge, there is a need for a communication system that includes penetrable spectral components. In the current spread spectrum / narrowband system, mutual interference cannot coexist with other narrowband users in the same spectrum (in the case of CDMA, the user's spectrum is overlapped, but the output control is strictly controlled). Must be done). This is because other users experience very heavy interference, and at the same time, our communication system also suffers from such users. Also, high speed links typically operate on microwave carriers that are easily blocked by terrain and inclusions. Such systems are composed of components (eg, antennas) that have a fairly flat frequency response over the bandwidth used, and thus have no effect on the waveform. Also, these systems assume the presence of multiple cycles (several cycles to multiple cycles) of the carrier between transitions (eg, zero crossings) of the modulating waveform.

Such a conventional narrow band modulation system (including a conventional spread spectrum system using direct spread and frequency hopping) is narrow because the spectrum of the modulation waveform reflects only 10 to 20% of the carrier frequency at the maximum. It is considered a band and its bandwidth is a narrow frequency range between F _l (lowest frequency) and F _h (highest frequency) containing 90% of the energy. The center frequency is F _c, and this is (F _l + / F _h 2)
If we call it "carrier frequency", the usable bandwidth of the UWB system exceeds 100%, which is not possible with the conventional "narrowband system". The present invention, which enables the decomposition of multipath (reflected signal) and the maintenance of a high data rate while operating at a low frequency that can be penetrated, has been realized by the recognition of this fact by the present inventors.

By the recognition of these phenomena by the present inventor, the present invention has been realized which enables multipath decomposition and high data rate maintenance while operating at a low frequency. This combination has significant advantages as low frequencies penetrate high loss media and minimize reflections from objects due to their relatively large wavelength. On the other hand, in contrast, conventional systems typically have a bandwidth of less than 10%, resulting in poor resolution at low frequencies.

Conventional UWB systems have relied on the generation and reception of 1-2 impulses of relatively low duty cycle short impulses. For example, deRosa (US Pat. No. 2,671,896), Robbins (US Pat. No. 3,662,316), Morey (US Pat. No. 3,806,7).
95), Ross and Mara (US Pat. No. 5,337,054).
, And Fullerton and Kowie (US Pat. No. 5,67,671).
No. 7,927) is an example, and 1 ns level impulse is 1 to 10 MH
It emits at a rate of z and has a duty cycle of 100: 1 to 1000: 1.

As currently recognized, two problems arise at such low duty cycles. First, the large peaks make it difficult (or almost impossible) to efficiently generate large average power. For example, its peak voltage is a low voltage (1
． 8V) above the breakdown voltage of CMOS, which limits standard low cost packaging. And the second problem is that this high peak destroys the "ore detector" receiver which is sensitive to the time domain space.

On the other hand, the waveform used in the present invention, in contrast, is a shape-modulated wavelet (ie, a short and spatially compact impulsive electromagnetic wave) with an energy envelope resembling a single smooth Gaussian pulse. It consists of a sequence of wavelets, and by transmitting this high duty cycle waveform, both of the above two problems are solved. The waveforms required for this can easily be generated by low voltage components, the transmitted energy of which is spread in both time and frequency over all areas and consequently looks like noise. JTFA (joint
In analysis procedures such as time-frequency analysis), time is the X axis and frequency is the Y axis.
While plotting the image of signal energy well as an axis, the bright spots represent high energy at a particular time and frequency, often with varying relationships between frequency resolution and time resolution. For low duty cycle UWB systems, these images are displayed as vertical bars with some random spacing. On the other hand, in the case of other high duty cycle UWB systems, such as Fleming and Kushner (US Pat. No. 5,748,891), these images may be structured, for example, like moire patterns. Has a generalized view. On the other hand, when the bi-phase waveform according to the present invention is used, the transmission signal is displayed smoothly on these screens.

Also, in the case of the conventional system, pseudo random time intervals are used between constant (essentially the same) pulses for spread spectrum and information transmission, and information is transmitted by pulse position modulation (PPM). It is transmitted. However, as already recognized today, this type of communication is not the best for several reasons. For example, a demodulator may mistake a false time shift due to multipath reflections as a formal time shift due to a modulator, and PPM is not the best choice for multipath channels. On the other hand, in contrast, the present invention conveys information by changing the pulse shape, and all multipaths are in a stationary relationship with the information-carrying pulse, and are multi-modulated by data modulation. The path will not be confused.

[0016] Yet another reason why PPM is not the best one is the high error probability of the PPM detector in the case of noisy signals. Coherent BPSK and PPM
According to the analysis result of the conventional system using BPSK, when the channel bandwidth is the same and the data rate and the bit error rate (BER) are equal, BPSK can tolerate a larger noise by about 6 dB. ing. In the case of the present invention, even in its simplest form (in the case of a single wavelet coded by inversion or non-inversion), this same 6 dB advantage is ensured. One reason for this is that a single pulse is always shorter than the window in which the pulse can be detected at two positions. That is, in the case of the present invention, the width of a single pulse represents a time slot for carrying information, whereas in the case of the PPM system, the time slot for carrying information is about 1.4 to 2 It is expressed in pulse width. Yet another reason is that the detected voltage difference between "1" and "0" is smaller than the BPSK signal,
As a result, more signal power is needed to overcome the noise.

To avoid interference with other users in the same spectrum, whitening of the UWB system transmission spectrum is required. A further difference between the conventional system and the present invention is the interference created by the transmitter. In the case of the present invention, pulse position randomization may be performed for spectrum control (ie, for smoothing the output power spectrum), but the present invention does not require this and is preferred. It is also not used in the examples. Instead, the spectrum is smoothed by generating a random sequence of shape-modulated pulses so that the entire waveform appears random. The transmitter according to the invention transmits one "symbol" of the pulse-shaped sequence family, each "symbol" being able to carry multiple bits of information, a sequence of information comprising this "symbol" in a conventional system. Rather, it produces a more "white" waveform. As a result, the interference level caused by transmissions by the transmitter of the present invention is lower than conventional systems even when the two systems broadcast the same average power over essentially the same bandwidth. That is,
In combination with the above 6 dB advantage over PPM, the present invention can provide comparable communication rate and bit error rate (BER) at much lower interference levels.

Regarding the whitening of the transmission spectrum of the UWB system, it is necessary to consider the influence on the reception. In the case of the conventional UWB system, the time interval between pulses is jittered in order to whiten the transmission spectrum, but the jittering has a side effect, and the present invention avoids this. That is, to give an example, a sine wave sampled at random times appears as “noise”. Similarly, spectral peaks arriving at conventional UWB receivers (ie, tones, or quasi-sinusoids such as conventional narrowband radiation) appear as noise in the data samples. In order to communicate, the signal must be strong enough to overcome this "noise". On the other hand, in the preferred embodiment of the present invention,
It does not jitter the pulse positions and instead gives the pulses the same spacing with an accurate clock. As a result, the tones that arrive at the receiver and are captured will appear as patterns in the data samples. By recognizing, calculating, and subtracting this pattern, as described herein below, it is possible to eliminate most of the interference caused by the tone. Also, if the tone arriving at the receiver is above the Nyquist cutoff frequency (ie, it is higher than half the data sampling frequency), the frequency folds to produce a pattern. Is done. As a result, because the receiver operates with low noise, the present invention is capable of operating at much higher range and data rates than conventional UWB systems in high noise environments.

As is now recognized, it is desirable to achieve high data rates in channels where there are high levels of multipath, but in conventional systems,
There is a constraint due to intersymbol interference caused by multipath. For example, in the case of a building, an extremely bad multipath (that is, an interference echo signal) delayed by about 500 ns from the signal on the direct propagation path is generated. As a result, inter-symbol interference only becomes severe even if pulses with intervals smaller than 500 ns are transmitted in order to obtain a high data rate. On the other hand, the present invention solves this difficult problem by transmitting a symbol that conveys information of multiple bits. Since each symbol is itself spectrally white, which means its autocorrelation is a low sidelobe spike, the multipath is continuously resolved over the symbol period. Therefore, even if there is a high level multipath, a high data rate can be obtained without intersymbol interference.

A feature of the invention is that by transmitting one of the pulse shape families, each pulse can carry multiple bits of information without losing any of the advantages described above. For PPM systems, additional time slots are used to convey more than one bit per pulse, which results in more severe multipath effects as described above, as well as other problems. It becomes like this. Conventional UWB systems lack control over the waveform shape and cannot send multiple bits per pulse.

Further, in the UWB communication system according to the conventional technique, a highly accurate clock is required in order to reduce the time required to acquire synchronization. However, even with the presence of an accurate clock, acquisition of synchronization often takes tens of seconds, which reduces the achievable data rate and makes the device difficult to use. One of the features of the present invention is that the synchronization can be acquired quickly, the duration of which is typically on the order of milliseconds.

SUMMARY OF THE INVENTION In accordance with the title of this section, selected features of the invention are briefly described below. A more detailed description of the invention is the subject of the entire specification.

It is an object of the present invention to provide ultra wideband (UWB) high speed digital communication systems and support system components, as well as methods for directly generating short and spatially compact electromagnetic wavelets (ie impulses or energy packets). That is. The waveforms of these wavelets are generated, for example, using inverted or non-inverted copies of the wavelets to propagate in free space and convey information. In the system according to the invention it is possible to convey information by sending a sequence of these impulses, the relevant spectrum being constrained both by the choice of sequence coding (sequence coding) and by the wavelet shape (wavelet selection). . The waveform generated by the present invention can penetrate obstacles such as walls and media such as soil.

A piecewise continuous wave consisting of a sequence of shape-modulated wavelets (ie, short and spatially compact impulsive electromagnetic wavelets) each having a single smooth Gaussian pulse-like energy envelope. UWB generated directly
Construction of a high speed digital communication system and method is another object of the present invention.

It is a further object of the present invention to generate a piecewise continuous wave directly from a digital logic circuit, where the characteristic shape of the wavelet is the voltage rise time of a logic gate formed in an integrated circuit. Formed at least in part from the characteristic.

The waveshapes of these wavelets are tuned to propagate in free space and convey information, and these wavelets are generated as a weighted sum of substantially orthogonal master wavelets.

That is, in the case of two master wavelets, the transmission signal is as follows.

[0028]

[Equation 7]

Here, the information is stored in the wavelet coefficients a _i and b _i , the functions W ₁ and W ₂ represent the master wavelet functions, τ represents the time interval between the wavelets and i is in the data element sequence. Represents the index of a specific data element in.

The master quadrature wavelet may be, for example, the derivative of an odd and even Gaussian (ie Gaussian shaped pulse) as shown in the following equation:

[0031]

[Equation 8]

Here, r and s are odd and even, respectively, and set a relative bandwidth of the wavelet, t represents time, and k represents a scaling parameter of time. Also, the master wavelet may be the real and imaginary parts of the Rayleigh wavelet as shown in the following equation.

[0033]

[Equation 9]

Here, n is a parameter for setting the relative bandwidth, and as described above, t
Represents time and k is a time scaling parameter. The set of master wavelets can also consist of wavelets with different temporal scaling parameters k and / or relative bandwidth parameters r, s, and n.

It is a further object of the invention to convey more than one bit per wavelet by generating each wavelet from the master wavelet shape family. It is an object of the present invention to convey information by allowing the variation of wavelet coefficients (eg a _i and b _i ) between +1 and −1 to send and receive inverted or non-inverted copies of the wavelet family. Is.

It is a further object of the invention to control the transmitted spectrum while transmitting information by transmitting a sequence of these impulse wavelets, where the choice of sequence coding for shaping the spectrum. And both wavelet shapes are used.

It is possible to create a 2 ^P spectrally white sequence family, where one sequence is called a symbol, so that P + 1 bits of information are conveyed upon receipt of a single symbol (ie one sequence of the family). It is a further object of the invention. In order for the decoder for this family to operate with the complexity of P · 2 ^P level,
Creating a 2 ^P sequence code (ie, symbol) family is one of the objects of the present invention.

It is a further object of the present invention to improve the reception performance of UWB signals by utilizing the multipath phenomenon.

It is a further object of the invention to provide the aforesaid object in a device that can be manufactured in large part by means of large scale integrated circuit (LSI) semiconductor technology.

It is also a further object of the invention to provide piecewise continuous waveforms generated from the inherent rise time (or fall time, these are commonly referred to as “edges”) of semiconductor logic circuits. is there.

It is a further object of the invention to provide a scalable radio capable of communicating over a wide spectrum as a function of the speed of operation of semiconductor circuits.

Further, a feature of the present invention is that the processing capacity is 18 when a new semiconductor circuit is introduced.
According to Gordon-Moore's hypothesis that "it almost doubles every month", the communication efficiency of the present invention is also improved according to "Moore's law".

Further, in the present invention, the waveform is generated based on the rising and falling times (edge) of the logic circuit. Therefore, as the speed and performance of logic circuits increase over time, so does the bandwidth available for transmission according to the present invention, which results in data for channels with limited power spectral density realized by the present invention. The rate also rises.

The above and other objects and advantages of the present invention will be made clear by the following description. In this description, reference is made to the following drawings, which form a part of the specification, which are presented for purposes of illustration and are not intended to limit the invention to the preferred embodiments. Further, the following description does not represent all of the present invention, and the present invention is
It can be implemented with different arrangements according to the breadth of the invention.

(Detailed Description of the Preferred Embodiment) (Definition of UWB) From the viewpoint of energy diffusion, that is, from the viewpoint of resolution, the bandwidth and the center frequency can be treated independently. However, the term UWB coined by the DARPA Research Panel does not include the term "relative bandwidth" in its name, but its definition requires it.

To the inventor's knowledge, the reason for preferring such a bandwidth-based definition in relation to the center frequency is due to the following three desirable main characteristics. The first is resistance to scintillation and multipath fading. The only way to prevent scintillation, speckle, and multipath fading is to have a resolution approximately equal to the wavelength. And the second reason is the penetration of the material by wide bandwidth signals. To communicate at maximum data rates over high loss media (or through high loss media, or to perform full resolution radar imaging in high loss media), low frequency (penetration) and wide band ( Resolution) is needed, and if these are put together, a wide relative bandwidth is needed. In such cases, the losses at high frequencies are too great to use. These advantages are provided by a particularly wide relative bandwidth and are not available in narrowband systems, so the definition of UWB is based on relative bandwidth.

When B is the bandwidth, f _c is the center frequency, and f _h and f _l are high frequency and low frequency cutoffs (eg, −6 dB from peak), the specific bandwidth B _f is defined by the following equation. To be done.

B _f = B / f _c = (f _h −f _l ) / ((f _h + f _l ) / 2) (1)

A UWB system is one that has a specific bandwidth B _f in the range of 0.25 to 2.0, which, in the case of a UWB system, has approximately the same bandwidth and center frequency (another expression). If you do, the resolution and the wavelength match).

Analytical UWB Waveforms In the present invention, it is desirable to use an analytical transmit excitation waveform (ie, signal) that is the derivative of a Gaussian shaped waveform, for this reason. Has the optimal time resolution for a given occupied bandwidth. Also, these signals are
It is continuous and can discriminate infinitely. The present invention uses higher order derivatives,
The first derivative (Gaussian monocycle) shown in the time domain of FIG. 2A is defined by the following equation.

[0051]

[Equation 10] (2)

Here, A is the amplitude of the peak, 2t _p is the pulse width between the peaks, and the energy is represented by the following equation.

[0053]

[Equation 11] (3)

[0054]   The spectrum s (t) shown in FIG. 2B given by equation (4) is also Gaussian.

[0055]

[Equation 12] (4)

[0056]   The peak of the spectrum occurs at the following position.

Ω _max = ± 1 / t _p rad / s, that is, f _max = ± 1/2 IIt _p Hz

The bandwidth is determined by the upper and lower frequencies at a power level of −6 dB as follows.

F _lo = 0.3191057f _max , f _hi = 1.9216229f _max (5)

[0060]   The center frequency is as follows.

F _c ((f _lo + f _hi ) / 2) = 1.12f _max (6)

Since each derivative places a zero in the direct current (DC) of the power spectrum, there are many zero crossings in the time domain with higher order derivatives, which results in a narrow bandwidth and low frequency cutoff. The center frequency is pushed up.

(Calculation of Peak Power to Average Power Ratio (Crest Factor) of XI Bi-Phase Modulation According to the Present Invention) It is desirable to operate in class C or switching mode in order to obtain an efficient transmitter. It is also desirable to minimize peak power in order to operate in standard low voltage high speed CMOS. That is, it is desirable that the crest factor (peak power to average power ratio) is low. The crest factor of the waveform used in the present invention can be calculated as follows. First, the signal interval (that is, the pulse repetition frequency (PR) is set so as to avoid large energy overlap between adjacent pulses.
F)) is defined. If α = 3.0342 and | t |> αt _p , then | s
(T) | <] can be represented by 0.05. The minimum pulse amplitude is 2 from the peak amplitude
If it is set 6 dB lower, the maximum pulse repetition interval (PRI
) T and the pulse repetition frequency (PRF) F _{prf_max} are as follows.

T = 2αt _p , F _{prf_max} = 1 / T (7)

[0065]   Then, the average power can be calculated as follows.

[0066]

[Equation 13] (8)

Since the peak power is A ² , the peak-to-average ratio is 2.519. As a reference for comparison, if the amplitude of the sine wave A, has an average power of peak power and A ^2/2 of A ^2, the peak-to-average ratio is 2.

This analysis shows that the bi-phase signals used in the present invention are similar in efficiency to continuous wave CW tones, and this low crest factor provides significant advantages. That is, unlike the conventional PPM system, in the case of the transceiver, receiver or transmitter according to the present invention, the antenna can be directly attached to the pin on the low voltage CMOS IC. Further, the low pulse voltage and not a large spike allows easy control of crosstalk between traces on the circuit board.

(Spread Spectrum) While the conventional system operates using a carrier wave, in the present invention, VSB is used.
It uses one form of technology and no carrier. The spectral characteristics will be derived below. First, let the basic pulse waveform s (t) be as described above, and let the user code h (t) be a length N _c consisting of equally spaced positive and negative impulses indexed by n according to the following equation. It will be expressed by the sequence of.

[0070]

[Equation 14] (9)

Here, τ _c is a code change interval. Applying the mapping {0,1} → {-1,1} such that the data d (t) is represented by one equally spaced positive and negative impulse per data bit indexed by k. Binary data shall be encoded by compression, forward error correction, and whitening scrambler. Furthermore, it is assumed that there are N _b code impulses per data bit (
That is, a chip composed of N _b impulses constitutes a single bit). For example, when d _k = [1, -1,1,1, -1,1, -1, -1, ...], the following is obtained.

[0072]

[Equation 15] (10)

Here, τ _d = N _b τ _c is a data change interval. A combination of a code and data will be represented as follows.

[0074]

[Equation 16] (11)

[0075]   Therefore, the radiation waveform x (t) becomes the following convolution.

[0076] x (t) = g (t) (X) s (t) (12)

Since convolution in time is a multiplication in the frequency domain, the transmitted spectrum is

[0078] X (ω) = G (ω) S (ω) (13)

Since G (ω) is whitened, the transmission spectrum is essentially the spectrum of the basic impulse waveform s (t). In the simplest case where the code length and the chip are the same length (ie, N _c = N _b ), g (t) = h (t) (X) d (t), so the transmission spectrum is As easy as.

[0080] X (ω) = H (ω) D (ω) S (ω) (14)

Once again, the transmission spectrum resembles that of the basic impulse waveform s (t), since both h (t) and d (t) are sufficiently white. Therefore, direct sequence spread spectrum (DSSS) and code division multiple access (CDMA)
) Identical codes commonly used in systems (eg Kasami, Gold,
Walsh-Haddamard, PN, etc.) can be used as part of the present invention.

(Information Theory Benefiting from Shannon's Theoretical Channel Capacity Limit) From the viewpoint of information theory and regulation, it is technical that the UWB waveform of the present invention is suitable for applications of short distance and high data rate. The rationale is based on the following Shannon's channel capacity formula.

[0083] C = Blog (1+ (S / N)) (15)

This equation relates the bit rate C to the channel bandwidth B and the signal-to-noise ratio (SNR), and mainly two pieces of information can be obtained from the equation (15). First, at low SNR levels, the log function is approximately linear, and if the power is doubled, the data rate is also roughly doubled. However, when the SNR is high, the log function becomes non-linear abruptly, greatly increasing the power. However, the data rate will only increase slightly. Complex modulation schemes (ie, schemes that provide more than 1 bit / Hz) require high SNR and high power must be used to improve SNR, but in this unfavorable log relationship It can be concluded that obtaining high data rates in narrowband systems is fundamentally difficult because power is converted to data rates.

The second main observation relates to the way in which the power of a signal is defined.
In the case of UWB systems, there is a regulatory limit for radiation of P _o = watts / Hz. Therefore, the signal power is proportional to the bandwidth, and Shannon's formula is as follows.

[0086]

[Equation 17]

The results of this analysis show that the data rate is proportional to the bandwidth. It also shows that the data rate is proportional to the power when using the low SNR modulation technique, which is a feature of the invention. Since the bandwidth exceeds the required data rate, it is combined with a CDMA code to trade the data rate for power reduction or range extension. By using codes of different lengths, the UWB system according to the present invention can be controlled entirely in software while in the range of 100 Kbits / sec to 100 Mbits / sec (and ranging from a few users to many users). It has expandability.

(Intrinsic Compatibility with WPAN) UWB has a competitive advantage in short-range, high-data-rate applications such as WPAN, but in long-distance such as mobile phone systems. For low data rate applications, it does not retain the same advantage. That is, for some reason, conventional systems retain their superiority outdoors. The first reason is that the conventional mobile phone system has a very large power, which is extremely important for overcoming the 1 / R ² channel loss. In the case of the UWB system, since it has to coexist without interfering with other users, there are severe restrictions in terms of power. This makes natural shielding by buildings ideal for indoor UWB applications, while outdoor broadcast applications are the least suitable for maximizing interference.

The second reason is that in the case of a narrow band system, high antenna gain (a directional array and parabola optimized for a narrow band) can be enjoyed in addition to the above-mentioned large power. UWB system antenna gain is FCC (or equivalent regulation in other countries)
Therefore, instead of limiting the transmission power, the peak V / m is 3m at all angles.
It has the opposite effect because it is limited to. That is, even if the transmission antenna gain is increased, U
In the case of WB systems, it is not possible to operate over long distances, which would only reduce the power requirements of low level transmitters. Also, the UWB system cannot benefit from using the most proven antenna structure (eg, log period, spiral, etc.) because it is decentralized.

Thirdly, the advantage of UWB systems in the decomposition of closely spaced multipaths, which is common in indoor environments, is not effective in the case of outdoor systems because the spread of the multipath delay is much larger. Absent. And the narrow band RAKE processing is (U
Not applicable indoors (advantageous for WB), but can be applied outdoors. That is, the superior ability of UWB technology to provide high data rates and very good performance in very closely spaced multi-paths is demonstrated in short range, high data rate, very low power applications such as WPAN. However, UWB technology can enjoy a competitive advantage in indoor applications where this is possible.

Advantages of Bi-Phase UWB Waveform Modulation According to the Present Invention Modems included in the present invention are fully coherent bi-phase baseband direct spreaders designed primarily for implementation in low power CMOS circuits. It can also be expressed as spread spectrum wireless technology. This implementation is in contrast to the video impulse pair and pulse position modulation (PPM / time hopping) implemented in conventional systems and some of the advantages of the invention are listed below.

1) By bi-phase modulation, as described in the next section, 3-6 dB depending on the preconditions as compared with PPM (time hopping) in an environment without multipath.
Brings the advantage of. Further, since multipath often appears in data modulation of PPM system, etc., when this is taken into consideration, an even greater advantage is realized.

2) For bi-phase modulation, using the lowest spectrum results in optimal material penetration while achieving a given multipath decomposition and data rate. That is, assuming that bi-phase modulation and PPM have equal data rates, both modulation techniques require the same time to encode 1 bit. However, in the case of PPM, two pulses must be fit within the same time window that stores one biphase pulse (see FIGS. 3A, 3B, 4A, and 4B). The pulse will have a bandwidth (i.e., twice as wide) as half the PPM pulse. Half bandwidth is equal to half the center frequency in the case of UWB systems, which allows the bi-phase modulation to operate at lower frequencies with good penetration, giving a superior data rate advantage through high loss material. Brought to bi-phase modulation.

3) Bi-phase modulation achieves optimum multipath resolution. Bi-phase systems have a 1-bit time window that matches the pulse width, whereas the 1-bit time window of PPM is twice the pulse width. In the case of PPM systems, multipath obscures the pulse position and is thus sensitive to multipath reflections, which actually appear in the data modulation domain.
On the other hand, in contrast, in the case of the bi-phase system, “1” and “0” are set.
Use the exact same propagation path and are resolved within a time window of a single pulse width, so that the effects of multipath are optimally minimized. 5A and FIG.
See B.

4) Similar to 3) above, bi-phase modulation is not sensitive to the number of zero crossings in the pulse waveform. In the case of PPM systems, the energy of "1" is applied in the "0" position at any phase as a function of where the extra zero crossings are, so that the extra power generated by filters, multipaths, antennas etc Confused by a naive zero crossing. In contrast, phase detectors in bi-phase systems, in contrast, continuously detect phase-encoded data because the phases are all maintained by a linear process (ie, filters, multipath, antennas, etc.). To do.

5) The peak power to average power ratio for bi-phase modulation is less than 3 (for reference, it is 2 for a sine wave). This results in an efficient "switching" transmitter, which is naturally compatible with inexpensive low voltage CMOS implementations. The antenna requires only 1V peak-to-peak voltage, which is within the performance range of low voltage CMOS.

6) In the case of bi-phase modulation, the jittering requirement is reduced, so that the wireless jittering requirement for clock generation can be satisfied by CMOS. In the case of PPM, it is necessary to accurately control arbitrary time positions at high speed (that is, from pulse to pulse), which requires a series of wide band circuits for accumulating jitter. In contrast, in a bi-phase system, only stable clocks (cheap, high-Q, low-jitter clocks) are needed because the pulses occur at regular intervals.

(Theoretical Demonstration of Superiority of Biphase over PPM) The superiority of biphase modulation over PPM will be mathematically demonstrated below. Both modulation techniques have time-packed pulses that produce high peak RF with equal bandwidth and average power, PPM pulses do not overlap, and bi-phase modulation has a 3 dB advantage in data rate. Suppose we need 3 dB less peak power than PPM. First, consider non-coherent PPM,
As a result, it is pointed out that the performance deteriorates due to the correlation coefficient. Next, the optimum PPM
Consider an impractical implementation of. Again, assume that both modulation techniques have time-packed pulses that produce a high PRF with equal bandwidth and average power, but in this case, the PPM pulse has a cross-correlation value. It is assumed that they overlap so as to be minimized. As a result of this examination, it is found that the optimum PPM is inferior in performance to the bi-phase modulation by 0.9 dB and needs a peak power as large as 0.8 dB. And the last comparative example is a transmitter with a low PRF, that is, a low duty cycle transmitter with equal bandwidth, average power and peak power for both modulation techniques. Modulation is PP
It is shown to be 3 dB better than M. The analysis is then concluded by comparing modulation techniques that take into account the practical issues of spectral whitening, and the differences in whitening techniques provide a significant implementation advantage for bi-phase systems over PPM systems.

In the following analysis, the influence of random multipath is not taken into consideration. In practice, due to the appearance of multipaths in the modulation domain of PPM systems, it is expected that biphase systems will have an additional advantage in the case of multipath channels.

[0100]   (Construction of theory)   The transmitted signal of a typical UWB system can be modeled as follows.

X (t) = V _t s (t, b) (17)

Here, t is time, b is a bit value (bε {0,1}), V _t is the amplitude of the transmission waveform, and s (t, b) is the energy-based waveform. . Therefore,
The signal alphabet is {s (t, 0), s (t, 1)}, which has the following dot product (ie, cross-correlation).

[0103] <S (t, 0), s (t, 0)> = 1 <S (t, 1), s (t, 1)> = 1 <S (t, 0), s (t, 1)> = ρ (18)

These cross-correlations represent the output of the peak-sampled coherent detector (ie, an ideal matched filter receiver). The transmission model described above applies to both bi-phase and pulse position modulation (PPM) techniques, where for bi-phase modulation s = (t, 0) =-s (t, 1) and ρ = <s (t. , 1)>
, S (t, 1)> = − 1, and in the case of pulse position modulation, ρ = 0, assuming that the two pulses do not overlap.

Assuming an additive white Gaussian noise (AWGN) channel, the received signal is

R (t) = V _t s (t, b) + n (t) (19)

Here, n (t) is AWGN having an average of 0 and a standard deviation of σ. Received signal r
The probability of erroneously identifying bit b of (t) is as follows.

[0108]

[Equation 18] (20)

This formula 20 was calculated in 1989 by Holtline Heart and Winston (Holt,
Rinehart, and Winston) company. P. By BP Lathi "
Modern digital and analog commun
ication systems ”, and John Wiley and Sons in 1968.
ley and Sons, Inc.). L. HL Van Trees, "Detection, Estimation, and Modulation Part 1 (Detection, Estimation, and Modulation)
theory Part 1) ”, and the entire contents of each are incorporated herein by this reference. Here, Q () is an error function defined as follows.

[0110]

[Formula 19] (21)

The error probability equation 20 can be used to study the relative performance of bi-phase and pulse position modulation.

3A and 3B show signal waveforms of the biphase system. In the case of bi-phase, ρ = −1, and the error probability is

[0113]

[Equation 20] (22)

[0114]   In the following section, equation (16) is compared with various PPM transmission schemes.

(Comparison of PPM and Bi-Phase-Case 1) There are several methods for comparing these two modulation forms. For example, it can be assumed that the peak power or the average power is the same, and the average power is significant from a theoretical point of view, but the mounting efficiency is determined by the peak power. Further, the performance comparison depends on the correlation coefficient of the PPM signal waveform, and it is necessary to select whether the data rate is the same or the bandwidth is the same. As mentioned above,
If the system data rates are equal, the bi-phase pulse can get half the bandwidth of the PPM pulse (ie, twice the duration). As a result, the data rate advantage over high loss material is brought to the bi-phase.

In this case 1, the bandwidths of the bi-phase and the PPM system are equal (that is,
Pulse widths are equal), and the PPM waveforms are assumed not to overlap. As a result, the data rate of the PPM system is half the data rate of the bi-phase system because the two bits of the bi-phase are fitted into the same time window as the one bit of the PPM. The PPM waveform is shown in FIGS. 4A and 4B.
The time axis of is double the time axis of the biphase waveform shown in FIGS. 3A and 3B. Bi-phase systems use two times the power of a PPM pulse to maintain an equal average power because they are sending two pulses during the duration of one PPM pulse. These two pulses are orthogonal (p = 0) and the error probabilities are:

[0117]

[Equation 21] (23)

Identical results are obtained from equations (22) and (23). Therefore, the bi-phase and PPM signal technologies have the same error probability, but the bi-phase technology has a data rate of P.
It is twice as high as the PM data rate, which is equal to a 3 dB advantage in power.

Note that the peak power required by the PPM is 3 dB higher than the peak power required for the bi-phase signal. That is, based on peak power instead of average power, bi-phase modulation is 6 dB better than PPM.

(Comparison of PPM and Bi-Phase-Cases 2 and 3) While the above is a valid comparison criterion, this analysis can be adjusted to evaluate a particular implementation. For example, using non-coherent detection (Case 2),
Due to the video filter setting time, ρ> 0, and the performance of PPM is slightly worse. On the other hand, by overlapping PPM pulses (Case 3
It is also possible to optimize the performance of the coherent detector and minimize the correlation coefficient ρ.
However, the optimum form of PPM is typically not used because the filter and antenna make the performance sensitive to phase jitter and pulse spreading. It also requires a sampler with sufficient bandwidth to capture the peak of the UWB pulse.

However, for completeness, FIGS. 5A and 5B show a preferred pulse constellation for a given waveform. In the case of this configuration, ρ = −0.6183, and the waveform is 1.2 times longer than that required for biphase modulation. Therefore, the amplitude is √1.
The average power is kept the same by increasing by 2. As a result, the error probability is as follows.

[0122]

[Equation 22] (24)

The SNR of this optimum PPM is 0.1486 dB inferior to the bi-phase modulation, and the data rate is 0.7712 dB inferior to the bi-phase modulation. That is, bi-phase modulation is 0.9298 dB better than this optimal PPM configuration. Furthermore, the peak power requirement for bi-phase modulation is 0.7712 dB less than the peak power requirement for PPM. That is, based on the peak power,
Bi-phase modulation is 1.7 dB better than the optimal PPM.

(Comparison of PPM and Bi-Phase-Case 4) As a final comparison example of PPM and bi-phase signal, both systems have the same pulse width, the same peak power, the same average power, the same bandwidth, Take the case with the same data rate and non-overlapping PPM pulses. Therefore,
The duty cycle and PRF are low and equivalent in both techniques. In this case, ρ = 0 for PPM and ρ = −1 for bi-phase, so that the error probabilities are:

[0125]

[Equation 23] (25)

Therefore, in this scenario, bi-phase has a 3 dB advantage over PPM.

(PPM Duty Cycle and PRF Constraints) The bi-phase system whitens the spectrum by modulating the data with a whitening / spreading code and has no effect on the realizable PRF. on the other hand,
In contrast, PPM achieves spectral whitening by jittering the pulse position in time, which allows the UWB PPM system to operate at low duty cycles. Must work. For example, the PPM system has a nominal 1/2 ns pulse and a nominal 10 MHz PRF.
Operates at (1/2 ns on, 99.5 ns off). Therefore, the peak-to-average ratio of 2
To obtain an average power of 1 mW at 00, the transmit pulse should have a peak power of 200 mW (+
23 dB) (equivalent to 9 V peak-to-peak sine wave for 50 Ω). On the other hand, in contrast, the bi-phase system of the present invention operates with a peak-to-average ratio of less than 3 (for reference, the peak-to-average ratio of a continuous sine wave is 2.) and thus the radio according to the present invention. In the case of, only 3mW of peak power (4.8dB) (equal to 1.1V peak-to-peak sine wave for 50Ω) is needed to transmit 1mW of average power. Absent. That is, it is clear that the peak power requirement of PPM is significantly higher than that of an equivalent bi-phase system. It is important to note that the 1.1 V swing of the high bandwidth has a low voltage. The high bandwidth 9V swing required by the 18u CMOS component requires an external snap recovery or tunnel diode and an external microwave circuit.

UWB Phenomenon Advantages There are four physical elements that increase the importance of UWB in solving a particular class of communication and remote detection problems. It is scattering phenomena, penetration depth, resolution and bandwidth coupling, and interference patterns. These elements in ultra-wideband implementation are summarized below.

(Scattering Phenomenon) Scattering from an illuminated object is dominated by the size-to-wavelength (λ) ratio of the object. If the size of the object is less than λ / 4, then the object is in the Rayleigh region, in which case the waves interact with the object very little and little scattering occurs. Further, there is no directivity in scattering, and the amplitude of scattering is usually proportional to f ² (f is frequency). Then, if the size of the object is in the range of λ / 2 to 6λ, the object is in the resonance region, in which case the amplitude of the scattering oscillates with the frequency sweep and widens at the peak. The scattered energy is usually directional, but in several broad beams. Then, when the object is larger than 6λ, the vibration phenomenon is attenuated and the object is said to be in the optical region. The scattered energy becomes many beams and becomes highly directional. The amplitude of a single beam is typically in the form of F ^a , where a = −1, −0.5, 0, 0.5, or 1, depending on the shape and features of the object causing the scattering. For UWB radars that utilize the frequency spectrum, the frequency spectrum is used to evaluate "a" to identify the scattering mechanism.

These three regions are important for radar and communication, because as the frequency decreases, the scattering lobes widen and scattering by objects is reduced or stopped. That is, in the case of radar, this phenomenon (decrease or stop of scattering due to decrease in frequency) reduces clutter. On the other hand, in the case of a communication system, this phenomenon reduces the density of multipath reflections and reduces multipath variations due to wide scattering lobes.

(Penetration Depth) The penetration depth into the high loss material / medium is proportional to λ. Therefore, for example, a very low frequency is essential when detecting or communicating with an obstacle buried in the ground, and similarly when communicating through a wall or floor, Also, low frequencies are more suitable. According to the measurement results, for example, the attenuation due to the concrete wall is about 10 fdB / m (where f is a unit of GHz) (for example, John Orlando (Plenum Press) of New York in 1997 (
John Aurand) "Ultra Wide Band Short Pulse Electromagnetics Vol. 3 (Ultra-Wideband Sh
ort Pulse electromagnetics 3) ”(supervised Baum et al) 239
~ 246 "Measurement of short pulse propagation through concrete wall (Measurement
Of Short Pulse Propagation Through Concrete Walls) ". All contents are included in the present specification by the present citation. See also Figure 1). Therefore, the lowest possible frequency must be used to penetrate these materials, but the best resolution is required for multipath reflection resolution and object imaging. That is, the optimum device for communication and imaging through concrete walls is one that operates at the lowest frequencies possible and provides the best resolution at those low frequencies.

(Bandwidth and Resolution) The time resolution and the frequency resolution are inversely proportional to each other. That is, a wide band is required to obtain high-definition time resolution, and conversely, a long time scale is required to obtain high frequency resolution. Similarly, to notch the frequency spectrum, time sidelobes must be included in the time domain waveform. The only way to get a wide band at low frequencies and consequently a fine time resolution is to have a wide relative bandwidth which is an attribute of UWB.

Multipath Channel Model of UWB (ie UWB Interference Characteristics) If the relative bandwidth is small enough to allow multiple cycles within the pulse (compressed / matched filtered), then the interference process Is chaotic (there are multiple peaks and zeros). Again, this phenomenon reverts to the UWB definition of high relative bandwidth. When a plurality of waveforms having a time shift smaller than the pulse width arrive, an interference pattern is always generated. 6A to 6C, a UWB signal having a 1 GHz bandwidth and a narrow band signal having a 1 GHz bandwidth (that is, the center frequency is 10 GHz).
This phenomenon is shown by comparing with. FIG. 6C shows the summed inverted and non-inverted pulses, but time-shifted so that they are clearly separated. The UWB waveform is superimposed on the X-band waveform (which has a large number of cycles), and FIGS. 6A and 6B show how the two pulses shift from exactly overlapping to non-overlapping. The output from the detector is shown. There are multiple fades with the narrowband signal, and it is clear that variations occur due to the short time scale. On the other hand, in contrast, the detected UWB
The signal is slowly varying with no deep zeros or multiple peaks.

By reducing the scintillation and the fluctuation rate, the U in the communication system is reduced.
Since the WB system can operate with a low link margin, the tracking of a plurality of signal propagation paths by simple hardware and the application of RAKE processing are also possible. In the case of radar systems, most of the fast multiple fades (eg radar scintillation) that are characteristic of narrowband systems are mitigated by the UWB system and there is no multiple lobe interference pattern. For example, speckles are present in normal narrowband SAR (Synthetic Aperture Radar) images (John McCorkle, published April 1992).
Written "Early Results from the Arm
y Research Laboratory Ultra Wide Bandwidth Foliage Penetration SAR) "S
See PIE 1942 (ISBN 0-8194-1178-7 / 93) pages 88-92. All the contents are included in the present specification by this reference). Speckle is an interference pattern generated by a plurality of added time shift waveforms,
In some cases, the peaks are expanded by adding in the same phase, and in other cases, they are added in different phases and canceled. However, in the case of addition of single-cycle waveforms that slide in time with respect to each other, speckles do not exist in the UWB SAR image because multiple zeros and peaks are not generated even if they are added unless they are accurately matched.

(Advantages on Wireless Implementation: Effect of UWB Propagation Physical Characteristics on Wireless Implementation) It is recognized by the present inventor that the various aspects related to UWB propagation described above operate in a building or other high clutter area. It is important in optimizing a communication system built to do so. However, from the viewpoint of indoor channel measurement and system development, there are three main issues that affect wireless engineering as follows.

1) A large number of resolvable propagation path lengths exist between the transmitter and the receiver, and communication can be performed using each of these propagation path lengths. Since these propagation path lengths can be decomposed in time, the SNR can be improved by using them in combination.

2) One bit is represented by one pulse that does not have multiple cycles of RF energy, so it is not necessary to derive the “phase” of a particular multipath, and when a peak is detected The phase is also detected.

3) It is unlikely that the multipath will satisfy the unique condition and cancel (or cause a fade) the UWB signal on the specific propagation path length. It is even less likely that a deep fade will occur at the same time for all channel lengths that exist between the transmitter and receiver, in contrast, there are likely to be multiple channel lengths that result in a strong signal. .

As a result of 1) above, it is desirable for the present invention to include one or more RAKE channels. As a result of 2), the deconvolution needed to undo (or undo) the channel model is not needed and the circuitry and power handling hardware required to implement those RAKE channels is simple. That is, since the signal of each propagation path length is decomposed,
The channel model is provided directly without DSP, the only processing required is weighting and addition. As a result of 3), the radio of the present invention has a low R compared to narrow band radio.
It can operate at F power. Since the zero point of multipath is not deep, the link margin in the worst case can be significantly reduced. And the link margin in the average case is also low because the indoor propagation loss follows the standard free space 1 / R ² curve instead of the 1 / R ^3.5 curve.

FIG. 7 shows the measurement results of loss versus distance on the 7th floor of a 12-story reinforced concrete office building. The transmitted signal emanates from one room and passes through a large structural concrete pillar surrounded by metal cabinets, copiers and printers,
Enter the meeting room through the door (at the dotted line). The transmitter was physically moved to the respective distance and the sequence was acquired / tracked by the receiver. In this sequence, the receiver starts communication after detecting the best propagation path (ie, maximum signal). FIG. 7 is a simple plot of received power level versus distance measured by the A / D converter used to make the bit decision.

The difference between this plot and the plot for a narrowband system is notable. 2.4GH
The same plot for an indoor system of z shows deep fades with random inches spacing. Narrowband systems operate in unresolved multipath interference fields, resulting in fast fades and peaks as the receiver moves. Designers can use multiple antennas to add additional degrees of freedom, but at the expense of radio complexity. A 2.4 GHz indoor channel is typically modeled as having an average received power versus distance that is proportional to 1 / R ^3.5 , schematically representing multipath fading (1 / R ^{3. 5} and standard free space 1 / R ² curves are included for reference). On the other hand, in the case of the UWB system according to the invention, by contrast, the time slot (ie the propagation path) can almost always be detected (in this case the direct propagation path is clear or the multipath signal is (Added in the same phase), a smooth channel-loss function is shown.

There are two observations that deserve attention. The first is that the average channel loss is not inferior to 1 / R ² and is actually superior to it. If the transceivers are randomly arranged, the loss is likely to be less than 1 / R ² . The second point of interest is that even if the received power falls, it does not fall so deeply (only a few dB). Due to these characteristics, the radio according to the present invention is
It can operate at lower power and is more robust than narrowband wireless.

FIG. 8 is the channel sounding performed by the implementation of the invention with all filters mounted. The X-axis is based on the maximum signal indicated by X = 0. The distance on the X-axis indicates the offset distance (distance from X = 0) of the alternative propagation path length. Since the X axis is periodic, 5m is also -1m. The A / D value indicates the signal strength in a predetermined propagation path length. A / D values below 128 are negative, indicating that the pulse is inverted, and A / D values above 128 are positive, indicating that the pulse is not inverted. Normally, the signal does not invert after an even number of bounds in the propagation path, but it does invert when it does an odd number of bounds. It is clear that there are multiple channels with very useful signals, especially those located at X = 0 and X = 2.8. This vibration is due to the filtering applied to the zero-point specific frequency. When this system is operated without a filter, the channels are much sharper. The addition of the filter reduces multi-path robustness, but not to the extent that it resists strong RF interference (RFI). It is noted that embodiments of the present invention utilize all of the physical elements described above, resulting in performance advantages not otherwise obtainable by the embodiments of the present invention. The two plots are clearly shown. Figure 9
A shows how the receiver according to the present invention decomposes multiple multipath signal propagation paths, and FIG. 9B shows that the narrowband signal deeply fades when multipath occurs.

Dynamic Range Although an 8-bit TLV5580 A / D can be used in one embodiment according to the present invention, there are some issues that impact the dynamic range requirements. (1) Observed RFI levels are fairly constant, typically varying by only +/− 3 dB in the general office space area in the afternoon. This is not as volatile as you might imagine, it is Gaussian but not white. (2) A 2.4 GHz microwave oven or other tone competes with noise throughout the 2 GHz receiver band.
Therefore, the sensitivity is less than expected. (3) In the RFI embodiment according to the present invention, the main interference sources are notched in both the analog and digital domains. In a simulation of a UWB radar showing an RFI power extraction of 20 dB with an 8-bit A / D value, it is improved to 36 dB with a 32-bit floating point value. Therefore,
It can be seen that the A / D dynamic range has advantages over simple modeling.

(Mitigation of Interference and Notch Filter) A feature of the present invention is that reception that enables the operation of a 900 MHz or 1.8 MHz mobile phone within 1 foot from the receiving antenna without affecting the communication distance and BER. The notch mechanism of the machine front end is used. In one embodiment, an active stub is used that produces a mature impulse response to the time-modulated UWB pulse and its time-delayed inversion according to the present invention to self-cancel the multi-period signal. Other functionally equivalent circuits implementing adaptive or static notching can be used as well. The present invention uses analog and digital interference mitigation methods, and these notch mechanisms are also applicable to the transmit waveform to prevent interference to the GPS receiver. The transmitted waveform is affected by this mechanism, but as with the receiver filter, the degradation in system performance is negligible.

This communication system is based on a combination of code arrangements of replicas obtained by time-shifting and inverting a short RF wavelet, not a modulated carrier, and is different from a conventional narrowband transmission system (including a conventional spread spectrum system). It sets the line.

FIG. 10 shows a schematic plot of frequency versus energy, UW
B plots, conventional narrowband, and spread spectrum communication signal plots are compared. The conventional narrow band communication signal NB3 occupies a relatively small spectrum bandwidth of about 6 MHz, and in the case of the spread spectrum communication signal SS2, the occupied bandwidth is wider but the spectrum density is smaller. On the other hand, the UWB signal UB1 occupies a remarkably large bandwidth and has a spectral energy density of SS2 or NB.
It is smaller than any of the three. Compared to other pulsed waveform systems, the UWB signal according to the invention is innovative because it takes advantage of the shape, direction and grouping of the wavelets to generate the transmitted waveform. This feature allows the system to smoothly change the peak-to-average power ratio of both the spectral and time domain waveforms, as well as generate a large collection of codes for use in encoding various baseband signals.

Typical pulse widths according to the invention are on the order of 100-1000 picoseconds, and bandwidths are respectively around 8 GHz to 1 GHz, but of course larger value ranges can be supported as well. FIG. 11 shows a normal biphase pulse having a width of 500 ps according to the present invention. The pulses in this FIG. 11 are not sinusoidal, but rather discrete time domain signals and are not periodic, but these discrete pulses are transmitted at millions of pulses per second.

FIG. 12A is a time domain graph of randomized signal pulse time coding, where the time code itself is generated to look like noise. The power spectrum corresponding to the time domain signal sequence shown in FIG. 12A is shown in FIG. 12B. As shown in the figure, the spectral density of the transmitted signal is spread over 5 GHz, and the power spectral density is very low, so that it does not become an interference source for the conventional narrow band communication system. Thus, the UWB system according to the invention can be used simultaneously with a narrowband signal without a noticeable negative effect on the narrowband system. Depending on the combination of the short pulse used and the encoding technique,
The signal energy is spread over such a wide frequency band, where almost no energy appears in the band of any narrowband user, which results in the UWB transmitted signal being below the detection threshold of conventional narrowband receivers. -ing

The system of the present invention uses signals with sufficient spatial resolution to resolve obstacles that cause multipath fading in conventional systems, and uses a variety of obstacles,
Information can be transmitted at very high data rates through walls, tunnels, buildings, and other obstacles.

Multipath occurs when multiple time delayed and attenuated copies of the transmitted waveform arrive at the receiver at the same time. FIG. 13A shows an example in which electromagnetic waves are reflected or penetrated by various obstacles such as trees and cause a multipath phenomenon.
As shown in FIG. 13A, the signal interacts with the tree and is reflected by the direct return propagation path (propagation path 1) with a certain size (shown as propagation path 1 in FIG. 13B).
In another reflection propagation path (propagation path 2) shown in FIG. 13B, it is delayed with respect to the propagation path 1 in time and the phase is inverted from the signal of the propagation path 1. In the case of urban environments, such phenomena are caused to a large extent by reflection and propagation by architectural features such as walls, floors, ceilings, and windows.

If the distance resolution of the receiver is large compared to the difference of multipaths, constructive and destructive interference that degrades system performance occurs. This is multipath fading. On the other hand, as in the case of the present invention, when the receiver decomposes multipath components, no interference occurs and the multipath components can be used to improve system performance.

The range resolution of the receiver is approximately inversely proportional to the bandwidth of the transmitted signal. Therefore, 10
In the case of MHz system, it has the following range resolution.

R _res = c / BW = (3 × 10 ⁸ m / sec) / 10 MHz = 30 m

Similarly, a 100 MHz system has a resolution of 3 m (10 ft) and a 1 GHz system has a resolution of 0.3 m (1 ft). Architectural features (such as walls and floors) are separated by a distance of 10 feet, so a system that resolves them well requires 1 foot resolution.

The pulse code used in the system of the present invention is composed of a replica of the underlying short wavelet pulse that is bi-phase modulated and time-shifted. The receiver uses a matched filter (correlation process) to compress the transmitted pulse code.
As a result, the width of the compressed code is nominally the width of a single pulse. As a result, it is sufficient to avoid the negative effects of multipath and RA in the time domain.
The one foot level spatial resolution that allows KE processing has been imparted to the radio of the present invention.

As an example, the cross-correlation filtering operation used in the present invention is shown in FIG. The bi-phase template waveform 1402 is either stored in memory or regenerated directly at the receiver for mixing with the incoming RF signal 1401 at the mixer 1403. Various time domain wave shape morphologies are known and are kept in a fixed alphabet (usually only two shapes for binary systems and four for systems with 2 bits per channel symbol). Is the shape)
. For example, in the case of a binary communication system, using the bi-phase communication architecture of the present invention, "1" is represented by a bi-phase pulse with a specific phase setting and "0".
"Has the same shape as the pulse used to represent a" 1 "but with a phase (ie direction)
Is represented by the inverted pulse. It is likewise possible to use more phases and amplitudes to provide a larger number of bits per channel symbol as in the case of QPSK and M-ARY signals.

The output of mixer 1403 is an output signal 1404, which is applied to integrator 1405 to accumulate overlapping energy. The output of this integrator 1406 is used by the detection circuit to determine whether a 1 or a 0 (a larger amount of data when using a multi-level signal is sent).

A feature of the present invention is the ability to use coherent pulse integration, which results in the storage of energy in adjacent pulses to increase processing gain and to produce strong communication links with relatively small energy densities. You can In addition, the redundancy of the number of pulses representing a single bit can be coherently re-acquired at the receiver, thus providing the ability to incorporate processing gain into the signal by encoding the particular symbol to be transmitted into a predetermined number of pulses. . For example, for a 10 Mbps system transmitting 10 Kbps, one data bit spreads over 1,000 pulses. As a result,
An additional processing gain of 30 dB (10 log 1000) is provided. Therefore, the total processing gain is the result of addition of the respective gains of the duty cycle, symbol repetition, and pulse integration. In this example, the pulse integration gain is 30 dB and the duty cycle gain is 23 dB out of the total processing gain 53 dB. Is. As a result,
As shown in FIG. 15, the time-modulated UWB according to the present invention can be used to combine communication signals, radar applications, positioning applications, and imaging systems.

In the following, the time and frequency domain issues associated with the embodiments of the present UWB communication system will be briefly discussed.

A waveform with good bandwidth and derivative properties is a Gaussian monocycle and is formed by taking the first derivative of the Gaussian waveform. The Gaussian Monocycle
It has the following sloped form of the Gaussian envelope.

[0162]

[Equation 24] (26)

The pulse width is a function of the constant k and controls the decay rate of the Gaussian envelope. FIG.
Shows the case where the constant k was chosen to produce a pulse width of about 1.5 nanoseconds. Equations (28)-(30) relate pulse width to bandwidth.

In addition to the Gaussian first derivative described above, other wavelets containing a single zero-mean impulse waveform can be used as well. Furthermore, it is also possible to modulate the impulse waveform to include an added combination of two master wavelets. In this case, those master wavelets are orthogonal wavelets, one is the even derivative of the Gaussian shaped pulse and the second master wavelet is the odd derivative of the Gaussian shaped pulse.

The circuits that generate these signals include avalanche transistors, step recovery diodes (SRDs) in comb filter circuits, high speed logic and transistors. In the exemplary embodiment, the system uses a pulse forming network of high speed discrete logic with signal conditioning. This logic is
For example, it can be mounted on a CMOS (usually a single chip).

(Frequency Domain Representation) An important characteristic of UWB signals is the relationship between pulse width and bandwidth. Generally, the narrower the pulse in time, the wider the frequency band in which energy is spread. Thus, with a fixed energy per Hz, a wide bandwidth pulse will transmit more energy than a narrow bandwidth one. From this viewpoint, the shorter the pulse width is, the better the pulse width is. Other criteria that influence the choice of pulse width include the resulting band propagation characteristics. Usually, frequencies lower than the high frequency band (HF (3 to 30 MHz), VHF (30 to 300)
MHz) and UHF (300-1000 MHz)) are said to have better material (building and foliage) penetration. As an option for indoor operation, VHF
And the frequency band spanning the UHF band is excellent. This is a trade-off of energy diffusion for propagation model and engineering feasibility.

Consider the Gaussian monocycle of (26) to link pulse width to bandwidth. The expression in the frequency domain is also Gaussian, and is given by the following equation.

[0168]

[Equation 25] (27)

The spectrum of the 1 GHz Gaussian monocycle spanning the VHF and UHF bands is shown in FIG. Both the time width and the bandwidth are controlled by the parameter k, and the working frequency is best described by the next peak of the power spectrum.

[0170]

[Equation 26] (28)

The bandwidth is determined by the upper and lower frequencies with the power level reduced by 6 dB (-6 dB) as follows.

F _lo = 0.3191057f _max f _hi = 1.9216229f _max (29)

[0173]   The center frequency is as follows.

F _c = (f _lo + f _hi ) /2=1.12f _max (30)

If √k = 2.5 × 10 ⁹ , then f _max = 560 MHz and the bandwidth is 1 GH
z, the pulse width is 1.5 nanoseconds. For a typical UWB signal, power is spread over a band that exceeds 100% of the operating frequency.

Codes for Integration and Channelization Transceivers according to the present invention use pulse codes for integration gain, channelization, whitening, and notch filtering. This pulse-coded transmitted signal consists of a shifted (time-hopping) and inverted (bi-phase) copy of the underlying ultra-wideband short pulse.

This time shift and phase reversal is achieved by a combination of analog and digital circuits such as fixed and programmable delays, phase inverters (hybrid tees), splitters and combiners, GaAs switches, and digital circuits that generate control. To be realized.

The template of code h (t) that is bi-phase time hopping can be expressed as the addition of weighted and shifted impulses as follows.

[0179]

[Equation 27] (31)

Here, T _n is the relative position of chip n, N _c is the length of the code, and the code coefficient is h _n ε {−1,1} in the case of biphase (antipodal) operation. Is. Code h (t
) Is the frequency domain representation of

[0181]

[Equation 28] (32)

The pulse code is a collection of replicas (shifted and weighted by ± 1) of the underlying pulse from the short pulse of equation (26) and the code template of equation (31) as follows: It is formed.

[0183]

[Equation 29] (33)

The frequency domain representation of the pulse code shown in equation (33) is as follows.

[0185] P (ω) = H (ω) S (ω) (34)

Treating the data as equidistant impulse sets, they have the following time and frequency domain representations.

[0187]

[Equation 30] (35)

Temporarily limiting the system to only bi-phase modulation of the pulse code of equation (33), the transmitted signal becomes:

[0189]

[Equation 31] (36)

[0190]   It has the following spectrum:

[0191] X (ω) = D (ω) P (ω) = D (ω) H (ω) S (ω) (37)

Assuming the data is white or is whitened by the process, the spectrum of the transmitted wave will be that of the coded pulse of equation (33). With proper selection of the pulse code delay and weight, the pulse spectrum and bandwidth are preserved, essentially resulting in the short pulse transmit waveform of equation (26). That is, the spectrum is nominally white over the operating bandwidth.

The UWB receiver used in the present invention faces an out-of-sync transceiver clock, an out-of-sync transceiver code, a clock that is not exactly the same frequency or dissociates as a function of time, temperature, and relative position. Will be done. The problem of synchronizing the transceiver clocks is, in short, finding an unknown time delay that corresponds to the maximum value of the correlation. Since the present invention uses a time-modulated signal and a correlation detector, the peak of correlation is a desirable observation point. Since the time-modulated pulse is narrow in time, the peak of the observed correlation is also very narrow. As already observed today, this narrowness gives the UWB communication system of the present invention the ability to operate in a multipath environment (by being able to decompose multipath).

The use of time-modulated and coded pulse sequences and correlation receivers, as well as the ability to resolve multipaths different from the use of multipaths, provides overall superior performance over other UWB and narrowband systems. The opportunity to win is provided. The improvement effect is to combine the energies of multiple signal propagation paths with each other.
It is considered that this is due to the coherent synthesis of the information of a plurality of signal propagation paths.

(Sliding Correlator DLL (Delay Locked Loop)) Usually, the entire correlation function is not executed in a single sequence. That is, after forming a single period of the correlation function, the slides are made in time series. Therefore, a sliding correlator is realized. The sliding correlator is used for clock acquisition, clock tracking, data detection, and maximum signal position scanning. This position changes dynamically due to moving objects in the environment, such as mobile applications, or movement between the transmitter and receiver. The sliding correlator forms the dot product of the received signal and the local code at different relative time delays (multiplying the signals and then integrating).

One way to obtain the clock (determine the unknown delay) is to increment the time delay by the code length while looking for the inner product with the largest absolute value, once the delay is selected, the sign of the inner product And magnitude are used as bit detection statistics. And
Clock tracking is implemented by a delay locked loop (DLL). Since the correlation is symmetric, the receiver can track the transmitted clock using the difference between the lead and delay dot products, and when the receiver is synchronized, this difference will be zero. Otherwise, the difference will be positive or negative depending on the timing advance or delay.

Radio Frequency Interference From a UWB system perspective, noise in an urban environment is dominated by narrowband interference. Since the UWB signal is “short in time” and “long in frequency”, it is easy to distinguish it from the conventional narrow band signal which is “long in time” and “short in frequency”. FIG.
Shows the power spectral density at the Virginia Research Institute in Alexandria. Taking advantage of these signal type differences, we develop RFI (Radio Frequency Interference) extraction algorithms to improve UWB transmission. 19A and 19B show the results for the collected data of the developed RFI extraction algorithm. Again, RFI extraction can be performed by a radio front end that exhibits an impulse response that matches the transmitted wavelet combined with the inverted and time-shifted copy of the wavelet.

FIG. 20 shows a bi-phase ultra wideband transmitter 200 according to the present invention for transmitting a bi-phase time modulated UWB signal to a bi-phase time modulated coherent receiver 2001.
It is a block diagram of 0. This transmitter 2000 accepts data,
The encoder 2003 executes a source and channel encoding process on the data. In this encoder 2003, channel encoding is performed forward to optionally reduce the redundancy of raw data by conventional source encoding techniques and to improve the signal performance of data transmitted over a wireless communication link. Performs other source encoding schemes such as error correction and differential encoding. The output of the encoder 2003 is the digital encoder 2005.
And a length and a digital code for encoding the data supplied from the encoder 2003 with a specific user code for an individual user or service are generated. The length and the type of the code are, for example, Walsh-Hadamm.
It may be a conventional code such as those used in CDMA systems such as urd sequences. The output of the digital encoder 2005 is supplied to the biphase wavelet selector and the analog encoder 2007. The selector and encoder 2007 maps each output of the digital encoder 2005 to a particular analog encoded time sequence of wavelets. The bi-phase wavelet selector can select different wavelets from the set of wavelet candidates such as the positive and negative Gaussian wavelets described above. It is also possible to use higher level signals so that an array of signal systems with a large number of wavelets per group can be used. The analog encoder temporally disperses the biphase wavelet selected by the wavelet selector. The output of the selector and encoder 2007, after being provided to a preconditioning circuit 2009 capable of performing pass-filtering and cleaning of signals such as those disclosed in previously referenced US patent application Ser. No. 09 / 563,292, It is transmitted from the transmission antenna 2011 via the amplifier 2010.

The receiver 2001 receives the ultra wideband transmission signal 2013 from the transmitter 2000. The energy received at antenna 2013 (which may be any of the antennas described above) is passed through low noise amplifier 2015 to radio frequency interference RFI extractor 2017.
Applied to. This RFI extractor reduces the effects of narrow band noise prior to performing the correlation and detection processing. Then, the output of the RFI extractor 2017 is supplied to the RAKE 2021 including the n-arm. This RAKE receives the synchronization information from the synchronizer 2019 and supplies its output to the correlator 2025. Then, the correlator 202
5 outputs the output of RAKE 2021 to the biphase wavelet generator 202.
Correlation reception is provided by associating with 3. The output is provided to the detector 2027 upon detection of the peak of the correlation, and after the individual symbols have been identified, the decoder 2
It is output via 029. RAKE 2021 includes at least one arm for each received multipath component and an optional search channel for adaptively selecting an optimal set of multipath components.

A block diagram of transmitter 2000 is shown in FIG. The function of this transmitter is to generate the waveform of equation (36) above (again, shown below).

[0201]

[Equation 32] (38)

In this equation, the transmitted waveform is defined as the convolution of the underlying pulse s (t) with the data stream d (t) with the code h (t).

The transmitter receives data from the data input 115. Mathematically binary data d (
t) is represented by an equidistant impulse stream, one for each data bit indexed by k. FIG. 23 is an example of data d _k = [1,0,1,1,0,1,0] as one positive and negative direction impulse per data bit, {0,1} → {−1. , 1} is applied.

In this embodiment, two conditions are set for the data. The first condition is that the data has been previously whitened (ie scrambled so that it is not associated). This means that the data has been processed so that the spectrum is nominally flat (ie white). The reason for imposing this constraint is that the spectrum of the transmission signal (38) is a product of the pulse S (ω), the code H (ω), and the data D (ω) by the convolution theorem as follows.

[0205] X (ω) = D (ω) H (ω) S (ω) (39)

Both the code and the pulse are generated at the transmitter such that H (ω) and S (ω) are nominally flat over the bandwidth of the system, so the spectrum of the output is similar to that of the data. There is. That is, if the data is white for an interval, the output will also be white for the interval. If the data are constant, eg all ones, then d (t) will be a matrix of periodic unit impulses T _c , so that in frequency bins spaced by f _c = l / T _c Consider that it results in a line spectrum and all transmitted energy is concentrated in multiples of f _c , resulting in interference with narrow band users at those frequencies.

Referring to FIG. 21, the whitening encoder 117 applies this data scrambling process to the data from the data input 115. In addition, forward error correction (FEC) such as a combination of interleaving, Reed-Solomon block code, and convolutional code can be applied at this stage. In addition, FEC
S. was published by Prentice-Hall in 1983. S. Lin and D.L. Costello, Jr. "
Error Control Coding: Fundamentals and App
(ISBN 0-13-283796-X) and Prentice Hall (P) in 1988.
B. Sklar, published by rentice-Hall) "Digital Communications (Di
gital communication) "(ISBN 0-13-211939-0), and the contents of both of these books are incorporated herein by reference.

The second condition is that the encoder 119 differentially encodes the data. The reason for imposing this restriction is that it can be arbitrarily inverted depending on the transmission waveform or environment. As a result, the receiver does not know whether the inverted or non-inverted wavelet was transmitted, but it knows if the two wavelets are the same or the opposite, and differential coding allows the receiver to distinguish between adjacent pulses. You can recover the data in a related way. By differentially encoding the data prior to transmission, the phase information and pulse position (two degrees of freedom) can be used to generate the pulse code.

In this embodiment, the data is transmitted by the bi-phase modulation of the code, not the modulation of the carrier. The code generator 111 generates a circular stream of pseudo-random bits representing a digital code. Each cycle the sync signal 112 triggers the differential encoder 119 so that the data modulation on line 120 is aligned. Exclusive-OR gate 114 modulates the digital code on 113 and produces a modulation code sequence on line 121. Different seed variables can be input from the load port 131 to generate a user-specific code.

FIG. 22 shows another block diagram of a circuit having modulation capability.
These circuits generate a data modulated code independent of the pulse waveform as follows.

[0211]

[Expression 33] (40)

This equation shows that after generating the pulse code h (t) for each bit d _{k of the} data stream, the entire code is multiplied by d _k . The effect is to invert or not invert the code h (t) because d _k is −1 or 1, respectively. Most of these code generation and modulation are processed digitally, and can be implemented by a process by software installed on an ASIC or CPU. In these circuits, modulo 2 addition (negative exclusive OR) replaces conventional multiplication and {1, -1} is mapped to {1.0}. FIG. 23 shows a data stream having a time interval of T _c (or T _n ) between code symbols.

The coefficient h _{n of the} digital code is a binary sequence of 1s and 0s occurring at time T _n from the start of the code. A sample 7-length code is shown in FIG. The coefficients in this case are evenly spaced Tp seconds apart, so Tn = nTp. Figure 2
5 shows a data modulation code as a result of applying the equation (40) to the data and code in FIGS. 23 and 24.

Referring to the block diagram of FIG. 22 and equation (40), the code coefficients are stored in memory 40 and addressed by counter 30. This counter 30
Produces a code index n and counts from 0 to Nc-1, which is the length of the code following clock pulse 2. Counter 30 is incremented once for each coefficient of code at intervals of T _n seconds. Then, each time the counter is incremented, the corresponding coefficient is ejected from the memory on line 13.

In equation (40), one codeword is applied to each data bit. These code words are synchronized by the parallel-serial conversion register 10 and the word counter 42. Upon completion of counting a single codeword, counter 30 asserts on line 41 which signal register 10 outputs one bit. When asserted, line 41 also signals the word counter 42 to advance the counting of shifted out data bits. This counter is programmed to count up to the bitwise width N _p of the (whitened and differentially encoded) data word. This counter asserts on line 12 which signal register 10 will load another data word after N _l advancements. For example, it is possible to calculate the code coefficient during operation by a linear feedback shift register.

Referring to FIG. 21 again, the code coefficient and the data bit are exclusive OR (XOR).
The code-modulated data y (t) presented to gate 114 is generated to produce line 1
21 is presented in the pulse forming network. Switch 107 is line 121
In response to the modulation code of, the positive (non-inverted) or negative (inverted) wavelet is selectively output on 108. Positive wavelet generator 103 and negative wavelet generator 104 generate wavelets in response to transmit clock signal 102 and provide outputs on lines 105 and 106, respectively. This wavelet shape is
Selected by external control logic via line 132.

Circuits that generate short pulses include avalanche transistors, step recovery diodes (SRDs) in comb filter circuits, and high speed discrete logic and transistors. In an embodiment according to the present invention, a circuit consisting of individual logic gates and passive delay lines is used to generate short pulses. 26 is a schematic diagram of a differential ECL implementation for generating wavelets approximating a Gaussian first derivative, and FIG. 27 is a timing diagram corresponding to FIG.

FIG. 26 shows a differential ECL implementation of a wavelet generation circuit. Clock 300 provides clock pulses to buffer 304 and inverting buffer 306 via lines 302 and 303. And the respective buffers 304 and 30
6 passes clock pulses through delay lines 310 and 308, respectively. Delay line 308 has a longer delay effect (X) than line 310. The outputs from these delay lines are coupled to AND gate 316 via lines 312 and 314.
Is supplied to. The output of this AND gate is output to the buffer 32 via line 318.
0 and a second set of inverting buffers 322 to provide delay lines 310 and 3
It is passed to a further delay line set 324 with the same delay (optional) as 08. Each output of this delay line is split to provide non-inverting and inverting versions along with its counterparts via buffers 332, 334, and 336. The outputs of the respective buffers are the lines 340, 342, 129,
And 346 to adder circuits 348 and 350.

Each line 340, 342, 129, and 346 has a pulse 360 shown in FIG.
, Etc. to provide different components that are positive or negative halves of individual pulses such as.
The outputs of the adder circuits 348 and 350 are the lines 352 and 354, respectively.
Is supplied to the switch 356 via. In the control circuit 358, the switch 356 is the first
Adder circuits 348 and 350 are provided to supply a biphase waveform of a predetermined phase "0" when set to 1 and to supply the opposite biphase wavelet when the switch is in the opposite situation of "1". The switching process for switching the output of is executed. FIG. 28 corresponds to FIG. 26 and includes 700, 702, 704, 706, 71.
0, 711, 712, 714, 716, 107, 726, 722, 720, 72
4, 728, 734, 748, 750, 752, 756, 758, 759, 76
0, 762, 777, 776, 774, 772, 770, 768, 766, 76
Each element of 4, 778, 780, 781, 782, 786, 784, and 788 corresponds to a similar structure shown in FIG. However, as shown, four channels are implemented and also include a shape selector line 739 used to switch different input signals with switches 736, 738, and 737.
By controlling the switches in this way, different pulse shapes can be used and polyphase signals and M-ary signals can be used.

FIG. 29A shows the timing diagram of FIG. 28 with the switches selected to produce the Gaussian second derivative. The switch 784 selects the polarity of the transmission wavelet according to the data modulation code of 786. Switches 736, 738, and 737
(FIG. 28) is driven to select the 0th, 1st, or 2nd derivative of Gauss. In the block diagram of FIG. 28, signal 2 is delayed by different line lengths and input to ports 104 and 105 of AND gate 106. The timing diagrams shown in Figures 27 and 29A show ideal waveforms for clarity, but the actual "rise and fall" time of the device produces the illustrated "filtered" output waveform. To be done. Usually, the transmitter and receiver waveform functions are not the same, and the wavelet shape used in the receiver is usually the derivative or Hilbert transform of the shape used in the transmitter.

A feature of the present invention is the ability to concatenate codes, and it is possible to implement two codes of different technologies. Referring to FIG. 21, the analog code generator 109 is implemented with a wideband microwave component, which does not have semiconductor technology limitations (although it is possible to use a semiconductor embodiment). Therefore, the transmitter according to the invention is able to generate codes consisting of both digital and analog parts. The digital code h _d (t) and the analog code h _a (t) are given by the following equations.

[0222]

[Equation 34] (41)

The effect of formulating equation (40) is to replace h (t) with a coupling code as follows.

[0224]

[Equation 35] (42)

The system can be operated with digital code only or analog code only. The purpose of the analog code is to produce a waveform with rapidly occurring pulses that can be conveniently processed digitally. The replica of the analog code replaces each pulse of the digitally generated code on a high-definition time scale. A code suitable for this analog part has a low autocorrelation side lobe like the Barker code. The reason for this is that the sidelobe structure of the joint code stores all combinations of main and sidelobes of the component code. On the other hand, if a circuit that can directly generate a wavelet from a digital code is used, the analog code is unnecessary, and FIG. 29B shows such a circuit.

FIG. 29B is a block diagram of an exemplary direct pulse forming integrated circuit 2900. One example of this integrated circuit is CMOS, and pulse forming integrated circuit 29
00 generates a piecewise continuous output waveform at a predetermined multiple or fraction of the clock pulse rate. The data input to this integrated circuit is supplied to the built-in inversion control circuit 2902. The inversion control circuit 2902 may be implemented as a separate logic gate. A user-specific code sequence of N code chips is input to the "code input" pin. This N code chip is buffered in the register 2901 and is registered in the register 2901 at a rate proportional to the clock rate (multiple or fraction thereof).
Is output from the clock. In one embodiment, a preselectable number of code chips is applied to the data (to set a predetermined number of data bits in the wavelet).

The data and output of register 2901 is applied to inversion control device 2902,
If the data bit is in the first logic state, the code sequence is inverted, and if the data bit is in the second logic state, the code sequence is passed as is. When using a multi-level wavelet (for example, a wavelet selected from N sets (N is a number of 3 or more)), a transmission wavelet can be selected by data of adjacent bits. In the present invention, a (inverted or non-inverted) code sequence is applied to a piecewise continuous wavelet generator 2903, where a logic transition edge in the logic circuit is used to invert control circuit 290.
A wavelet corresponding to the logic level pulse supplied from 2 is generated. Since the implementation of this embodiment operates in a limited power spectral density environment, the low CMOS voltage level on output pin 2904 can be directly supplied to antenna 2905.

FIG. 30 shows a schematic block diagram of a circuit for generating an analog code. The analog code generator of FIG. 30 includes a combiner 101 as shown in the drawing.
6 and 1020 with four individual lines 1004, 1006, 1008,
And 1010 to provide a 1: 4 splitter 1016 and an input 1 supplied to it.
00 is included. The combiner output 1040 provides three pulses, as shown in FIG. 31, and the combiner 1020 output provides a single pulse on line 1012. The respective signals are synthesized by the non-inverting and inverting inputs of the adder circuit 1060, and four pulses are supplied to the line 1080 as a synthesized signal. As shown in FIG. 31, the third pulse is an inverting pulse.

FIG. 31 is a timing diagram related to FIG. The digitally modulated wavelet pulse code of signal 1000 is input to splitter 1002. Although the implementation of the present invention uses a passive power splitter, a resistive divider, active network, etc. could be used instead, depending on cost, size and power. Select line length L ₁ ~L ₄ power splitter output from 1004 to 1010, temporally delaying the pulse. The line length required to generate the delay T _n in (b) of the figure is as follows.

L _n = nε _r T _d , n = 1 ,. ． ． , 4

Here, ε _r is the propagation velocity in the medium, and requires a delay that is a multiple of T _d . Delays are usually not limited to common multiples. All pulses in non-inverted time slots are added in power combiner 1016, and all pulses in inverted time slots are added in power combiner 1020. The inverted pulse is then subtracted from the non-inverted set by hybrid 1060 and the difference is output on signal line 1080.

Referring again to FIG. 21, the signal is filtered 122 before final amplification and transmission.
It is also possible to suppress the frequency band energy outside the operating band of the receiver by performing band pass filter processing at. Further, the output signal is subjected to notch filter processing by the filter 124,
Other narrow band signal energies can be removed if desired. It should be noted that another way to generate a notch in the output spectrum is to configure the code h (t) so that the unwanted frequencies are attenuated by the pulse code p (t) = s (t) * h (t). It is to be. Antenna 128 is driven by amplifier 126 and broadcasts the wavelet sequence.

FIG. 32 shows an example of a programmable distributed analog code using a grating structure that allows the transmission lines to be implemented on a coupler and a printed circuit board. The line length is minimized according to the binary length formula.

FIG. 33 shows an example of a programmable distributed analog code that uses an inverting amplifier instead of a hybrid coupler to obtain programmable polarity.

FIG. 34 shows an example of a programmable distributed analog code using a tapped transmission line and active network suitable for embedding in a monolithic integrated circuit.

FIG. 35 illustrates an embodiment of a transceiver sharing the same distributed analog code hardware mechanism 924, where the distributed analog code structure is bidirectional.
7 illustrates an embodiment in which a common antenna is switched between a transmitter and a receiver. This feature is useful when embedding an amplifier within a programmable distributed analog code structure.

Turning now to the receiver part, the receiver is shown to have a sliding correlation for accommodating short pulse waveforms as shown in the two receiver embodiments of the block diagrams of FIGS. 36 and 37. Implements an instrument delay lock loop. The difference between these two receivers is the position where the analog code is inserted. That is, in the case of FIG. 36,
The analog code acts on the wavelet sequence entering the "LO" of the mixer / multiplier correlator, and in the case of Figure 37 the analog code is used as a compression matched filter. The analog code in the transmitter spreads a single pulse into a pulse sequence, whereas the analog code on the receiving side does the opposite, ie compresses the pulse sequence into a single pulse.

This receiver has two main operation modes: (1) acquisition and (2) tracking and detection. The sliding correlator DLL implemented in this system will be described below.

FIG. 36 is a receiver architecture used in the present invention. This receiver may be housed in a single device together with the transmitter of FIG. 21, for example, to implement the transceiver function. First, the received signal is received via the illustrated planar antenna 200. This antenna may be of the type disclosed in the aforementioned US patent application Ser. No. 09 / 563,292. The energy of the antenna 200 is supplied to a low noise amplifier 201 which receives an adjustment signal from an acquisition / tracking / detection / RFI / AGC / control circuit 203 via an automatic gain control (AGC) line 202 as shown. The output signal of the amplifier 201 is supplied to the RFI extraction circuit 203 which removes narrow band interference from the pass band of the receiver. The interference of interest is narrowband interference such as mobile phone transmissions and wireless communications within office or home environments. And
This output signal is supplied to splitter 205 via line 204. This splitter, which follows the RFI extraction circuit 203, produces the individual parts of the signal,
Capacitor 208 from line 206 via diode 209 and line 207
There is one path for the received signal shorted at. This signal functions as an RF-AGC line digitized by the A / D converter 236 before being supplied to the acquisition / tracking / RFI / AGC control circuit 234. The control circuit 234 identifies the magnitude of the received signal and other temporal and statistical characteristics associated with the signal in order to adjust the front end gain of the LNA 201 via line 202.

The individual delay lines 210, 211, 212 and the mixers 213-215 in the associated lines 219-221 from the splitter 259 correspond to the delay lines 310-312 and mixers 320, 392, 390 shown in FIG. This will be described later. The outputs of the respective mixers 213 to 215 are amplifiers 2 which feed the signals via lines 223 to 225 to integration and dump circuits 226 to 228.
16-218. The integration / dump circuits 226 to 228 are connected to the control line 2
Via 60, as shown, synchronous load code / length N digital code mechanism 2
A timing signal is received from an integration / dump timing circuit 229 which receives a control signal via 45. The output of the integration / dump circuit is the A / D converter 230-2.
Acquisition, tracking, detection, RFI, AGC control circuit 2 for detecting different symbols digitized by 32 and transmitted over a wireless UWB channel
34. Then, the detection data is supplied to the decoder 237 from the output port of the acquisition / tracking / detection / RFI / AGC control circuit 234, and after being differentially decoded, processed by the data whitening FEC decoder 238 in the I / O circuit 239. And output as output data 240.

The control function of the controller 234 includes the respective wavelet code generators 28.
In addition to providing output waveforms used by 2 and 264 to trigger wavelet generation in each RAKE arm, output control lines for adjusting the amount of phase delay in phase delay circuits 241, 262, and 280 are also included. Furthermore, the receiver of FIG.
Two arms of the AKE receiver are also included (although the arm fed by delay line 270 implements the search channel). The arm having an input port on line 281 is connected to a wavelet generator 282, which provides the output via line 283 to the integration A / D dump timing circuit 284.
This circuit feeds the integration and dump circuit 289 via line 286. The output of the integration / dump circuit 289 supplies a signal which is digitized by the A / D converter 287 which receives the sampling input via the line 285 as shown in the figure, and this signal is acquired / tracked via the line 288.・ Detection ・ RFI ・
It is supplied as feedback to the AGC control circuit 234. Similarly, integration / A /
The D-dump timing circuit 284 also supplies an input to the controller 234.
The generator 264, the line 271, the timing circuit 265, the line 272, the integration / dump circuit 266, the line 273, the A / D converter 267, and the line 274 provide the same function to the second arm of the RAKE (however, it is not necessary). , This arm can also be used as a search channel).

For example, the output of wavelet code generator 282 is provided to mixer 294 via line 293 and, as shown, the signal from splitter 205 provided via line 297, delay line 296, and line 295. To be mixed. In this way, it is possible to mix the output of the wavelet generator and the delayed version of the signal received via the antenna 200, the amount of delay being influenced by the delay line 296 and the phase delay 280 under the control of the controller 234. Receive. The output of mixer 294 on line 292 is then provided to amplifier 291 via line 290 to provide the signal to integration and dump circuit 289. Similarly, the signal for the second RAKE arm is provided from splitter 205 via line 298, delay line 270, mixer 269, line 275, and amplifier 268. The clock 235 supplies timing to the controller 234.

The main receiver wavelet code generator is supplied with the clock signal from the controller 234 and also receives via line 243 the load code for the length N digital code which is also used in the transmitter. This code is provided through the gate 246, as shown, and the generator 251 and lines 254 or 252 and 2
Control signal switch 255 for selecting positive or negative wavelet by 53
Is supplied to. The sync signal line 244 is connected from the encoding circuit 245 to the integration / dump timing circuit 229. Main reception wavelet code generator 242
Selects a similar wavelet code as in the case of the transmitter of FIG. 21, and the code is the analog code circuit 257 and the load line 25 as in the case of the transmitter.
It is further modified by 8.

The receiver embodiment of FIG. 37 is similar to that of FIG. 36, except that an analog code is added immediately after the RFI extraction process to implement the compression matched filter. In the case of the embodiment of FIG. 36, the analog code acts on the wavelet sequence input to the "LO port" of the mixer / multiplier correlator.

FIG. 38 is a delay locked loop (DLL) used in the receiver of this system.
It is a detailed block diagram of a circuit. Amplification, filtering, optional R
After FI extraction, the received signal is input to the DLL on line 300. Splitter 4
00 divides the received signal into three copies, each delayed by a different amount. L
+ Y, LY, and L line lengths are used to form these lead, delay, and on-time signals for the DLL, and the difference in these line lengths causes the on-time signal to be pulse code autocorrelated. A time delay selected at the time of acquisition has been realized, which is arranged at the maximum value of the function and symmetrically arranged the leading and delayed signals before and after it. What is called the "local oscillator" is in this case the received wavelet code on line 10r. Although unique to the present UWB high speed communication system, this signal is different from the transmit waveform to represent the transmit effect of the antenna. This signal is generated similar to the transmit system except that it drives the data line to a logic low state.
Bandpass and notch filtering similar to that used in the transmitter can be applied to the signal 10r in order to improve the noise discrimination capability, which has the effect of improving the integrity of the filter. Similar to the received signal, the 10r local code is also split into three by splitter 402, where the line length and path delay are kept the same for mixers 320, 390, and 392. This is so that the lead, delay, and on-time dot products formed by the mixer and gate integrator can all operate with the same control signal.

The process of acquisition is to find the time delay that maximizes the dot product of the on-time signal and the local code. Received signal 300 is delayed by on-time delay 312 and then input to the RF port of mixer 390 and the local code is applied to the LO port of the mixer. The resulting product is the on-time IF signal, which is input to the gate integrator 380. The integration control signal 403 for the integrator is synchronized by the controller 500 so that integration begins when the local code arrives at the input of the integrator. Then, when the local code ends, the controller 500 issues an encode command to the A / D converter 370 via the control line 404. This completes the formation of the inner product, and the value is output to the digital line 405. The controller 500 issues a dump signal to the integrator, clears the integrated value and prepares the integrator for the next dot product. This gate integrator can be made using ping-pong technology that allows continuous time gate integration.

FIG. 39 is a block diagram showing an implementation of a continuous time integrator. The digitized on-time dot product value is input to controller 500 (FIG. 38) via line 405 (FIG. 38). At the start of the acquisition process, controller 5
00 stores this value. Then, the controller 500 issues a forward command to the phase delay circuit 520 and increments the count of the number of forward steps already applied. Note that the controller also stores the number of advancements corresponding to the dot product just collected.

The effect of the delay circuit 520 (520a and 520b) is to slide the local code in time with respect to the received signal.

40 and 41 are block diagrams of circuits having the ability to slide a clock. The circuit of FIG. 40 operates by raising or lowering the frequency of the reference oscillator for a short period of time to slide the phase of the clock, while the one of FIG. 41 operates by directly programming the delay time. In either case, the start time of the code generated by the circuit that implements FIG. 22 is changed.
The controller 500 continuously advances the phase with respect to the on-time signal to obtain dot products and their associated advance counts. Then, each inner product formed is compared with the one stored immediately before, and if it is larger, it is replaced and its advance count is recorded. If the total amount of time sliding the local code equals the width of a single code, then the code has made one rotation, stopping the acquisition process after several rotations of the code.
The unknown phase between the transmitter and receiver oscillators and codes is interpreted as the maximum detected on-time dot product delay. At this point, the system switches to tracking and detection mode.

Referring back to FIG. 38, like the on-time signal, the inner products of the lead and delay are also formed by delays 310 and 311 and mixers 320 and 392, respectively. Integrating and dumping circuits 330 and 333 perform the integration of these signals, which are synchronized with the local code by controller 500 in a manner similar to the on-time signals. Once the dot products are formed, circuits 340 and 342 obtain their absolute values.

These differences are formed by the adder 350 and digitized by the A / D converter 360 based on the encode command from the controller. The timing of this encode command is set so that a sufficient time is ensured for the difference between the inner product of the lead and the delay to propagate to A / D.

The lead and delay signals are formed by symmetrical time delays before and after the on-time signal. Since the correlation function is also symmetrical, these values will be equal if the local code is synchronized with the received signal. Upon detecting a non-zero value, the controller
The phase is moved forward or backward to bring the error to zero. As a way to improve the SNR of this error signal, it is also possible to add many of these before making a forward or backward decision.

Detection is performed by window comparison of on-time dot products. Values greater than zero are mapped to one and less than zero are mapped to zero. Similarly, erase zones may be added. The detection data is subjected to differential decoding, white decoding, and forward error correction.

Referring to FIG. 36, the “search” channel, also referred to as the second RAKE arm, shown at elements 262-270, continuously scans for stronger multipath signals than the one (or more) currently in use. is doing. When 234 detects a stronger correlation peak, it adjusts the phase delay 241 or 280 to adjust the main channel (ie, the auxiliary RA).
KE channel) and track its larger peak. This process allows the system to operate in dynamic multipath conditions.

The first RAKE channel, shown at elements 280-297, is used and the second strongest signal is tracked to add the next strongest correlation to the main channel signal. This addition is the first in a time domain RAKE filter that takes advantage of multipath to improve BER performance. That is, although only one RAKE and one search channel are shown in FIG. 36, both RAKE channels can be used as RAKE channels, or more RAKE channels can be used to process various numbers of multipath signals. It is also possible to include (one RAKE channel per multipath signal).

As for the conventional RAKE receiver, in 1996, Prentice Hall (Prenti
ce-Hall) issued by V. V. Garg. And Wilkes
By "Wireless and Personal Communication
Systems) "(ISBN 0-13234626-5), pages 151-152, the entire contents of which are hereby incorporated by reference. The RAKE receiver has the ability to decompose into separate fading channels and cross-correlate the received signal with multiple time-shifted versions of the transmitted signal. As a result, the respective signals received via the different arms of the RAKE receiver can be combined by the diversity combiner to provide a single output.

FIG. 42 is a graph showing the amplitude of the signal output of the correlator in the receiver of FIG. 36 versus time. This correlator is formed in the control circuit 234 and, as shown in the figure, the peak of the correlation is very narrow in time, and therefore it is an effective indicator when the correlation is actually obtained.

22 to 24 describe another embodiment for supplying the aforementioned jittered clock signal.

Most communication systems (narrowband or other types of ultra-wideband communication systems)
In general, multipath causes performance degradation. The problem with such a system is that when there are many resolvable channel links between the carrier and the receiver, and the signal energy of different channels in a time-continuous signal overlaps with another,
That is, the signals are often added in different phases. This phenomenon can be seen, for example, in the comparison of FIGS. 46 and 47. Figure 46 shows this UWB
Figure 3 shows a time domain signal of a pulse-by-pulse discrete time modulated signal for the system. As shown in the figure, the pulse received from one propagation path (propagation path 1) has a different reception time (and possibly direction) from the pulse of the propagation path 2 which is regarded as an inverted propagation path. However, the time modulation characteristics of the UWB wavelet signal according to the present invention make it possible to decompose two different propagation paths, and once decomposed, it can be considered that the energies of the different propagation paths are related to each other. , RAKE
It is possible to overlap the energy of different propagation paths by
As a result, the energy can be coherently added to improve the signal reception performance.

On the other hand, FIG. 47 shows a problem associated with a conventional signal system using a narrowband signal having a time continuous signal (a plurality of carriers). As shown, the two trajectories in FIG. 47 merely show an example of a normal fade that occurs in a communication channel of a conventional narrow band system, and a very deep fade occurs in signal reception (300M).
Near Hz). With such deep fades, signal performance drops dramatically, resulting in a dramatic increase in bit error rate and degraded channel quality. For example, in the case of a system such as a mobile phone, the signal may disappear during a signal fade and the call may be disconnected.

One technique to overcome this problem is to increase the signal level of conventional transmission systems. However, doing this would have a significant impact on adjacent channel interference and interfere with the communication of other users of other systems.
That is, the feature of the present invention is that the multiple signal propagation paths can be decomposed, and therefore can be coherently combined with the direct communication propagation path, and as a result, expansion (that is, addition) by other multipath signal energy ) Is to actually use multipath in an indoor environment, such as an office environment, where it is possible to enhance the actual received signal to benefit.

FIG. 48 shows a method of incorporating the transceiver function according to the present invention into a “wireless ASIC” (ASIC is an abbreviation for application-specific integrated circuit). Wireless ASI
C performs the wireless transmission function to interface with different applications as part of the stacked protocol architecture.
This wireless ASIC performs signal generation, transmission, and reception functions as a communication service for an application that transmits and receives data, such as a wired IO port.
For example, the specific application 5100 is a video signal from a camera in an office environment and provides a data stream to a wireless ASIC via a home application program interface (API) 5101. The home API 5101 receives this data and formats it for use by the next component 5102 of the communication protocol stack (in this example, this component corresponds to Microsoft).
Universal Plug and Play (UPnP) or Sun Microsystems (Su
n Microsystem) JINI application). These features allow the home API to provide the input data stream used by the wireless ASIC.
The data acquired from the application 5100 via 5101 is formatted. And, as shown, this data is Media Access Control (MAC)
It is formatted by components and physical layer services that transform and provide data to the signal antenna.

In this flow, the wireless ASIC can be regarded as the base of a protocol stack similar to the OSI protocol stack. Thus, the wireless ASIC can serve as the basis for various physical layers to extend near field communication and can provide wireless communication capabilities to any device. That is, a wireless ASIC can be used to route information for various devices, such as home refrigerators and power meters, or digital devices such as computers and wireless telephony systems in office environments. When distributed in the home, it is not necessary to wire these various devices together.

An example of this wireless integration is shown in FIG. FIG. 49 shows various devices for home use such as games, home automation, security functions, audio devices such as radio, hall-house audio, voice applications such as telephony and teleconference, Internet applications, or video applications such as DVD applications. Shows the method used in. Each of these various devices may incorporate interconnects either by wire technology or using the wireless ASIC of the present invention. The residential gateway 5200 also receives and fuses information in accordance with the present invention and provides the wireless ASIC of the present invention with concentrated data to an external source, eg, via a cable connection, digital subscriber line, or wireless microwave link. It has a built-in transceiver. As a result, an inexpensive wireless infrastructure within the home or office is built, providing a mechanism to route distributed data between devices or to the outside world.

Although FIG. 52 shows a home or office environment, the present invention can be implemented in a relatively small device such as an application specific integrated circuit, and thus may be used in a vehicle or in a portable device such as a palm computing device. It can also be built into the device.

In addition to imaging, the wireless ASIC can also be used for ranging applications (for vehicle or mobile applications). In range finding applications, radio can be used to determine the distance from a particular transceiver by observing the return velocity and position of the reflection from a particular object. As a result, the wireless ASIC of the present invention can be incorporated as part of a home security system to provide a ranging sensor for intruder detection and monitoring.

In addition, the wireless ASIC is designed to trigger audible or visual alerts or take corrective action when a moving object is detected approaching other objects to dangerous levels. Alternatively, it can be built into a fixed structure that the vehicle approaches.

Wireless ASICs can also be used in test equipment for antenna testing and channel sounding as an inexpensive way to determine suitability of UWB for applicability to antenna testing and other communication scenarios. In the case of a communication system, a wireless ASIC may, for example, provide a telephony interface to a public switched telephone network, eg, a wireless security system, a wireless factory setting for process control, a wireless public for using a variety of different phones. It can also be built into a branch exchange or wireless local loop application.

The wireless ASIC can also be used as a wireless alternative to various high speed interfaces. For example, the wireless ASIC can be used as an interface for a local area network such as 100BaseT, Firewire (IEEE1394), and universal serial bus. Furthermore, wireless AS
The IC can provide a wireless link to facilitate communication in digital video recorders, video cameras, DVDs, MP3 players, general purpose computers, etc. As a result, wireless provides a simple and convenient mechanism for connecting to nearby devices and the Internet.

Further, this wireless ASIC is a home telephone network association (home P network).
NA), home plug (runs on the power line at 10 Mbit / s), and 1
UW spanning 0 or 100 BaseT Ethernet networks
It may be included as part of B WPAN. To the present inventor's knowledge, it is difficult and time-consuming to transmit information from an information recording device to another device because of the restrictions associated with conventional systems such as computers and digital video recorders. Thus, the wireless ASIC can be used to facilitate file transfer, such as digital still cameras (DSC), digital light projectors (DLP), gaming devices,
It can be used to convey voice over IP, MP3 audio applications, storming video, or personal digital assistant information.

The ASIC UWB radio according to the present invention can be used to extend a local area network with a high speed broadband personal area network.
In addition, the WPAN topology can be used to save subscription costs using free multimedia information sources. That is, the wireless ASIC can function as a replacement for subscription-based services such as wireless cellular. Using conventional interface, digital camera to kiosk data transfer, camera to camera transfer, camera to printer data transfer, camera to PC data transfer,
Alternatively, the wireless ASIC can provide a function such as transfer from the camera to the set-top box for later viewing on a display device such as a television monitor.

It will also be appreciated by the inventor that a transceiver using a wireless ASIC according to the present invention can function not only in conjunction with the wireless system according to the present invention, but also in multiple modes of operation. For example, one embodiment of the present invention includes a receiver mechanism that receives signals transmitted according to the "Bluetooth" specification.
For example, “Bluetooth System Specification (Sp
ecification of the Blue Tooth System) version 1.0B Volume 1 "Core Specification" and Volume 2 "Profiles Speci"
fication) ”. All the contents are included in this specification by this citation. To this end, in order to provide "universal" wireless operation, U
The present invention incorporates a transceiver that operates not only in the WB operation mode but also in other wireless communication modes.

In another embodiment of the present invention, a transceiver (ie, wireless AS) is provided in a portable computing device such as a wireless telephone, personal digital assistant, or palmtop computer.
IC) is embedded. As a result, an individual using the portable computing device can quickly and easily transfer information stored on the portable computing device to another transceiver (or receiver). For example, mobile communication devices may include wireless access mechanisms, calendar-related mechanisms (eg, sending one user's calendar content to another user to identify mutually agreeable dates and times for certain events), map distribution mechanisms (office or A convenient method of explaining how to reach a predetermined place such as home to another individual), a personal profile distribution mechanism (similar to transmission of electronic business card information), etc. can be provided.

In one configuration, an office environment is established by a wired network that includes wireless transceivers that cooperate with UWB bi-phase pulses according to the present invention to receive the information transmitted by them. As a result, the data collection radio can be used as part of a personal area network (PAN). That is, the wireless ASIC
When the information originates from one of the two, the network with the various data collection transceivers distributed in the room or building rebroadcasts its energy for reception by other devices.

Furthermore, this communication network enables interconnection of adjacent buildings. That is, the networks with broadcast transceivers present in the room are interconnected by wirelines (typically FDDI networks). For example, a wireless transmission from a portable wireless ASIC inside a particular room or building is received, and the data received by the ASIC radio is transmitted over the wireline to an adjacent office space (typically, for example, in another town office, Some rooms have at least one wireless transceiver that sends to some offices located in different states. Once the transceiver receives the data on the other side, the transceiver rebroadcasts the energy by the ultra-wideband transmission according to the present invention for reception by other devices at the remote location.
This provides a wireless "portal area" that conveys information from one location to another.

Alternatively, in the case of a public area such as an airport, for example, rather than providing a specific telephone room (in this case, an individual must use a private telephone system provided as an airport facility) ), The airport may be provided with a wireless transceiver network for exchanging information with a user who uses the personal communication device for transmitting and receiving bi-phase communication information according to the present invention. As a result, the present invention facilitates the use of UWB communications, where an individual does not go to a particular location to communicate with a communication partner in a nearby area (with possible obstructions between them). Communication status is maintained.

One advantage of using UWB bi-phase wideband communication according to the present invention is that it has a much larger number of multiple access users compared to conventional cellular and other multiple access technologies that share a particular radio frequency spectrum. is there. For example, the UWB code provided by the present invention is 100 times longer than the conventional split spectrum code. Thus, millions of unique addresses can be assigned to different individuals in a relatively small space. As a result, all these millions of unique addresses can be used at the same time, unlike restrictive connections by round robin or spread spectrum systems.

Another feature of the bi-phase modulation system according to the present invention is that a significantly large code space is required for searching, and the signal appears smoothly in both space and time, which makes it difficult to eavesdrop. To be provided. Furthermore, there are no edges to search.

FIG. 50 illustrates a processor system 801 in which embodiments of the controller and application interface of the present invention may be implemented. The processor system 801 includes a bus 802 or other communication mechanism for transmitting information, and a processor 803 for information processing connected to the bus 802. Further, the processor system 801 is connected to a bus 802 and stores random access memory (RAM) for storing information and instructions executed by the processor 803 and other dynamic storage devices (eg, dynamic RAM (DRAM), static RA).
Main memory 804 such as M (SRAM) and synchronous DRAM (SDRAM)
Is also included. This main memory 804 can also be used to store temporary variables and other intermediate information during execution by the processor 803. Further, the processor system 801 may include a read only memory (ROM) 805 connected to the bus 802 for storing static information and instructions for the processor 803 or other static storage device (eg, programmable ROM (PROM), Erasable PROM (
EPROM)), and electronically erasable PROM (EEPROM)).

Further, the processor system 801 includes a magnetic hard disk 807 connected to the bus 802 and a removable medium drive 808 (eg, floppy (registered trademark) disk drive, read-only compact disk drive, read / write compact disk). Also included is a disk controller 806 that controls one or more storage devices that store information and instructions, such as drives, compact disc jukeboxes, tape drives, and removable magneto-optical drives. The storage device must have an appropriate device interface (eg SCSI (smal
l computer system interface), IDE (integrated device electronics),
E-IDE (enhanced-IDE), DMA (direct memory access), or ultr
a-DMA) can be used to add to the processor system 801.

The processor system 801 is a special purpose logic device (eg, ASI).
C (application specific integrated circuit) or configurable logic device (eg SPLD (simple programmable logic device), CPLD (comp
lex programmable logic device) and FPGA (field programmable gate a
rray)) can also be included.

Further, the processor system 801 is connected to the bus 802, and a CRT (cathode
A display controller 809 may be included that controls a display 810 that displays information to a computer user, such as a ray tube) or LCD display. This computer system includes a processor 803 for communicating information with a computer user, such as a keyboard 811 and pointing device 812.
It includes an input device provided by. For example, pointing device 812 may be a mouse, trackball, or pointing stick that conveys direction information and command selections to processor 803 and controls cursor movement on display 810. In addition, the printer can output a printed list of data structures / information stored and / or generated by the processor system 801.

The processor system 801 is responsive to the processor 803 to communicate with the main memory 80.
4. One or more sequences of one or more instructions stored in memory, such as four, are executed to carry out some or all of the steps of the invention. Such instructions may be read into main memory 804 from another computer-readable medium, such as hard disk 807 or removable media drive 808. It also uses one or more processors in a multi-processing array to
It is also possible to execute a sequence of instructions stored within. In alternative embodiments, hard-wired circuitry may be used in place of (or in combination with) software instructions. That is, the embodiments are not limited to a particular combination of hardware circuitry and software.

As mentioned above, the processor system 801 is for holding instructions programmed according to the present disclosure and for storing data structures, tables, records, or other data described herein. It includes at least one computer readable medium or memory. Examples of the computer-readable medium include compact discs, hard discs, floppy discs, tapes, magneto-optical discs, PROMs (EPROM, EEPR).
OM, flash EPROM), DRAM, SRAM, SDRAM, or other magnetic media, compact disc (eg CD-ROM), or other optical media,
Punched card, paper tape, or other physical medium with a hole pattern, carrier (
(Described below) or other computer-readable medium.

The invention is stored on any computer readable medium or any combination thereof for controlling computer system 801 and driving a device to implement the invention such that processor system 801 is a human user. Includes software that allows you to interact with (eg, print production personnel). Such software includes, but is not limited to, device drivers, operating systems, development tools, and application software. Such computer readable media further includes the computer program product of the present invention for performing all or part of the processing performed in an implementation of the present invention (when the processing is distributed). .

The computer encoding device of the present invention is a script, an interpretable program, a dynamic link library (DLL), a Java class, a complete executable program, etc., but not limited to, interpretable or executable. Any coding mechanism may be used. In addition, performance, reliability, and / or
Alternatively, parts of the process of the invention can be distributed to improve cost.

The term “computer-readable medium” as used herein refers to any medium that provides executable instructions to processor 803.
The computer readable medium can take many forms, including, but not limited to, non-volatile media, volatile media, and transmission media. Non-volatile media include, for example, optical disks such as hard disk 807 and removable media drive 808, magnetic disks, and magneto-optical disks. Volatile media also includes dynamic memory, such as main memory 804. Transmission media includes coaxial cables, copper wire and fiber optics, including the wires that comprise bus 802. Transmission media can also take the form of acoustic or light waves, such as those generated during radio wave and infrared data communications.

Various forms of such computer readable media may be involved in carrying out one or more sequences of one or more instructions for execution to processor 803. For example, the instructions may first be stored on a magnetic disk of a remote computer. The remote computer loads those instructions, which implement all or part of the invention, into a dynamic memory at a remote location and sends them by modem over a telephone line (or other communication channel, wired or wireless). The local modem of processor system 801 then receives the data over the telephone line and uses an infrared transmitter to convert the data to an infrared signal. Then, the infrared detector connected to the bus 802 receives this data carried by the infrared signal and outputs it on the bus 802. The bus 802 stores data in the main memory 80.
4 from which the processor 803 obtains and executes the instructions. It should be noted that the instructions received in the main memory 804 may optionally be stored in the storage device 807 or 808 before or after processing by the processor 803.

The processor system 801 also includes a communication interface 813 connected to the bus 802. The communication interface 813 allows a bidirectional UW to a network link 814 connected to another communication network 816 such as a personal area network (PAN) 815 or the Internet.
B data communication connection is provided. For example, this UWB communication interface 81
Reference numeral 3 may be a network interface card to be attached to the packet switched UWB compatible PAN. Alternatively, as another example, the communication interface 813 is
UWB connectable ADS providing data communication connection to corresponding type of communication line
L (asymmetrical digital subscriber line) card, ISDN (integrated s)
ervices digital network) card, or a modem.

Network link 814 typically provides data communication through one or more networks to other data devices. For example, the network link 814 is
An RF connection can be provided to another computer via the personal area network 815 (eg, PAN) or via a device operated by a service provider that provides communication services via the communication network 816. In the preferred embodiment, local network 814 and communication network 816 preferably use electronic, electromagnetic, or optical signals that carry digital data streams. Signals over the various networks that carry digital data to and from processor system 801, and over network link 814 and through communication interface 813, are exemplary forms of carrier waves carrying information. Processor system 801 can send and receive data, including program code, through networks 815 and 816, network link 814, and communication interface 813.

FIG. 51 is a block diagram of an embodiment of a transceiver according to the present invention. The transceiver uses an antenna 5101 for transmitting and receiving UWB signals according to the invention, which antenna 5101 is a transmit / receive (T / R) switch 510.
Connected to 3. In the case of transmission, the T / R switch 5103 is controlled by the T / R switching control signal and is connected to the transmitter arm of the transceiver (upper path shown in FIG. 51). On the other hand, when receiving, the T / R switch 5103
Controlled by the / R switch line, the energy from antenna 5101 is connected to the receive signal path starting from Rx filter 5109.

In the transmission operation mode, the first mixing tracking arm 5113 (MTA1
) Transmission mode operation is enabled. In the transmission operation mode, the waveform to be transmitted is passed from the transmission line to the transmission filter 5107.
It is an adaptive filter that adjusts the spectral energy of the output signal to be transmitted. Then, the output of the transmission filter 5107 is, as shown in FIG. 51, a power control device 5105 which receives a control signal from the remote controller via the control line UWB control.
Passed to. This power control device 5105 adjusts the transmission power of all the energy to be transmitted. In the normal operation mode, the transceiver shown in FIG. 51 operates in an environment where the power spectral density is adjusted. That is, the power control device 5105
Suppresses the power so that the power spectral density in a specific part of the band satisfies the power spectral density requirement. The output of the power control device 5101 is radiated from the antenna 5101 after passing through the T / R switch 5103 and the wireless UW according to the present invention.
A B waveform is generated.

On the other hand, in the reception operation mode, the energy from the antenna 5101 is T /
It is passed to the reception filter 5109 via the R switch 5103. This reception filter 5109 is an adaptive filter, and has narrow band interference (the one called "RFI" above).
Reduce the effect of. After filtering by this receive filter 5109, the received signal is passed to an automatic gain control (AGC) circuit 5111 which adjusts the overall level of signal energy prior to subsequent processing. The control signals for the reception filter 5109 and the AGC process are “RxF control” and “AGC control” as shown in the figure.
It is supplied from the signal line. Then, the output of the AGC device 5111 is divided into 3
It is distributed to three different mixing tracking arms 5113, 5115, and 5117. Each of these three mixing tracking arms can be independently controlled so that the same processing is performed so that each mixing tracking arm can detect and receive different multipath components of the received signal. That is, each mixing tracking arm 5113, 5115, and 5117 includes the following components. First, a 160 MHz clock is applied to each mixing tracking arm 5113, and other reference signals are also supplied based on the processing content. Then, a clock control signal, a code / phase control signal that becomes a digital control signal, and a pulse forming network control bus including a predetermined number of control lines are also input to the mixing tracking arm. These mixing tracking arms associate the received signal energy with a local version of the signal energy for subsequent detection processing. When operating in the receive mode of operation, the mixing enable signal configures the mixing tracking arm to operate in the receive mode of operation. During operation in the reception operation mode, the analog output signal is supplied to the A / D converter 5119 and the tracking signal is supplied to the tracking A / D converter 5121 as illustrated. Similar processing is performed in the mixing tracking arms 5115 and 5117 by the A / D converters 5123, 5125, 5127, and 5129.

FIG. 52 shows a block diagram of the mixing tracking arm 5113 of FIG. The block diagram of FIG. 52 is shown in FIGS.
D, FIG. 54A-54D, FIG. 55A-55C, FIG. 56, and the corresponding signal waveforms shown in FIG. Referring to FIG. 52, the pulse forming network 5207 includes a clock controller 5209, as shown.
And a UWB bi-phase pulse according to the present invention under the control of the code supplied from the code FIFO 5211 and the PFN data controller 5213. The clock controller 5209 receives the 160 MHz input signal reference and the clock control signal. Typically, the output of clock controller 5209 is a chip clock operating at 1.6 GHz, which is applied to both pulse forming network 5207 and pulse forming network data controller 5213. A code / phase control signal is applied to the code FIFO, which signal is specific to the particular user attempting to receive. The code FIFO simply feeds the data to the pulse forming network 5207 on a set of N (shown as 33) signal lines. The pulse forming control network 5213 is also provided with control inputs that determine which subset of codes will eventually be output from the pulse forming network 5207. Then, the code clock generated by the pulse forming network 5207 is delayed by a predetermined amount and supplied as the output signal write data and the write track, as shown in the figure. The series of pulses provided by the pulse forming network 5207 are provided to mixers 5203 and 5205, as described below.

As shown in FIG. 52, the incoming RF signal is divided into three individual signals, “lead”, “delay”, and “main”. The lead signal is applied to the negative input of adder 5202. The "main" signal is delayed by 30 picoseconds with delay 1 (although other delays could be used in the preferred embodiment). The third input is then delayed by delay 2 for 60 picoseconds (although it may be delayed by varying amounts), formed as a "delayed" signal and then applied to the positive input of adder 5202. It The adder 5202 is enabled by the tracking enable signal as in the tracking mixer 5205 as shown, and the operation of the mixer 5203 is enabled by the mixer enable control signal. Thus, as shown in FIGS. 53A-55C, the lead signal is 30 picoseconds ahead of the main signal and the delayed signal is 30 picoseconds behind the main signal. The output of this adder 5202 is
It is a collection of delayed and leading signals and will be referred to as "delay-leading".

54A-54D show the signal after it has been passed through the mixer and processed by the LO waveform. As shown in FIG. 54A, the LO waveform is probably out of time with the main signal. Therefore, as shown in FIG. 54B, the output mixer 5203
Indicates that a significant amount of energy is collected below zero V. Similarly, the LO signal and the error signal (delay-lead) are shown in FIG. 54C, and these signals are applied to mixer 5205. The output of this mixer 5205 is "ER
A signal labeled "R-MIX" is provided, which is shown in Figure 54D.
The output of mixer 5203 is provided by an output data signal called "ON-DATA". The output of tracking mixer 5205 is a signal called "tracking".

FIG. 55A shows an event where energy is not evenly distributed around zero V due to a peak-to-peak pulse width of 130 picoseconds and the local oscillator signal is too slow, and there is a negative DC bias. On the other hand, FIG. 55B shows the event where the local oscillator is directly aligned in time with the RF signal so that the total energy is concentrated to zero. FIG. 55C illustrates an event where the local oscillator is fast and there is a DC shift in the positive voltage direction.

FIG. 56 shows the mixer output when the tracking area is high, low, or correct. The mixer output is combined with the alignment error in picoseconds shown in FIG. If there is no matching error (error channel output shows zero V), maximum main channel output is obtained. However, if the alignment error deviates from zero picoseconds, the main channel output will fall below the maximum possible detection voltage. That is, the LO signal is the main R
The performance of the communication system is not optimal because it is not exactly aligned with the F channel.

FIG. 57 shows that when a match error above zero V or below zero V is observed, L
The PFN data controller 5213 instead asserts control via the control line "Sign / Phase" to guide the LO output of the PFN 5207 (Fig. 52) towards the optimal protection where O exactly matches the main channel. Is shown.

FIG. 58 is a block diagram showing the architecture of a transmitter according to the present invention for generating piecewise continuous UWB bi-phase transmission waveforms. The code generation mechanism (5800) generates a code unique to a particular user and passes the code along with the data to the pulse forming network 5801. The pulse forming network 5801 uses semiconductor technology to generate a piecewise continuous bi-phase UWB signal according to the present invention, the output of which is amplified by an amplifier 5803 and transmitted from an antenna 5804. Since the common clock is input to both the code generation circuit 5800 and PFN5801, both devices operate in synchronization.

The uppermost waveform (digital logic signal) in FIG. 59 indicates the code 110 input to the PFN5801. Although PFN5801 is formed on an integrated circuit, it may be in the form of ECL, CMOS, bi-CMOS, or silicon germanium. This PFN5801 generates a pulse for each data signal,
This pulse, as shown, is the "edge" 5910-591 of the digital logic.
Generated using 2. The lower two traces, labeled "ECL" and "new CMOS", also show the rise times inherent in the semiconductor circuits used to generate these waveforms. In the case of ECL, a rise time (edge) of 330 picoseconds is observed and a fall time of 330 picoseconds is also observed. Meanwhile, the new C
When using a MOS device, the rise and fall times of 100 picoseconds (or
(At least less than 200 picoseconds) has been observed.

To the inventor's knowledge, technological advances should continually reduce the rise and fall times of current and next-generation components. Therefore, by generating a piecewise UWB signal using the edge of the semiconductor logic circuit according to the present invention,
The bandwidth of the communication system according to the present invention will be improved as the temporal performance of the semiconductor device is continuously improved. FIG. 60 shows this, the slow EC of FIG.
Two L-waveforms and two spectrum graphs corresponding to the edges of the newer CMOS logic shown in FIG. 59 are shown. The spectrum corresponding to the slow ECL logic is
It has a predetermined bandwidth B ₁ , while it has three times the bandwidth in the case of fast new CMOS technology. Furthermore, by using piecewise continuous waveforms (see the bottom two trajectories) as shown in FIG. 59 to generate piecewise continuous waveforms, the performance of radios implementing the present invention has been shown to be Gordon Moore. ), It will rise with the passage of time. According to Gordon Moore's Law, every 18 months an integrated circuit doubles in speed and halves in size. Therefore, the wireless implementation according to the present invention is also realized by generating a waveform by such an integrated circuit in the present invention.
Every month it can be twice as fast and half the size. Unlike the conventional radio in which the performance is improved by the log of the processing power, in the case of the present invention, the performance is linearly improved as the IC performance is improved.

Those of ordinary skill in the art will appreciate that the mechanisms and processes described herein may be implemented using a conventional general-purpose microprocessor programmed according to the disclosure herein. It will also be apparent to those skilled in the art that skilled programmers can easily prepare suitable software coding based on the present disclosure.

Accordingly, the invention also includes computer-based products that include instructions that are provided on a storage medium and that can be used to program a computer to perform processes in accordance with the invention. This storage medium can be any type of disk such as floppy disk, optical disk, CD-ROM, magneto-optical disk,
ROM, RAM, EPROM, EEPROM, flash memory, magnetic or optical cards, or any type of medium suitable for storing electronic instructions are included, but are not limited to.

Obviously, many modifications and variations of the present invention are possible in light of the above disclosure. It is therefore to be understood that the invention may be practiced other than as specifically described herein within the scope of the appended claims.

[Brief description of drawings]

[Figure 1]   6 is a graph showing the amount of RF signal attenuation versus frequency as it propagates through different materials.

[FIG. 2A]   4 is a time waveform of a biphase pulse according to the present invention.

FIG. 2B   3 is a frequency waveform of a biphase pulse according to the present invention.

FIG. 3A is a time waveform of a bi-phase UWB waveform representing a “1” information bit according to the present invention.

FIG. 3B is a time waveform of a bi-phase UWB waveform representing “0” information bits according to the present invention.

FIG. 4A   9 shows a pulse position modulation (PPM) waveform showing an information bit of "1".

FIG. 4B   9 shows a pulse position modulation (PPM) waveform showing an information bit of "0".

FIG. 5A is a graph of a PPM waveform in the best case with “1” information bits.

FIG. 5B is a graph of a PPM waveform in the best case with “0” information bits.

FIG. 6A is a graph comparing a UWB signal and a narrowband signal at the same phase overlap.

FIG. 6B is a graph comparing a UWB signal and a narrowband signal in overlaps with different phases.

FIG. 6C   3 is a signal cycle waveform of both a narrow band signal and a UWB signal.

FIG. 7 is a plot of power versus distance showing a comparison of measured power and received signal power of both theoretical and jamming theoretical power.

FIG. 8 is a signal level vs. distance plot for a particular UWB transmission according to the present invention to show “sounding” of a particular channel.

9A is an amplitude versus time waveform of a signal resolvable by a UWB signal of the present invention in terms of a channel in which multipath is present. FIG.

FIG. 9B is a plot of signal level versus frequency for a fading channel induced by multipath in a “traditional narrowband” communication system.

FIG. 10 is an exemplary energy versus frequency plot comparing the bandwidth of each of a narrowband signal, conventional spread spectrum communication, and the UWB communication spectrum according to the present invention.

FIG. 11   It is an exemplary bi-phase signal used as a waveform for information transmission of the present invention.

FIG. 12A is an amplitude vs. time plot showing a position alignment method of a signal wavelength sequence according to the present invention for providing an aligned pulse sequence as part of a symbol set according to the present invention.

FIG. 12B is a plot of power spectral density versus frequency of a random noise signal measured in the present invention.

FIG. 13A is a diagram showing how multipath reflection occurs due to an obstacle in the communication channel.

FIG. 13B shows an example of a time waveform generated by the multipath generated in the situation of FIG. 13A.

FIG. 14   3 illustrates a detection circuit that can be implemented according to the present invention.

FIG. 15 is a diagram showing a method of merging four different functions with time modulation used in the present invention.

FIG. 16   3 is a graph of the amplitude of a biphase pulse used in the present invention versus time.

FIG. 17   3 is a frequency plot of signals used in the present invention.

FIG. 18 is a power vs. frequency plot of the power spectrum for the 0-1 GHz frequency band in an exemplary embodiment.

FIG. 19A shows a time domain signal prior to performing radio frequency interference (RFI) extraction in accordance with the present invention.

FIG. 19B shows a time domain signal after performing radio frequency interference (RFI) extraction in accordance with the present invention.

FIG. 20 is a block diagram of a UWB bi-phase communication system according to the present invention.

FIG. 21 is a block diagram of a transmitter section of a UWB communication transceiver according to the present invention.

22 is a block diagram of a circuit used to generate the receiver data modulation digital code of FIG. 21. FIG.

FIG. 23 is a plot of time showing an example of a data stream that is convolved with a wavelet function and code stream by the transmitter of the present invention.

FIG. 24 is a plot of an example of equidistant codestreams convolved with a wavelet in a transmitter according to the present invention.

FIG. 25 is a plot of the convolved data and codestream of FIGS. 23 and 24, respectively.

FIG. 26 is a schematic block diagram of a wavelet generator configured to generate a pseudo derivative orthogonal Gaussian (DOG) wavelet shape for use in a transmitter according to the present invention.

FIG. 27   FIG. 27 is a timing diagram of the wavelet generator shown in FIG. 26.

FIG. 28   FIG. 7 is a circuit diagram of a wavelet generator capable of selecting a Gaussian Nth derivative shape.

29A is a timing diagram of the wavelet generator shown in FIG. 28 having a Gaussian shape of the second derivative selected for use in the present invention.

FIG. 29B   It is a circuit that directly generates a waveform without an analog code.

FIG. 30 is a block diagram of an example of a simple analog code for use in the present invention.

FIG. 31 is a timing diagram of waveforms used to generate an analog code of the type shown in FIG.

FIG. 32   FIG. 6 is a circuit diagram of a programmable distributed analog code according to the invention.

FIG. 33 is a schematic diagram of a programmable distributed analog code that uses an inverter instead of a hybrid coupler to enable programming.

FIG. 34 is a circuit diagram of a programmable distributed analog code using a tap transmission line.

FIG. 35 illustrates a common antenna switching mechanism between a transmitter and a receiver in an embodiment of a transceiver according to the present invention.

FIG. 36   3 is a block diagram of a receiver unit according to the present invention.

FIG. 37   It replaces the receiver shown in FIG.

FIG. 38   3 is a circuit implementing a delay locked loop according to the present invention.

FIG. 39   6 is a circuit implementing a continuous time integrator for use in a receiver according to the present invention.

FIG. 40 is a block diagram of a programmable delay phase shift calculator using a direct digital synthesizer according to the present invention.

FIG. 41 is a block diagram of an example of a programmable delay using a concatenated programmable one-shot circuit.

FIG. 42 is a graph of amplitude versus time of a correlation signal of a transmitted code showing that it can operate in a severe multipath environment with a high level of spatial resolution.

FIG. 43 is a circuit for generating a clock using a jitter code stored in a memory according to the present invention.

FIG. 44   FIG. 44 shows an alternative embodiment of the circuit shown in FIG. 43.

FIG. 45 is another alternative embodiment for jittering a clock using a cryptographic data sequence.

FIG. 46 shows a time plot of a UWB signal according to the invention when multipath occurs in a communication channel.

FIG. 47 shows a frequency plot of a conventional signal when multipath occurs in a communication channel.

FIG. 48 shows a protocol stack using a wireless ASIC according to the present invention for using an embodiment according to the present invention in an application for generating data and communicating with external sources.

FIG. 49 illustrates an embodiment of a transceiver according to the invention used to facilitate communication between home and office equipment for communicating with an access provider via a gateway.

FIG. 50   3 is a block diagram of a system level controller according to the present invention.

FIG. 51 is a block diagram illustrating an embodiment of a transceiver according to the present invention including a plurality of mixing tracking arms (MATs) used to take advantage of the multipath phenomenon for the receiver of the transceiver.

52 is an enlarged block diagram of one of the mixing tracking arms of FIG. 51. FIG.

53A shows a time waveform of a UWB waveform passing through a particular part of the block diagram shown in FIG.

53B shows a time waveform of a UWB waveform passing through a specific part of the block diagram shown in FIG. 2. FIG.

FIG. 53C shows a time waveform of a UWB waveform passing through a particular part of the block diagram shown in FIG.

53D shows a time waveform of a UWB waveform passing through certain different parts of the block diagram shown in FIG.

FIG. 54A   53 shows a further time waveform of the mixing tracking arm shown in FIG. 52.

FIG. 54B   53 shows a further time waveform of the mixing tracking arm shown in FIG. 52.

FIG. 54C   53 shows a further time waveform of the mixing tracking arm shown in FIG. 52.

FIG. 54D   53 shows a further time waveform of the mixing tracking arm shown in FIG. 52.

FIG. 55A   53 shows a further waveform of the mixing tracking arm shown in FIG. 52.

FIG. 55B   53 shows a further waveform of the mixing tracking arm shown in FIG. 52.

FIG. 55C   53 shows a further waveform of the mixing tracking arm shown in FIG. 52.

FIG. 56 shows mixer output with high, low, and correct tracking regions.

FIG. 57   4 is a plot of post-integrator output versus tracking error according to the present invention.

FIG. 58 is a schematic block diagram showing that the performance improvement realized by the present invention improves as the semiconductor switching speed increases.

FIG. 59 illustrates a method of generating a piecewise continuous waveform from the “edge” of a semiconductor logic gate and the effect of the transition speed of the logic gate on the pulse width of a UWB waveform generated using a concatenated edge according to the present invention. The various time waveforms shown are shown.

FIG. 60 shows a plot of the spectral distribution of two signals produced using slow ECL logic and new digital logic with fast rise and fall times.

[Procedure amendment]

[Submission date] March 27, 2003 (2003.3.27)

[Procedure Amendment 1]

[Document name to be amended] Statement

[Correction target item name] 0200

[Correction method] Change

[Contents of correction]

A block diagram of transmitter 2000 is shown in FIG. The function of this transmitter is to generate the waveform of equation (36) above (again, shown below).

[Procedure Amendment 2]

[Document name to be amended] Statement

[Correction target item name] 0258

[Correction method] Change

[Contents of correction]

43 to 45 describe another embodiment for supplying the aforementioned jittered clock signal.

[Procedure 3]

[Document name to be amended] Statement

[Name of item to be corrected] 0292

[Correction method] Change

[Contents of correction]

In the transmission operation mode, the first mixing tracking arm 5113 (MTA1
) Transmission mode operation is enabled. In the transmission operation mode, the waveform to be transmitted is passed from the transmission line to the transmission filter 5107.
It is an adaptive filter that adjusts the spectral energy of the output signal to be transmitted. Then, the output of the transmission filter 5107 is, as shown in FIG. 51, a power control device 5105 which receives a control signal from the remote controller via the control line UWB control.
Passed to. This power control device 5105 adjusts the transmission power of all the energy to be transmitted. In the normal operation mode, the transceiver shown in FIG. 51 operates in an environment where the power spectral density is adjusted. That is, the power control device 5105
Suppresses the power so that the power spectral density in a specific part of the band satisfies the power spectral density requirement. The output of the power control device 5105 is radiated from the antenna 5101 after passing through the T / R switch 5103, and the wireless UW according to the present invention is output.
A B waveform is generated.

─────────────────────────────────────────────────── ─── Continued front page    (81) Designated countries EP (AT, BE, CH, CY, DE, DK, ES, FI, FR, GB, GR, IE, I T, LU, MC, NL, PT, SE), OA (BF, BJ , CF, CG, CI, CM, GA, GN, GW, ML, MR, NE, SN, TD, TG), AP (GH, GM, K E, LS, MW, MZ, SD, SL, SZ, TZ, UG , ZW), EA (AM, AZ, BY, KG, KZ, MD, RU, TJ, TM), AE, AG, AL, AM, AT, AU, AZ, BA, BB, BG, BR, BY, BZ, C A, CH, CN, CR, CU, CZ, DE, DK, DM , DZ, EE, ES, FI, GB, GD, GE, GH, GM, HR, HU, ID, IL, IN, IS, JP, K E, KG, KP, KR, KZ, LC, LK, LR, LS , LT, LU, LV, MA, MD, MG, MK, MN, MW, MX, MZ, NO, NZ, PL, PT, RO, R U, SD, SE, SG, SI, SK, SL, TJ, TM , TR, TT, TZ, UA, UG, UZ, VN, YU, ZW (72) Inventor Loffhart, Martin             Maryland 20770, United States,             Green Belt, Green Way Center             -Drive 7501, Suite 760, D             Xtreme Spectrum, Incorporated             Itted F term (reference) 5J064 AA01 BA16 BC04 BC06 BC16                       BC18 BC21 BD02                 5K022 EE02 EE14 EE21 EE31

Claims (127)

[Claims] 1. A device for transmitting data, comprising wavelet generation means having predetermined time domain and frequency domain characteristics, and code generation means connected to the wavelet generation means for generating a group of wavelets according to a predetermined code. And data encoding means for changing the polarity of the wavelet as a function of the data to be transmitted. 2. The apparatus according to claim 1, wherein said wavelet generating means comprises means for generating a wavelet containing a Gaussian monocycle. 3. The apparatus of claim 2, wherein the wavelet has a pulse width in the range of about 100 to about 2000 picoseconds. 4. The wavelet generation means is about 10 GHz to about 500 MH.
An apparatus according to claim 3 for generating wavelets in the z range. 5. The wavelet has a pulse width of 1000 picoseconds and 1 GHz.
The device of claim 2 having a frequency of. 6. A communication system configured to transmit information in a series of impulse waveforms over a wireless communication channel and controlling an impulse waveform shape for each impulse waveform of the series of impulse waveforms. A transmitter having a controller capable of independently controlling each impulse waveform shape, and a receiver configured to receive the series of impulse waveforms and extract the information. And the system. 7. The system of claim 6, wherein the impulse waveform shape of each of the series of impulse waveforms is a wavelet, and the wavelet is a single zero-mean impulse waveform. 8. The system of claim 6, wherein the controller includes a modulator configured to modulate the impulse waveform shape. 9. The system of claim 6, wherein the controller is configured to control the impulse waveform shape to represent multiple bits of information per impulse waveform. 10. The system of claim 7, wherein the controller is configured to control the impulse waveform by weighting the sum of two master wavelets. 11. The system of claim 10, wherein the two master wavelets are orthogonal wavelets. 12. The two master wavelets include a first master wavelet and a second master wavelet, each of the first and second master wavelets being an even-ordered Gaussian-shaped pulse mathematically represented by the following equation. And an odd-order derivative, where r is an even integer, s is an odd integer, t is a predetermined unit of time, and α is an impulse scale factor. . [Equation 1] 13. The system of claim 12, wherein each of r and s is 10 or less. 14. The two master wavelets include a first master wavelet and a second master wavelet, the first master wavelet being a real part of a Rayleigh wavelet, and the second master wavelet being an imaginary part of the Rayleigh wavelet. The system of claim 11, wherein said Rayleigh wavelet is mathematically represented by the equation: [Equation 2] 15. The system of claim 14, wherein n is less than 10. 16. The two master orthogonal wavelets include a first master wavelet and a second master wavelet, the first master wavelet being a derivative of a Gaussian shaped pulse and the second master wavelet being a derivative of the Gaussian shaped pulse. The system of claim 11, which is a Hilbert transform of the derivative. 17. The system of claim 10, wherein the controller is configured to apply weights in the range of -1 to +1 to the two master wavelets. 18. The controller comprises a modulated wavelet encoded sequence composed of a predetermined number of said series of impulse waveforms ranging from 1 to N2, which is transmitted as a modulated wavelet group to convey said subset of information. The system of claim 6, further comprising an encoding mechanism configured to: 19. The encoding mechanism may arrange the modulated wavelet encoded sequence as one of a set of candidate encoded sequences such that the subset of information is transmitted in at least one of the candidate encoded sequences. The system of claim 18, configured. 20. The system of claim 6, wherein the receiver includes at least one of a delay locked loop and a phase locked loop configured to track the position of the series of impulse waveforms during signal reception. 21. The receiver comprises a tapped transmission line comprising a predetermined number of taps separated by a discrete distance interval; and adding energy applied to each of the taps and outputting a combined signal. 7. The system of claim 6, comprising: 22. The summing network comprises each tap amplifier having a gain set to provide an amplified output signal whose level corresponds to an amplified output signal provided by another amplifier in the summing network. Claim 21 including
The system described. 23. The system of claim 22, wherein the summing network includes summing amplifiers configured to sum the amplified output signals provided by each of the tapping amplifiers. 24. The summing network includes a switch for each amplifier, each amplifier including a normal output and an inverting output, and each switch is either a normal output signal or an inverting output signal depending on an operating state of the switch. 23. The system of claim 22, connected to the normal and inverting outputs of an amplifier corresponding to the switch. 25. The system of claim 21, wherein the summing network is formed on a monolithic integrated circuit. 26. The system of claim 6, wherein the receiver includes a radio frequency interference extraction circuit configured to suppress narrowband interference that is not part of the signal. 27. The receiver comprises an input port and an output port, and is configured to generate an inverted and a non-inverted copy sequence of an impulse waveform applied to the input port in a predetermined pattern at the output port. The system of claim 6 including a pulse spreader / compressor network. 28. The spreader / compressor network comprises a tapped transmission line having a predetermined number of taps, a predetermined time shift associated with each of the taps for outputting a combined signal, the tapped transmission line being connected to the taps. A summing network configured to add signal energy applied to each of the taps according to weights; an inverting and non-inverting output for each tap, and a matched delay between an input and the inverting and non-inverting output An amplifier network including amplifiers with matched gains, and a predetermined number of controllable switches connected to the amplifiers for each tap, the inverting and non-inverting to implement a switched output for each tap. A switching network configured to controllably output one of the inverting outputs and the spreader / compressor different 28. The system of claim 27, further comprising: a combining network configured to generate the combined signal output by adding respective signals of the switched outputs of each tap via a path. 29. The spreader / compressor is configured to receive a splitter input, a 0 degree output, and a 180 degree output splitter, and a signal of the 0 degree output and the 180 degree output of the splitter, and an input. A crossover ladder network having a predetermined number of output taps for outputting an inverted copy of an impulse waveform and a non-inverted copy of the input impulse waveform, each output tap having a different delay between the inputs, and the impulse from each tap. A set of switches configured to select one of the inverted and non-inverted copies of the waveform, and a summing ladder configured to accept a signal at the output of each of the switch sets and generate a composite output signal The system of claim 27, comprising a network. 30. The delay of the signal by each path of the spreader / compressor is such that a single impulse waveform applied to the spreader / compressor is an impulse with equal spacing on the composite signal output. 29. The system of claim 28, having equal increment intervals. 31. The crossover ladder network is a forward type wherein all stages of the logic output network except the last stage are arranged to drive two other logic output networks until a predetermined number of output taps are achieved. A series of logical output networks, each logical output network having a left 0 degree splitter and a right 0 degree splitter with a left input and a right input and a left output and a right output, respectively, and a subsequent stage of the logic output network. Means that the left output of the right 0 degree splitter crosses over with the right output of the left 0 degree splitter, and the left output of the left splitter and the left output of the right splitter form a left logical output network of the next stage. And the right output of the left splitter and the right output of the right splitter are both connected to the right logical output network of the next stage. 30. The system of claim 29 connected to be connected to. 32. The spreader / compressor comprises a ladder comprising splitters each having an input and a predetermined number of output taps, each splitter configured to drive at least two splitters by cables of different lengths; A network of amplifiers comprising inverting and non-inverting outputs and an input connected to the output taps of the splitter ladder, and connected to each amplifier of the amplifier network between the inverting and non-inverting outputs of the amplifiers. A network of switches configured to select, and at least 2 from another combiner having a predetermined set of inputs connected to said switch network and located upstream relative to the direction of signal flow in the ladder. Includes a synthetic output stage containing a series of combiners arranged to accept one input. 28. The system of claim 27, further comprising a combiner ladder. 33. The different length cables have respective lengths L _n equal to at least one of a _n and a _n +2 ⁿ D, where a _n is a predetermined time for the nth stage. 33. The system of claim 32, wherein the delay is a delay, n is an integer representing a ladder stage, n? 0, and D is a time delay between subsequent output pulses. 34. The communication system of claim 6, wherein the controller includes a wavelet generator configured to generate an impulse waveform shape for each of the series of impulse waveforms. 35. The wavelet generator is adapted to construct the waveform shape by adding a time-shifted copy of a programmable number of narrow unipolar pulses with a programmable weight between +1 and −1. The communication system of claim 34 configured. 36. The wavelet generator comprises a positive narrow spike and
A digital logic gate arranged to form a negative narrow spike; a length arranged to delay a predetermined portion of the positive narrow spike and the negative narrow spike by a predetermined amount. 36. Different transmission lines and an adder configured to add the narrow spike in the positive direction to the spike in the negative direction after a time delay to form at least half of the waveform shaping cycle. The communication system described. 37. The wavelet generator comprises a positive narrow spike,
A digital logic gate arranged to form a negative narrow spike, a length of the digital logic gate arranged to delay a predetermined portion of the positive narrow spike and the negative narrow spike by a predetermined amount. A different transmission line and a selector switch programmably operable to form said waveform shape as at least one of a Gaussian monocycle derivative, a Gaussian higher order derivative, a delayed and inverted or non-inverted pulse. 36. The communication of claim 35, further comprising: an adder configured to combine the output signal of the selector switch to add a selected positive narrow spike and a negative narrow spike with a selected delay. system. 38. The transmitter is configured to route the series of impulse waveforms to a single antenna and a single antenna during a transmit mode of operation, and the single antenna during a receive mode of operation. 7. A communication system according to claim 6, comprising a transmit / receive switch configured to route a received signal from one antenna to the receiver. 39. The impulse configured to spread the impulse waveform shape into a group of transmit impulse waveforms when operating in a transmit mode of operation and provided by the transmit / receive switch during a receive mode of operation. 39. The system of claim 38, further comprising a programmable spreader and compressor configured to receive a waveform group and compress the impulse waveform group into a collective impulse waveform. 40. The receiver comprises a RAK comprising a plurality of isolated receive paths.
The system of claim 6, including an E architecture. 41. The receiver includes, as one of the separated reception paths, a search channel configured to detect a stronger multipath signal than is received by other separated reception channels. The system of claim 40. 42. The receiver separates the other separation if it is determined that one of the other separated reception paths is receiving a weaker multipath signal than the stronger multipath signal. 42. The system of claim 41 configured to switch one of the received received signals to the stronger multipath signal. 43. A wireless communication method, the step of transmitting information in a series of impulse waveforms over a wireless communication channel, the step of independently controlling each impulse waveform shape of said series of impulse waveforms. Receiving the sequence of impulse waveforms and extracting the information. 44. The method of claim 43, wherein the controlling step comprises controlling the impulse waveform shape of each of the series of impulse waveforms into a single zero-mean impulse waveform wavelet. 45. The method of claim 43, wherein the controlling step comprises modulating the impulse waveform shape. 46. The method of claim 43, wherein the controlling step comprises controlling the impulse waveform shape to represent multiple bits of information per impulse waveform. 47. The method of claim 44, wherein the controlling step comprises controlling the impulse waveform by performing a weighting of the sum of two master wavelets. 48. The method of claim 47, wherein the weighting step comprises weighting the sum of the two master wavelets, the two master wavelets being orthogonal wavelets. 49. The weighting step includes weighting a first master wavelet and a second master wavelet of the two master wavelets, wherein the first and second master wavelets are mathematically represented by the following equations. 44. The method of claim 43, wherein the Gaussian shaped pulses are even and odd derivatives, respectively, where r is an even integer, s is an odd integer, and t is a predetermined unit of time. , Α is the impulse scale factor). [Equation 3] 50. The method of claim 44, wherein the weighting step comprises the step of weighting the first master wavelet and the second master wavelet when each of the r and s is 10 or less. 51. The step of controlling includes weighting a sum of the two master wavelets including a first master wavelet and a second master wavelet, the first master wavelet being a real part of a Rayleigh wavelet, 48. The method of claim 47, wherein the second master wavelet is the imaginary part of the Rayleigh wavelet and the Rayleigh wavelet is mathematically represented by the equation: [Equation 4] 52. The method of claim 46, wherein the weighting step comprises weighting the first master wavelet and the second master wavelet if n is less than 10. 53. The controlling step includes controlling the impulse waveform shapes of the two master wavelets including a first master wavelet and a second master wavelet, the first wavelet being a derivative of a Gaussian shaped pulse. 44. The method of claim 43, wherein the second wavelet is a Hilbert transform of the derivative of the Gaussian shaped pulse. 54. The method of claim 47, wherein the controlling step comprises applying weights in the range of -1 to +1 to the two master wavelets. 55. Forming a modulated wavelet encoded sequence consisting of a predetermined number of a series of impulse waveforms ranging from 1 to N2; and transmitting the modulated wavelet encoded sequence to convey a subset of the information. 44. The method of claim 43, further comprising: transmitting as a group of. 56. The forming step includes arranging the coded sequence as one of a set of candidate coded sequences so that the subset of information is transmitted in at least one of the candidate coded sequences. 55. The method according to 55. 57. The method of claim 43, wherein the receiving step comprises tracking the position of the series of impulse waveforms by at least one of a delay locked loop and a phase locked loop. 58. The receiving step comprises delaying the series of impulse waveforms on a tapped transmission line comprising a predetermined number of taps separated by discrete distance intervals, and comparing the energy applied to each of the taps. 45. The method of claim 44, including the step of adding and outputting a combined signal. 59. The summing step comprises summing the energy applied to each of the taps with an amplifier having a gain set to provide a corresponding amplified output signal at each signal level. The method described. 60. The method of claim 58, wherein the summing step includes selecting either a normal output signal or an inverted output signal from each of the amplifiers. 61. The method of claim 58, wherein the adding step comprises adding the energies on a monolithic integrated circuit. 62. The method of claim 43, wherein the receiving step comprises suppressing narrowband energy before extracting the information from the series of impulse waveforms. 63. The method of claim 43, wherein the receiving step includes the step of generating a sequence of inverted copies of the impulse waveform and non-inverted copies of the impulse waveform. 64. The step of applying applies signal energy from each tap of a tapped transmission line to taps arranged to convey a predetermined time shift and weight to the impulse waveform, Delaying the signal energy at each tap to provide a matched delay and matched gain between the input and output ports of the amplifier used to amplify the signal at each tap; 64. The method of claim 63, further comprising: selecting one of a non-inverting output and a non-inverting output; and adding each signal after the selecting step to produce a composite output signal. 65. The method of claim 64, wherein the adding step comprises delaying each of the signals by an equal incremental delay so that a single impulse is produced as the composite signal output. 66. The method of claim 43, wherein the controlling step includes the step of generating an impulse waveform shape for each of the series of impulse waveforms. 67. The controlling step includes the step of constructing the waveform shape by adding a time-shifted copy of a programmable number of narrow unipolar pulses with a programmable weight between +1 and -1. The method of claim 66. 68. Routing the series of impulse waveforms to a single antenna during a transmit mode of operation; routing received signals from the single antenna to a receiver during a receive mode of operation. 44. The method of claim 43, further comprising: 69. Spreading the impulse waveform shape into a group of impulse waveforms in the transmission operation mode and transmitting the impulse waveform group; and collecting the impulse waveform groups in the reception operation mode. 69. The method of claim 68, further comprising: compressing into. 70. The method of claim 43, wherein the receiving step comprises the step of receiving the series of impulse waveforms via a plurality of separate receive paths of a RAKE receiver. 71. The method of claim 70, wherein said receiving step comprises the step of searching one of said separated reception paths for a stronger multipath signal than those received by other reception paths. 72. The step of receiving further comprises separating the other isolated receive path if it is determined that one of the other isolated receive paths is receiving a weaker multipath signal than the stronger multipath signal. 72. The method of claim 71 including the step of switching one of the received receive paths to said stronger multipath signal. 73. A wireless communication system, means for transmitting information in a series of impulse waveforms over a wireless communication channel; and means for independently controlling each impulse waveform shape of the series of impulse waveforms, Means for receiving the sequence of impulse waveforms and extracting the information. 74. The system of claim 73, wherein the control means includes means for controlling the impulse waveform shape of each of the series of impulse waveforms to be a single zero-mean impulse waveform wavelet. 75. The system of claim 73, wherein the control means includes means for modulating the impulse waveform shape. 76. The system of claim 73, wherein said control means includes means for controlling said impulse waveform shape to represent multiple bits of information per impulse waveform. 77. The system of claim 74, wherein said control means includes means for controlling said impulse waveform shape by means for weighting the sum of two master wavelets. 78. The system of claim 77, wherein the weighting means includes means for weighting the sum of the two master wavelets, the two master wavelets being orthogonal wavelets. 79. The weighting means includes the step of weighting the first master wavelet and the second master wavelet of the two master wavelets, wherein the first and second master wavelets are mathematically represented by the following equations. 79. The system of claim 78, wherein each is an even and odd derivative of a Gaussian shaped pulse, wherein r is an even integer, s is an odd integer, and t is a predetermined unit of time. , Α is the impulse scale factor). [Equation 5] 80. The system of claim 79, wherein the weighting means comprises the step of weighting the first master wavelet and the second master wavelet when each of the r and s is 10 or less. 81. The control means comprises weighting a sum of the two master wavelets, the two master wavelets including a first master wavelet and a second master wavelet, the first master wavelet being Rayleigh. 78. The system of claim 77, wherein the system is a real part of a wavelet, the second master wavelet is an imaginary part of the Rayleigh wavelet, and the Rayleigh wavelet is mathematically represented by: [Equation 6] 82. The weighting means sets the first weight when the n is less than 10.
82. The system of claim 81, comprising weighting a master wavelet and the second master wavelet. 83. The control means includes the step of controlling the impulse waveform shapes of the two master wavelets including a first master wavelet and a second master wavelet, wherein the first master wavelet is a derivative of a Gaussian pulse. 80. The system of claim 79, wherein the system is a function and the second master wavelet is a Hilbert transform of the derivative of the Gaussian shaped pulse. 84. The system of claim 77, wherein said control means comprises applying weights in the range of -1 to +1 to said two master wavelets. 85. Means for forming a modulated wavelet encoded sequence consisting of a predetermined number of said series of impulse waveforms ranging from 1 to N2; and modulating said modulated wavelet encoded sequence to convey said subset of information. 74. The system of claim 73, further comprising: means for transmitting as a wavelet group. 86. The forming means comprises arranging the coded sequence as one of a set of candidate coded sequences such that the subset of information is transmitted in at least one of the candidate coded sequences. 85. The system according to item 85. 87. The system of claim 73, wherein said receiving means includes the step of tracking the position of said series of impulse waveforms by at least one of a delay locked loop and a phase locked loop. 88. The receiving means delays the series of impulse waveforms on a tapped transmission line comprising a predetermined number of taps separated by individual distance intervals, and energy applied to each of the taps. 75. The system of claim 74, including means for adding and outputting a combined signal. 89. The summing means comprises summing the energy applied to each of the taps by an amplifier with a gain set to provide a corresponding amplified output signal at each signal level. The system described. 90. The system of claim 89, wherein said adding means includes the step of selecting either a normal output signal or an inverted output signal from each of said amplifiers. 91. The system of claim 88, wherein said adding means includes the step of adding the energies on a monolithic integrated circuit. 92. The system of claim 73, wherein the receiving step includes the step of suppressing narrowband energy prior to extracting the information from the series of impulse waveforms. 93. The system of claim 73, wherein said receiving means includes the step of generating a sequence of inverted copies of the impulse waveform and non-inverted copies of the impulse waveform. 94. The means for generating includes applying signal energy from each tap of a tapped transmission line to a tap arranged to convey a predetermined time shift and weight to the impulse waveform. The system delays the signal energy at each tap to provide a matched delay and matched gain between the input and output ports of an amplifier used to amplify a signal at each tap; 94. The system of claim 93, further comprising: means for selecting either the inverting or non-inverting output of each of the amplifiers; and means for adding the respective signals after the selecting means to produce a composite output signal. . 95. The system of claim 94, wherein said summing means comprises delaying said respective signals by equal incremental delays so that a single impulse is produced as said composite signal output. 96. The system of claim 73, wherein said control means includes the step of generating said impulse waveform shape for each of said series of impulse waveforms. 97. The control means comprises constructing the waveform shape by applying a time-shifted copy of a programmable number of narrow unipolar pulses with programmable weights between +1 and -1. The system of claim 96. 98. Means for routing the series of impulse waveforms to a single antenna during a transmit mode of operation; and means for routing a received signal from the single antenna to a receiver during a receive mode of operation. 74. The system of claim 73, further comprising: 99. Means for spreading the impulse waveform shape into groups of impulse waveforms in a transmission operation mode, means for transmitting the impulse waveform group, and a group of impulse waveform groups in a reception operation mode. 99. The system of claim 98, further comprising: means for compressing into a waveform. 100. The system of claim 73, wherein said receiving means includes the step of receiving said series of impulse waveforms via a plurality of separate receive paths of a RAKE receiver. 101. The system of claim 100, wherein said receiving means includes means for searching one of said separated receive paths for a multipath signal that is stronger than those received by the other receive paths. 102. The receiving means is one of the other separated receiving paths.
Means for switching one of the other separated receive paths to the stronger multipath signal if it is determined that one is receiving a weaker multipath signal than the stronger multipath signal. The system of claim 101. 103. A computer program product comprising a computer readable medium for suppressing radio frequency interference in an ultra wideband communication system, the computer program product comprising a series of data transmitted from a transmitter to a receiver over a wireless communication channel. A radio frequency interference extraction mechanism configured to suppress narrowband interference that is not part of a spread spectrum signal comprising a series of impulse waveforms generated by a controller of the transmitter that controls the impulse waveform shape of the impulse waveform. The radio frequency interference mechanism is configured to remove direct current from the spread spectrum signal, and a direct current rejection mechanism configured to divide the predetermined bandwidth containing the spread spectrum signal into a set of bins. Configured frequency mapping mechanism and adjacency Comparing a predetermined energy of a bin with an energy level in each of the bin sets, and a comparison mechanism configured to reduce the energy level in the bin if the energy level is greater than the predetermined energy level, Computer program product characterized by including. 104. In an ultra wideband transmitter configured to transmit a signal stored in a spread spectrum signal comprising a series of impulse waveforms to a receiver, shaping each impulse waveform of the series of impulse waveforms. An impulse waveform generator configured to have at least a subset of the series of impulse waveforms, the impulse waveform generator having an impulse waveform shape independently controllable by the impulse waveform generator, and the series of impulses. An encoder configured to arrange the waveforms in a predetermined array, the impulse waveform subset of the series of impulse waveforms including at least a portion of the information; Ultra wideband transmitter. 105. The transmitter of claim 104, wherein another portion of the information is stored within the predetermined array of the series of impulse waveforms. 106. The impulse waveform shape of each of the series of impulse waveforms is a single zero-mean impulse waveform wavelet.
The listed transmitter. 107. The transmitter of claim 104, wherein the impulse waveform generator comprises a modulator configured to modulate the impulse waveform shape. 108. The transmitter of claim 104, wherein the impulse waveform generator is configured to control the impulse waveform shape to represent multiple bits of information per impulse waveform. 109. The encoder is configured to form a modulated wavelet encoded sequence composed of a predetermined number of the series of impulse waveforms ranging from 1 to N2, the modulated wavelet encoded sequence comprising: 105. The transmitter of claim 104, which is transmitted as a group of modulated wavelets to convey the at least a portion of the information. 110. An ultra wideband receiver configured to receive information stored in a spread spectrum signal comprising a series of impulse waveforms from a transmitter, comprising:
A receiver front end configured to convert the spread spectrum signal to an electrical signal, wherein the signal is stored in respective shapes and arrays of at least a subset of the impulse waveform; And a sliding correlator configured to detect said information stored in said series of impulse waveforms. 111. The receive of claim 110, wherein the sliding correlator comprises at least one of a delay locked loop and a phase locked loop configured to track the position of the series of impulse waveforms upon signal reception. Machine. 112. The receiver front end includes a tapped transmission line having a predetermined number of taps separated by individual distance intervals, and energy applied to each of the taps to output a combined signal. 112. The receiver of claim 110, comprising: a summing network configured in. 113. The summing networks each for each of the taps having a gain set to provide an amplified output signal whose level corresponds to an amplified output signal provided by another amplifier in the summing network. 113. The receiver of claim 112 including an amplifier of. 114. The receiver of claim 113, wherein the summing network includes summing amplifiers configured to sum the amplified output signals provided by each of the tapping amplifiers. 115. The summing network includes a switch for each amplifier,
Each amplifier includes a normal output and an inverting output, and each switch has a normal output and an inverting output of the amplifier corresponding to the switch so as to provide either the normal output signal or the inverting output signal according to the operating state of the switch. 114. The receiver of claim 113 connected to. 116. The receiver of claim 112, wherein the summing network is formed on a monolithic integrated circuit. 117. The receiver of claim 110, further comprising a radio frequency interference extraction circuit configured to suppress narrowband interference that is not part of the spread spectrum signal. 118. A pulse spreader having an input port and an output port, the pulse spreader configured to generate a sequence of inverted and non-inverted copies of an impulse waveform applied to the input port at the output port in a predetermined pattern. The receiver of claim 110, further comprising a / compressor network. 119. The spreader / compressor network comprises a tapped transmission line comprising a predetermined number of taps and a predetermined time shift associated with each of the taps to output a combined signal connected to the taps. And a summing network configured to add the signal energy applied to each of the taps according to a weight, the summing network comprising a matched delay and a matched gain between the input and the inverting and non-inverting outputs. An amplifier network including an amplifier having inverting and non-inverting outputs for each tap, and a predetermined number of controllable switches connected to the amplifier for each tap, implementing a switched output for each tap. A switching network configured to controllably output either the inverted or non-inverted output. Network and a synthetic network configured to generate said combined signal output by adding respective signals from the switched output for each tap, said respective signals passing through different paths within said spreader / compressor. 120. The receiver of claim 118, including: 120. The spreader / compressor is configured to have a splitter input, a 0 degree output, and a 180 degree output, and a signal from the 0 degree output and the 180 degree output of the splitter. A crossover ladder network having a predetermined number of output taps for outputting an inverted copy of the input impulse waveform and a non-inverted copy of the input impulse waveform, each output tap having a different delay between the inputs, and output from each tap A set of switches configured to select between an inverted and a non-inverted copy of the impulse waveform that is configured to: accept a signal at each output of the switch set and generate a composite output signal 120. The receiver of claim 118, further comprising: a summing ladder network. 121. The delay of the signal by each path in the spreader / compressor is such that a single impulse waveform applied to the spreader / compressor results in a sequence of equally spaced impulses on the composite signal output. 120. The receiver of claim 119, having equal incremental spacing to be produced. 122. The crossover ladder network comprises a series of logic output networks arranged such that all stages of the logic output network except the last stage drive two other logic output networks until a predetermined number of output taps are achieved. A forward logic output network, each logic output network having a left 0 degree splitter and a right 0 degree splitter having a left input and a right input and a left output and a right output, respectively, and a subsequent stage of the logic output network , The left output of the right 0 degree splitter crosses over with the right output of the left 0 degree splitter, and the left output of the left splitter and the left output of the right splitter are connected to a left logical output network of the next stage. The right output of the left splitter together with the right output of the right splitter in the right logical output network of the next stage. 121. The receiver of claim 120 connected to be connected to. 123. The spreader / compressor comprises a ladder of splitters each having an input and a predetermined number of output taps, each ladder configured to drive at least two splitters by cables of different lengths. A network of amplifiers having inverting and non-inverting outputs and an input connected to an output tap of the splitter ladder, and connected to each amplifier of the amplifier network to select one of the inverting and non-inverting outputs of the amplifier A network of switches configured as such, and at least two inputs from another combiner having a set of inputs connected to the switch network and located upstream relative to the direction of signal flow in the ladder. A synthetic output stage containing a series of combiners arranged to accept The receiver of claim 118, further comprising a combiner ladder including. 124. The cables of different lengths have lengths a and a + 2nL at the output of the first splitter of the splitter ladder, lengths b and b + 2n-1L at the next stage of the splitter ladder, and the splitter, respectively. The last stage of the ladder is provided with lengths z and z + 20, respectively.
23. The receiver according to 23. 125. The receiver of claim 110, wherein the receiver front end comprises a RAKE architecture with a plurality of separate receive paths. 126. The receiver of claim 125, wherein the plurality of separate receive paths includes a search channel configured to detect multipath signals that are stronger than those received by the other separate receive paths. 127. The receiver front end may further determine if one of the other separate receive paths is receiving a weaker multipath signal than the stronger multipath signal. 127. The receiver of claim 126, including a switch mechanism configured to switch one of the separate receive paths to said stronger multipath signal.