JP2003516069A - Method for deriving at least three audio signals from two input audio signals - Google Patents

Abstract

Various equivalent adaptive audio matrix arrangements are disclosed, each of which includes a feedback-derived control system that automatically causes the cancellation of undesired matrix crosstalk components in the matrix output. Each adaptive audio matrix arrangement includes a passive matrix that produces a pair of passive matrix signals in response to two input signals. A feedback-derived control system operates on each pair of passive matrix signals, urging the magnitudes of pairs of intermediate signals toward equality. Each control system includes variable gain elements and a feedback and comparison arrangement generating a pair of control signals for controlling the variable gain elements. Additional control signals may be derived from the two pairs of control signals for use in obtaining more than four output signals from the adaptive matrix.

Description

Detailed Description of the Invention

[0001]

【Technical field】

The present invention relates to audio signals. In particular, the present invention relates to three or more streams of audio input signals (or "each signal" or "each channel") from each pair of audio input signals (or "each signal" or "each channel"). ) Is derived using the "adaptive" (or "active") audio matrix method "multidirectional" (
Or "multi-channel") audio decoding. The present invention is useful for recovering audio signals in which each signal is associated with a direction and combined by a coding matrix into a smaller number of each signal. Although the present invention is described in terms of such an intentional matrix encoding, the present invention need not be used in any particular matrix encoding and is a satisfactory orientation from material originally recorded for two-channel playback. It is also useful for producing effects.

[0002]

[Background technology]

Audio matrix encoding and decoding is well known in the prior art. For example, in "4-2-4" matrix encoding and decoding, four source signals, i.e. generally four fundamental directions (e.g. left, center, right and environment or left front, right front, left rear and right rear). , Etc.) is amplitude-phase matrix encoded into two signals.
After the two signals have been transmitted or stored, they are decoded with an amplitude-phase matrix decoder to recover an approximation of the four original signals. The decoded signal is an approximation because it suffers from the well-known disadvantage of crosstalk between decoded audio signals in a matrix decoder. Ideally, the decoded signal should be identical to the source signal with infinite boundaries between the signals. However, the crosstalk inherent in the matrix decoder results in only 3 dB of separation between signals associated with adjacent directions. An audio matrix whose matrix properties do not change is known as a "passive" matrix.

To overcome the crosstalk problem of matrix decoders, it is known in the prior art to improve the separation between the decoded signals and adaptively change the decoding matrix properties to more closely approximate the source signals. Has been. A well known example of such an active decoder is the Dolby Pro Logic decoder described in US Pat. No. 4,799,260. The patent is incorporated herein by reference in its entirety. The '260 patent refers to a number of prior art patents thereto, many of which describe various other types of adaptive matrix decoders.
Other prior art patents are US Pat. Nos. 5,625,696, 5,644,640 and 5,
, 504, 819, 5,428, 687 and 5,172,415. All of these patents are also incorporated herein by reference in their entirety.

Prior art adaptive technology matrix decoders are intended to reduce crosstalk in the regenerated signal and more closely duplicate the source signal, which the prior art does in each method. Many are complex and cumbersome, and the method does not recognize the desired relationship between the intermediate signals of the decoder that can be used to simplify the decoder and improve the accuracy of the decoder.

Accordingly, the present invention is directed to a method and apparatus for recognizing and using previously ununderstood relationships between intermediate signals of adaptive matrix decoders. Undesirable crosstalk components can be easily canceled by exploiting these relationships, especially by using an automatic self-cancellation device with negative feedback.

[0006]

[Means for solving the problem]

  According to one aspect of the invention, the invention comprises at least three audio signals from two input signals.
A method for deriving an audio output signal, in which two audio signals are responsive
Two input audio channels by a passive matrix that generates two pairs of audio signals.
Four audio signals are derived from the audio signal. That is, the derived audio
The first pair of signals represents each direction lying on the first axis (such as "left" and "right").
), Representing each direction in which the second pair of derived audio signals lies on the second axis ("
(Such as "center" and "environment"), the first and second axes are substantially perpendicular to each other.
You Each pair of derived audio signals comprises a respective first and second pair (respectively
Middle / left / right and center / environment pair) processed to produce an intermediate audio signal
The relative amplitudes of the audio signals are equal in each pair of inter-audio signals.
Will be forced. Derived audio where the first pair (left / right pair) of intermediate signals is generated
A first output signal (left output) representing the first direction lying on the axis of the signal pair (left / right pair)
Signal L_outIs a second pair of intermediate audio signals (center / ring) with the same polarity.
Boundary) is generated by combining at least one component of each. First pair (
Axis of the derived audio signal pair (left / right pair) where the intermediate signal (left / right pair) is generated
The second output signal (the right output signal R_outLike)
, At least each of the second pair (central / environmental pair) of opposite polarity, intermediate audio signals
It is generated by combining one component. Intermediate signal of the first pair (left / right pair) is emitted
The first one lying on the axis of the derived derived audio signal pair (center / environment pair)
Third output signal (center output signal C_outOr environmental output signal S_outAs
Na) is the first pair of intermediate audio signals (center / environment pair) with the same or opposite polarities.
Generated by combining at least one component of each of the. Optional, intermediate communication
The second pair of signals (center / environment) is generated, the pair of derived audio signals (center / ring)
The fourth signal (third output signal is the central output signal C) that represents the second direction lying on the axis of
_outThen environmental signal S_out, Or the third output signal is S_outThen C_outNo
Is generated by combining the third output signals with the same polarity,
If generated in opposite polarity or by combining the third output signal in opposite polarity
, The first pair of intermediate audio signals of the same polarity Generated by combining each at least one component (left / right pair).

The previously unrecognized relationship between the coded signals is that the undesired crossing of the decoded output signals is forced by equalizing the magnitude of each pair of intermediate audio signals. That is, the talk component is substantially suppressed. The principle does not require perfect equality to achieve substantial crosstalk cancellation. Such processing is easily and desirably performed by using a negative feedback device that causes automatic cancellation of unwanted crosstalk components.

The present invention includes each embodiment having an equivalent topology. In all of the above embodiments, each intermediate signal is derived from a passive matrix that acts on a pair of input signals and forces these intermediate signals to be equal. In embodiments using the first topology, the cancellation component of the intermediate signal is combined with the passive matrix signal (acting on the input signal or from another passive matrix) to produce the output signal. In embodiments using the second topology, the intermediate signal pair is coupled to the output signal.

Another aspect of the invention involves the derivation of an additional control signal that produces an additional output signal.

A first object of the present invention is to provide a wide variety of input signals by using a circuit that does not have a specific requirement for accuracy, which is unprecedented in the prior art, and a circuit element that does not require an abnormal complexity of a control path. Achieving a high degree of crosstalk cancellation that is measurable and perceptible under conditions.

Another object of the present invention is to achieve such high performance with simpler and less costly circuit elements than prior art circuits.

[0012]

BEST MODE FOR CARRYING OUT THE INVENTION

The passive decoding matrix is shown functionally and diagrammatically in FIG. The following is the input L _t and R _t (whole left and right) L _out = L _t (Equation 1) R _out = R _t (Equation 2) C _out = 1/2 * (L _t + R _t ) ( Equation 3) S _out = 1/2 * (L _t −R _t ) (Equation 4) As used in the above and other equations throughout this specification, the symbol “*” indicates multiplication.

The central output is the sum of each input and the environmental output is the difference of each input. Furthermore, both have arbitrary scaling (scale) and are halved for ease of explanation. Other scaled values are possible. C _out is obtained by acting on L _t and R _t with a scale factor of 1/2 to the linear combiner 2. The S _out output is obtained by acting on the linear combiner 4 L _t and R _t with scales +1/2 and -1/2 respectively.

The passive matrix of FIG. 1 thus produces two pairs of audio signals: the first pair L _out and R _out and the second pair C _out and S _out . In this example, the basic directions of the passive matrix are called left, center, right and environment. Each adjacent elementary direction lies on an axis at right angles to these direction labels, such that the left is adjacent to the center and the environment, the environment is adjacent to the left and right, and so on. The invention applies to any quadrature 2: 4 decoding matrix.

A passive matrix decoder derives an n audio signal from an m audio signal and has an invariant relationship (eg C _out is always 1/2 * (R _out + L _out ) in FIG. 1).
}, N is larger than m. In contrast, active matrix decoders derive n audio signals with a variable relationship. One way to construct an active matrix is to combine each signal-dependent signal component with each output signal of the passive matrix. For example, as shown functionally and diagrammatically in FIG. 2, four VCAs (Voltage Controlled Amplifiers) 6, 8, 10 deriving a variably scaled output of the passive matrix output.
And 12 are unchanging passive matrix outputs in linear combiners 14, 16, 18 and 20 (ie the two inputs themselves along with the two outputs of combiners 2 and 4).
Is summed up with. Each VCA is left passive matrix, right, since having their inputs derived respectively from the center and the environment, their gain g _l, g _r, may be referred to as g _c and g _{s (all} positive). The VCA output signal constitutes a cancellation signal and
To improve the directivity of the matrix decoder by suppressing crosstalk, the cancellation signal is combined with a passively derived output with crosstalk from each direction from which it is derived.

Note that in the device of FIG. 2, each path of the passive matrix is still present.
Each output is the sum of each passive matrix output and the two VCA outputs. The VCA outputs are selected and scaled to give the desired crosstalk cancellation for each passive matrix output, taking into account that crosstalk components occur at each output representing adjacent fundamental directions. It For example, the center signal has passively decoded left and right signal crosstalk and the environmental signal has passively decoded left and right signal crosstalk. Therefore, the left signal output should be combined with the cancellation signal component derived from the passively decoded center and environment signals, as well as the other four outputs. The manner in which each signal is scaled, polarized and combined in FIG. 2 provides the desired crosstalk suppression. By varying each VCA gain from zero to one (for the scaling example of FIG. 2)
Undesired crosstalk components of the passively decoded output can be suppressed.

[0017]   The device of FIG. 2 has the following formula:

L _out = L _t −g _c * (L _t + R _t ) −g _s * 1/2 * (L _t −R _t ) (Equation 5) R _out = R _t −g _c * 1/2 * _{(L t R t) + g} s * 1/2 * (L t -R t) ( equation _{6) C out = 1/2} * (L t + R t) -g l * 1/2 * L t -g r * 1/2 * R _t (Equation 7) S _out = 1/2 * (L _t −R _t ) −g _l * 1/2 * L _t + g _r * 1/2 * R _t (Equation 8)

If all VCA's have zero gain, the device is identical to the passive matrix. For any equivalent of all VCA's, the device of FIG. 2 is identical to the passive matrix except for constant scaling. For example, if all of the VCA has a gain of _{0.1, L out = L t -0.05} * (L t + R t) -0.05 * (L t -R t) = 0.9 * L t R out = R t _{-0.05 * (L t + R t} ) -0.05 * (L t -R t) = 0.9 * R t C out = 1/2 * (L t + R t) -0.05 * L t -0.05 * R t = 0.9 * _{1/2 * (L t +} R t) S out = 1/2 * (L t -R t) -0.05 * L t + 0.05 * R t = 0.9 * 1/2 * (L t -R t) result Is a passive matrix scaled to a scale of 0.9. Therefore, the exact value of the stationary (inactive) VCA gain, described below, is not critical.

Considering an example, for each of the basic directions (left, right, center and environment), each respective input signal is L _t only, R _t only, L _t = R _t (same polarity) and L _{t. t} = -R _t (
Opposite polarity) and the corresponding desired outputs are L _out only, R _out only, C _out only and S _out only. In each case, ideally only one output should give one signal and the rest should give nothing.

The test controls each VCA such that if one corresponding to the desired fundamental direction has a gain of one and the rest is much less than one, all but one desired. At the output, the VCA signal will cancel the unwanted input to the output. As previously mentioned, in the configuration of FIG. 2, the VCA output operates to cancel the crosstalk component in the adjacent fundamental direction (the passive matrix has crosstalk in it).

Thus, for example, if both inputs are provided with equal in-phase signals, then R _t = L _t = (for example) 1, resulting in g _c = zero and g _l , g _r and g _s are all zero and we have:

L _out = 1-1 * 1/2 * (1 + 1) -0 * 1/2 * (1-1) = 0 R _out = 1-1 * 1/2 * (1 + 1) + 0 * 1/2 * (1-1) = 0 C _out = 1/2 * (1 + 1) -0 * 1/2 * 1-0 * 1/2 * 1 = 1 S _out = 1/2 * (1-1) -0 * 1/2 * 1 + 0 * 1/2 * 1 = 0 The only output is from the desired _Cout . Similar calculations will prove that the same is true for signals from only one of the other three fundamental directions.

[0024]   Equations 5, 6, 7 and 8 may be equivalently written as:

L _out = 1/2 * (L _t + R _t ) * (1-g _c ) + 1/2 * (L _t −R _t ) * (1-g _s ) (Equation 9) C _out = 1 / _{2 * L t * (1-} g l) + 1/2 * R t * (1-g r) ( equation _{10) R out = 1/2} * (L t + R t) * (1-g c) -1 / 2 * (L _t −R _t ) * (1-g _s ) (Equation 11) S _out = 1/2 * L _t * (1-g _l ) −1 / 2 * R _t * (1-g _r (Equation 12) In this device, each output is the combination of two signals. L _out and C _out are of each gain of the sum and difference of the input signals and the sum and difference VCA (its inputs are derived from the central and environmental directions, the paired directions being at right angles to the left and right directions). Including both. C _out and S _out are the actual input signals and the respective gains of the left and right VCA (the inputs are derived from the left and right orientations, each paired direction being at right angles to the central and environmental directions). Including.

Considering the non-fundamental direction, R _t is provided with the same signal L _t of the same polarity but attenuated. This condition represents a signal placed somewhere between the left and center fundamental directions, and therefore should carry the outputs from L _out and C _out and little or no output from R _out and S _out .

For R _out and S _out , this zero output is achieved if the two terms are of equal magnitude but opposite polarities.

For R _out , the relationship to this offset is:

The magnitude of [1/2 * (L _t + R _t ) * (1-g _c ) = the magnitude of [1/2 * (L _t −R _t ) * (1-g _s ) (equation 13 ) For S _out , the corresponding relationship is as follows.

[1/2 * L _t * (1-g _l ) size = [1/2 * R _t * (1-g _r ) size (Equation 14) Pan between any two basic directions A consideration of the signal that has been dropped (or simply placed) will reveal the same two relationships. In other words, if each input signal represents a sound that is panned between any two adjacent outputs, then these loudness relationships are such that the sound emerges from the outputs corresponding to these two adjacent fundamental directions and the other two outputs are It will ensure that you do not convey anything. To substantially achieve that result, the magnitudes of the two terms in each equation 9-12 should be forced, ie, forced, to be equal. This can be achieved by seeking to keep the relative magnitudes of the two pairs of signals in the active matrix equal. That is, the magnitude of [(L _t + R _t ) * (1-g _c ) = the magnitude of [(L _t −R _t ) * (1-g _s ) (Equation 15) [L _t * (1-g _l) size _{= [R t * (1-} g r) of the desired relationship in size (represented by the formula 16) 15 and 16, formula except for the omission of scaling 13 and 14
Is the same as The polarities to which each signal is combined and their scaling can be processed when the respective outputs are obtained, as in combiners 14, 16, 18 and 20 of FIG.

The present invention is based on the discovery of equal amplitude magnitude relationships not previously appreciated, and is preferably based on the use of automatic feedback control to maintain these relationships, as described below. .

From the above discussion of cancellation of unwanted crosstalk signal components and the requirement for the fundamental direction, it can be concluded that the maximum gain for VCA should be unity for the scaling used in this description. When the gain of one VCA in the pair needs to rise from its quiescent value towards 1, the other of the pair can either stay in quiescent gain or move in the opposite direction. A convenient practical relationship is to keep the product of the same pair of gains constant.
By using an analog VCA where the dB gain is a linear function of the VCA control voltage, this will happen automatically if the control voltage is applied equally (but in reverse polarity) to the two of the pairs. Of course, the invention may be implemented digitally or in software rather than using analog components.

Thus, for example, given a static gain of 1 / a, the practical relationship between the two gains of each pair could be their product:

The values for g _l * g _r = 1 / a ² and g _c * g _s = 1 / a ² “a” will be in the range 10-20.

FIG. 3 shows a feedback derived control system for the left and right VCAs of FIG. 2 (6 and 12 respectively). It receives the input signals L _t and R _t , processes them to obtain intermediate signals L _t * (1-g _l ) and R _t * (1-g _r ) and compares the magnitudes of the intermediate signals. , Generate an error signal in response to any difference in magnitude. The error signal should reduce the magnitude difference through each VCA. One way to achieve such a result is to rectify the intermediate signals to derive their magnitude.
One of the added magnitude signal to the comparator, the comparator, for example, is that the increase of the L _t signal is to control the gain of each VCA polar such as to reduce the and g _r increases g _l . Each circuit value (or equivalent value in a digital or software embodiment) is chosen such that the quiescent amplifier gain is less than 1 if the comparator output is zero.

In the analog domain, a practical way to implement the comparison function is to convert the two magnitudes to the logarithmic domain and have the comparator subtract them rather than determine their ratio. Many analog VCA's have a gain that is proportional to the exponent of the control signal so that the logarithmic based control output of the comparator is inherently and expediently taken. However, in contrast, if implemented digitally, it would be more convenient to divide the two magnitudes and use the resulting one as a direct multiplier or divisor for the VCA function.

More specifically, as shown in FIG. 3, the L _t input is applied to the left VCA 6 and linear combiner 22, where it is scaled by +1. The left VCA6 output is applied to combiner 22 with -1 scaling (thus forming a subtractor),
Is applied to the full wave rectifier 24. The R _t input is applied to the right VCA 12 and the linear combiner 26, where it is scaled by +1. Right VCA 12 output is-
A scale of 1 (thus forming a subtractor) is applied to combiner 26 and the output of combiner 26 is applied to full wave rectifier 28. Each output of rectifiers 24 and 28 is applied to the non-inverting and inverting inputs of operational amplifier 30, respectively. The output of the op amp 30 provides a control signal, characterized by an error signal, which is applied to the gain control input of VCA 6 without inversion and to the gain control input of VCA 12 with polarity inversion. The error signal indicates the difference in the magnitude of the two signals that should be equalized in magnitude. This error signal is used to "point" each VCA in the correct direction to reduce the magnitude difference between each intermediate signal. The outputs to combiners 16 and 18 are taken from VCAs 6 and 12. Therefore, only the component of each intermediate signal is output coupler, i.e., added to L _t g _r and R _t g _l.

Due to the steady state signal conditions, the magnitude difference can be reduced to a negligible magnitude by providing sufficient loop (closed circuit) gain. However, it is not necessary to reduce the magnitude difference to zero or a negligible amount to achieve substantial crosstalk cancellation. For example, a loop gain sufficient to reduce the dB difference by a factor of 10 results in a crosstalk that is theoretically better than a 30 dB reduction at worst. For dynamic conditions, the time constant of the feedback controller should, in one aspect, be chosen to force it to be of equal magnitude to render it essentially inaudible for at least most signal conditions. is there. The details of time constant selection in the various configurations described are outside the scope of the present invention.

The circuit parameters give a negative feedback of about 20 dB and the VCA gain is unity.
It is desirable to be selected so as not to exceed. The VCA gain is from a small value (eg, 1 / a ² much less than 1) to a number of 1 (units for each scaling example described herein in connection with the apparatus of FIGS. 2, 4 and 5). Original) It changes within the following range. Due to the negative feedback, the device of Figure 3 operates to keep the signals entering the rectifier approximately equal.

If the gain is small, its exact value is not conclusive, so one of the gains in the pair is 1
Any relationship that forces the other gain to a small value whenever it rises toward will yield similar acceptable results.

The feedback derived control system for the central and environmental VCA (8 and 10 respectively) of FIG. 2 is substantially the same as the apparatus of FIG. 3 as already mentioned. However, instead of receiving L _t and R _t , their sum and difference are received, and VCA6 and VC
Its output from A12 (which constitutes the respective intermediate signal component) is applied to combiners 14 and 20.

Therefore, a high degree of crosstalk can be achieved under a wide variety of input signal conditions by using circuit elements having no special requirement for accuracy and at the same time utilizing a simple control path integrated in the signal path. Offsets can be achieved. The feedback derived control system is
Intermediate Audio Signal Operates to process each pair of audio signals from the passive matrix such that the relative amplitudes of the intermediate audio signals of each pair are forced to be equal.

The feedback derived control system, shown in FIG. 3, has two VCA 6 and VC so that each input to rectifiers 24 and 28 is forced to be equal.
Control the gain of A12. To the extent that these two terms are forced to be equal,
It depends on the characteristics of the rectifiers, their subsequent comparators 30, and the gain / control relationship of each VCA. The greater the loop gain, the closer the equality gets, but the forcing towards equality will occur irrespective of the characteristics of these factors (assuming that the signal polarities are of course designed to reduce the level difference). Conditions). In fact, the comparator does not have infinite gain and is implemented as a finite gain subtractor.

Given that the rectifiers are linear, ie their output is directly proportional to the magnitude of the input,
The output of the comparator or subtractor is a function of the signal voltage or current difference. If instead the rectifiers respond to the logarithm of their input magnitude, ie the level expressed in dB, the subtraction done at the comparator inputs is equivalent to taking a ratio of the input levels. This is beneficial as the result then does not depend on the absolute level of the signal but only on the difference of the signals expressed in dB.

Considering the source signal level in dB, which more closely reflects human perception,
Another issue is equality, which means that the loop gain is independent of magnitude, and thus the degree of forcing equality is also independent of magnitude. Of course, at some very low levels, the logarithmic rectifier will not work correctly and therefore there will be an input threshold below which the equalization constraint will cease. However, as a result, control can be maintained over 70 or more dB ranges without the need for extremely high loop gain for high input signal levels, but with potential problems for the stability of the resulting loop. .

Similarly, VCA 6 and 12 are directly or inversely proportional to their control voltage (ie, multiplier or divisor). This means that if the gain is small, a small absolute change in control voltage will be d.
It will have the effect of causing a large change in gain, represented by B. For example, with a maximum gain of 1 and a control voltage V _c varying from 0 to 10 volts, with a gain of A = 0.1 *, as required in this feedback derived control system configuration.
Consider a VCA that can be expressed as V _c . When V _c is close to its maximum value, a change of 100 mV (millivolt) from 9900 to 10000 mV is 20 * l
It gives a gain change of og (10000/99000), that is, about 0.09 dB.
When V _c is much smaller, the change of 100 mV from 100 to 200 mV is 20 * l
og (200/100), that is, a gain change of 6 dB is given. As a result, the response speed will change drastically depending on the actual loop gain and hence the magnitude of the control signal. Again, there may be issues with loop stability.

This problem can be eliminated by using a VCA whose dB gain is proportional to the control voltage or, using a different expression, whose voltage or current gain depends on the exponent or antilogarithm of the control voltage. . Then a small change in the control voltage, such as 100 mV, will give the same dB gain change wherever the control voltage is within that range. Such a device is readily available as each analog IC and its characteristics or approximations thereof are easily achieved in digital implementations.

Therefore, the preferred embodiment uses a logarithmic rectifier and an exponentially controlled variable gain amplification, and a more uniform equalization (considered in dB) for a wide range of input levels and two input signal ratios. To force.

In human hearing, the perception of direction is not constant with frequency, so a rectifier is entered to emphasize each frequency that contributes most to the human sense of direction and to treat frequencies that can lead to improper orientation. It is desirable to add some frequency weighting to the signal. Thus, in a practical embodiment, the rectifiers 24 and 28 of FIG. 3 are preceded by empirically derived filters to attenuate low and very high frequencies and provide a slowly rising response about the center of the audible range. give. These filters do not change the frequency response of the output signal, only the VCA gain of the control signal and the feedback derived control system.

A device equivalent to the combination of FIGS. 2 and 3 is shown functionally and diagrammatically in FIG. The difference from the combination of FIGS. 2 and 3 of the same figure is that instead of receiving the output signal from the passive matrix from which the cancellation components are derived, each output combiner responds to the L _t and R _t input signals with the passive matrix output. To generate a signal. The device gives the same result as given by the combination of FIGS. 2 and 3, provided that the addition coefficients are essentially the same in the passive matrix. FIG. 4 incorporates the feedback device described in connection with FIG.

More specifically, in FIG. 4, as in the passive matrix configuration of FIG.
The L _t and R _t inputs are first applied to the passive matrix including combiners 2 and 4. Similarly L _t input is passive matrix "left" output, is applied to one input of a linear combiner 34 with a scaling of the "left" VCA32 and +1. The left VCA 32 output is added to combiner 34 with -1 scaling (thus forming a subtractor).
. The R _t input, which is also a passive matrix “right” output, is applied to one input of the “right” VCA 44 and a linear combiner 46 with +1 scaling. The right VCA 44 output is added to combiner 46 with a -1 scaling (thus forming a subtractor). The outputs of combiners 34 and 46 are the signals L _t * (1-g _l ) and R _t * (1-g _r ), respectively, which keep these signals equal or force them to be equal. Is desirable. To achieve that result, these signals are preferably applied to a feedback circuit such as that shown in FIG. 3 and described in connection therewith. The feedback circuit then controls the gain of VCA 32 and 44.

Still referring to FIG. 4, the “center” of the passive matrix from the coupler 2
The output is applied to the "center" VCA 36 and one input of a linear combiner 46 with a scale of +1. The output of the central VCA 36 is added to the combiner 38 with a scale of -1 (
Therefore, it constitutes a subtractor). The passive matrix "environment" output from combiner 4 is applied to one input of "environment" VCA 40 and linear combiner 42 having a scale of +1. The environment VCA 40 output is added to combiner 42 with a -1 scaling (thus forming a subtractor). The outputs of combiners 38 and 42 are respectively the signals 1/2 * (L _t
+ R _t ) * (1-g _c ) and 1/2 * (L _t -R _t ) * (1-g _s ), it is either to keep these signals equal or to force them to be equal. desirable. To achieve that result, these signals are preferably applied to a feedback circuit such as that shown in FIG. 3 and described in connection therewith. The feedback circuit then controls the gain of VCA 38 and 42.

The output signals L _out , C _out , S _out and R _out are generated by combiners 48, 50, 52 and 54. Each combiner has two VCAs (each VCA) to provide a cancellation signal component and one or both input signals that cause each passive matrix signal component to be provided.
The CA output receives a component of each intermediate signal which is sought to be kept equal in magnitude). More particularly, the input signal L _t is the L _{o ut} coupler 48 on a scale of +1, in addition to the S _out combiner 52 with a measure of C _out combiner 50 to and +1/2 a measure of the + 1/2 To be The input signal R _t is scaled by +1 and the R _out combiner 54
To C _out combiner 50 with +1/2 scaling and S _out combiner 5 with -1/2 scaling.
Added to 2. The left VCA 32 output is applied to C _out combiner 50 with a -1/2 scale and similarly to S _out combiner 52 with a -1/2 scale. Right VCA44 output is-
It is added to C _out combiner 50 with a scale of 1/2 and to S _out combiner 52 with a scale of +1/2. The central VCA 36 output is applied to L _out combiner 48 with a -1 scale and to R _out combiner 54 with a -1 scale. The environment VCA 40 output is applied to L _out combiner 48 with a scale of -1 and to R _out combiner 54 with a scale of +1.

In various figures, eg, FIGS. 2 and 4, initially it appears that each cancellation signal does not conflict with each passive matrix signal (eg, within each cancellation signal a passive matrix signal is added). It will be noted that some are added to the combiner with the same polarity as). However, in operation, when the cancellation signal becomes significant, it will have a polarity that is not in conflict with the passive matrix signal.

A combination of FIGS. 2 and 3 and another device equivalent to FIG. 4 is shown functionally and diagrammatically in FIG. In the configuration of FIG. 5, each signal that should be kept equal has each output decoupler and each V
This signal is applied to the CA control feedback circuit. These signals include passive matrix output signals. In the device of FIG. 4, in contrast, each signal applied to the output combiner from the feedback circuit is the VCA output signal and excludes each passive matrix component. Therefore, in FIG. 4 (and in the combination of FIGS. 2 and 3) the passive matrix components must be explicitly combined with the output of the feedback circuit, whereas in FIG. 5 each output of each feedback circuit is It contains a passive matrix component and is sufficient by itself. It will also be noted that in the apparatus of FIG. 5, each intermediate signal output is applied to the output combiner, rather than each VCA output, each of which constitutes only one component of the intermediate signal. Nevertheless, FIG. 4 and FIG.
Each of the configurations (along with the combination of FIGS. 2 and 3) is equivalent, and the output coefficients from FIG. 5 are identical to those from FIG. 4 (and the combination of FIGS. 2 and 3) if the sum coefficients are accurate. .

[0056]   In FIG. 5, by processing the passive matrix output, equations 9, 10, 11 and
4 intermediate signals [1/2 * (L_t+ R_t) * (1-g_c)], [1/2 * (L_t
-R_t) * (1-g_s)], [1/2 * L_t* (1-g_l)] And [1/2 * R_t* (1-g _r )] Is then obtained and then added or subtracted to derive the desired output. Belief
The number also includes a rectifier of two feedback circuits, as already described in connection with FIG.
And a feedback circuit that equalizes the magnitude of each signal in the pair.
It is desirable to act to keep. As used in the configuration of FIG. 5, each frame of FIG.
The feedback circuit is connected to the combiner 22 and 26 rather than from the VCA 6 and 12.
Output to those output couplers.

Still referring to FIG. 5, the connections between couplers 2 and 4, VCAs 32, 36, 40 and 44 and couplers 34, 38, 42 and 44 are the same as in the apparatus of FIG. Also, in both devices of FIGS. 4 and 5, couplers 34, 38, 42 and 46 are shown.
Output of two combiner circuits 34 and 46 to generate a control signal for two feedback control circuits (VCA 32 and 44) to the first such circuit, and V
The outputs of combiners 38 and 42 are applied to such a second circuit) to generate control signals for CAs 36 and 40. In FIG. 5, the output of the combiner 34, L _t *
The (1- _gl ) signal is applied to the C _out combiner 58 with a scale of +1 and to the S _out combiner 60 with a scale of +1. The output of the combiner _{46, R t * (1-} g r) signal, the _{C out} combiner 58 with a measure of + 1, also applied to the _{S out} combiner 60 with a measure of -1. The output of combiner 38, the 1/2 * (L _t + R _t ) * (1-g _c ) signal, is +1 scaled to L _out combiner 56 and +1 scaled to R _out combiner 62. Added. The output of the combiner 42, the 1/2 * (L _t -R _t ) * (1-g _s ) signal, is scaled by +1 to the L _out combiner 56 and scaled by -1 to the R _out combiner. 62.

Unlike prior art adaptive matrix decoders where the control signal is generated from the input, the present invention preferably uses closed loop control, where the magnitude of the signal providing the output is measured for adaptation. And feedback is given. In particular, unlike prior art open loop systems, the desired cancellation of unwanted signals for non-fundamental directions does not depend on exact matching of signal and control path characteristics, and closed loop configurations greatly reduce the need for circuit element accuracy. Reduce.

Ideally, except for practical circuit imperfections, each of the “keep equal size” configurations of the present invention are arbitrary provided in the L _t and R _t inputs with known relative amplitudes and polarities. The source of
"Complete" in the sense that it gives a signal from the desired output and only a negligible signal from the other outputs. "Known relative amplitudes and polarity" means that the L _t and R _t inputs represents one of the positions between the basic direction for basic direction or adjacent.

Considering equations 9, 10, 11 and 12 again, it will be appreciated that the total gain of each variable gain circuit incorporating a VCA is a subtractor of the form (1-g). Each VCA gain varies from a small value to 1 but does not exceed 1. Correspondingly, the variable gain circuit gain (1-g) can vary from a value very close to 1 to zero. Therefore, FIG. 5 can be rewritten as FIG. 6 in which all VCAs and associated subtractors are replaced by only one VCA, the gain of which is in the opposite direction to that of the VCA of FIG. Change. Thus, all variable gain circuit gains (1-g) (e.g. implemented by a VCA with a gain "g" whose output is subtracted from the passive matrix output, as in FIGS. 2 and 3, 4 and 5). Is replaced by a corresponding variable gain circuit gain “h” (eg, implemented by an isolated VCA with gain “h” acting on the passive matrix output). Gain "(1-g)"
6 is equivalent to the gain "h" and the feedback circuit operates to maintain equality between the required signal pair magnitudes, the configuration of FIG. 6 is equivalent to that of FIG. And will give the same output. In fact, FIGS. 2 and 3, 4, 5, and 6
, That is, all disclosed configurations are equivalent to each other.

The configuration of FIG. 6 functions equally and exactly the same as all the previous configurations, but the passive matrix is implicitly invisible and potential. In the rest or unoriented state of the previous configuration, the VCA gain g drops to a small value. In the configuration of FIG. 6, when all VCA gains h rise to their maximum, ie 1 or close to them, a corresponding undirected condition occurs.

[0062] More particularly reference to Figure 6, "left" output of the same input signal L _t same passive matrix and are, in order to generate an intermediate signal L _t * h _l, a gain h _l "left" VCA64 Added to. Similarly, for the same passive matrix as the input signal R _t
Right "output, in order to generate an intermediate signal _{1/2 * (L t +} R t) * h c, gain _{h c}
Is added to the "right" VCA 66. Of passive matrix from combiner 4
The "environmental" output is applied to an "environmental" VCA 68 having a gain h _s to generate an intermediate signal 1/2 * (L _t -R _t ) * h _s . As described above, the h gain characteristic is (1
-G) The VCA gain h operates inversely to the VCA gain g so that it has the same gain characteristic.

Generation of Control Voltages The analysis of control signals detailed in connection with the embodiments described thus far provides a better understanding of the invention and from a pair of audio input signal streams, each associated with a direction, It is useful in explaining how the teachings of the present invention can be used to derive five or more audio signal streams.

In the following analysis, the results consider an audio source that is panned clockwise around the listener in a circle starting from behind and going back through the left, center, right. Will exemplify the results. The variable α is a measure of the angle (°) of the image with respect to the listener, where 0 ° is at the back and 180 ° is at the center front. The input magnitudes L _t and R _t are related to α by

[0065]

[Formula 1]

[Formula 2]

A one-to-one mapping between the parameter α and the magnitude and polarity of the input signal (
Function). That is, the use of α leads to more convenient analysis. If α is 90 °, then L _t is infinite and R _t is zero, that is, left only. If α is 180 °, L _t
And R _t have the same polarity (center front) and are equal. If α is 0 °, L _t and R _t are equal but opposite in polarity (center back). As discussed further below, the particular values of interest occur when L _t and R _t differ by 5 dB and have opposite polarities, which gives an α value of 31 ° on either side of zero. In fact, the left and right front speakers are
It is generally placed anterior to +/- 90 degrees with respect to the center (eg, +/- 30 to 45 degrees), so α does not actually represent an angle for the listener, but panning (
(Horizontal movement) is an arbitrary parameter. The figures described are on the horizontal axis (α =
The center of 180 ° represents the front of the center, and the left and right extremes (α = 0 and 360 °) represent the rear.

As already mentioned with respect to the description of FIG. 3, the convenient and practical relationship between the gains of the feedback-derived control system pair VCA keeps the product of each gain constant. With each VCA controlled exponentially such that one gain increases and the other decreases, it is automatically supplied to both pairs with the same control signal, as in the embodiment of FIG. Happens in a normal way.

[0068] represents an input signal in L _t and R _t, and set the product of the VCA gains g _l and g _r 1 / a ² and a sufficiently large loop gain forced into the resulting equality is completed Assuming that the feedback derived control system of FIG. 3 adjusts the VCA gain to satisfy the following equation:

[0069]

[Formula 3] Furthermore, g _l · g _r = 1 / a ² (Formula 19)

Clearly, first of all, the absolute amounts of L _t and R _t are inadequate. The result depends only on their ratio L _t / R _t , which is called X. Substituting g _r from the second equation into the first equation, we obtain the quadratic equation in the form of g _l with the answer (other roots of the quadratic equation do not represent the real system).

[0071]

[Formula 4]

Plotting g _l and g _r against the panning angle α yields FIG. 7. As expected, when the input represents only left (α = 90), g _l rises from a very low value in the back to a maximum of 1 and then returns to a low value for the center front (α = 180). . In the right half, _gl stays very small. Similarly and symmetrically, g _r is
Small except for the center of the right half of bread, 1 when α is 270 degrees (right only)
Rise to.

The above results are for a control system with L _t / R _t feedback derivation. The sum / difference feedback derived control system operates in exactly the same way and gives a plot of the sum gain g _C and the difference gain g _s as shown in FIG. Again, as expected, the sum gain rises to 1 in the center front and falls to a low value in the other, while the difference gain rises to 1 behind.

If the feedback derived control system VCA gain depends on the exponent of the control voltage, as in the preferred embodiment, then the control voltage depends on the logarithm of the gain. Therefore, from the above equations, L _t / R _t and the sum / difference control voltage, that is, the equation concerning the output of the comparator (comparator 30 of FIG. 3) of the control system which is feedback derived can be derived. FIG. 9 illustrates left / right and sum / difference control voltages, the latter being inverted (ie, effectively difference / difference in one embodiment where the maximum and minimum values of the control signal are +/− 15 volts).
Sum). Obviously, other scaling is possible.

The curves in FIG. 9 intersect at two points, one where the signal represents an image somewhere in the left rear of the viewer and the other somewhere in the front half. Due to the inherent symmetry of the curve, these intersecting points are exactly in the middle of each α value corresponding to the adjacent principal direction. In FIG. 9, they occur at 45 and 225 degrees.

The prior art (eg, US Pat. No. 5,644,640 of the present inventor James W. Fosgate) describes 2
It is shown that from one main control signal it is possible to derive a further (larger positive) or smaller (smaller positive) further control signal of the two. However, the prior art derives the main control signal in different ways and utilizes the resulting control signal in different ways. FIG. 10 illustrates a signal equal to the smaller of each curve in FIG. This derived control rises to the highest point (maximum amount) when α is 45 degrees, that is, the value where the two original curves intersect.

It may be undesirable for the highest point of the derived control signal to rise to that highest point exactly at α = 45 degrees. In a practical embodiment, it is desirable that the derived basic direction representing the left rear part be closer to the rear part, ie have a value less than 45 degrees. Cancel one or both of the left / right and sum / difference control signals (add or subtract constants) so that their curves cross at the desired value of α before taking a more positive or more negative effect. ) Or by scaling the exact position of the highest point can be moved. For example, FIG. 11 shows the same behavior as FIG. 10, except that the sum / difference voltage is scaled by 0.8, so that the current highest point occurs at α = 31 degrees. Indicates.

Exactly in the same way, the maximum point corresponds to the right rear part of the listener by comparing the inverted left / right control with the inverted sum / difference control and using similar cancellation or scaling. A second new control signal that occurs at a given position is the desired and given α (eg, zero 31
It can be derived at 360-31, or 329 degrees), which is symmetrical to the left rear on the opposite side. It is the left / right inversion of FIG.

FIG. 12 shows the effect of adding these derived control signals to each VCA so that the most positive value gives a gain of one. Just as the left and right VCA's give a gain of up to 1 in the left and right fundamental directions, one signal is at each given position (α = 31 degrees on either side of the zero in this example). When placed, these derived left rear and right rear VCA gains rise to 1 but remain very small for all other positions.

Similar results can be obtained with a linearly controlled VCA. The curves for the main control voltage versus the panning parameter α are different, but will intersect at each point that can be chosen by proper scaling or cancellation. Therefore, additional control voltages for each particular image position other than the initial four basic directions can be derived with less actuation.
Obviously, you invert the control signal so that less (more negative) than greater (
It is possible to derive a new control signal by taking the more positive one.

[0081]   Move the main control signal intersection before taking the greater or lesser
Through modification to the same signal, instead of offsetting or scaling
Alternatively or additionally, non-linear actuation could alternatively be configured. The modification is L _t And R_tAlmost any desired ratio of magnitude and relative polarity of (each input signal)
It will be clear that it allows the generation of further control signals, which have the highest point.

Adaptive Matrix with Five or More Outputs FIGS. 2 and 4 show that in addition to canceling (eliminating) unwanted crosstalk, the passive matrix can have adaptive cancellation terms. In these cases, four VCA
There are four possible cancellation terms derived through, and there is one source, corresponding to the dominant output from one of the four outputs (left, center, right and behind) in one of the four fundamental directions. For each VCA reached the highest gain (typically 1). The system is complete in the sense that one signal panned between two adjacent fundamental directions provides little or no output other than the outputs corresponding to the two adjacent fundamental directions.

This principle can be extended to active systems with 5 or more outputs. In such cases, the system is not "perfect", but the results are not audibly impaired by crosstalk, since the unwanted signals can still be sufficiently canceled. See, for example, the output matrix of FIG. FIG. 13 is a functional and schematic view of a portion of an active matrix according to the present invention, which helps to explain how more than four outputs can be obtained. FIG. 14 shows the derivation of the six cancellation signals that can be used in FIG.

First, referring to FIG. 13, there are six outputs. That is, left front (L _out ),
Center front (C _out ), right front (R _out ), center rear (or environment) (S _out ), right rear (RB _out ) and left rear (LB _out ). For the three front and environmental outputs, the starting passive matrix is identical to that of the four-output system above (direct input L _t , 1/2 scaled to give the central front and L applied to the linear combiner 80). _{The combination of t} plus R _t, the combination of L _t minus R _t , scaled by 1/2 to give the central posterior and added to the linear combiner 82 and the direct input R _t ). There are two additional rear outputs. That is, a left rear and back rear, a measure of 1, respectively L and _t and scale of -b of R _t and a linear combiner 84 is added and that the respective scaled -b L _t and scaling 1 Left rear and rear rear corresponding to different combinations of inputs with the formulas LB _out = L _t -b * R _t and RB _out = R _t -b * L _t , which are produced by adding R _t and R _t to the linear combiner 86. Is. Where b is generally 1
It is a positive coefficient of less than, for example, 0.25. Note that symmetry is not essential to the invention but is expected in any practical system.

In FIG. 13, in addition to each passive matrix term, the output linear combiner (88, 90,
92, 94, 96 and 98) are the multiple active cancellation terms (lines 100, 102, 104, 106, 108, 110) as required to cancel the passive matrix output.
, 112, 114, 116, 118, 120 and 122). These terms consist of the input and / or combination of each input multiplied by the gain of each VCA (not shown) or the combination of each input with the input multiplied by the gain of each VCA. As already mentioned, each VCA is controlled so that for one basic input condition the gain of each VCA rises to 1 and is essentially smaller for each of the other conditions.

The configuration of FIG. 13 has six basic directions given by inputs L _t and R _{t of} limited relative magnitude and polarity, each of which results in a signal from only the appropriate output. However, at the other five outputs the signals should essentially cancel. Two
For input conditions that represent a signal panned between two adjacent elementary directions, the outputs corresponding to these elementary directions should provide signals, but the remaining outputs should provide little or no. Therefore, for each output there are some cancellation terms in addition to the passive matrix (actually three or more are shown in FIG. 13), which are unnecessary for each input corresponding to each of the other fundamental directions. Will correspond to the output. actually,
The arrangement of FIG. 13 uses the central rear S _out output (and thus the combiner 8) so that the central rear is merely a halfway pan between the rear left and rear right rather than the sixth fundamental direction.
2 and 94 are removed) can be modified.

[0087]   6 possibilities for either the 6-output system of FIG. 13 or its 5-output alternative
There is a canceling signal. That is, left / right and sum / difference feedback derived control systems
4 derived through two pairs of VCA that are each part of the
See also the embodiment of FIG. 14 described below) controlled left rear and right rear V
There are two additional signals derived via CA. The gain of 6 VCA is shown in Fig. 7 (
Left rear g_lAnd right rear g_r), FIG. 8 (sum g_cAnd the difference g_s) And FIG. 12 (left rear g _lb And right rear g_rb). The cancellation signal is an undesired cross signal, as described below.
With each coefficient calculated or otherwise selected to minimize talk
Summed with passive matrix terms.

By considering the input signal and VCA gain for all other fundamental directions, we arrive at the required cancellation mixing factor for each fundamental output, but these VC
Remember that the A-gain rises to 1 only for the signal in the corresponding fundamental direction and falls off from 1 very rapidly as the image moves further away.

Therefore, in the case of the left output, it is necessary to consider the center front, only the right, the right rear, the center rear (not the true basic direction in the case of 5 outputs), and the left rear.

Let us consider the left output L _out in detail for the five-output variant of FIG. It contains terms from the passive matrix L _t . If L _t = R _t and g _c = 1 then in order to cancel the output when the input is centered, just as in the 4-output system of FIG. 2 or 4, the term −1 / 2 * g _c * (L _t + R _t ) is required. If the input is center-rear or somewhere between center-rear and right-front (and thus includes right-rear), then FIG.
Or exactly the term -1 / 2 * g _c * (L _t -R _t ) is required as in the 4 output system of 4. When the input represents left back, the gain g _lb it takes a signal from the left rear VCA vary as shown in FIG. 12. This can clearly give a significant cancellation signal only when the input is in the left rear area. The left rear VCA can be thought to be somewhere between the left front represented by L _t only and the center rear represented by 1/2 * (L _t −R _t ), so the left rear VCA It should be expected to act on the signal combination of.

Various mixed bonds can be used, but left and difference VCA, ie _gl * L _t
And by using the sum of each signal already penetrating 1/2 * g _s * (L _t −R _t ), the coupling is not exact, but the position of each signal panned in the left rear region And gives a better offset for each of these pans as well as the base left rear itself. This left rear position, considered to be midway between the left and the back, both have a finite value less than 1 for both g _l and g _s . Therefore, the following expression is expected for L _out . L _out = [L _t ] -1 / 2 * g _c * (L _t + R _t ) -1 / 2 * g _s * (L _t -R _t ) -x * g _lb * ((g _l * L _t + g _s * 1/2 * (L _t -R _t )) (Equation 21)

[0092]   The coefficient x is empirical or, if the sound source is in the area to the left rear basic direction, the accurate VC
It can be derived from consideration of A-gain. Term [L_t] Is the passive matrix term. Each item 1/2
* g_c* (L_t+ R_t), -1 / 2 * g_s* (L_t-R_t) And 1/2 * x * g_lb* ((g_l* L_t+ g_s* 1/2 * (L _t -R_t)) Is the output audio signal L_outTo obtain the linear combiner 88 (FIG.
) L_tRepresents a cancellation term (see FIG. 14) that can be combined with As already mentioned, FIG.
More than two (100 and 102) crosstalk cancellation terms
There can be power.

[0093]   R_outThe following equations for are derived similarly or by symmetry. R_out= [R_t] -1 / 2 * g_c* (L_t+ R_t) + 1/2 * g_s* (L_t-R_t) -1 / 2 * x * g_rb* ((g_r* R_t-g _s * (L_t-R_t)) (Equation 22)

[0094]   [R_t] Terms are passive matrix terms. -1 / 2 * g_c* (L_t+ R_t), 1/2 * g_s* (L_t-R
_t) And -1 / 2 * x * g_rb* ((g_r* R_t-g_s* (L_t-R_t)) Is the output audio signal R_out R of the linear combiner 98 (FIG. 13) to obtain_tCancellation term that can be combined with (see FIG. 14)
Represents). As already mentioned, from the two (120 and 122) shown in FIG.
There can be more than two crosstalk cancellation term inputs.

The central forward output C _out is a 4-output system in addition to the passive matrix term 1/2 * (L _t + R _t ) -1 / 2 * g _l * L _t and -1 / 2 * g _r * R Includes left and right offset terms as for _t . That is, C _out = 1/2 (L _t + R _t ) -1 / 2 * g _l * L _t * -1 / 2 * g _r * R _t * (Equation 23) For left rear, center rear or right rear Explicit offsetting terms are not needed, as they are valid pans between the left and right anterior through the back (four output environment) and have already been offset. The term [1/2 (L _t + R _t )] is a passive matrix term. Term
-1 / 2 * g _l * L _t and -1 / 2 * g _r * R _t are applied to inputs 100 and 102 to obtain the output audio signal C _out and the scale of linear combiner 90 (FIG. 13). L _t
And R _t represent a cancellation term (see FIG. 14) that can be combined with R _t .

[0096]   The initial passive matrix for the left rear output is, as already mentioned, L_t-b * R _t Is. G for left-only input_lWhen = 1, the necessary offset term is therefore clear.
Light-g_l* L_tIs. G for right-only input_rWhen = 1, the offset term is +
b * g_r* R_tIs. L for center forward input_t= R_tAnd g_cIf = 1, L_t-
b * R_tUnwanted output from (1-b) * g_c* 1/2 * (L_t+ R_t) Can be offset.
The right rear offset term is R_outHave the same optimized coefficient y as the term used for
-G_rb* (g_r* R_t-1 / 2 * g_s* (L_t-R_t)), Which is again empirically reached
Alternatively, it can be calculated from the VCA gain for left or right rear conditions. Therefore, LB_out= [L_t-b * R_t] -g_l* L_t+ b * g_r* R_t-(1-b) * g_c* 1/2 * (L_t+ R_t) -y * g_rb* (g_r* R_t-g_s* 1/2 *
(L_t-R_t)) (Equation 24) Similarly, RB_out= [R_t-b * L_t] -g_r* R_t+ b * g_l* L_t-1 (1-b) * g_c* 1/2 * (L_t+ R_t) -y * g_lb* (g_l* L_t-g_s* 1/2
* (L_t-R_t)) (Equation 25)

With respect to Equation 24, the term [L _t -b * R _t ] is a passive matrix term, and the terms -g _l * L _t, + b * g _r * R _t , -1 / 2 * (1-b ) * g _c * (L _t + R _t ) and -Y * g _rb * (g _r * R _t -g _s * 1/2 * (L _t -R _t )) obtain the output audio signal LB _out 13 represents a cancellation term (see FIG. 14) that can be combined with the linear combiner 92 (FIG. 13) L _t -b R _t . As previously mentioned, there can be more than two crosstalk cancellation term inputs, more than the two (108 and 110) shown in FIG.

With respect to Equation 25, [R _t −b * L _t ] is a passive matrix term and the components −g _r * R _t , b * L _t *
g _l , -1 / 2 * (1-b) * g _c * (L _t + R _t ), and -y * g _rb * (g _l * L _t + g _s * 1/2 * (L _t -R _t )) Represents the cancellation term (see FIG. 14) that may combine the output audio with R _t -b * L _t in the linear combiner 96 (see FIG. 1) to derive the signal RB _out . As already mentioned, 2 shown in FIG.
There may be more than two (116 and 118) crosstalk cancellation terms.

In practice, all coefficients require adjustment to offset the finite loop gain and other drawbacks of the feedback derived control system (which do not give exactly equal signal levels exactly), requiring six Other combinations of cancellation signals can be used.

These principles can, of course, be extended to embodiments having 6 or more outputs. Further control signals are derived by further using scaling, cancellation or non-linear processing of the two main control signals from the left / right and sum / difference feedback parts of the feedback derived control system to derive the desired other of α. It may be possible to generate an additional cancellation signal via the VCA where the gain rises to a maximum for a given value of. A synthesis process that considers each output in the presence of each signal in each of the other fundamental directions will in turn provide the appropriate terms and coefficients to produce the additional output.

Referring to FIG. 14, a left matrix signal output from the L _t input, a right matrix signal output from the R _t input, each linear combiner 132 having L _t and R _t of scaled +1/2 as inputs. And the input signals L _t and R to the passive matrix 130, which produce the environmental outputs from the linear combiner 134, respectively, whose input is the scaled +1/2 and −1/2 L _t and R _t , respectively. _t is added. The basic directions of the passive matrix are “left”, “center”, “right” and “environment”. Adjacent base directions are on axes that are at right angles to each other, with respect to these direction slogans (labels), the left is adjacent to the center and the environment, the environment is adjacent to the left and right, and so on.

The left and right passive matrix signals are applied to a first pair of variable gain circuits 136 and 138 and associated feedback derived control system 140. The central and environmental passive matrix signals are applied to a second pair of variable gain circuits 142 and 144 and associated feedback derived control system 146.

The “left” variable gain circuit 136 includes a voltage controlled amplifier (VCA) 148 with a gain _gl and a linear combiner 150. The VCA output is subtracted from the left passive matrix signal at combiner 150, resulting in an overall gain of the variable gain circuit of (1- _gl ) and an output of the variable gain circuit at the combiner output, which constitutes the intermediate signal ( 1- _gl ) *
It is to be in L _t. VCA148 output signal constituting a cancellation signal becomes g _l * _{L t.}

[0104] "right" variable gain circuit 138 includes an amplifier (VCA) 152 which is a voltage controlled with a gain g _r and a linear combiner 154. VCA output is subtracted from the right passive matrix signal in combiner 154, the total gain of the variable gain circuit becomes (1-g _r) and, constituting an intermediate signal, the output of the variable gain circuit at the combiner output ( 1-g _r) *
It is made to be R _t . VCA152 output signal g _r * _{R t} constitutes a cancellation signal. (
The 1-g _l ) * R _t and (1-g _r ) * L _t intermediate signals constitute the first pair of intermediate signals. The relative magnitude of this first pair of intermediate signals is preferably forced to be equal. This is accomplished by the associated feedback control system 140, described below.

The “middle” variable gain circuit 142 includes a voltage controlled amplifier (VCA) 156 with a gain g _c and a linear combiner 158. The VCA output is subtracted from the central passive matrix signal at combiner 158 to give a total gain of the variable gain circuit of (1-g _c ), and the output of the variable gain circuit at the combiner output, which constitutes the intermediate signal, is one. / 2 * (
1-g _c ) * (L _t + R _t ). The VCA 156 output signal 1/2 * g _c * (L _t + R _t ) constitutes the cancellation signal.

[0106]   The “environment” variable gain circuit 144 has a gain g_rAnd voltage control with linear combiner 162
A controlled amplifier (VCA) 160 is included. VCA output is environmentally passive at combiner 162
Subtracted from the matrix signal, the total gain of the variable gain circuit is (1-g_s)become
And the output of the variable gain circuit at the output of the combiner, which constitutes the intermediate signal, is 1/2 * (
1-g_s) * (L_t-R_t). VCA160 output signal 1/2 * g_s* (L _t -R_t) Constitutes the cancellation signal. 1/2 * (1-g_c) * (L_t+ R_t) And 1/2 * (1-g_s)
* (L_t-R_tThe intermediate signal constitutes the second pair of intermediate signals. This second pair of phases of the intermediate signal
It is desirable that the relative sizes are forced to be equal. This is described below
, Associated feedback control system 146.

A feedback derived control system 140 associated with the first pair of intermediate signals includes filters 164 and 166 that receive the outputs of combiners 150 and 154, respectively.
including. Each filter output is applied to a logarithmic rectifier 168 and 170 that rectifies the input to produce a logarithm. The rectified and logarithmic output is applied in reverse polarity to a linear combiner 172, which constitutes a subtractor at the input, and its output is applied to a non-inverting amplifier 174 (devices 172 and 174 are of the magnitude of FIG. 3). Corresponding to the comparator 30). Subtracting the logarithmized signal provides the comparison function. As already mentioned,
This is a practical way to implement the comparison function in the analog domain. In this case, V
CAs 148 and 152 are of the kind that inherently take the antilogarithm of their inputs, and thus take the antilogarithm of the control output of a logarithmically based comparator. Amplifier 174
The outputs of the two form the control signals for VCA 148 and 152. As already mentioned, if implemented digitally, it is more convenient to divide the two magnitudes and use the resulting one as a direct multiplier for the VCA function. As already mentioned,
Filters 164 and 166 are empirically derived and attenuate low frequencies and very high frequencies to provide a gradual rising response over the middle of the audible range. These filters do not change the frequency response of the output signal, only the control signal and VCA gain of the feedback derived control system.

The feedback derived control system 146 associated with the second pair of intermediate signals includes filters 176 and 178 that receive VCAs 158 and 162, respectively. Each filter output is applied to a log rectifier 180 and 182 that rectifies each input to produce its logarithm. The rectified and logarithmic output is applied in reverse polarity to a linear combiner 184, which constitutes a subtractor at the input, and its output is applied to a non-inverting amplifier 186 (devices 184 and 186 are of the magnitude of FIG. 3). Corresponding to the comparator 30). The feedback derived control system 146 is the control system 140.
Works the same way as. The output of amplifier 186 constitutes the control signal for VCA 158 and 162.

[0109]   Additional control signals are feedback derived control systems 140 and 146.
Derived from. The control signal of the control system 140 has first and second scaling, cancellation.
Etc. are used for functions 188 and 190. The control signal of the control system 146 is the first
And functions 192 and 194 for second scaling, offsetting, etc. Function 188, 1
90, 192 and 194 are one or more polarity reversals, amplitude cancellations, amplitude scales
And / or the above non-linear processing. Similarly, from the above description,
To generate additional control signals used for the one VCA 200 and the right rear 202
Less or more of the outputs of functions 188 and 192 and functions 190 and 194.
Many do so with less or more features 196 and 198
Each can be taken. In this case, a left rear cancellation signal and a right rear cancellation signal are generated.
An additional control signal is derived in the above manner to provide a control signal suitable for
Be done. The input to the left rear VCA 200 is the left of the linear combiner 204 and the environmental cancellation signal.
Is obtained by additively combining. The input to the right rear VCA 202 is a line
Obtained by subtractively combining the right and environmental cancellation signals of the form combiner 204
. Instead, and less desirably, the inputs to VCA 200 and 202 are:
From left and environmental passive matrix outputs and right and environmental passive matrix outputs respectively
Can be derived. The left rear VCA 200 has a left rear cancellation signal g_lb* 1/2 * (g_l* (L_t−
R_t)). The right rear VCA 202 receives the right rear cancellation signal g._rb1/2 * (g_r* R_t+ G _s (L_t-R_t)).

FIG. 15 is a circuit configuration diagram showing a practical circuit embodying each aspect of the present invention. Resistor values are shown in ohms. Where not indicated, capacitor values are given in microfarads.

The “TL074” in FIG. 15 is a Texas Instruments quadrature low noise JFET-input (high input impedance) general purpose operational amplifier intended for high fidelity and audio preamplifier applications. Details of the device are widely available in the published literature.
The data sheet is on the internet below << http://www.ti.com/sc/docs/products
You can find it in /analog/tl074.html >>.

“SSM-2120” in FIG. 15 is a semiconductor integrated circuit intended for audio use. It contains two VCAs and two level detectors, allowing logarithmic control of the gain or attenuation of each signal depending on the level magnitude of the signal sent to the level detector. Details of the device are widely available in the published literature. The data sheet can be found on the internet at << http://www.analog.com/pfd/1788_c.pdf >>.

The following table associates the terms used in this document with the labels on the VCA output of FIG. 15 and the labels on the vertical bus.

[0114]

[Table 1]

The labels in FIG. 15 regarding the lines that reach the output matrix resistors are intended to convey their function rather than the source of each signal. So, for example, a few top lines leading to the left front output are:

[0116]

[Table 2]

In FIG. 15, the matrix itself provides for the inversion of any term (such as U2C), regardless of the polarity of the VCA term. Further, “servo” in FIG. 15 refers to a feedback derived control system as described herein.

The present invention may be implemented by analog, hybrid analog / digital and / or digital signal processing, where each function is implemented in software and / or digital and / or firmware. Analog terms such as VCA, rectifier, etc. are intended to include their equivalents. For example, in a digital embodiment VCA is implemented by multiplication or division.

[Brief description of drawings]

FIG. 1 is a functional block diagram of a prior art passive decoding matrix useful in understanding the present invention.

FIG. 2 is a functional block diagram of a prior art active decoding matrix useful in understanding the present invention, in which a passively scaled output of each type is a linear combiner of each type. It is summed with the invariant passive matrix output.

FIG. 3 is a functional block diagram of a feedback derivation control system according to the present invention for the right and left VCAs, the sum and difference VCAs of FIG. 2 and other VCA embodiments of the present invention.

FIG. 4 is a functional block diagram of an apparatus according to the invention equivalent to the combination of FIGS. 2 and 3, in which the output combiner receives a cancellation component from a passive matrix. Instead, it produces a passive matrix signal component in response to the L _t and R _t input signals.

FIG. 5 is a functional block diagram according to the invention showing an apparatus equivalent to the present combination of FIGS. 2 and 3. In the configuration of FIG. 5, the signals that should be kept equal are output decoupler and VC
This signal is applied to the feedback circuit for A control. That is, the output of the feedback circuit contains passive matrix components.

FIG. 6 is a functional block diagram according to the present invention showing a device equivalent to the present combination device of FIGS. 2 and 3 and FIGS. 4 and 5; In this figure, the variable gain circuit gain (1-g) provided by the VCA and subtractor is replaced by a VCA whose gain varies in the opposite direction of each VCA in the VCA and subtractor configuration. In this embodiment, the passive matrix component is potential. In other embodiments, the passive matrix component is overt.

FIG. 7 is an idealized plot of left and right VCA gains g _l and g _r of an L _t / R _t feedback derived control system (vertical axis) against panning angle α (horizontal axis). It is a graph.

FIG. 8 is an idealized graph plotting the sum and difference VCA gains g _c and g _s of a sum / difference feedback derivation control system (vertical axis) against the panning angle α (horizontal axis). is there.

FIG. 9 shows that the maximum and minimum values of the control signal are +/- with respect to the panning angle α (horizontal axis).
FIG. 6 is an idealized graph plotting left / right and inverse sum / difference control voltages for a scale of −15 volts (vertical axis).

FIG. 10 is an idealized graph plotting the smaller of the curves of FIG. 9 (vertical axis) against the panning angle α (horizontal axis).

FIG. 11: The smaller one of the curves in FIG. 9 (vertical axis) with respect to the panning angle α (horizontal axis) when the sum / difference voltage is scaled by 0.8 before taking the smaller one of the curves Is an idealized graph that plots.

FIG. 12 is an idealization plot of left rear and right rear VCA gains g _lb and gr _b of a left rear / right rear feedback derived control system (vertical axis) against panning angle α (horizontal axis). It is the graph which was done.

FIG. 13 is a functional block diagram of a portion of an active matrix decoder according to the present invention in which six outputs are derived.

FIG. 14 is a functional block diagram showing the derivation of six cancellation signals for a 6-output active matrix decoder as in FIG.

FIG. 15   It is a circuit block diagram which shows the practical circuit which embodies each aspect of this invention.

─────────────────────────────────────────────────── ─── Continued front page    (81) Designated countries EP (AT, BE, CH, CY, DE, DK, ES, FI, FR, GB, GR, IE, I T, LU, MC, NL, PT, SE, TR), OA (BF , BJ, CF, CG, CI, CM, GA, GN, GW, ML, MR, NE, SN, TD, TG), AP (GH, G M, KE, LS, MW, MZ, SD, SL, SZ, TZ , UG, ZW), EA (AM, AZ, BY, KG, KZ, MD, RU, TJ, TM), AE, AG, AL, AM, AT, AU, AZ, BA, BB, BG, BR, BY, B Z, CA, CH, CN, CR, CU, CZ, DE, DK , DM, DZ, EE, ES, FI, GB, GD, GE, GH, GM, HR, HU, ID, IL, IN, IS, J P, KE, KG, KP, KR, KZ, LC, LK, LR , LS, LT, LU, LV, MA, MD, MG, MK, MN, MW, MX, MZ, NO, NZ, PL, PT, R O, RU, SD, SE, SG, SI, SK, SL, TJ , TM, TR, TT, TZ, UA, UG, UZ, VN, YU, ZA, ZW

Claims (34)

[Claims] 1. A method of deriving at least three audio output signals from two input audio signals, wherein four audio signals are derived from the two input audio signals, and the four audio signals are two audio signals. In a passive matrix generating two pairs of audio signals in response to
A second pair of derived audio signals represent respective directions located on an axis, the second pair of derived audio signals representing respective directions located on a second axis, the first and second axes being substantially perpendicular to each other, Processing each of said each pair of derived audio signals to generate a respective first and second pair of intermediate audio signals, the magnitude of the relative amplitude of said audio signal of each pair of intermediate audio signals being Generating a first output signal representative of a first direction located on the pair axis of the derived audio signal from which the first pair of intermediate signals is generated, the first output signal being generated to be equal. One output signal of the same polarity generated by at least combining at least one component of each of the second pair of intermediate audio signals, the derived audio from which the first pair of intermediate signals is generated. Belief A second output signal representative of a second direction located on the axis of said pair of said second output signals having opposite polarities and at least one component of each of said second pair of intermediate audio signals. Generating a third output signal representative of a first direction located on the axis of the derived audio signal from which the second pair of intermediate signals is generated, the third output signal being generated by the third output signal. An output signal having the same polarity, produced by at least combining at least one component of each said first pair of intermediate audio signals, said second pair of intermediate signals being derived derived audio signals Generating a fourth output signal representative of a second direction located on the pair of axes, the third signal being coupled to the third output signal with the same polarity or at least coupled with the same polarity. To If so produced, a method of deriving an audio output signal, comprising at least combining with at least one component of each of said first pair of intermediate audio signals of the same polarity. 2. Generating a first output signal comprises combining each component of said second pair of intermediate audio signals with a passive matrix representing said first direction, said component being said passive matrix audio. Constructing a cancellation signal that opposes the signal and producing a second output signal comprises combining each component of the second pair of intermediate audio signals with a passive matrix representing the second direction, said component Constructing a cancellation signal that opposes the passive matrix audio signal, producing a third output signal, combining each component of the first pair of intermediate audio signals with a passive matrix representing the third direction. And wherein the component constitutes a cancellation signal that opposes the passive matrix audio signal and selectively produces a fourth output signal. And combining each component of the first pair of intermediate audio signals with a passive matrix representing the fourth direction,
The method of claim 1, wherein the components constitute a cancellation signal that opposes the passive matrix audio signal. 3. The method of claim 2, wherein matrix audio signals respectively representing the first, second, third and optionally fourth directions are generated by the passive matrix. 4. A passive matrix audio signal, each representing said first, second, third and optionally fourth direction, is generated by a plurality of linear combiners, said linear combiners also being said passive matrix audio. The method of claim 2, wherein a signal is combined with one of the components of each signal. 5. The method of claim 1, wherein the respective output signals are generated by combining each pair of the intermediate signals. 6. The method of claim 1, 2 or 5, wherein the processing comprises feeding back each pair of intermediate audio signals used to control the relative amplitude of the respective pair of intermediate audio signals. Or one way. 7. The process includes applying each derived audio signal to a respective variable gain circuit, wherein the gain of each variable gain circuit associated with each pair of derived audio signals is at each pair. 7. The method of claim 6, wherein the method is controlled in response to the amplitude of the output of the variable gain circuit. 8. Each variable gain circuit includes a voltage controlled amplifier (VCA), the V
The CA has a gain g combined with a subtractive combiner, the resulting variable gain circuit gain being (1-g), wherein the cancellation signal is taken from the output of the voltage controlled amplifier. Method 7 9. Each variable gain circuit includes a voltage controlled amplifier (VCA), the V
8. The method of claim 7, wherein CA has a gain g, the resulting gain of the variable gain circuit is g, and the cancellation signal is taken from the output of the voltage controlled amplifier. 10. The method of claim 7, wherein the gain of each variable gain circuit is low for static input signal conditions and the signal output is substantially the signal generated by the passive matrix. 11. The method of claim 7, wherein the gain of each variable gain circuit is high for stationary input signal conditions and the signal output is substantially the signal generated by the passive matrix. 12. The gain of the variable gain circuit associated with each pair of derived audio signals is controlled by applying the output of the respective variable gain circuit of the pair to a magnitude comparator. 8. The method of claim 7, wherein the instrument generates a control signal that controls the gain of the variable gain circuit. 13. Each of the magnitude comparators controls the gain of the variable gain circuit associated with the pair of derived audio signals, and for some input signal conditions, the output of one variable gain circuit. 13. The method of claim 12, wherein the gain of the variable gain circuit with the increased output is decreased as the magnitude of V increases with respect to the other. 14. Each of the magnitude comparators controls the gain of the variable gain circuit associated with the pair of derived audio signals, and for some input signal conditions, the output of one variable gain circuit. 14. The method of claim 13, wherein the gain of the variable gain circuit that does not have the increased output is substantially unchanged as the magnitude of V increases with respect to the other. 15. Each of the magnitude comparators controls the gain of the variable gain circuit associated with the pair of derived audio signals, and for some input signal conditions, the output of one variable gain circuit. 14. The method of claim 13, wherein the product of the gains of the variable gain circuit is made substantially constant as the magnitude of V increases with respect to the other. 16. Each of the magnitude comparators controls the gain of the variable gain circuit associated with the pair of derived audio signals, and for some input signal conditions, the output of one variable gain circuit. 13. The method of claim 12, wherein the gain of the variable gain circuit with the increased output is increased as the magnitude of V increases with respect to the other. 17. Each of the magnitude comparators controls the gain of the variable gain circuit associated with the pair of derived audio signals, and for some input signal conditions, the output of one variable gain circuit. 17. The gain of the variable gain circuit that does not have the increased output is likewise substantially unchanged as the magnitude of V increases with respect to the other.
the method of. 18. Each of the magnitude comparators controls the gain of the variable gain circuit associated with the pair of derived audio signals, and for some input signal conditions, the output of one variable gain circuit. 17. The method of claim 16, wherein the product of the gains of the variable gain circuit is likewise made substantially constant as the magnitude of R increases with respect to the other. 19. The dB gain of the gainable circuits is a function of the voltage of those circuits, each magnitude comparator has an infinite gain, and the output of each variable gain circuit is a magnitude comparator through a rectifier. And the rectifier provides an output signal proportional to the logarithm of its input. 20. Each rectifier is preceded by a filter having a response, the response attenuating low and very high frequencies to provide a response that rises slowly over the middle of the audible range. the method of. 21. One or more additional control signals are derived from the two control signals that control the variable gain circuit associated with each pair of passive matrix audio signals, each one or both control signals being derived. By altering and generating an unaltered control signal and a lesser or greater of the altered control signals or a lesser or greater of the two altered control signals, 13. The method of claim 12, further comprising causing additional control signals to be derived. 22. The method of claim 21, wherein one or both of the control signals are modified by polarity reversing, amplitude cancellation, amplitude scaling and / or non-linear processing of the respective signals. 23. One or more additional variable gain circuits further receiving as inputs the combination of the plurality of cancellation signals or the combination of the two passive matrix signals, the one or more One or more additional control signals control each of the one or more additional variable gain circuits, the input signal being the first
And the gain of the circuit is increased to a maximum when representing a direction other than the direction located on the second axis, the one or more ones each being one of the one or more additional control signals. 22. The method of claim 21, further comprising: generating one or more additional cancellation signals by controlling the additional variable gain circuit of. 24. Each of at least five passive matrix audio signals,
Combining at least five output signals by combining two or more of said plurality of cancellation signals and said one or more additional cancellation signals, said cancellation signals opposing each passive matrix audio signal, 24. The passive matrix audio signal is substantially canceled by the cancellation signal when the input audio signal represents a signal associated with a direction other than the direction represented by the passive matrix audio signal. Method. 25. The magnitude of the audio signal of the first pair of intermediate audio signals is:     [(L_t+ R_t) * (1-g_c)] Or equivalently, [(L_t-R_t) *
h_c] Size and     [(L_t-R_t) * (1-g_s) Or equivalently, [(L_t-R_t) * H _s ]],     The magnitude of the other pair of intermediate audio signals is:     [(L_t* (1-g_l)] Or equivalently, [L_t* (H_l)] Size
as well as     [(R_t* (1-g_r)) Or equivalently, [R_t* (H_r)]
Represented by   Where L_tAnd R_tIs the passive matrix L_t+ R_tGenerated by
A pair of audio signals, L_t-R_tIs the passive matrix L_t+ R _t Is another pair of audio signals generated by_c) And h_cIs
, L of the passive matrix_t+ R_tIs the gain of the variable gain circuit associated with the output
, (1-g_s) And h_sIs the L of the passive matrix_t-R_tRelated to output
The gain of the variable gain circuit, which is (1-g_l) And h_lOf the passive matrix
L_tIs the gain of the variable gain circuit associated with the output, (1-g_r) And h_rIs the above
R of passive matrix_t13. The gain of the variable gain circuit associated with the output,
the method of. 26. A method of deriving at least three audio signals, each of which is associated with a direction, from two input audio signals, said passive matrix comprising two pairs of passive audio signals in response to said two input audio signals. Generating a plurality of passive matrix signals including a matrix audio signal, the first pair of passive matrix signals representing a direction on a first axis, and the second pair of passive matrix signals on a second axis. And processing each of said pairs of passive matrix audio signals to generate respective first and second pairs of intermediate audio signals, wherein said first and second axes are substantially perpendicular to one another. The equalization of the relative amplitude magnitudes of the audio signals of each pair of intermediate audio signals is forced, Deriving a plurality of cancellation signals from the pair and generating at least three output signals by combining each of the at least three passive matrix audio signals with two or more of the plurality of cancellation signals, the cancellation signals being As opposed to each passive matrix audio signal, the passive matrix audio signal is substantially canceled by the cancellation signal when the input audio signal represents a signal associated with a direction other than the direction represented by the passive matrix audio signal. A method for deriving an audio signal, the method comprising: 27. The method of claim 26, wherein the processing comprises feeding back each pair of intermediate audio signals used to control the relative amplitude of each pair of intermediate audio signals. 28. The processing includes applying each of the two pairs of passive matrix signals of a passive matrix audio signal to a respective variable gain circuit, each circuit having a gain g associated with a subtractive combiner. 28. The method of claim 27 including a voltage controlled amplifier (VCA), wherein the resulting variable gain circuit gain is (1-g) and the cancellation signal is taken from the voltage controlled amplifier. . 29. The gain of the variable gain circuit associated with each pair of passive matrix matrix audio signals is provided to a magnitude comparator that produces a control gain that controls the gain of the variable gain circuit. 29. The method of claim 28, controlled by applying the output of a variable gain circuit. 30. The output of each variable gain circuit of each pair is applied to a magnitude comparator via a rectifier, the rectifier providing a signal proportional to the logarithm of their inputs, the comparator 30. The method of claim 29, having infinite gain, wherein the VCA dB gain is a linear function of their control voltages. 31. Deriving one or more additional control signals from the two control signals that control the variable gain circuit associated with each pair of passive matrix audio signals, the one or more additional control signals. By modifying one or both control signals and generating an unmodified control signal and a modified control signal or two lesser or greater of the two unmodified control signals, 30. The method of claim 29, further comprising causing each of the one or more additional control signals to be obtained. 32. The method of claim 31, wherein one or both of the control signals is modified by polarity reversing, amplitude cancellation, amplitude scaling and / or non-linear processing of the respective signals. 33. One or more additional variable gain circuits further receiving as inputs the combination of the plurality of cancellation signals or the combination of two passive matrix signals, the one or more One or more additional control signals control each of the one or more additional variable gain circuits, the input signal being the first
And the gain of the circuit is increased to a maximum when representing a direction other than the direction located on the second axis, the one or more ones each being one of the one or more additional control signals. 32. The method of claim 31, further comprising generating the one or more additional cancellation signals by controlling the additional variable gain circuit of. 34. Each of at least five passive matrix audio signals,
Combining at least five output signals by combining two or more of said plurality of cancellation signals and said one or more additional cancellation signals, said cancellation signals opposing each passive matrix audio signal, 34. The passive matrix audio signal is substantially canceled by the cancellation signal when the input audio signal represents a signal associated with a direction other than the direction represented by the passive matrix audio signal. Method.