JP2003333900A - Controller for synchronous motor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To drive a wide band with stable efficiency in a synchronous motor, and to reduce a torque ripple at a low speed and at a high-torque occurring time. <P>SOLUTION: A controller for the synchronous motor refers to a q-axis current command value and a rotor speed detected value to control the motor having a main winding wound around a stator made of a soft magnetic material and at least one or more sub-windings, winding-superposing values of the q-axis current and a d-axis current of the main winding and sub-winding are calculated by a winding superposing value generator, and hence the motor is driven by the main winding + sub-winding or only the main winding in response to the winding superposing value. The main winding is handled as for a high-speed rotation and the sub-winding as for auxiliary winding for a high-speed winding, and the motor is driven by a power amplifier having a small capacity at a high-speed drive time. When there is application which needs torque at a low- sped drive time, the motor is driven by the power amplifier having a large capacity of the sub-winding. Thus, the simultaneous drive of the power amplifier for the main winding + sub-winding power amplifier can be performed to obtain large torque. <P>COPYRIGHT: (C)2004,JPO

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a synchronous motor control device, and more particularly to a synchronous motor control device having a plurality of stator windings.

2. Description of the Related Art Conventionally, synchronous motors have been used for various purposes such as machine tools. The rotation of the synchronous motor is controlled by the control device, and a block diagram of the control device for such a synchronous motor is shown in FIG.

The synchronous motor 10 includes a speed controller 1 and an IV
Command calculator 6, power converter 8, rotor position detection means 1
It is controlled by a control device including a position-speed calculator 5, a phase calculator 4, and the like. In FIG. 7, SVC is a speed command value, SIQC0 is a q-axis (torque) current command value, SI
DC0 is the d-axis (field) current command value, SVU, SVV, S
VW is the voltage command value of U phase, V phase, W phase, SPH, respectively
S is a phase value, SPD is a rotor position detection value, and SVD is a rotor speed detection value.

The speed controller 1 amplifies the deviation between the speed command value SVC and the rotor speed detection value SVD from the host controller by an amplifier and sets it as a torque command value, which is represented by a PI controller. It is a vessel. In the speed controller 1, the torque command value or the q-axis current command value SIDC0 and the d-axis current command value SIDC calculated with reference to the rotor speed SVD.
0 is output. The IV command calculator 6 is a current-voltage (IV) calculator, and outputs the q-axis current command value SIDC0 and the d-axis current command value SIDC0 output from the speed controller 1.
By referring to the phase value SPHS output from the phase calculator 4 as the U-axis, V-phase, and W-phase d-axis and q-axis current command values each having a phase difference of 2π / 3, and then vectoring each phase A combined current value of the U-phase, V-phase, and W-phase is obtained by the combined operation.
Although not particularly shown, the detected current value of each phase obtained by the current detection means provided for detecting the current applied to the electric motor 10 is fed back to the current-voltage (IV) calculator 6. The voltage command values SVU, SVV, SVW for each phase are calculated by calculating the deviation from the current command value by a calculator such as a PI calculator. The power converter 8 outputs a voltage based on the voltage command values SVU, SVV, SVW of each phase output from the current-voltage (IV) calculator 6 to the electric motor 1
By applying 0, a current flows through each phase winding of the electric motor 10. Normally, the rotor position detecting means 11 is directly connected to the shaft of the electric motor 10 and the rotor position detection value SPD is set to the position-.
Output to the speed calculator 5. The position-speed calculator 5 sets the rotor position detection value SPD to the rotor speed detection value SVD by performing a calculation process such as differentiation. The phase calculator 4 is based on the rotor position detection value SPD obtained from the rotor position detecting means 11, and the phase value SPHS corresponding to each q-axis current command value and each d-axis current command value of the U phase, V phase, and W phase. As current-voltage (I-
V) Output to the calculator 6.

A typical controlled synchronous motor is a permanent magnet (PM) type in which a permanent magnet is attached to the rotor surface, and a permanent magnet insertion (IPM) in which the permanent magnet is covered with a soft magnetic material.
Examples include a mold and a reluctance (RM) type in which the rotor is composed only of a soft magnetic material. By controlling the d-axis (field) current, which is the field component, any of the synchronous motors can rotate up to a region above the rated speed, but in particular, it is a reluctance type synchronous motor using a soft magnetic material as a rotor. In the case of electric motors and permanent magnet insertion type synchronous motors, a voltage drop occurs due to winding inductance as the number of revolutions (frequency) increases, and torque decreases due to the limitation of the power supply voltage. (A decrease in the dq axis inductance ratio) causes a torque decrease, making it difficult to obtain a stable output in a high speed region.

Therefore, in order to obtain stable controllability in the high speed region, the number of windings of the motor is reduced to improve the torque characteristic in the high speed region by sacrificing the torque obtained in the low speed region as a method of lowering the inductance, or There is used a method of improving characteristics in a high speed range by dividing the high speed range and the low speed range by a known technology such as Y-Δ or Y-Y having a different number of windings.

Further, a reluctance type permanent magnet interpolating type synchronous motor utilizing reluctance force is caused by the rotor structure and the shape of the teeth portion of the rotor in the direction of rotation viewed from the stator winding on the surface of the rotor. Inductance change occurs, and this causes a change in energy stored in the inductance, which causes torque ripple.

When the torque ripple is used for the feed shaft of a machine tool, it causes problems because it appears as stripes on a work piece due to ripples, and as noise and vibration when driving a motor.

[0008]

In the case of a reluctance type permanent magnet interpolating type synchronous motor utilizing reluctance force, in order to realize a wide band which stably rotates up to high speed rotation, the winding inductance is usually Since it is larger than that of a magnet type synchronous motor, the power factor tends to decrease and the drive voltage tends to increase. The decrease in power factor causes a phenomenon in which the reactive power component increases and the efficiency of the motor decreases, which also causes the unstable control.

Further, a reluctance type permanent magnet insertion type synchronous motor utilizing reluctance force is caused by the rotor structure and the shape of the stator tooth portion in the rotation direction when viewed from the stator winding on the surface of the rotor. Inductance changes occur, which causes changes in the energy stored in the inductance, which causes torque ripple.

An object of the present invention is to realize an efficient wide-band drive in a synchronous motor, particularly a reluctance type or permanent magnet interpolating type synchronous motor utilizing reluctance force, with a low torque ripple at low speed rotation or high torque generation. An object of the present invention is to provide a control device for a synchronous motor, which can obtain stable controllability and improve efficiency.

[0011]

The above object can be achieved by the following means.

According to the present invention, a main winding wound around a stator made of a soft magnetic material, at least one sub winding wound around the stator, and a current flowing through the winding. Is a controller for controlling a synchronous motor including a mover that operates in accordance with the above, and a winding weight value generator that calculates a winding weight value based on command information for driving the motor. And a dq current distributor that calculates a current command value for each winding based on the winding weight value.

Thus, depending on the winding weight value, the main winding +
It is possible to drive the motor only with the sub winding or the main winding. In this case, the main winding is used for high-speed rotation with a reduced number of windings, and the auxiliary winding is used as an auxiliary winding for high-speed winding. Since it does not exist, the capacity of the power amplifier connected to the main winding can be small.
Further, when there is a use requiring torque during low speed driving, by connecting a large capacity power amplifier to the auxiliary winding, it is possible to drive simultaneously with the main winding power amplifier, and a larger torque can be obtained. During high-speed driving, a dedicated winding (main winding) is used, so there is little impedance voltage drop due to inductance, and reactive power drops, so stable control can be obtained.

Further, the present invention further comprises a phase calculator for calculating a phase shift amount θshift corresponding to the winding weight value, and it is preferable that a current whose phase is shifted by the phase shift amount is supplied to each winding. Is. Phase calculation is performed on the currents applied to the main winding and the sub winding according to the weight values of the q-axis current and the d-axis current of the main winding and the sub winding calculated by the winding weight generator. Is calculated so that there is a phase difference between the current command values of the respective windings.

As a result, the same effect as when the windings are wound in a short pitch or the magnetic pole centers of the rotor are arranged at unequal pitches can be obtained, and the torque ripple can be reduced.

As described above, according to the present invention, efficient and stable control at the time of high speed driving can be obtained, and control for reducing the torque ripple of the electric motor becomes possible.

[0017]

BEST MODE FOR CARRYING OUT THE INVENTION A control device for a synchronous motor according to an embodiment of the present invention will be described below with reference to the drawings. Unless otherwise specified, the elements and signals having the same symbols and numbers as those in the conventional technique shown in FIG. 7 have the same functions and performances.

The control device for a synchronous motor of the present invention controls the synchronous motor shown as an example in FIG. This electric motor
It is a flux barrier type reluctance synchronous motor that uses only reluctance force without using permanent magnets. The stator 21 has a shape in which a plurality of stator pole teeth 28 extend in the center direction from a ring-shaped yoke portion 27, and a slot 23 between the stator pole teeth 28 has a main winding 22 A and a sub winding 22.
B is paid. A rotor 29 is arranged inside the stator pole teeth 28 of the stator 21. The rotor 29 is fixed to the shaft 25 and is composed of a rotor constituting body 24 made of a soft magnetic material and a slit-shaped nonmagnetic material portion 26 (air, resin, etc.) provided therein. The example shown is
This is the case where there is one main winding and one sub winding.

FIG. 2 is a control block diagram of the synchronous motor controller according to the present embodiment. This control device includes a speed controller 1, a dq current distributor 2, a winding weight value generator 3, a phase calculator 4d, a position-speed calculator 5, an IV command calculator 6, 7, and a power converter. It is configured to include 8, 9 and the like. The processing performed by this control device will be described.

The speed controller 1 inputs a speed command value SVC from a host controller (not shown) and a rotor speed detection value SVD from a position-speed calculator 5, and based on a difference between them, q
The axis current command value SIQC0 and the d-axis current command value SIDC0 are calculated.

The winding weight value generator 3 inputs the rotor speed detection value SVD and the q-axis (torque) current command value SIQC0, and based on these inputs, the main winding 22A and the sub winding 22B.
Calculate the winding weight value SMODE of
The ODE is output to the dq current distributor 2 and the phase calculator 4d.

The calculation of the winding weight value SMODE performed by the winding weight value generator 3 will now be described. The winding weight value generator 3 stores two internal functions having the characteristics shown in FIGS. 3 (a) and 3 (b). The internal function shown in FIG. 3A is an internal function that receives the q-axis current command value SIQC0 as an input and outputs a coefficient SK1 corresponding to its absolute value | SIQC0 |. The output coefficient SK1 has an absolute value | SIQC0 | of 0 or more and a threshold value SIQC
In the range below OT, the constant value is 0.5, and the absolute value | S
In the range where IQC0 | exceeds the threshold SIQCOT,
It is a value that linearly increases corresponding to the absolute value | SIQC0 |. The internal function shown in FIG. 3B is an internal function that inputs the rotor speed detection value SVD and outputs the coefficient SK2 corresponding to its absolute value | SVD |. The coefficient SK2 output has an absolute value | SVD | of 0 or more and a threshold value SVD.
In the range of T or less, the constant value is 1.0, and the absolute value | SV
In a range in which DT | exceeds the threshold value SIQCOT, the absolute value | SVDT | is a value that decreases as it increases.

The rotor speed detection value S is added to these internal functions.
Based on the coefficients SK1 and SK2 obtained by inputting VD and the q-axis (torque) current command value SIQC0, the winding weight value SM
ODE is calculated. The winding weight value SMODE is a value obtained by multiplying the coefficients SK1 and SK2 as shown in the equation (1).

[0024]

[Equation 1]     SMODE = SK1 * SK2 ... (1)

The characteristic of the winding weight value SMODE is determined by the above-mentioned two internal functions. In this embodiment, when torque is required, the q-axis (torque) current command value SIQ
The characteristics are such that it increases when C0 is large, and decreases when it rotates at a high rotational speed, that is, when the rotor speed SVD is large and a large torque is not required.
That is, the winding weight value SMODE is an index indicating the torque required by the electric motor. In the present embodiment, the winding weight value SMODE described above is set, but the function of the winding weight value SMODE is not limited to that shown in FIG. 3 and can be set arbitrarily. That is, it may be set in consideration of how to use the electric motor, that is, what kind of current is passed through the main winding and the sub winding. Coefficients SK1, S
By adjusting the threshold values SKIQCOT and SVDT for each of K2, the utilization ratio of the main winding and the auxiliary winding is adjusted,
The capacity of the power amplifiers 8 and 9 and the characteristics of the electric motor 10 can be set to desired characteristics. Also, if these stored functions can be rewritten by the operator from the outside,
When manufacturing the electric motor, the function can be adjusted according to the characteristics of each electric motor.

Next, the dq current distributor 2 uses the q-axis current command value SIQC0, the d-axis current command value SIDC0, and the winding weight value SMODE to determine the main winding q-axis current command S.
IQC1, d-axis current command SIDC1, and auxiliary winding q-axis current command SIQC2, d-axis current command SIDC2 are calculated.

The arithmetic processing performed by the dq current distributor 2 will be described in detail. The dq current distributor 2 has a configuration shown in FIG.
Two functions having the characteristics shown in (a) and (b) are stored. For these functions, the winding weight value SMODE is input, and the main winding utilization rate SKIC1 and the sub winding utilization rate S are respectively entered.
Outputs KIC2. The function that outputs the main winding utilization rate SKIC1 is always a constant value 1 for the winding weight value SMODE.
SKIC1 is output. In addition, the auxiliary winding utilization rate SKIC
The function that outputs 2 outputs 0 when the winding weight value SMODE is less than or equal to the predetermined threshold SMODET, but outputs the threshold SMOD.
Above ET, a value that linearly increases corresponding to the winding weight value SMODE is output.

The winding weight value SMODE is input to these functions, and SKIC1 which is the utilization factor of the main winding and SKIC2 which is the utilization factor of the auxiliary winding are calculated. Then, these utilization rates SKIC1, SKIC2 and the q-axis current command value SIQ
Based on C0 and the d-axis current command value SIDC0, the current command values SIQC1, SIDC1, SIQC2, and SIDC2 of the d-axis current and the q-axis current applied to the main winding and the auxiliary winding are
It is calculated by the following equations 2 and 3.

[0029]

[Equation 2]   [SIQC1, SIDC1] = SKIC1 [SIQC0, SIDC0]                                                   (2)

[0030]

[Equation 3]   [SIQC2, SIDC2] = SKIC2 [SIQC0, SIDC0]                                                   ... (3)

Since SKIC1 is always 1, the current command values SIQC1 and SIDC1 for the main winding are the same as the original current command values SIQC0 and SIDC0 and are always used. In addition, the utilization factor for the sub winding is the winding weight value SMO
When DE is less than or equal to the threshold value SMODET, the current command values SIQC2 and SIDC2 of the sub winding are 0, the sub winding is not used, and the winding weight value SMODE is the threshold value SMO.
When DET is exceeded, the current command value SIQC2, S of a predetermined value
Outputs IDC2 and uses the sub winding. These calculated current command values SIQC1, SIDC1, SIQC2, S
The IDC 2 is input to the current-voltage (IV) command calculators 6 and 7.

The phase calculator 4d receives the output SPD of the position detecting means and the winding weight value SMODE, and calculates a phase shift amount θshift from the winding weight value SMODE.
Based on the rotor position SPD, which is the output of the position detecting means, and the phase shift amount θshift, phase calculation is performed according to equations (4) and (5). The rotor position SPD is multiplied by a numerical value of 1/2 of the number of motor poles, and an arbitrary magnetic pole position adjustment value: α is added. This magnetic pole position adjustment value: α is used to adjust the relative position of the magnetic pole and the phase of the sinusoidal current or voltage applied to the electric motor. Phase value SPHS1 and phase value SP
HS2 has the relationship shown in Expression (5a) or Expression (5b), and the phase value SPHS1 and the phase value SPHS2 always have a phase difference according to the phase shift amount θshift. (However, the phase shift amount θshift may be 0 as described later, and there is no phase difference at that time) The phase value SPHS1 and the phase value SPHS2 are respectively the IV command calculators 6 and 7.
Is output to.

[0033]

## EQU00004 ## SPHS1 = MOD [(P / 2) * SPD + .alpha., 2.pi.] (4) where P: motor pole number, .alpha .: magnetic pole position adjustment value, and the function MOD [A, B] is A This is a function that finds the remainder when B is divided by B.

[0034]

[Equation 5]   SPHS2 = SPHS1 + θshift (5a)     Or   SPHS2 = SPHS1-θ shift (5b)

Next, the calculation processing of the phase shift amount θshift used in the calculation of the above equation (4) will be described. The function shown in FIG. 5A is stored in the phase calculator 4d. This function is a function that inputs the winding weight value SMODE and outputs the phase shift amount θshift. If the winding weight value SMODE is less than or equal to a predetermined threshold value SMODE1, θsh
ift is 0, and threshold SMODE1 to threshold S
It increases in the range of MODE2 and becomes a constant value when the threshold value SMODE2 or more. The phase shift amount θshift is 0 when the winding weight value SMODE is small, that is, when the required torque is small or when the rotor speed SVD is large. Conversely, when the winding weight value SMODE is large, that is, when the required torque is large or when the rotor speed SVD is small, the value of the phase shift amount θshift is a large value. The phase shift amount θshift is calculated, and the phase value SPHS1 and the phase value SPHS2 calculated according to Expression (4), Expression (5a), and Expression (5b) are output to the IV command calculators 6 and 7. .

The main-winding current-voltage (IV) command calculator 6 receives the current command values SIQC1, S from the dq current distributor 2.
IDC1 is input, and also the phase value SPHS1 obtained from the phase calculator 4d is input, based on these, phase distribution is performed, and then vector combination calculation is performed, and processing such as current-voltage (IV) command conversion is performed. The three-phase voltage command SV
U1, SVV1, and SVW1 are calculated. Similarly, the auxiliary winding current-voltage (IV) command calculator 7 inputs the current command values SIQC2 and SIDC2 from the dq current distributor 2 and also outputs the phase value SPHS2 from the phase calculator 4d. The three-phase voltage commands SVU2, SVV2, and SVW2 are input and calculated.

Three-phase voltage command SVU for main winding at this time
1, SVV1, SVW1 and three-phase voltage command SVU for auxiliary winding
The relationship between 2, SVV2 and SVW2 is shown in FIG. For convenience of explanation, the figure shows a main winding voltage command value waveform 52 and an auxiliary winding voltage command value waveform 53 of the same phase (for example, U phase). As can be seen from FIG. 5B, the main winding voltage command and the auxiliary winding voltage command are out of phase by the phase shift amount θshift. In this embodiment, the effect of reducing the torque ripple is obtained by providing this phase shift.

The power converter 8 for the main winding performs voltage amplification control based on the three-phase voltage commands SVU1, SVV1 and SVW1 and applies the amplified voltage to the main winding 22A of the electric motor 10 for main winding. An electric current is applied to the line 22A. Similarly, the power converter 9 for the auxiliary winding also has the three-phase voltage commands SVU2, SV.
Voltage amplification control is performed based on V2 and SVW2, the amplified voltage is applied to the sub winding 22B of the electric motor 10, and a current is passed through the sub winding 22B. A magnetic flux is generated in the main winding 22A and the sub winding 22B by the current supplied from the power converters 8 and 9. FIG. 6 is an operation vector diagram of magnetic flux generated by the main winding 22A and the sub winding 22B. D- of the main winding 22A
The q axis (d1, q1 axis) and the dq axis (d
(2, q2 axes) are respectively current-commanded, and their magnetic fluxes are combined with each other to generate a magnetic flux φ0 acting on the rotor. The main winding and the sub winding often have different numbers of turns, and therefore, the d- and q-axis inductances of the main winding and the sub winding are different from each other. Therefore, when setting the respective functions of the winding weight value generator 3, the dq current distributor 2, and the phase calculator 4d, the current command considers the difference in the inductance, and the magnetic flux acting on the rotor is desired. It is necessary to adjust and set the function so that the phase becomes.

In the control device for the synchronous motor of the present embodiment described above, the q-axis current command value SIQC0 and the speed detection value S.
Based on the command information for driving the electric motor such as VD,
A winding weight value generator 3 for calculating a winding weight value, and a dq current distributor 2 for calculating a current command value for each winding based on the winding weight value and a current command value. . Therefore, there is an effect that the current flowing through the two windings can be appropriately set according to the command value given to the electric motor and the condition of the electric motor. For example, when the winding weight value is large as in the present embodiment, if the current is set to flow not only in the main winding but also in the sub winding, a reluctance type permanent magnet insertion using reluctance force is performed. Even in the case of the synchronous motor of the type, it is possible to realize a wide band capable of stably rotating up to a high-speed rotation and control it efficiently. This is the rotor speed SVD
Is low and requires torque (= q-axis (torque) current command value S
When IQC0) is large, it means that driving can be performed with the sum of power amplifier capacities connected to the main winding and the auxiliary winding, and high torque can be obtained. On the contrary, when the rotor speed SVD is low and the torque is not required so much, it is driven only by the main winding and is driven by the main winding (= high speed winding) and the power amplifier for the main winding wire, which is wasteful. Power is not generated.

The change in the direction of the rotor of the magnetic energy stored in the inductance between the stator teeth made of soft magnetic material and the rotor appears as a torque ripple, and the generated torque is particularly large (stator- When the magnetic energy between the rotors is large), the magnetic energy stored in the inductance also increases, and the change in the magnetic energy in the rotation direction also increases, so that the torque ripple also increases.

In order to solve this problem, the embodiment is provided with a phase calculator for calculating the phase shift amount θshift corresponding to the winding weight value, and a current whose phase is shifted by this phase shift amount is applied to each winding. It's flowing. By applying a phase shift amount θshift to the current in the main winding and the auxiliary winding, the stator −
There is an effect that it can be reduced by shifting and canceling the phase of the inductance change ripple in the rotation direction between the rotors.

Further, the known technique of manufacturing the stator winding by the short pitch winding and the technique of shifting the center pitch angle of the magnetic poles of the rotor have the same effect, but these techniques are also available. In the technology, once created, the phase shift amount θsh
ift cannot be changed. On the other hand, in the present embodiment, the amount of phase shift θshift can be set arbitrarily, so that setting it for each electric motor makes it easier to cope with torque ripples that differ due to individual differences in manufacturing of the electric motor.
Also, when setting this phase shift amount θshift,
Only when the winding weight value SMODE is low, that is, when the rotor speed SVD is high, the frequency component of the torque ripple deviates from the operating band of the control device or the electric motor, so that the phase of the current is shifted and the magnetic flux efficiency is increased. Therefore, the phase shift amount θshift may be set to 0.

In the above description, the reluctance type synchronous motor is described as an example, but the present invention is not limited to this, and can be applied to other types of rotating machines. It can also be applied to a linear synchronous motor. If the sub winding and the power amplifier for the sub winding are set to perform the regenerative operation, the magnetic flux generated by the main winding in the sub winding can be used for power generation, and the generated power can be supplied to the power source. It can also be regenerated to efficiently use energy.

[0044]

As described above, according to the present invention, the current flowing through the two windings can be set appropriately. As a result, even in the case of a reluctance type or permanent magnet insertion type synchronous motor that utilizes reluctance force, it is possible to realize a high frequency band that stably rotates up to high speed and to control it efficiently.

Further, it becomes possible to obtain a control device for a synchronous motor which can obtain stable controllability with a low torque ripple when a low speed rotation or a high torque is generated.

[Brief description of drawings]

FIG. 1 is an example of a synchronous motor to which a control device for a synchronous motor of the present invention is applied.

FIG. 2 is an explanatory diagram showing an embodiment of a control device for a synchronous motor of the present invention.

FIG. 3 is an explanatory diagram showing a function example of a winding weighting value generator of the synchronous motor control device of the present invention.

FIG. 4 is an explanatory diagram showing a function example of a dq current distributor of the synchronous motor control device of the present invention.

FIG. 5 is an explanatory diagram showing current phase shifts of a main winding and a sub winding of the control device for a synchronous motor of the present invention.

FIG. 6 is an explanatory diagram showing a vector diagram of a control device for a synchronous motor of the present invention.

FIG. 7 is an explanatory diagram showing a form of a conventional controller for a synchronous motor.

[Explanation of symbols]

1 speed controller, 2 dq current distributor, 3 winding weight value generator, 4 phase calculator, 4d phase calculator (phase shift function), 5 position-speed calculator, 6 current-voltage (I
-V) calculator 1, 7 current-voltage (IV) calculator 2,
8 power converter 1, 9 power converter 2, 10 synchronous motor, 11 rotor position detecting means, SMODE winding weighting value.

Claims (2)

[Claims] 1. A main winding wound around a stator made of a soft magnetic material, at least one auxiliary winding also wound around the stator, and an electric current flowing through the winding. A controller for controlling a synchronous motor including a mover, and a winding weight value generator that calculates a winding weight value based on command information for driving the motor, A dq current distributor that calculates a current command value for each of the windings based on the winding weight value, and a controller for the synchronous motor. 2. The control device for a synchronous motor according to claim 1, further comprising a phase shift amount θshif corresponding to the winding weight value.
A synchronous motor control device comprising a phase calculator for calculating t, which supplies a voltage command to each winding in which the phase is shifted by a phase shift amount and causes a current to flow.