JP2003101603A - Low if system of receiver - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a low IF system of receiver, which can reduce interference components passing each sub band filter by constituting the low IF system of receiver, using a band division system of A/D converter. SOLUTION: This low IF system of receiver stores in advance the coefficient errors between band-dividing filters 16-0, 16-1,..., 16-M-1 which are analog parts, in a digital memory 22a and digitally demodulates them by a digital demodulator 22.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a low IF type receiver, and more particularly to a low IF type which uses a complex coefficient filter bank to suppress DC offset and image frequency signals.
Regarding the method receiver.

[0002]

2. Description of the Related Art In recent years, there has been an increasing demand for broadband wireless communication systems with the spread of networks. Further, in order to realize a multi-mode / multi-band terminal, a direct conversion receiver is being studied.

FIG. 10 is a diagram showing the structure of a conventional direct conversion receiver. The direct conversion method eliminates the intermediate frequency (IF) stage, etc., and the signal received via the antenna 51 is band-amplified by the amplifier filter 52, mixed by the mixer 53 with the local oscillation signal from the local oscillator 54, and directly based. Converted to band signal. The converted signal passes through the amplifier filter 55 and is
It is input to the / D converter 56. The output of the A / D converter 56 is digital signal processed and demodulated. This method is I
Since the F filter is not used, it is easy to widen the band of the receiver, and the flexibility of the receiver is increased, and at the same time, one chip can be realized. However, in the direct conversion method, since the received frequency and the local frequency are equal, D
The problem of C offset has been pointed out. The DC offset is a problem that a part of the local signal leaks from the input side of the mixer, returns to the mixer again, and self-mixes with the local signal to cause an offset in the DC component. Alternatively, a part of the local signal is emitted to the outside through the antenna and the reflected wave is received and mixed with the local signal.

Low I to mitigate the problem of DC offset
The F method has been proposed (J. Crolsand MSJ Steya
ert, "Low-IF Topologies for High-Performance Analo
g Front Ends of Fully Intefrated Receivers, "IEEE
Trans. On Circuits and Systems, vol.45, no.3, pp.2
69-282, March 1998.).

FIG. 11 is a diagram showing the configuration of a conventional low IF type receiver. A signal received via the antenna 61 passes through a bandpass filter (BPF) 62 and an LNA 63, is mixed by a mixer 64 with a local oscillation signal from a local oscillator 65, and is converted into a low IF signal. The converted signal passes through the IF filter 66 and the amplifier 67 and is input to the A / D converter 68. The output of the A / D converter 68 is digital signal processed (DSP) by the demodulator 69 and demodulated. In the low IF method, since the desired signal does not exist near DC, the DC offset does not interfere with the desired signal. But D
C offset and image frequency signal A / D
In order to convert and remove, it is necessary to simultaneously convert the DC offset and the image frequency signal and the desired signal into a digital signal, so that an A / D converter with a high conversion speed is required.

The improvement tendency of the A / D converter is about 1.5 in 8 years.
Considering the tendency that the resolution is one bit and the resolution is reduced by one bit every time the conversion speed is doubled, the improvement rate of the conversion speed equivalent to the improvement rate of the signal processing speed is not seen.

As a method of improving the conversion speed of an A / D converter, a method of dividing a signal band using a filter bank has been proposed (R. Khoini-Poorfard, LB Lim, a.
nd DA Johns, "Time-Interleaved Oversampling A / D
Converters: Theory and Practice, "IEEE Trans. On
Circuits and Systems-II: Analog and Digital Signal
Processing, vol.44, no.8, pp.634-645, August 199
7.). By dividing the band and performing A / D conversion in parallel, the conversion speed required for each A / D converter can be relaxed. A optimized for each sub-band signal at the same time
By applying GC, it is possible to relax the resolution requirement of the A / D converter.

[0008]

However, in this A / D conversion system, each complex coefficient sub-band filter is composed of a switched capacitor filter (SCF) or the like, and an error occurs in the coefficient of the filter and the stop band is generated. Characteristics deteriorate. Then, the DC offset and the image frequency signal may interfere with the desired signal.

In view of the above problems, the present invention configures a low IF type receiver by using a band division type A / D converter, and at that time, an interference component passing through each subband filter is eliminated. An object of the present invention is to provide a low IF type receiver that reduces the number of receivers.

[0010]

SUMMARY OF THE INVENTION A low IF type receiver of the present invention is a band division analog filter for dividing a low IF signal into signals of a plurality of bands, and an analog / digital conversion of the output of the band division analog filter. A / D converter,
A known signal supply means for supplying a low IF known signal to the low IF stage; a memory for storing a signal related to the A / D converter output when the known signal is supplied by the known signal supply means; A digital demodulator for reading the contents of the memory and demodulating the output of the A / D converter.

Further, the known signal supplying means uses a local oscillator for converting the received signal frequency into a low IF, so that the local oscillator can be used both for receiving frequency conversion and for storing filter characteristics. it can.

Further, the band division analog filter is
The filter bank configuration enables high-speed processing.

Further, the band division analog filter is
With the wavelet configuration, it is possible to flexibly deal with changes in filter division characteristics.

Further, the band division analog filter is
With the complex coefficient filter configuration, the DC component and the image frequency can be suppressed and the band of the divided filter can be effectively used.

The number of band divisions of the band division analog filter is N (N is an integer of 2 or more), and hatd = hatRd, where hatd is an output signal amplitude of each band (N, 1). ) Matrix, d: (N, 1) matrix having the input signal amplitude of each band as a component, and (N, N) matrix hatR ^-1 , which is an inverse matrix to hatR, is stored in the memory, and
By providing the decorrelator that multiplies the output of the A / D converter by hatR ⁻¹ , it is possible to remove the interference caused between the frequency signals orthogonal to each other due to the error of the analog filter.

[0016]

BEST MODE FOR CARRYING OUT THE INVENTION Preferred embodiments of the present invention will be described in detail below with reference to the accompanying drawings.

FIG. 1 is a diagram showing the configuration of a low IF type receiver according to the first embodiment of the present invention. The low IF type receiver according to the first embodiment includes an antenna 11, a BPF 12,
LNA (low noise amplifier) 13, mixer 14, local oscillator 15, band division filters 16-0, 16-1, ..., 16
-M-1, amplifiers 17-0, 17-1, ..., 17-M-
1, decimator 18-0, 18-1, ..., 18-M-
1, A / D converters 19-0, 19-1, ..., 19-M-
1, up sampler 20-0, 20-1, ..., 20-M
-1, band synthesis filter 21-0, 21-1, ..., 21
-M-1, and a demodulator 22 having a memory 22a.

The signal received via the antenna 11 is B
It is converted into a low IF signal by the mixer 14 through the PF 12. The converted signal is input to the complex coefficient band division filters 16-0, 16-1, ..., 16-M-1 from H ₀ to H _M-1 . Band division filters 16-0, 16-1,
, 16-M-1 separates the desired signal from the DC offset and image frequency signals. The input signal is 1 / N by the decimator 18-0, 18-1, ..., 18-M-1.
To be downsampled. As a result, the A / D converter 19
-0, 19-1, ..., 19-M-1 conversion speed is 1 / N
To Further, the resolution of the A / D converters 19-0, 19-1, ..., 19-M-1 can be changed according to the signal received in each subband. A / D converter 19-
0, 19-1, ..., 19-M-1 sampled signals are upsamplers 20-0, 20-1,
..., converted to N samples by 20-M-1,
Complex coefficient band synthesis filters 21-0, 21-1, ..., 2
The signals of the respective bands are combined by 1-M-1, and the demodulator 22
Demodulated by.

The filter bank includes a DFT (Digital Four
ier Transform) filter bank, CQF (Conjugate Q)
uadrature Filter) Many proposals such as filter banks have been made. Especially when M = 2, QMF (Quadrature Mir)
bank is known. Assuming that z conversion of the input signal x (n) and the output signal hatx (n) is X (z) and hatX (z),

[0020]

[Equation 1] Express.

[0021]

[Equation 2] To get The QMF filter bank has H ₁ (z) = H ₀ (−z), (2.2) F ₀ (z) = H ₀ (z), (2.3) F ₁ (z) = −H ₁ (z) (2.4) At this time, the aliasing avoidance condition H ₀ (z) F ₀ (z) + H ₁ (z) F ₁ (z) = 0 (2.5) is satisfied, and the all-pass condition H ₀ (−z) F ₀ (z) + H ₁ ( −z) F ₁ (z) = 2z ^−L (2.6) is approximately satisfied. Where L is the order of the filter.

FIG. 2 is a diagram showing the frequency response of a real coefficient QMF filter bank. In order to suppress the DC component of the received signal and the image frequency signal, the real number coefficient is converted into a complex coefficient by the following conversion formula. _{h 0 (n) = exp (} jnφπ) h r (n), (2.7) h 1 (n) = exp (jn (φ ± 1) π) h r (n). (2.8) where phi is the filter It is the frequency shift amount of the impulse response.

FIG. 3 shows the QM when the π / 4 frequency is shifted.
It is a figure which shows the frequency response of an F filter bank. By thus frequency-shifting, the DC component and the image frequency signal can be suppressed and the desired frequency band can be evenly divided.

Usually, such a band division filter is composed of a switched capacitor. The characteristics of the switched capacitor filter include an error due to parasitic capacitance and the like.

FIG. 4 is a diagram showing the configuration of the low IF type receiver according to the first embodiment of the present invention in the case of two divisions. That is, here, M = 2 and N = 2. Antenna 31,
BPF32, LNA33, mixers 34-0, 34-1,
Local oscillator 35, band division filters 37-0, 37-
1, amplifiers 38-0, 38-1, decimator 39-0,
39-1, A / D converters 40-0 and 40-1, upsamplers 41-0 and 41-1 and band synthesis filter 42-.
0, 42-1 and the demodulator 47 are the antenna 11, the BPF 12, the LNA 13 and the mixer 1 shown in FIG. 1, respectively.
4, local oscillator 15, band division filters 16-0, 16
-1, ..., 16-M-1, amplifiers 17-0, 17-1,
..., 17-M-1, decimator 18-0, 18-1,
..., 18-M-1, A / D converters 19-0, 19-1,
..., 19-M-1, upsampler 20-0, 20-
1, ..., 20-M-1, band synthesis filters 21-0, 2
It corresponds to 1-1, ..., 21-M-1, and the demodulator 22. Here, the error is estimated from the desired signal band H ₁ by the error estimation 44, the filter coefficient is estimated from the average 43 of the DC offset pass band H ₀ , and the desired signal is output from the adder 46.

First, the error of the complex coefficient is estimated. The estimation is performed in a state where the input ends of the mixers 34-0 and 34-1 are grounded via a resistor and no signal is input from the antenna 31.
At this time, the local signal from the local oscillator 35 is fed to the mixer 3
When input to 4-0 and 34-1, DC offset occurs. Using this, the error of the complex coefficient is estimated.

The known complex baseband signals sampled by the band division filters 37-0 and 37-1 are barr.
(n)

[0028]

[Equation 3] Here, * indicates a complex conjugate, and m is an index indicating a band to be divided. Also, adc {} indicates A / D conversion. h
aty _m to (n) n = n, ... , determined to n + L-1.

[0029]

[Equation 4] The filter coefficient hatHm is further estimated and stored in the memory 22a.

Tl is the coefficient of the band division filter including the error
Let dhm (n). It is assumed that a DC offset passes through tldho (n) and a desired signal passes through tldh ₁ (n). Let barr (n) be the received signals sampled by the band division filters 37-0 and 37-1.

[0031]

[Equation 5] Here, δho represents the error of the filter ho. tldyo (n) is down-sampled, A / D converted, and then up-sampled.

[0032]

[Equation 6] Assuming that the received signal r (n) contains a large DC offset, the estimated value of y _U0 is

[0033]

[Equation 7] The DC offset is estimated from the equation (3.19).

[0034]

[Equation 8] tldy ₁ (n) is down-sampled, A / D converted, and then up-sampled.

[Equation 9] The interference component due to the DC offset is removed as follows.

[0035]

[Equation 10] The desired signal is taken out as follows.

[0036]

[Equation 11] FIG. 5 is a diagram showing the relationship between the frequency shift amount of the filter and the BER (bit error rate). “ADC” is the case where the DC offset component is removed by the complex coefficient subband filter after A / D converting the received signal as it is. In this case, the A / D conversion speed twice as high as that of other methods is required. “No compensation” is a case where the DC offset is removed by the complex coefficient analog subband filter, but the error compensation is not performed. “Invention” is the case where the proposed error compensation method is used. In either case, -4π / 16 ~
The BER decreases around −6π / 16 [rad / sample]. When shifted to more than −6π / 16 [rad / sample], the DC offset is mixed in the filtered signal,
BER becomes high. If it is shifted to −3π / 16 [rad / sample] or less, the image frequency signal is mixed with the filtered signal, so that the BER is also increased.

FIG. 6 shows the relationship between BER and Eb / No.
Is. Eb / No means “bit energy (power) vs.
"Noise power density", which means signal interference per information bit
Sound power ratio (S / N or SNR: Signal to Noise powe
r Ratio). "No DC offset" is DC off
The case without a set indicates the theoretical value of QPSK.
Propose by comparing “no compensation” and “invention”
BER can be greatly improved by using the error compensation method
Recognize. However, BER = 10 compared to “ADC” ^-2To
In addition, the characteristic deteriorates by about 1 dB. This is decimation
Aliasing component of DC offset
This is because it mixes with the output after filtering.

FIG. 7 is a diagram showing the relationship between the BER and the resolution of the A / D converter. As shown in FIG. 7, when the resolution of the A / D converter is 10 bits or less, the DC offset is suppressed by the complex coefficient subband filter in advance and then the A / D converter is used.
The BER is lower when D conversion is performed. On the other hand, the resolution is 1
For 2 bits or more, it is shown that the BER is smaller when the A / D converter is used to convert the digital signal and then the DC offset is removed by digital signal processing with a small filter coefficient error.

FIG. 8 is a diagram showing the structure of a low IF type receiver according to the second embodiment of the present invention. The low IF type receiver according to the second embodiment includes an antenna 71, a BPF 72,
LNA 73, mixers 74-1, 74-1, local oscillator 7
5, delay devices 76-0, 76-1, ..., 76-N-2, decimators 77-0, 77-1, ..., 77-N-1, analysis filter bank 78, amplifiers 79-0, 79-
1, ..., 79-N-1, A / D converters 80-0, 80-
, ..., 80-N-1, and a decorrelator 81 having a memory 81a.

The signal received via the antenna 71 is B
After passing through the PF 72, it is RF-amplified by the LNA 73 and converted by the mixer 74 into two low IF signals having a quadrature relationship. The converted signals are delayed by the delay units 76-0, 76-1, ...
After being sequentially delayed by 76-N-2, the decimator 7
., 77-N-1, downsampled to 1 / N and input to the analysis filter bank 78. As a result, the A / D converters 80-0, 80-
The conversion speed of 1, ..., 80-N-1 is set to 1 / N. The analysis filter bank 78 divides the input signal into a plurality of frequency components and outputs signals of each subband. The signals of each subband are amplifiers 79-0, 79-1, ..., 79-N.
A / D converters 80-0, 80-1,
, 80-N-1 is converted into a digital signal, and the decorrelator 81 removes mutual interference components from the input signal.

Here, the frequency component of the transmission signal is d = {d
Suppose we have ₀ , d ₁ , ..., d _N-1 } ^T. At this time, the received signal is represented as follows.

[0042]

[Equation 12] Here, ΔT is the sampling interval, and N is the number of inputs to the analysis filter bank 78. When there is no error in the coefficient of the analysis filter bank 78, the kth output is as follows.

[0043]

[Equation 13] However, the amplification degree of the amplifiers 79-0, 79-1, ..., 79-N-1 is set to 1 for simplification.

In the equation (2), the influence of the coefficient error of the analysis filter and the noise is neglected. In reality, the coefficient of the analysis filter contains an error, and the received signal contains noise.

[0045]

[Equation 14] Here, c _nk represents the error of the n-th coefficient of the filter that extracts the k-th band, and η represents thermal noise. When the formula (3) is displayed in matrix,

[0046]

[Equation 15] Further, the decimated signal is sent to each A / D converter 80.
Quantization noise is added when passing through −0, 80-1, ..., 80-N−1. Let this be ζ = {ζ ₀ , ζ ₁ , ..., ζ _{N -1} } ^T
And

[0047]

[Equation 16] In the present embodiment, the coefficient of the analysis filter bank 78 is estimated in advance by using the DC offset, the design value and the estimated value are compared, the cross-correlation matrix between the frequency-divided signals is calculated, and the cross-correlation is calculated. Compute the inverse of the matrix,
It is stored in the memory 81a. In the decorrelator 81,
The hatR inverse matrix is converted into A / D converters 80-0, 80-1,
The output of 80-N-1 is multiplied. Ie

[0048]

[Equation 17] The output of the decorrelator includes each frequency signal component d _k and a signal in which the thermal noise η and the quantization noise ζ are affected by the cross correlation.

Here, the error rate characteristics of the conventional receiver that does not use the decorrelator will be calculated. The interference component from the desired signal and other frequency bands received from the equation (10) is expressed as follows.

[0050]

[Equation 18] Here, Re {a} represents the real part of the complex number a, and [A] _k is N
X1 means the k-th element of the matrix A. Also, the variance of noise due to the influence of thermal noise is calculated from equation (8).

[0051]

[Formula 19] Where σ ² is the variance of thermal noise. Therefore

[0052]

[Equation 20] Further, assuming that the absolute value of the maximum input value of the A / D converter is A _M , the resolution is Bq bits, and the first 1 bit is used as a positive or negative sign, the quantization noise ζ _k is [− (A _M / 2 ^Bq ) , (A _M /
2 ^Bq )].

D _k is the BPSK (Binary Phase Shift Keyi
ng) signal, the conditional error rate is calculated as follows.

[0054]

[Equation 21] Here, [A] k, k means the (k, k) th element of the N × N matrix A.

[0055]

[Equation 22] When the variable function is expanded and the error function is expanded,

[0056]

[Equation 23] here

[0057]

[Equation 24] Therefore, the error rate of the receiver with compensation is

[0058]

[Equation 25] Here, the characteristics of the decorrelator type error compensation method of the present embodiment will be analyzed. The variance of the noise component is calculated in equation (10). About thermal noise

[0059]

[Equation 26] As for the quantization error, assuming that the variances of the quantization errors generated in each A / D converter are equal and λ ^2.

[0060]

[Equation 27] Given in. Quantization noise has a uniform distribution, but is modeled as Gaussian noise by being multiplied by the decorrelator coefficient. Therefore, the SNR (SN ratio) in the kth frequency band is obtained as follows.

[0061]

[Equation 28] Here, [A] kk means the (k, k) th element of the N × N matrix A. If d _k is a BPSK-modulated signal and the noise component is Gaussian, the BER is obtained as follows.

[0062]

[Equation 29] Here, as a simulation, each coefficient of the analysis filter bank 78 is generated according to a uniform distribution [-
1000 when adding an error of 0.01, 0.01]
Calculate the average error rate of 0 times.

It is assumed that the desired signal is BPSK-modulated, transmitted, received, and then down-converted to IF = 3π / 8 [rad / sample]. The communication path is assumed to be a white Gaussian noise path. Assuming coherent reception, one bit of the given resolution of the A / D converter is used to represent a positive or negative sign, and A _M = 2hatd _k or A _M = 2s _k (d)
And As for the image frequency signal, the BPSK signal is -3π /
It is assumed that it is received at 8 [rad / sample]. It is assumed that the power of the image frequency signal is 60 [dB] higher than the desired signal.

FIG. 9 is a diagram showing the relationship between BER and Eb / No in the second embodiment. Here, the resolution of the A / D converter is set to 10 [bit]. As shown in the figure, in the case of the conventional receiver that does not use the decorrelator, the BER characteristic shows the floor due to the coefficient error of the filter, and it hardly changes with the increase of Eb / No. This is because the image frequency signal interferes with the desired signal due to coefficient error.
On the other hand, when the decorrelator type compensation method is used, B
The ER characteristic shows almost the theoretical value of BPSK. This is because the decorrelator removes the interference of the image frequency signal.

The present invention is not limited to the above embodiment.

In the above-described embodiment, the known signal is produced by using the local signal by grounding the input of the mixer through the resistor. However, a local oscillator for generating the known signal is separately provided and supplied. You may do it.

Wavelets may be used instead of the filter bank. That is, instead of dividing into a large number of bands at once, for example, 2
You may make it divide | segment hierarchically so that division may be performed repeatedly. By doing so, the fineness of division can be switched by, for example, switching which short-circuits a predetermined layer to be divided.

[0068]

As described above, according to the present invention, the coefficient error of the band division filter which is the analog section can be compensated by the digital signal processing. This gives A
The requirements for the conversion speed and resolution of the / D converter can be relaxed, and it is possible to support a wideband communication system. Furthermore, parallel signal processing is enabled, and a low power consumption reception configuration can be realized.

[Brief description of drawings]

FIG. 1 is a diagram showing a configuration of a low IF system receiver according to a first embodiment of the present invention.

FIG. 2 shows the frequency response of a real coefficient QMF filter bank.

FIG. 3 is a diagram showing a frequency response of a QMF filter bank when the frequency is shifted by π / 4.

FIG. 4 is a diagram showing a configuration of a low IF type receiver of the first embodiment in the case of two divisions.

FIG. 5 is a diagram showing a relationship between a frequency shift amount of a filter and BER in the first embodiment.

FIG. 6 is a diagram showing a relationship between BER and Eb / No in the first embodiment.

FIG. 7 is a diagram showing a relationship between BER and resolution of an A / D converter in the first embodiment.

FIG. 8 is a diagram showing a configuration of a low IF type receiver according to a second embodiment of the present invention.

FIG. 9 is a diagram showing a relationship between BER and Eb / No in the second embodiment.

FIG. 10 is a diagram showing a configuration of a conventional direct conversion receiver.

FIG. 11 is a diagram showing a configuration of a conventional low IF type receiver.

[Explanation of symbols]

11, 31, 71 antenna 17, 38, 79 Amplifier 18, 39, 77 decimator 20,41 Up sampler 21, 42 band synthesis filter

Claims (6)

[Claims] 1. A band-division analog filter for dividing a low-IF signal into signals of a plurality of bands, an A / D converter for analog / digital converting the output of the band-division analog filter, and a known low-IF signal A known signal supplying means for supplying to the IF stage, a memory for storing a signal related to the output of the A / D converter when the known signal is supplied by the known signal supplying means, And a digital demodulator for demodulating an A / D converter output.
F system receiver. 2. The low IF type receiver according to claim 1, wherein the known signal supplying means uses a local oscillator for converting a received signal frequency into a low IF. 3. The band division analog filter has a filter bank configuration.
The low IF method receiver described. 4. The band division analog filter has a wavelet configuration.
The low IF method receiver described. 5. The low IF type receiver according to claim 1, wherein the band-division analog filter has a complex coefficient filter configuration. 6. The number of band divisions of the band division analog filter is N (N is an integer of 2 or more), and hatd = hatRd, where hatd is an output signal amplitude of each band (N, 1). ) Matrix d: an inverse matrix hatR to the (N, N) matrix hatR that is an (N, 1) matrix having the input signal amplitude of each band as a component
The low IF system receiver according to claim 1, further comprising a decorrelator for storing ^-1 in the memory, and further for multiplying the output of the A / D converter by hatR ^-1 .