JP2002315327A - Flyback converter with improved power factor - Google Patents

Abstract

PROBLEM TO BE SOLVED: To solve the problems of causing high cost and significantly lowering energy efficiency, in spite of a proposal of a means for solving the following problem: distortion of input current had been large and its power factor had been low in a capacitor input type power supply unit for the AC voltage input. SOLUTION: Windings for improving power factor, extracting power from an input capacitor, and taking out output power are provided in a plurality of yokes of a core at a transformer for the power conversion in a flyback type power supply unit, the transformer generates DC magnetic saturation, and a power rate taken out from the respective windings in changed based on the degree of the DC magnetic saturation to control the input current, while keeping the output voltage to a constant level, thus it is possible to improve the power factor at low cost.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an AC input and DC output flyback type power supply.

[0002]

2. Description of the Related Art FIG. 12 shows a typical circuit of a flyback type power supply of a capacitor input type. FIG. 13 shows the waveforms of the input AC voltage Vac and the input current J6 of the power supply device. In this power supply device, the alternating current of the input AC power supply 11 is rectified by a full-wave rectifier circuit including diodes D11, D12, D13, and D14, and is immediately input to the input side smoothing capacitor C1. Therefore, the waveform of the input current J6 is distorted as shown in FIG. 13 and has a poor power factor.

Several methods have been proposed for improving the power factor. FIG. 14 is a circuit example of a flyback type power supply that improves the input power factor. This power supply device uses one switching element Q1 to improve the power factor and control the output voltage.
Has been realized. In this power supply device, the input power is passed through a two-stage flyback converter in order to obtain the output power, so that there is a problem that the efficiency becomes low. Further, since two transformers for converting power are required, there is a problem that the cost is increased correspondingly. Other methods for improving the input power factor of a flyback type power supply have been proposed, but it has not been easy to achieve both low cost and high efficiency.

[0004]

SUMMARY OF THE INVENTION The present invention provides a method of improving the input power factor at low cost and high efficiency in an AC input flyback switching power supply.

[0005]

SUMMARY OF THE INVENTION In a transformer for converting power in a flyback type power supply, magnetic saturation is generated in a part of a core, and stored in the transformer depending on the degree of the magnetic saturation. By changing the ratio of the amount of energy used for output and the amount used for power factor improvement, the power factor can be improved with high efficiency at low cost.

[0006]

FIG. 1 is a circuit diagram of a first embodiment of the present invention. The first embodiment of the present invention is according to claim 2. In FIG. 1, an AC power supply 11 is a power supply of an AC voltage to be input to the power supply device, and a load 12 is a load of the power supply device. The power supply device according to the first embodiment of the present invention includes a controller 2, a transformer 31, voltage detectors 41 and 42,
Current detector 44, input side smoothing capacitor C1, output side smoothing capacitor C2, capacitor C3, diode D1
1, D12, D13, D14, D21 and a switching element Q1. The switching element Q1 turns on when the switching element control voltage VG is higher than a certain positive voltage, and turns off when the switching element control voltage VG is lower than the positive voltage.

FIG. 2 is a sectional view of a transformer 31 according to a first embodiment of the present invention.
1, 312 and windings L1, L2, L3. Between the yoke Y11 and the core 312 of the core 311 and the core 31
A gap is provided between one yoke Y12 and the core 312. The yoke Y13 of the core 311 and the core 312
No gap is provided between them, and a part of the yoke Y13 is thinner, so that magnetic saturation is more likely to occur in the yoke Y13 than in other parts of the cores 311 and 312.

FIG. 3 shows some signal waveforms in the first embodiment of the present invention. The switching element control voltage VG, the yoke Y11 of the transformer 31,
The respective magnetic fluxes F11, F1 in Y12, Y13
2 and F13 and waveforms of respective currents J1, J2, J3 flowing through the windings L1, L2, L3.

A time section P11 is a time section in which the switching element Q1 is on and no magnetic saturation occurs in the transformer 31, and a time section P12 is a time section following the time section P11. A time section in which magnetic saturation occurs in the transformer 31 and a time section P13 are a time section following the time section P12, and a time section and a time section in which magnetic saturation occurs in the transformer 31 with the switching element Q1 turned off. P14 is time section P
13 is a time section in which the switching element Q1 is in an off state, no magnetic saturation occurs in the transformer 31 and the current J3 flows, and a time section P15 is a time section following the time section P14. Switching element Q1
Is a time section in which the switch is off.

The operation of the first embodiment of the present invention is as follows. The AC voltage input from the AC power supply 11 is converted into a pulsating voltage by a full-wave rectifier circuit including diodes D11, D12, D13, and D14.

On the other hand, the input-side smoothing capacitor C1 is stored at a certain voltage, and the voltage across the input-side smoothing capacitor C1 is equal to diodes D11, D12, D13, D14.
Is higher than the peak value of the output voltage of the full-wave rectifier circuit composed of

Now, consider a case where the switching element Q1 is turned on only during the time intervals P11 and P12 in FIG. 3 from the state where the switching element Q1 is turned off. When the switching element Q1 is turned on, a current starts to flow through the winding L2. At this time, no current flows through the winding L1 and the winding L3. During the time section P11, no magnetic saturation occurs, and the gap between the yoke Y11 and the core 312 causes
Most of the magnetic flux in the yoke Y12 circulates through the yoke Y13, and little magnetic flux flows in the yoke Y11. During the time section P11, the current flowing through the winding L2 increases linearly with time, and accordingly, the magnetic flux in the yoke Y13 also increases linearly with time.

At the end of the time section P11, magnetic saturation begins to occur in a part of the yoke Y13. Since the switching element Q1 remains on, the time interval P1
During the period 2, the magnetic flux in the yoke Y12 continues to increase with time. At this time, the magnetic flux flowing through the yoke Y13 does not change much, and the magnetic flux starts flowing through the yoke Y11. During the time interval P12, the self-inductance of the winding L2 is reduced due to the magnetic saturation in the yoke Y13.
11, the rate of increase of the current flowing through the winding L2 with respect to time in the time section P12 is larger than that in the time section P11.

Next, assume that the switching element Q1 is turned off at the end of the time section P12. Here, a case is considered where the following relationship is established between the number of turns NL1 of the winding L1 and the number of turns NL3 of the winding L3.

[0015]

(Equation 1)

However, Vin is a diode D11, D1
VC1 is a voltage across the input-side smoothing capacitor C1, Vo is a voltage across the load 12, and VF is a forward direction of the diode D21. It is a voltage drop.

When this condition is satisfied, in the time section P13 when the switching element Q1 is turned off,
A current J1 as shown in FIG. 3 flows through the winding L1, and the input side smoothing capacitor C1 is charged. The length of this time section P13 is inversely proportional to the difference between the pulsating voltage Vin and VC1. At this time, the amount of energy extracted from the winding L1 is proportional to the square of the length of the time section P12. In the time section P13, due to the influence of the leakage magnetic flux from the winding L2, the winding L
3, a slight current flows as shown in FIG.

When the time section P13 ends and the magnetic saturation of the transformer 31 disappears, no current flows through the winding L1, and a current starts flowing through the winding L3. At this time, the winding L
3 is determined only by the cores 311 and 312 of the transformer 31, and the amount of energy extracted from the time interval P12
Has no bearing on the length of

The current flowing through the winding L3 is a diode D2
1, is smoothed by the output-side smoothing capacitor C2, and supplied to the load 12.

The current flowing through the winding L1 is a diode D1
The input current of this power supply device is passed through a full-wave rectifier circuit composed of D1, D12, D13, and D14. However, as shown in FIG. 3, the current flowing through the winding L1 contains many high-frequency line segments. The smoothed AC current is supplied from the AC power supply 11.

The controller 2 has two functions of controlling the voltage supplied to the load 12 to a desired value and controlling the input current from the AC power supply 11 of the power supply device.

The voltage supplied to the load 12 is detected by a voltage detector 41, and the signal is sent to the controller 2. The energy extracted from the winding L3 in one cycle of the on / off of the switching element Q1 is substantially constant when the on time of the switching element Q1 is longer than a certain level. The voltage supplied to the load 12 detected by 41 is compared with its target value, and based on the comparison result, the frequency at which the switching element Q1 is turned on and off is changed, and the voltage supplied to the load 12 is We are trying to approach that target value.

The method of controlling the input current from the AC power supply 11 of the power supply device is performed by adjusting the ON time when the switching element Q1 is turned ON / OFF. As described above, if the time in the time section P12 in FIG. 3 is long, the input current increases. Further, as described above, the input current also changes depending on the output voltage of the full-wave rectifier circuit including the diodes D11, D12, D13, and D14 and the voltage VC1.

Since the value of VC1 can be estimated from other measured values, the output voltage of the full-wave rectifier circuit constituted by the diodes D11, D12, D13 and D14 is detected by the voltage detector 4.
2, the time for turning on the switching element Q1 is determined by the detected value.

In the first embodiment of the present invention, the time for turning on the switching element Q1 is not directly determined, but the maximum value of the current J2 corresponding to the time for turning on the switching element Q1 is calculated. Then, the current J2 is detected by the current detector 44, and the switching element Q1 is turned off when the current J2 reaches the calculated maximum value after the switching element Q1 is turned on. By doing so, it is possible to reduce the influence on the input current due to factors such as an estimation error of the self-inductance of the winding L2.

In the first embodiment of the present invention, the winding L3 of the transformer 31 is wound around the yoke Y12.
It may be wound around the yoke Y13. At this time, a winding for recovering energy due to leakage magnetic flux generated near the winding L2 may be provided in the yoke Y12.

In the first embodiment of the present invention, the yoke of the core of the transformer 31 is a yoke Y11, a yoke Y13,
Although they are arranged in the order of the yoke Y12, they may be arranged in the order of the yoke Y11, the yoke Y12, and the yoke Y13.
Further, the yokes Y11, Y12, Y13 do not have to be arranged in a straight line.

In the first embodiment of the present invention, magnetic saturation in the transformer 31 occurs in a part of the yoke Y13. However, magnetic saturation may occur in the entire yoke Y13. Or it may occur in other parts of the core. Further, magnetic saturation may occur in a plurality of portions of the core.

In the first embodiment of the present invention, VC1
Is not measured, a means for detecting the value of VC1 may be provided, and the value detected by VC1 may be used to calculate the time for turning on the switching element Q1.

In the first embodiment of the present invention, the winding L
Although the time during which the switching element Q1 is turned on by detecting the current flowing through the switching element Q2 is determined, the time during which the switching element Q1 is turned on without detecting the current flowing through the winding L2 may be determined.

In the first embodiment of the present invention, although the input current from the AC power supply 11 is not detected,
1 or the value of a variable corresponding thereto is detected, and the feedback control is performed by changing the time for turning on the switching element Q1 so that the input current from the AC power supply 11 is proportional to the voltage of the AC power supply 11. May be hung.

In the first embodiment of the present invention, the switching element Q1 is determined based on the output voltage of the full-wave rectifier circuit composed of the diodes D11, D12, D13 and D14.
Is calculated, but even if the time during which the switching element Q1 is turned on is fixed, a certain effect of power factor improvement can be obtained, so that the instantaneous voltage of the AC power supply 11 and the diodes D11, D12, D13 can be obtained. , D14 may be constant without detecting the output voltage of the full-wave rectifier circuit.

In the first embodiment of the present invention, the winding L
A filter for smoothing the high-frequency component of the current flowing through 1
Diodes D11, D12, D
13 and D14, closer to the AC power supply 11 than the full-wave rectifier circuit.
It is also possible to adopt a configuration in which the diode is provided closer to the winding L1 than the full-wave rectifier circuit constituted by D12, D13, and D14, and a diode for preventing a backflow of current is provided at the output of the filter. By doing so, the diode D11,
D12, D13 and D14 can be used at low speed.

FIG. 4 is a circuit diagram of a second embodiment of the present invention. The second embodiment of the present invention relates to claim 3. In FIG. 4, an AC power supply 11 is a power supply of an AC voltage to be input to the power supply device, and a load 12 is a load of the power supply device. The power supply device according to the second embodiment of the present invention includes:
Controller 2, transformer 32, voltage detectors 41, 4
2, 43, current detector 44, input side smoothing capacitor C
1, output side smoothing capacitor C2, capacitor C4, diodes D11, D12, D13, D14, D21, D3
1, D32 and D33, and a switching element Q1.
The switching element Q1 has a switching element control voltage V
It turns on when G is higher than a certain positive voltage, and turns off when G is lower.

FIG. 5 is a sectional view of a transformer 32 according to a second embodiment of the present invention.
1, 322 and windings L1, L2, L3. Between the yoke Y22 and the core 322 of the core 321 and the core 32
A gap is provided between the first yoke Y23 and the core 322. The yoke Y21 of the core 321 and the core 322
No gap is provided between them, and a part of the yoke Y21 is thin, so that magnetic saturation is more likely to occur in the yoke Y21 than in other parts of the cores 321 and 322.

FIGS. 6 and 7 show some signal waveforms in the second embodiment of the present invention, in which the switching element control voltage VG and the yokes Y21, Y22 and Y23 of the transformer 32 are respectively shown. Magnetic flux F
The waveforms of the currents J1, J2, J3 flowing through the windings 21, F22, F23 and the windings L1, L2, L3 are shown. 6 and 7 show waveforms under different conditions.

FIG. 6 shows diodes D11, D12 and D1.
3 shows the waveform of a signal when the pulsating voltage Vin, which is the output voltage of the full-wave rectifier circuit constituted by D14, is lower than a certain threshold voltage Vth.
Is a time section in which no magnetic saturation occurs in the transformer 32 when the switching element Q1 is on, and a time section P22 is a time section following the time section P21, and magnetic saturation occurs in the transformer 32 when the switching element Q1 is on. The time section in which the occurrence occurs, the time section P23, is a time section following the time section P22, and the time section, in which the switching element Q1 is off and magnetic saturation occurs in the transformer 32, the time section P24 corresponds to the time section P23. The time interval that follows,
When the switching element Q1 is turned off and no magnetic saturation occurs in the transformer 32 and the current J3 flows, a time section P25 is a time section following the time section P24, and the switching element Q1 is turned off. It is a certain time section.

FIG. 7 shows that the pulsating voltage Vin is equal to the threshold voltage Vt.
5 shows the waveform of the signal at a time higher than h. Time section P26 is a time section in which the switching element Q1 is on and no magnetic saturation occurs in the transformer 32, and time section P27 follows the time section P26. The time section is a time section in which magnetic saturation occurs in the transformer 32 when the switching element Q1 is on, and the time section P28 is a time section following the time section P27.
Is a time section in which magnetic saturation occurs in the transformer 32 in a state in which the transformer is off, and a time section P29 is a time section following the time section P28, and magnetic saturation occurs in the transformer 32 in a state in which the switching element Q1 is off. The time section P20 in which the current J3 flows is a time section following the time section P29, and is a time section in which the switching element Q1 is in the off state.

The threshold voltage Vth is a voltage determined by the following equation.

[0040]

(Equation 2)

Where VC1 is the input side smoothing capacitor C
1, NL1 is the number of turns of the winding L1, NL
2 is the number of turns of the winding L2.

The operation of the second embodiment of the present invention is as follows. An AC voltage input from the AC power supply 11 is converted into a pulsating voltage Vin by a full-wave rectifier circuit including diodes D11, D12, D13, and D14.

On the other hand, the input side smoothing capacitor C1 is stored at a certain voltage, and the voltage across the input side smoothing capacitor C1 is higher than the peak value of the pulsating voltage Vin.

First, the pulsating voltage Vin is changed to the threshold voltage Vth.
The operation in the case of lower than this will be described.

Now, consider a case where the switching element Q1 is turned on only during the time intervals P21 and P22 in FIG. 6 from the state where the switching element Q1 is turned off. When the switching element Q1 is turned on, a current starts to flow through the winding L2 because the pulsating current voltage Vin is lower than the threshold voltage Vth. At this time, no current flows through the winding L1 and the winding L3. During the time section P21, magnetic saturation does not occur, and most of the magnetic flux in the yoke Y22 circulates through the yoke Y21 because of the gap between the yoke Y23 and the core 322, and little magnetic flux flows in the yoke Y23. During the time section P21, the current flowing through the winding L2 increases linearly with time, and accordingly, the yoke Y2
The magnetic flux at 1 also increases linearly with time.

At the end of the time section P21, magnetic saturation begins to occur in a part of the yoke Y21. Since the switching element Q1 remains ON, the time interval P2
During the period 2, the magnetic flux in the yoke Y22 continues to increase with time. At this time, the magnetic flux flowing through the yoke Y21 does not change much, and the magnetic flux starts flowing through the yoke Y23. During the time interval P22, the self-inductance of the winding L2 is reduced due to the magnetic saturation in the yoke Y21.
21 and the rate of increase of the current flowing through the winding L2 with respect to time in the time section P22 is larger than that in the time section P21.

Next, it is assumed that the switching element Q1 is turned off at the end of the time section P22. Here, the number 1 is set between the number of turns NL1 of the winding L1 and the number of turns NL3 of the winding L3.
Let us consider a case where the relationship is established. At this time, in the time section P23 in which the switching element Q1 is turned off, a current J1 as shown in FIG. 6 flows through the winding L1, and the input side smoothing capacitor C1 is charged. This time section P2
The length of 3 is inversely proportional to the difference between VC1 and pulsating voltage Vin. At this time, the amount of energy extracted from the winding L1 is proportional to the square of the length of the time section P22. Time section P
At 23, a slight current flows through the winding L3 due to the influence of leakage magnetic flux from the winding L2 as shown in FIG.

When the time interval P23 ends and the magnetic saturation of the transformer 32 disappears, no current flows through the winding L1, and a current starts flowing through the winding L3. At this time, the winding L
3 is determined only by the cores 321 and 322 of the transformer 32, and is determined by the time interval P22.
Has no bearing on the length of

The current flowing through the winding L3 is a diode D2
1, is smoothed by the output-side smoothing capacitor C2, and supplied to the load 12.

Next, the pulsating voltage Vin is changed to the threshold voltage Vth.
The operation in the case of higher than this is described.

Now, let us consider a case where the switching element Q1 is turned on only during the time intervals P26 and P27 in FIG. 7 from the state where the switching element Q1 is turned off. When the switching element Q1 is turned on, the pulsating current Vin is higher than the threshold voltage Vth, so that the current starts to flow through the winding L1. At this time, no current flows through the winding L2 and the winding L3. During the time section P26, magnetic saturation does not occur, and most of the magnetic flux in the yoke Y22 circulates through the yoke Y21 because of the gap between the yoke Y23 and the core 322, and little magnetic flux flows in the yoke Y23. During the time section P26, the current flowing through the winding L1 increases linearly with time, and accordingly, the yoke Y2
The magnetic flux at 1 also increases linearly with time.

At the end of the time section P26, magnetic saturation begins to occur in a part of the yoke Y21. Since the switching element Q1 remains ON, the time interval P2
During the period 7, the magnetic flux in the yoke Y22 continues to increase with time. At this time, the magnetic flux flowing through the yoke Y21 does not change much, and the magnetic flux starts flowing through the yoke Y23. During the time interval P22, the self-inductance of the winding L1 is reduced due to the magnetic saturation in the yoke Y21.
26, and the rate of increase of the current flowing through the winding L1 with respect to time in the time section P27 is larger than that in the time section P26.

Next, assume that the switching element Q1 is turned off at the end of the time section P27. Here, the number 1 is set between the number of turns NL1 of the winding L1 and the number of turns NL3 of the winding L3.
Let us consider a case where the relationship is established. At this time, in the time period P28 during which the switching element Q1 is turned off, a current J1 flows through the winding L1 as shown in FIG. 6, and the input-side smoothing capacitor C1 is charged. This time section P2
The length of 8 is inversely proportional to the difference between VC1 and pulsating voltage Vin. At this time, the amount of energy extracted from the winding L1 is proportional to the square of the length of the time section P27. Time section P
At 28, a slight current flows through the winding L3 due to the finite magnetoresistance of the yoke Y23 as shown in FIG.

When the time period P28 ends and the magnetic saturation of the transformer 32 disappears, no current flows through the winding L1, and a current starts flowing through the winding L3. At this time, the winding L
3 is determined only by the cores 321 and 322 of the transformer 32, and the time interval P27
Has no bearing on the length of

The current flowing through the winding L3 is the diode D2
1, is smoothed by the output-side smoothing capacitor C2, and supplied to the load 12.

The current flowing through the winding L1 is shown in FIGS.
As shown in FIG. This high-frequency component is removed by the capacitor C4, and the smoothed current passes through the full-wave rectifier circuit composed of the diodes D11, D12, D13, and D14, and passes through the AC power supply 11 of the power supply device.
From the input current.

The controller 2 has two functions of controlling the voltage supplied to the load 12 to a desired value and controlling the input current from the AC power supply 11 of the power supply device.

The voltage supplied to the load 12 is detected by the voltage detector 41, and the signal is sent to the controller 2. The energy extracted from the winding L3 in one cycle of the on / off of the switching element Q1 is substantially constant when the on time of the switching element Q1 is longer than a certain level. The voltage supplied to the load 12 detected by 41 is compared with its target value, and based on the comparison result, the frequency at which the switching element Q1 is turned on and off is changed, and the voltage supplied to the load 12 is We are trying to approach that target value.

The method of controlling the input current from the AC power supply 11 of the power supply device is performed by adjusting the ON time when the switching element Q1 is turned ON / OFF. As described above, if the time in the time section P22 in FIG. 6 or the time section 27 in FIG. 7 is long, the input current increases. Further, as described above, the input current also changes according to the voltage difference between VC1 and the pulsating voltage Vin.

Therefore, the absolute value of the input current from the AC power supply 11 of the power supply device is detected by the current detector 44, and the absolute value of the input AC voltage Vac is detected by the voltage detector 42 as the pulsating voltage Vin. Is the pulsating voltage Vi
The feedback control is performed while adjusting the ON time of the switching element Q1 so as to be substantially proportional to n.

At this time, the voltage VC1 across the input-side smoothing capacitor C1 rises or falls depending on the value of the coefficient to be multiplied by the pulsating voltage Vin when determining the target value of the absolute value of the input current and the load condition. Therefore, VC1 is connected to the voltage detector 4
While detecting by 3, the above-mentioned coefficient is adjusted so that the value of VC1 becomes substantially constant.

The first embodiment of the present invention in the second embodiment of the present invention
The advantage over the embodiment of the present invention is that the flow of energy when the instantaneous value of the absolute value of the input AC voltage Vac is higher than a certain level is not
From the input side smoothing capacitor C1 and there is no extraction of energy from the input side smoothing capacitor C1, so that the efficiency of the power supply device can be improved. Further, the capacitance required for the input-side smoothing capacitor C1 is also small.

In the second embodiment of the present invention, the yoke of the core of the transformer 32 is a yoke Y21, a yoke Y23,
Although they are arranged in the order of the yoke Y22, they may be arranged in the order of the yoke Y21, the yoke Y22, and the yoke Y23.
Further, the yokes Y21, Y22, Y23 do not have to be arranged in a straight line.

In the second embodiment of the present invention, magnetic saturation in the transformer 32 occurs in a part of the yoke Y21. However, magnetic saturation may occur in the entire yoke Y21. Or it may occur in other parts of the core. Further, magnetic saturation may occur in a plurality of portions of the core.

In the second embodiment of the present invention, VC1
Of the switching element Q1.
Is used to calculate the time for turning on the switching element Q1, but the time for turning on the switching element Q1 without detecting the value of VC1 may be calculated. When the value of VC1 rises, the threshold voltage V
Since th also increases, the power supplied to the input-side smoothing capacitor C1 decreases, and self-negative feedback is applied.

In the second embodiment of the present invention, the absolute value of the input current from the AC power supply 11 is detected, and the detected value is used to calculate the time for turning on the switching element Q1. It is also possible to calculate the time during which the switching element Q1 is turned on without detecting the amount related to the input current from the AC power supply 11, and control the input current from the AC power supply 11 in an open loop.

In the second embodiment of the present invention, VC1
And the absolute value of the input current from the AC power supply 11 are detected, and the detected values are used to calculate the time for turning on the switching element Q1, but the value of VC1 and the input from the AC power supply 11 are used. The switching element Q is determined based on only the value of the pulsating voltage Vin detected without detecting the amount related to the current.
The time for turning 1 on may be calculated.

In the second embodiment of the present invention, VC1
, The value of Vin and the absolute value of the input current from the AC power supply 11 are used to calculate the time for turning on the switching element Q1, but the detected value is used to turn on the switching element Q1. The time to perform may be constant irrespective of those values. Even in this case, a certain power factor improvement is possible.

In the second embodiment of the present invention, a part of the core of the transformer 32 causes magnetic saturation for an arbitrary value of Vin, but when the value of Vin is less than a certain value. May not be magnetically saturated.

FIG. 8 is a circuit diagram of a third embodiment of the present invention. The third embodiment of the present invention relates to claim 4. 8, an AC power supply 11 is a power supply of an AC voltage to be input to the power supply device, and a load 12 is a load of the power supply device. The power supply device according to the third embodiment of the present invention includes:
Controller 2, transformer 33, voltage detectors 41, 4
2, 43, current detector 44, input side smoothing capacitor C
1, output side smoothing capacitor C2, capacitor C4, diodes D11, D12, D13, D14, D21, D2
2, D31, D32, D33, D34, D35, and a switching element Q1. The switching element Q1 turns on when the switching element control voltage VG is higher than a certain positive voltage, and turns off when the switching element control voltage VG is lower than the positive voltage.

FIG. 9 is a sectional view of a transformer 33 according to a third embodiment of the present invention.
1, 332 and windings L1, L2, L3, L4, L5. Between the yoke Y31 of the core 331 and the core 332,
Gaps are provided between the yoke Y32 and the core 332 of the core 331 and between the yoke Y34 and the core 332 of the core 331. No gap is provided between the yoke Y33 of the core 331 and the core 332. In addition, a part of the core 332 is thinner, and magnetic saturation is more likely to occur in that part than in other parts of the cores 331 and 332. The number of turns of the winding L4 is smaller than the number of turns of the winding L3.

FIGS. 10 and 11 show some signal waveforms in the third embodiment of the present invention. The switching element control voltage VG, the yokes Y31, Y32, Y33 and Y34 of the transformer 33 are shown. F31, F32, F33, F34 and winding L
Currents J1, J2, J flowing in L1, L2, L3
3 is shown. 10 and 11 show waveforms under different conditions.

FIG. 10 shows diodes D11, D12, D
13 and D14 show the waveform of a signal when the pulsating voltage Vin, which is the output voltage of the full-wave rectifier circuit, is higher than the threshold voltage Vth defined by the equation (2). A time section in which no magnetic saturation occurs in the transformer 33 when the switching element Q1 is on and a time section P32 is a time section following the time section P31.
3, a time section in which magnetic saturation occurs, time section P33
Is a time section following the time section P32, a time section in which magnetic saturation occurs in the transformer 33 when the switching element Q1 is off, a time section P34 is a time section following the time section P33, and the switching element Q1 In the off state, no magnetic saturation occurs in the transformer 33 and the current J3
, The time section P35 is the time section P3
4, which is a time section in which the switching element Q1 is off.

FIG. 11 shows a signal waveform when the pulsating voltage Vin is lower than the threshold voltage Vth defined by the equation (2). In the time section P36, the switching element Q1 is in an on state. The time section, the time section P37, is a time section following the time section P36, in which the switching element Q1 is off and the current J3 flows.
The time section P38 is a time section following the time section P37, and is a time section in which the switching element Q1 is turned off and the current J3 does not flow.

The operation of the third embodiment of the present invention is as follows. An AC voltage input from the AC power supply 11 is converted into a pulsating voltage Vin by a full-wave rectifier circuit including diodes D11, D12, D13, and D14.

On the other hand, the input side smoothing capacitor C1 is stored at a certain voltage, and the voltage across the input side smoothing capacitor C1 is higher than the peak value of the pulsating voltage Vin.

First, the pulsating voltage Vin is changed to the threshold voltage Vth.
The operation in the case of higher than this is described.

Now, consider a case where the switching element Q1 is turned on only during the time intervals P31 and P32 in FIG. 10 from the state where the switching element Q1 is turned off. When the switching element Q1 is turned on, the pulsating current Vin is higher than the threshold voltage Vth.
The current starts to flow through. At this time, the winding L2, the winding L3,
No current flows through winding L4. During the time section P31, no magnetic saturation occurs, and the yoke Y32 and the core 332
, Most of the magnetic flux in the yoke Y31 and the magnetic flux in the yoke Y34 circulate through the yoke Y33, and little magnetic flux flows in the yoke Y32. During the time section P31, the current flowing through the winding L1 increases linearly with time, and accordingly, the magnetic flux in the yoke Y31 also increases linearly with time. The current flowing through the winding L5 also increases linearly with time.

At the end of time section P31, core 3
Magnetic saturation starts to occur in a part of the part 31 which is thin. Since the switching element Q1 remains on, the magnetic flux in the yoke Y31 continues to increase with time during the time section P32. At this time, the magnetic flux flowing from the yoke Y31 through the yoke Y32 does not change much,
Magnetic flux starts to flow through the yoke Y32. During the time section P32, the self-inductance of the winding L1 is
1 due to magnetic saturation in a part of the tapered portion, which is smaller than during the time interval P31,
The rate of increase of the current flowing through the winding L1 with respect to time in 2 becomes larger than during the time section P31.
On the other hand, the magnetic flux circulating from the yoke Y34 through the yoke Y33 is not affected by magnetic saturation, so that the self-inductance of the winding L5 does not change. Therefore, the current flowing through the winding L5 also increases linearly with time in the time section P32 as in the time section P31.

Next, assume that the switching element Q1 is turned off at the end of the time section P32. Here, a case is considered where the following relationship is established between the number of turns NL1 of the winding L1 and the number of turns NL3 of the winding L3.

[0081]

(Equation 3)

At this time, in the time period P33 during which the switching element Q1 is turned off, a current J1 as shown in FIG. 10 flows through the winding L1, and the input side smoothing capacitor C1 is charged. At the same time, a current J3 as shown in FIG. 10 also flows through the winding L3, and the current is supplied to the load side. The length of this time section P33 is inversely proportional to the difference between VC1 and the pulsating voltage Vin. At this time, the amount of energy extracted from the winding L1 is proportional to the square of the length of the time section P32. In the time section P33, a slight current flows through the winding L4 due to the influence of the leakage magnetic flux from the winding L2 as shown in FIG.

When the time section P33 ends and the magnetic saturation of the transformer 33 disappears, no current flows through the winding L1. A current flows through the winding L3 as shown in FIG. At this time, the amount of energy extracted from the winding L3 is close to the sum of the amount determined by the cores 331 and 332 of the transformer 33 and the amount injected into the transformer 33 from the winding L5 during the time section P31 and the time section P32. Become.

The current flowing through the winding L3 is equal to the diode D2
The current flowing through the winding L4 is rectified by the diode D22, is smoothed by the output-side smoothing capacitor C2, and is supplied to the load 12.

Next, the pulsating voltage Vin is changed to the threshold voltage Vth.
The operation in the case of lower than this will be described. Ripple voltage Vin
When the voltage is lower than the threshold voltage Vth, it is assumed that the transformer 33 operates so that magnetic saturation does not occur.

Now, consider a case where the switching element Q1 is turned on only during the time interval P36 in FIG. 11 from the state where the switching element Q1 is turned off. When the switching element Q1 is turned on, the pulsating voltage Vin becomes the threshold voltage V
Therefore, the current starts to flow through the winding L1 and the winding L2. At this time, no current flows through the winding L3, the winding L4, and the winding L5. Since the pulsating voltage Vin is low,
Accordingly, the current flowing through the winding L1 during the time section P36 decreases, and the magnetic flux passing through the yoke Y31 also decreases. Due to the absence of magnetic saturation and the gap between the yoke Y32 and the core 332, most of the magnetic flux in the yoke Y31 circulates through the yoke Y33, and little magnetic flux flows through the yoke Y32. During the time section P36, the current flowing through the winding L1 and the current flowing through the winding L2 increase linearly with time, and accordingly, the magnetic flux in the yoke Y33 also increases linearly with time. To go.

Next, assume that switching element Q1 is turned off at the end of time section P36. Here, the number 3 between the number of turns NL1 of the winding L1 and the number of turns NL3 of the winding L3.
Let us consider a case where the relationship is established. At this time, in the time period P37 during which the switching element Q1 is turned off, the current J1 flowing through the winding L1 decreases rapidly as shown in FIG. 11, and the input-side smoothing capacitor C1 is hardly charged. On the other hand, the current J3 flowing through the winding L3 flows as shown in FIG. At this time, the amount of energy extracted from the winding L3 is proportional to the square of the length of the time section P36.

The current flowing through the winding L3 is equal to the current of the diode D2.
1, is smoothed by the output-side smoothing capacitor C2, and supplied to the load 12.

The current flowing through the winding L1 contains many high-frequency components as shown in FIGS. This high-frequency component is removed by the capacitor C4, and the smoothed current becomes an input current from the AC power supply 11 of the power supply device through a full-wave rectifier circuit including diodes D11, D12, D13, and D14.

The controller 2 has two functions of controlling the voltage supplied to the load 12 to a desired value and controlling the input current from the AC power supply 11 of the power supply device.

The voltage supplied to the load 12 is detected by the voltage detector 41, and the signal is sent to the controller 2. The energy extracted from the winding L3 in one cycle of turning on and off the switching element Q1 depends on the time during which the switching element Q1 is on, but is adjusted by the frequency at which the switching element Q1 is turned on and off. The controller 2 compares the voltage supplied to the load 12 detected by the voltage detector 41 with the target value, and based on the comparison result, determines the switching element Q
The frequency for turning on / off 1 is changed so that the voltage supplied to the load 12 approaches its target value. The time during which the switching element Q1 is on also changes as described later, but its influence on the voltage supplied to the load 12 is eliminated by using closed loop control.

The method of controlling the input current from the AC power supply 11 of the power supply device adjusts the on-time when the switching element Q1 is turned on and off when the pulsating voltage Vin is higher than the threshold voltage Vth. It is done by doing. In the case of the third embodiment of the present invention, the input current increases when the time in the time section P32 in FIG. 10 is long. Further, as described above, the input current also changes depending on the voltage of the difference between the pulsating voltage Vin and VC1.

Therefore, the absolute value of the input current from the AC power supply 11 of the power supply device is detected by the current detector 44, and the absolute value of the input AC voltage Vac is detected by the voltage detector 42 as the pulsating voltage Vin. Is the pulsating voltage Vi
Feedback control is performed while adjusting the time for turning on the switching element Q1 so as to be substantially proportional to n.

The voltage VC1 across the input-side smoothing capacitor C1 rises or falls depending on the load value and the value of the coefficient to be multiplied by the pulsating voltage Vin in determining the target value of the absolute value of the input current at that time. Therefore, VC1 is connected to the voltage detector 4
While detecting by 3, the above-mentioned coefficient is adjusted so that the value of VC1 becomes substantially constant.

When the pulsating current Vin is lower than the threshold voltage Vth, the input current from the AC power supply 11 of the power supply device is determined by the input AC voltage Vac and the input-side smoothing capacitor C1.
Is almost determined by the voltage VC1 at both ends and the load current. However, since the absolute value of the input current is smaller as the absolute value of the input AC voltage Vac is smaller, there is no major problem, and the time for turning on the switching element Q1 is fixed. ing.

The first embodiment of the present invention in the third embodiment of the present invention.
An advantage of the third embodiment and the second embodiment of the present invention is that the efficiency of the power supply device can be increased. In the third embodiment of the present invention, when the instantaneous value of the absolute value of the input AC voltage Vac is higher than a certain level, the flow of energy is from the AC power supply 11 to the load 12 and from the AC power supply 11 to the input-side smoothing capacitor C1, There is no extraction of energy from the input side smoothing capacitor C1. In addition, when the instantaneous value of the absolute value of the input AC voltage Vac is lower than a certain level, the flow of energy is
And from the input side smoothing capacitor C1 to the load 12,
There is almost no energy supply to the input side smoothing capacitor C1. Therefore, the efficiency of the power supply device can be increased, and the capacitance required for the input-side smoothing capacitor C1 is small.

In the third embodiment of the present invention, the yoke of the core of the transformer 33 is a yoke Y21, a yoke Y23,
Although the yokes Y22 are arranged in this order, the yokes Y31 and Y32 may be in the reverse order, and the yokes Y33 and Y34 may be in the reverse order. In addition, yoke Y3
1, Y32, Y33, and Y34 need not be arranged in a straight line.

In the third embodiment of the present invention, the magnetic saturation in the transformer 32 is generated in a part of the core 331. However, the magnetic saturation may be generated in a part of the core 332. Good. Further, magnetic saturation may occur in a plurality of portions of the core.

In the third embodiment of the present invention, VC1
And the absolute value of the input current from the AC power supply 11 are detected, and the detected values are used to calculate the time for turning on the switching element Q1, but the value of VC1 and the input from the AC power supply 11 are used. The switching element Q is determined based on only the value of the pulsating voltage Vin detected without detecting the amount related to the current.
The time for turning 1 on may be calculated.

In the third embodiment of the present invention, VC1
, The value of Vin and the absolute value of the input current from the AC power supply 11 are used to calculate the time for turning on the switching element Q1, but the detected value is used to turn on the switching element Q1. The time to perform may be constant irrespective of those values. Even in this case, a certain power factor improvement is possible.

In the third embodiment of the present invention, the diode D35 is used in the circuit of the power supply device, but this is for recovering the energy of the leakage magnetic flux, and the diode D35 may not be provided.

In the third embodiment of the present invention, although the transformer 33 has the winding L5, the winding L5 may not be provided.

In the third embodiment of the present invention, the transformer 33 has the winding L4, but this recovers the energy of the leakage magnetic flux, and the winding L4 may not be provided. The energy recovered by the winding L4 may be returned to the input-side smoothing capacitor C1.

[0104]

As described above, by using the present invention, a flyback type power supply having a high power factor can be realized using one switching element and one transformer.
It is economical. In addition, since the power is converted by one transformer, the conversion loss can be suppressed to a low level, and a flyback power supply with improved power factor and high efficiency can be realized.

[Brief description of the drawings]

FIG. 1 is a circuit diagram of a first embodiment of the present invention.

FIG. 2 is a sectional view of a transformer according to the first embodiment of the present invention.

FIG. 3 is a signal waveform according to the first embodiment of the present invention.

FIG. 4 is a circuit diagram of a second embodiment of the present invention.

FIG. 5 is a sectional view of a transformer according to a second embodiment of the present invention.

FIG. 6 is a signal waveform according to the second embodiment of the present invention.

FIG. 7 is a signal waveform according to the second embodiment of the present invention.

FIG. 8 is a circuit diagram of a third embodiment of the present invention.

FIG. 9 is a sectional view of a transformer according to a third embodiment of the present invention.

FIG. 10 shows a signal waveform according to a third embodiment of the present invention.

FIG. 11 shows a signal waveform according to a third embodiment of the present invention.

FIG. 12 is a circuit diagram of a conventional technique.

FIG. 13 shows waveforms of an input voltage and an input current in a circuit according to the related art.

FIG. 14 is a circuit diagram of a conventional technique.

[Explanation of symbols]

11 AC power supply 12 Load 2 Controller 31, 32, 33, 35, 36 Transformers 311, 312, 321, 322, 331, 332,.
Core 41, 42, 43 Voltage detector 44 Current detector C1 Input-side smoothing capacitor C2 Output-side smoothing capacitor C3, C4 Capacitor D11 , D12, D13, D14, D21, D22, D
31, D32, D33, D34, D35... Diode F11, F12, F13... Magnetic flux F21, F22, F23... Magnetic flux F31, F32, F33, F34. J3, J4, J5, J6 ... currents L1, L2, L3, L4, L5 ... windings P11, P12, P13, P14, P15 ... time sections P20, P21, P22, P23, P24, P25, P
26, P27, P28, P29 ... time section P31, P32, P33, P34, P35, P36, P
37, P38 time section Q1 switching element VG switching element control voltage Vac input AC voltage Vin pulsating voltage Vth threshold voltage Y11 ... Yoke Y21, Y22, Y23 ... Yoke Y31, Y32, Y33, Y34 ... Yoke

Claims (4)

[Claims] 1. A flyback converter having AC power as input and DC power as output, wherein at least one of the transformers for converting power has a core having three or more yokes, and There is a case where a winding is provided for one or more yokes, and magnetic saturation is generated with respect to a magnetic flux generated by a current flowing through the winding. When extracting the energy stored in the plurality of windings, the ratio of the energy extracted from each of the windings depends on the degree of the magnetic saturation, and by making the degree of the magnetic saturation constant or changing, A power factor improving flyback converter having a function of adjusting an input current of the flyback converter. 2. The power factor improving flyback converter according to claim 1, further comprising a rectifier circuit, a transformer, a switching element, and an input-side smoothing capacitor, wherein the transformer includes a first yoke, a second yoke, and a third yoke. May have a first winding and a second winding, and may cause magnetic saturation in a part of the transformer core, and the first winding is wound around the first yoke. , The second winding is wound around a second yoke, the rectifier circuit has a first output terminal and a second output terminal, receives an input AC voltage, and outputs a first output signal. And outputting a pulsating voltage converted between the terminal and the second output terminal, wherein the input side smoothing capacitor has a first terminal and a second terminal,
A first winding of the transformer is connected between a first output terminal of the rectifier circuit and a first terminal of the input-side smoothing capacitor, and a second winding of the transformer and the switching element are connected in series. The connected circuit is connected between a first terminal and a second terminal of the input side smoothing capacitor, and a second terminal of the input side smoothing capacitor is connected to a second output terminal of the rectifier circuit. A flyback converter with improved power factor. 3. The power factor improving flyback converter according to claim 1, wherein the rectifier circuit, the transformer, the switching element, the input side smoothing capacitor, the first rectifier element, the second rectifier element, the third rectifier element, The transformer has a first connection point and a second connection point, and the transformer includes a core having a first yoke, a second yoke, and a third yoke, a first winding, and a second winding.
And a third winding may cause magnetic saturation in a part of the core of the transformer, and the first winding and the second winding are both wound around a second yoke, The third winding is wound around a first yoke, and the rectifier circuit has a first output terminal and a second output terminal, receives an input AC voltage, and receives a first output terminal. And a second output terminal. The input side smoothing capacitor has a first terminal and a second terminal, and a first winding of the transformer is provided. A wire is connected between a first output terminal of the rectifier circuit and a first connection point, and a first rectifier element is connected between the first connection point and a first connection point of the input side smoothing capacitor.
And a circuit in which a second rectifying element and a second winding of the transformer are connected in series is connected between a first terminal of the input-side capacitor and a second connection point. The third rectifying element is connected between the first connection point and the second connection point,
The switching element is connected to a second connection point and a second output terminal of the rectifier circuit, a second terminal of the input-side smoothing capacitor is connected to a second output terminal of the rectifier circuit,
A power factor improving flyback converter, wherein a third winding of the transformer is provided to extract power of an output of the converter from the transformer. 4. The power factor improving type flyback converter according to claim 1, wherein the rectifier circuit, a transformer, a switching element, an input side smoothing capacitor, a first rectifier element, a second rectifier element, a third rectifier element, A transformer having a first connection point and a second connection point, wherein the transformer includes a core having a first yoke, a second yoke, a third yoke and a fourth yoke, a first winding, a second winding; A first winding wound on a first yoke and a second winding on the first yoke, wherein the first winding has a winding and a third winding, which may cause magnetic saturation in a part of the core of the transformer.
Is wound around a second yoke, the third winding is wound around a third yoke, the rectifier circuit has a first output terminal and a second output terminal, And outputs a pulsating voltage converted between a first output terminal and a second output terminal. The input-side smoothing capacitor includes a first terminal and a second terminal. A first winding of the transformer is connected to a first output terminal of the rectifier circuit and a first output terminal of the rectifier circuit.
, The first rectifying element is connected between the first connecting point and the first terminal of the input side smoothing capacitor, and the second rectifying element and the second winding of the transformer are connected. A circuit in which the lines are connected in series is connected between a first terminal and a second connection point of the input side capacitor, and a third rectifying element is connected between the first connection point and the second connection point. The switching element is connected to a second connection point and a second output terminal of the rectifier circuit, a second terminal of the input-side smoothing capacitor is connected to a second output terminal of the rectifier circuit, 3. The power factor improving flyback converter according to claim 1, wherein the third winding is provided to extract electric power of the output of the converter from the transformer.