JP2002232502A - Signal generating circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To solve a problem of a conventional radio unit conducting direct modulation and direct demodulation that cannot have realized a 90-degree phase shifter when the frequency of transmission reception gets higher. SOLUTION: The signal generating circuit is configured such that two oscillators 100, 101 and two mixers 103a, 103b are used and after output signals from the one oscillator are distributed to the mixers with a phase difference of 90-degrees, the mixers mix signals outputted from the other local oscillator and a desired local oscillation LO signal is obtained.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a modulator / demodulator used for mobile communication, and more particularly to a quadrature modulator or a quadrature demodulator suitable for a radio communication device.

[0002]

2. Description of the Related Art As a method for reducing the size of a radio section of a mobile communication device, a direct conversion method for directly converting the frequency of a baseband signal and an RF (high frequency) signal using a direct quadrature modulator and a direct quadrature demodulator is known. It has been known.
Operates by the conventional, direct conversion method,
FIG. 9 shows a configuration example of a radio using a direct quadrature modulator and a direct quadrature demodulator. In FIG. 9, 100 is a local (L
O) a local oscillator for oscillating a signal, 200 an antenna for transmitting and receiving radio waves, 201 a bandpass filter, 202
Is an antenna switch for switching between transmission and reception, 203 is a low noise amplifier, 204 is a quadrature demodulator, 205a and 205b are baseband signal output terminals, 206 is a power amplifier,
07 is a quadrature modulator, and 208a and 208b are baseband signal input terminals.

First, the operation at the time of reception will be described. The RF signal received by the antenna 200 is input to the antenna switch 202 after unnecessary frequency components are suppressed by the band-pass filter 201. At the time of reception, the antenna switch 202 is connected to the low-noise amplifier 203 side, and the RF signal is amplified by the low-noise amplifier 203 and then input to the quadrature demodulator 204. The LO (local) signal output from the local oscillator 100 has the same frequency as the RF signal, and the RF signal is down-converted by being mixed with the LO signal inside the quadrature duplexer 204 to output the baseband signal. The signals are distributed to the baseband signals of the in-phase component and the quadrature component from the terminals 205a and 205b and output respectively.

Next, at the time of transmission, the in-phase component and quadrature component baseband signals input from the baseband signal input terminals 208a and 208b respectively are transmitted to the quadrature modulator 207.
Is mixed with the LO signal and up-converted into an RF signal. The RF signal is amplified to a desired power by the power amplifier 206, and then output from the antenna 200 via the antenna switch 202 and the band pass filter 201.

Next, the internal circuit of the quadrature demodulator 204 is shown in FIG.
Explanation will be made using 0. 10, the same or corresponding parts as those in FIG. 9 are denoted by the same reference numerals, 104a and 104b are mixers, 105a and 105b are baseband signal output terminals, and 106 is an RF signal input terminal.

[0006] RF input from RF signal input terminal 106
The signal is distributed inside quadrature demodulator 204 and
R output from the local oscillator 100 at 4a and 104b
It is mixed with the LO signal having the same frequency as the F signal. This LO signal is output from the local oscillator 100 by 9
The signal is divided into two using the 0-degree distributor 102 and the limiter amplifier 1
After removing the amplitude error at 07a and 107b, the mixer 1
04a and 104b, and baseband signal output terminals 105a and 105b output in-phase and quadrature-component baseband signals, respectively.

On the other hand, in the internal circuit of the quadrature modulator 207, 105a and 105b may be considered as baseband signal input terminals and 106 as an RF signal output terminal in FIG. Omitted.

By the way, in the radio equipment of the direct conversion system described above, the LO frequency and the transmission / reception frequency must be the same. At this time, due to the large power modulated signal transmitted from the power amplifier 206 during transmission,
There is a problem that the oscillation frequency of the local oscillator 100 is disturbed and the modulation accuracy is reduced.

In order to solve such a problem, a configuration including an LO signal generation circuit as shown in FIG. 11 is known. In the figure, the same parts as those described above are denoted by the same reference numerals, and description thereof will be omitted. 101 is a local oscillator,
210 is a mixer.

In this configuration, two local oscillators 100,
The signal output from 101 is mixed by mixer 210 to obtain a desired LO frequency that is the same frequency as the RF signal. Thereby, the transmission / reception frequency of the radio device and the oscillation frequency of the local oscillator can be offset, and the oscillation frequency of the local oscillator can be stabilized.

[0011]

However, in the above-described configuration, when the frequency used for transmission and reception increases, 9
There is a problem that a 0 degree phase shifter cannot be realized.

FIG. 12 shows a configuration of a 90-degree phase shifter which is an RC phase shifter using widely used resistors and capacitors. As shown in the figure, in the RC phase shifter, the input terminal 2
After the signal input from 13 is divided into two, one is
The other is input to a circuit composed of the series capacitance 212a and the parallel resistance 211a, and the other is input to a circuit composed of the series resistance 211b and the parallel capacitance 212b. Here, the capacity 212
a, the capacitance value of 212b is C, the resistance value of the resistors 211a, 211b is R, the frequency of the input signal is f,

[0013]

If the resistance value and the capacitance value are set so that f = 1 / (2πRC) (Equation 1), 9
A signal having a phase difference of 0 degrees is output to the output terminals 214a and 214b.
Output from

However, as can be seen from (Equation 1),
Both the resistance value and the capacitance value need to be reduced as the frequency of the input / output signal increases, and it becomes impossible to realize a high-precision 90-degree phase shifter due to variations in circuit elements and the influence of temperature deviation.

In particular, when a modulator / demodulator is realized by a semiconductor integrated circuit, the dispersion increases when the difference between the contact resistance between the wiring layers and the desired resistance value decreases, and when the capacitance value decreases, the floating in the substrate or wiring increases. Since the capacitance and the like cannot be ignored, it is difficult to realize a high-precision 90-degree phase shifter.

There is also a type in which a half frequency divider is used as a 90-degree phase shifter, but there is a problem that current consumption increases as the frequency increases. Further, with a half frequency divider, the frequency divider cannot follow at higher frequencies such as millimeter waves and quasi-millimeter waves.

In the direct quadrature modulation method, power control is often performed in the baseband system. However, when the power of the baseband signal is reduced, the power of the LO signal leaking to the RF signal output terminal 106 is reduced by the modulation signal. Power becomes relatively small. The leakage of the LO signal power is called carrier leakage, and needs to be suppressed to a sufficiently small value so as not to deteriorate the reception characteristics.

In view of these conventional problems, the present invention provides a signal generation circuit, a quadrature modulator, and a signal generation circuit having a configuration for offsetting a local oscillation frequency from a transmission / reception frequency and capable of easily realizing a 90-degree phase shifter even at a high frequency. It is an object to provide a quadrature demodulator.

[0019]

According to a first aspect of the present invention, a quadrature demodulator or a quadrature modulator is input to a quadrature demodulator or quadrature modulator together with a signal to be demodulated or modulated. And a 90-degree divider for receiving an input of a first oscillation signal and distributing the same into an in-phase component and a quadrature component; and receiving an input of a second oscillation signal. Receiving a first mixer that multiplies the in-phase component signal output from the 90-degree divider with the second oscillation signal to perform frequency conversion, and receives the input of the second oscillation signal, A second mixer that multiplies the quadrature component signal output from the degree divider by the second oscillation signal to perform frequency conversion, and includes an output of the first mixer and a second mixer. Output as the local signal. A signal generating circuit for.

Further, the second invention (corresponding to claim 2)
In the present invention, the first oscillation signal is supplied from a first local oscillator, and the second oscillation signal is supplied from a second local oscillator.

The third invention (corresponding to claim 3)
Is that the first oscillation signal and the second oscillation signal are
This is the present invention as described above, which is supplied from the same third local oscillator.

A fourth aspect of the present invention (corresponding to claim 4)
Are a first limiter amplifier provided between the 90-degree distributor and the first mixer, and a second limiter amplifier provided between the 90-degree distributor and the second mixer And the second local oscillator, the first mixer and the second
And the gain control amplifiers respectively provided between the mixers.

The fifth invention (corresponding to claim 5)
Is the present invention as described above, comprising at least one frequency divider or multiplier provided between the third local oscillator and the 90-degree divider.

The sixth invention (corresponding to claim 6)
Is the present invention as described above, comprising at least one frequency divider or multiplier provided between the third local oscillator and the first mixer and the second mixer.

Further, a seventh aspect of the present invention (corresponding to claim 7)
Are a first limiter amplifier provided between the 90-degree distributor and the first mixer, and a second limiter amplifier provided between the 90-degree distributor and the second mixer And the at least one divider or multiplier and a first
And the gain control amplifier provided between the second mixer and the second mixer.

The eighth invention (corresponding to claim 8)
A phase comparator for comparing the phase of the in-phase signal and / or the quadrature signal output from the 90-degree divider with a phase of a preset reference signal, and filtering an output from the phase comparator And a loop filter that feeds back to the third local oscillator to form a phase locked loop between the first mixer, the second mixer, and the third local oscillator. .

The ninth invention (corresponding to claim 9)
A signal input to the first mixer and / or a signal input to the second mixer from an output of the first mixer and / or an output of the second mixer,
A phase comparator for comparing a phase of a preset reference signal with a reference signal, and a loop filter for filtering an output from the phase comparator and feeding back the output to the second local oscillator or the third local oscillator, The present invention is the above-described invention in which a phase locked loop is formed between a first mixer and the second mixer and the second local oscillator or the third local oscillator.

According to a tenth aspect of the present invention (corresponding to claim 10), a signal and / or a signal input to the first mixer from the output of the first mixer and / or the output of the second mixer are provided. A filter for extracting a difference frequency component between two frequency components of the signal input to the second mixer; a phase comparator for comparing an output from the filter with a phase of a preset reference signal; A loop filter for filtering an output from a comparator and feeding back the output to the second local oscillator or the third local oscillator, wherein the first mixer and the second mixer and the second local oscillator or In the present invention, a phase locked loop is formed with the third local oscillator.

Further, according to an eleventh aspect of the present invention (corresponding to claim 11), the 90-degree distributor includes a resistor R and a resistor R.
The present invention is the C phase shifter according to the present invention.

In a twelfth aspect of the present invention (corresponding to claim 12), the 90-degree distributor is a half frequency divider.

According to a thirteenth aspect of the present invention (corresponding to claim 13), the signal generating circuit according to any one of the first to twelfth aspects is multiplied by an output of the first mixer and an RF signal. A quadrature demodulator comprising: a third mixer for obtaining a baseband signal of an in-phase component by using a third mixer for multiplying an output of the second mixer by the RF signal to obtain a baseband signal of a quadrature component It is.

According to a fourteenth aspect of the present invention, there is provided a signal generation circuit according to any one of the first to twelfth aspects, wherein a baseband signal having an in-phase component with the output of the first mixer is provided. And a sixth mixer for obtaining the RF signal based on an output of the second mixer and a baseband signal of a quadrature component. It is a vessel.

According to a fifteenth aspect of the present invention (corresponding to claim 15), the oscillation frequency is set so that the difference frequency between the first local oscillator and the second local oscillator becomes constant. The present invention is the above-described present invention including a cutoff unit for cutting off a frequency component.

The present invention as described above, for example,
Using two local oscillators and two mixers, an output signal from one oscillator is distributed by 90 degrees, and then mixed with a signal output from the other local oscillator to obtain a desired LO frequency. .

Further, as another example, the present invention has one local oscillator and two mixers, and mixes a signal obtained by distributing the output from the local oscillator by 90 degrees with the other signal output from the local oscillator. By doing so, a desired LO frequency is obtained.

[0036]

Embodiments of the present invention will be described below with reference to the drawings.

(Embodiment 1) FIG. 1 shows a configuration of a quadrature modulator or a quadrature demodulator according to a first embodiment of the present invention. In the figure, 100 and 101 are local oscillators, 102 is a 90-degree distributor, 103a, 103b, 1
Reference numerals 04a and 104b denote mixers, 105a and 105b denote baseband signal input terminals or baseband signal output terminals, 106 denotes RF signal output terminals or RF signal input terminals, 107a and 107b denote limiter amplifiers, and 108 denotes a gain control amplifier.

Here, the operation of the present embodiment will be described using a quadrature modulator as an example. In the case of the quadrature demodulator, the basic operations are the same, except that the flows of the baseband signal and the RF signal are reversed. This is the same in the following embodiments.

Baseband input terminals 105a, 105b
The baseband signals input from the
The signals are mixed with the LO signal at 04a and 104b. This LO signal has the same frequency as the transmission frequency. After the output signal of the local oscillator 100 is distributed by the 90-degree distributor 102, the LO signal is passed through limiter amplifiers 107a and 107b to remove an amplitude error, and 103a,
At 103b, the signal is generated by mixing and up-converting the output of the local oscillator 101 amplified by the gain control amplifier 108.

Therefore, since the phase difference between the LO signals input to the mixers 104a and 104b is maintained at 90 degrees, a signal subjected to quadrature modulation is output from the RF signal output terminal. That is, according to this configuration, the local oscillator 10
The signal output from the local oscillator 101 is mixed with the signal output from the local oscillator 101 after the signal output from the local oscillator 101 is distributed by the 90-degree splitter 102. A high-precision 90-degree distributor can be realized while keeping the oscillation frequency low.

As an example, when the 90-degree divider 102 is constituted by an RC phase shifter, if the desired LO signal frequency is 5 GHz, the oscillation frequency of the local oscillator 100 is 2 GHz and the oscillation frequency of the local oscillator 101 is 3 GHz. It can be realized by setting. At this time, the resistance value and the capacitance value of the RC phase shifter are about 160Ω and about 0.5 pF, respectively, which are values that can be formed on the semiconductor integrated circuit. Furthermore, since the oscillation frequencies of local oscillators 100 and 101 are offset with respect to the transmission frequency of 5 GHz, disturbance of the oscillation frequency due to the transmission signal is less likely to occur.

When a 1/2 frequency divider is used as the 90-degree divider, for example, the oscillation frequency of the local oscillator 100 is set to 2 G
Hz and the oscillation frequency of the local oscillator 101 may be set to 4 GHz.

Further, the present embodiment has a function of controlling the gain of the quadrature modulator by the gain control amplifier 108. FIG. 2 shows LO injection level versus conversion gain characteristics of the mixers 104a and 104b. As shown in FIG.
The conversion gains of 04a and 104b change according to the LO injection level. Accordingly, the LO injection level of the mixers 103a and 103b (in this case,
When the output from the local oscillator 101 is changed, the output levels of the mixers 103a and 103b change, and the conversion gains of the mixers 104a and 104b change accordingly.

According to the present embodiment, when the output power of the quadrature modulator is reduced by the gain control function,
The power of the LO signal input to 4a and 104b is also reduced, and as a result, the leakage of the LO signal can be suppressed.
Further, since the output from the local oscillator 101 is distributed after passing through the gain control amplifier, the amplitude error between the quadrature component and the in-phase component can be reduced.

(Embodiment 2) FIG. 3 shows a configuration of a quadrature modulator or a quadrature demodulator according to a second embodiment of the present invention. And the description is omitted. Further, in the figure, reference numeral 120 denotes a 1/2 frequency divider which is a 90-degree divider, 1
Reference numeral 21 denotes a 1 / N frequency divider.

The operation of the quadrature modulator according to the embodiment of the present invention as described above is as follows.

The output from the local oscillator 100 is branched,
One is divided by a 1 / N frequency divider 121, and the other is further branched and output to mixers 103a and 103b.

The output of the local oscillator 100 that has been frequency-divided by the N-frequency divider 121 is divided into 9 by the frequency-divider 120.
After being distributed to 0 degrees and having their amplitude errors removed by limiter amplifiers 107a and 107b, mixers 103a,
At 04b, up-conversion is performed by mixing with the other signal that has been branched and output from the local oscillator 100, and is output as an LO signal.

According to the present embodiment, the same effect as in the first embodiment can be obtained with one local oscillator, so that a quadrature modulator or a quadrature demodulator with small size and low current consumption can be realized.

As an example, if the desired LO frequency is 5 GHz,
If the frequency division number of the frequency divider 121 is 2, the local oscillator 100
Has an oscillation frequency of 4 GHz.

In FIG. 3, a 1/2 frequency divider is used as the 90-degree divider, but an RC phase shifter may be used. As an example of the case where the half frequency divider 120 is replaced with an RC phase shifter, a desired LO frequency is set to 5 GHz and the frequency divider 1
Assuming that the frequency division number of 21 is 4, the oscillation frequency of local oscillator 100 is 4 GHz.

(Embodiment 3) FIG. 4 shows a configuration of a quadrature modulator or a quadrature demodulator according to a third embodiment of the present invention. The same or similar parts as those in the circuit of FIG. The reference numerals are used and the description is omitted. Further, in the figure, reference numeral 122 denotes a multiplier, and reference numeral 108 denotes a gain control amplifier.

The operation of the quadrature modulator according to the embodiment of the present invention as described above is as follows. Local oscillator 1
00 is divided into two parts. One part is divided by the divider 121 and then divided by 90 degrees by the half divider 120. The amplitude error is removed by the limiter amplifiers 107a and 107b. The signals are input to the mixers 103a and 104b. The other output is multiplied by the multiplier 122 and then input to the mixers 103a and 104b via the gain control amplifier 108. In mixers 103a and 104b, these two signals are mixed and up-converted, and
An O frequency is obtained.

According to this configuration, the output of the quadrature modulator can be varied based on the same principle as that of the first embodiment, while suppressing the leakage of the LO signal when the output power is small. Further, by using the multiplier 122, the local oscillator 10
Without increasing the oscillation frequency of 0 or using a local oscillator having a lower oscillation frequency, a high LO
Frequency can be obtained.

As an example, if the frequency division number of the frequency divider 121 is 2 and the frequency multiplier of the frequency multiplier 122 is 2, the LO frequency of 9 GHz can be obtained with the oscillation frequency of the local oscillator 100 of 4 GHz.

Although a 1/2 frequency divider is used as the 90-degree divider in FIG. 4, an RC phase shifter may be used, and the frequency divider 121 is replaced with a multiplier. Needless to say, the same effect can be obtained.

(Embodiment 4) FIG. 5 shows a configuration of a quadrature modulator or a quadrature demodulator according to a fourth embodiment of the present invention. The same or similar parts as those in the circuit of FIG. The reference numerals are used and the description is omitted. Further, in the figure, 130 is a voltage controlled oscillator, 131 is a phase comparator, and 132 is a loop filter.

In this embodiment, the limiter amplifier 107 is used to stabilize the oscillation frequency of the local oscillator 100.
a and a part of the output from 107b are input to a phase comparator 131, and the phase comparison result with the reference signal is
2 has a phase-locked loop configuration that feeds back to the voltage-controlled oscillator 130.

According to the present embodiment, the phase comparator 131
Is converted to a lower frequency by the frequency dividers 120 and 121, so that a part of the output of the voltage controlled oscillator 130 is directly input to the phase comparator 131, compared with a method of configuring a normal phase locked loop. The current consumption of the phase comparator can be reduced.

In the configuration shown in FIG. 5, both outputs of the limiter amplifiers 107a and 107b are connected to the phase comparator 131.
However, one of the outputs may be fed back. In this case, the phase comparator 1
Although the input level at 31 decreases, the same effect can be obtained with a simple configuration. Also, the 1 / N frequency divider 12
1 may be fed back to the phase comparator 131.

(Embodiment 5) FIG. 6 shows a configuration of a quadrature modulator or a quadrature demodulator according to a fifth embodiment of the present invention. The same or similar parts as those in FIG. The reference numerals are used and the description is omitted. Also, in the figure, 130 and 135 are voltage controlled oscillators, 131 and 13
7 is a phase comparator, 132 and 138 are loop filters, 1
36 is a filter.

In this embodiment, the voltage controlled oscillator 130
Constitutes a normal phase locked loop composed of a phase comparator 131 and a loop filter 132.

On the other hand, the voltage controlled oscillator 135 is
Of the frequency components included in the outputs 03a and 103b, the difference frequency component between the oscillation frequencies of the voltage controlled oscillators 130 and 135 is extracted by the filter 136, and the phase comparator 137
To form a phase locked loop by feeding back to the voltage controlled oscillator 135 via the loop filter 138.

According to the present embodiment, the phase comparator 137
Since the input frequency can be reduced, current consumption can be reduced.

Also, as in the configuration example shown in FIG. 7, even in the configuration of the second embodiment using one voltage-controlled oscillator (local oscillator 100), the difference between the input frequencies from the outputs of mixers 103a and 103b can be obtained. By extracting frequency components, a similar effect can be realized.

In FIGS. 6 and 7, the mixer 103a,
Although both outputs of 103b are fed back to the phase comparator, either output may be fed back.

(Embodiment 6) FIG. 8 shows a configuration of a quadrature modulator or a quadrature demodulator according to a sixth embodiment of the present invention. The same or similar parts as those in the circuit of FIG. The reference numerals are used and the description is omitted. Further, in the figure, reference numeral 140 denotes a band rejection filter.

In this embodiment, the local oscillator 10
The oscillation frequencies f1 and f2 of 0 and 101 are

[0069]

F1 = (fRF-fIM) / 2 (Equation 2)

[0070]

F2 = (fRF + fIM) / 2 (Equation 3) Here, fRF is the transmission / reception frequency, fIM
Is the cut-off frequency of the band cut filter 140.

The LO signals of the mixers 104a and 104b are
The outputs of the local oscillators 100 and 101 are output to mixers 103a, 1
This is obtained by mixing and up-converting at 03b. At this time, a difference frequency component is generated together with the frequency component of the sum of the two frequencies. Here, the oscillation frequencies of the local oscillators 100 and 101 are set as shown in (Equation 2) and (Equation 3).
Respectively

[0072]

F1 + f2 = fRF (Equation 4)

[0073]

F2−f1 = fIM (Equation 5), and the difference frequency component fIM can be set to a constant value. Therefore, the fIM component included in the outputs of the mixers 104a and 104b is blocked by the band rejection filter 140 and is not output from the RF signal output terminal 106.

In a radio communication device, unnecessary frequency components radiated from an antenna interfere with other radio communication devices, and thus need to be suppressed to a sufficiently small value. Thus, the fIM component can be effectively suppressed. In FIG. 8, the band cutoff filter 104 is used as a circuit for cutting off the fIM component, but other means may be used.
A similar effect can be obtained by configuring the matching circuit of the RF port 4b so as to cut off fIM.

In each of the above embodiments, the local oscillator 100 corresponds to the first local oscillator or the third local oscillator according to the present invention, and the local oscillator 101 corresponds to the second local oscillator according to the present invention. Is equivalent to The mixer 103a is the first mixer of the present invention,
3b corresponds to the second mixer of the present invention. The limiter amplifier 107a corresponds to the first limiter amplifier of the present invention, and the limiter amplifier 107b corresponds to the second limiter amplifier of the present invention. Also, the mixer 104a
Corresponds to the third mixer or the fifth mixer of the present invention,
The mixer 104b is the fourth mixer or the sixth mixer of the present invention.
This is equivalent to a mixer. Further, the band-pass filter 201 corresponds to a blocking unit of the present invention.

In each of the above embodiments, the description has been made assuming that the LO signal obtained by the signal generation circuit of the present invention is used for the quadrature modulator or the quadrature demodulator. The modulator or the quadrature demodulator can be used in any of a direct conversion system and a heterodyne system. However, it is more desirable to use it in a direct conversion wireless device.

As described above, according to the present invention, as an example, using two local oscillators and two mixers,
After distributing the output signal from one oscillator by 90 degrees, mixing with the signal output from the other local oscillator to obtain a desired LO frequency, a high-precision 90-degree phase shifter can be easily realized, and It is possible to realize a quadrature modulator and a quadrature demodulator in which disturbance of the oscillation frequency of a local oscillator due to a transmission signal is suppressed.

Further, a signal having one local oscillator and two mixers, wherein the output from the local oscillator is distributed by 90 degrees,
Quadrature modulation that easily realizes a high-precision 90-degree phase shifter by obtaining the desired LO signal frequency by mixing the output signal of the local oscillator and suppresses disturbance of the oscillation frequency of the local oscillator due to the transmission signal And a quadrature demodulator can be realized with one local oscillator.

[0079]

As described above, according to the present invention, a high-precision 90-degree phase shifter can be used even when a high-frequency LO signal is output, and the oscillation frequency of the local oscillator due to the transmission signal can be reduced. A signal generation circuit used for a quadrature modulator or a quadrature demodulator that suppresses disturbance can be realized.

[Brief description of the drawings]

FIG. 1 is a block diagram illustrating a configuration of a quadrature modulator or a quadrature demodulator according to a first embodiment of the present invention.

FIG. 2 is a graph showing a relationship between a LO injection level of a mixer and a conversion gain.

FIG. 3 is a block diagram showing a configuration of a quadrature modulator or a quadrature demodulator according to a second embodiment of the present invention.

FIG. 4 is a block diagram showing a configuration of a quadrature modulator or a quadrature demodulator according to a third embodiment of the present invention.

FIG. 5 is a block diagram showing a configuration of a quadrature modulator or a quadrature demodulator according to a fourth embodiment of the present invention.

FIG. 6 is a block diagram showing a configuration of a quadrature modulator or a quadrature demodulator according to a fifth embodiment of the present invention.

FIG. 7 is a block diagram showing a modification of the quadrature modulator or the quadrature demodulator according to the fifth embodiment of the present invention.

FIG. 8 is a block diagram showing a configuration of a quadrature modulator or a quadrature demodulator according to a sixth embodiment of the present invention.

FIG. 9 is a block diagram showing a configuration of a wireless communication device using direct modulation and direct demodulation;

FIG. 10 is a block diagram showing a configuration of a conventional quadrature modulator or quadrature demodulator.

FIG. 11 is a block diagram showing a configuration of a quadrature modulator or a quadrature demodulator using two local oscillators.

FIG. 12 is a circuit diagram showing a configuration of an RC phase shifter;

[Explanation of symbols]

101, 101 Local oscillator 102 90-degree distributor 103a, 103b, 104a, 104b Mixer 105a, 105b Baseband signal input terminal or baseband signal output terminal 106 RF signal output terminal or RF signal input terminal 107a, 107b Limiter amplifier 108 Gain control Amplifier

 ──────────────────────────────────────────────────続 き Continuing on the front page (72) Inventor Mitsuru Tanabe 1 Yukicho, Takatsuki-shi, Osaka, Japan Matsushita Electronics Industrial Co., Ltd. In-house (72) Inventor Kenshi Fukuda 1 Yukicho, Takatsuki-shi, Osaka, Japan Matsushita Electronics Industrial Co., Ltd. F-term ( Reference) 5K004 AA01 BA02 BD01

Claims (15)

[Claims] A quadrature demodulator or a signal generation circuit for generating a local signal to be input together with a signal to be demodulated or modulated to a quadrature modulator. A 90-degree divider that distributes the components into a quadrature component and a second oscillation signal, and the in-phase component signal output from the 90-degree divider is multiplied by the second oscillation signal. A first mixer that performs frequency conversion, and receives the input of the second oscillation signal, multiplies the quadrature component signal output from the 90-degree divider with the second oscillation signal, and performs frequency conversion. And a second mixer that performs the output of the first mixer and the output of the second mixer as the local signal. 2. The signal generation circuit according to claim 1, wherein the first oscillation signal is supplied from a first local oscillator, and the second oscillation signal is supplied from a second local oscillator. 3. The signal generation circuit according to claim 1, wherein the first oscillation signal and the second oscillation signal are supplied from the same third local oscillator. 4. A first limiter amplifier provided between the 90-degree distributor and the first mixer, and a second limiter amplifier provided between the 90-degree distributor and the second mixer. 3. The signal generation circuit according to claim 2, further comprising a limiter amplifier, and a gain control amplifier provided between the second local oscillator, the first mixer, and the second mixer. 4. 5. The signal generation circuit according to claim 3, further comprising at least one frequency divider or multiplier provided between the third local oscillator and the 90-degree divider. 6. The signal according to claim 3, further comprising at least one frequency divider or multiplier provided between said third local oscillator and said first and second mixers. Generation circuit. 7. A first limiter amplifier provided between the 90-degree distributor and the first mixer, and a second limiter amplifier provided between the 90-degree distributor and the second mixer. 7. The signal generation circuit according to claim 6, further comprising: a limiter amplifier, and a gain control amplifier provided between the at least one frequency divider or multiplier and a first mixer and a second mixer. 8. A phase comparator for comparing the phase of the in-phase signal and / or the quadrature signal output from the 90-degree divider with a phase of a preset reference signal, And a loop filter for filtering an output and feeding back to the third local oscillator, wherein a phase-locked loop is formed between the first and second mixers and the third local oscillator. Three
3. The signal generation circuit according to claim 1. 9. A signal input to the first mixer and / or a signal input to the second mixer from an output of the first mixer and / or an output of the second mixer in advance. A phase comparator for comparing the phase of the reference signal with a set reference signal; and a loop filter for filtering an output from the phase comparator and feeding back the output to the second local oscillator or the third local oscillator. 4. The signal generation circuit according to claim 2, wherein a phase-locked loop is configured between the first mixer and the second mixer and the second local oscillator or the third local oscillator. 5. 10. A signal input to the first mixer and / or a signal input to the second mixer from an output of the first mixer and / or an output of the second mixer. A filter for extracting a difference frequency component between frequency components; a phase comparator for comparing an output from the filter with a phase of a preset reference signal; and a second local unit for filtering an output from the phase comparator. An oscillator or a loop filter for feeding back to the third local oscillator, wherein a phase locked loop is formed between the first mixer and the second mixer and the second local oscillator or the third local oscillator. The signal generating circuit according to claim 2 or 3, wherein the signal generating circuit is configured. 11. The signal generation circuit according to claim 1, wherein said 90-degree divider is an RC phase shifter having a resistance and a capacitance. 12. The signal generating circuit according to claim 1, wherein said 90-degree divider is a half frequency divider. 13. A signal generating circuit according to claim 1, further comprising: a third mixer for multiplying an output of said first mixer by an RF signal to obtain a baseband signal of an in-phase component. A quadrature demodulator including a third mixer that obtains a quadrature component baseband signal by multiplying the output of the second mixer by the RF signal. 14. The signal generating circuit according to claim 1, wherein: a fifth mixer that obtains an RF signal based on an output of the first mixer and a baseband signal having an in-phase component; A quadrature modulator comprising: a sixth mixer that obtains the RF signal based on an output of the second mixer and a baseband signal of a quadrature component. 15. The apparatus according to claim 14, further comprising: an oscillation frequency setting unit that sets an oscillation frequency so that a difference frequency between the first local oscillator and the second local oscillator is constant, and shuts off the difference frequency component. Quadrature modulator.