JP2001177585A - Transmitter using non-linear distortion compensation circuit by negative feedback system - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a negative feedback amplifier which controls a phase without opening/closing a Cartesian loop and observing an in-phase component and an orthogonal component, which follows a phase change even if the phase in the loop is changed by temperature fluctuation and a secular change, stably operates the phase characteristic of the loop, stabilizes a transmission operation characteristic and prevents the deterioration of a spurious characteristic in a digital radio machine having a power amplifier distortion compensation system by the Cartesian loop. SOLUTION: A phase difference between the base band signal being a reference and a signal inputted to a modulator is compared. When the phase difference exists, the phase characteristic of the loop is stably kept, the oscillation phenomenon of the loop and the occurrence of spuriousness are suppressed and a stable operation is conducted by controlling the phase of the negative feedback loop by a phase controller and a phase shifter.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a transmitter, and more particularly, to a negative feedback circuit for performing nonlinear distortion compensation and a phase control method used in the transmitter.

[0002]

2. Description of the Related Art Linear digital modulation method, for example, 16QAM
(Quadrature Amplitude Modulation) and π / 4 shift QPS
In a wireless system using K (Quadrature Phase Shift Keying) or the like, nonlinear distortion compensation of a power amplifier is indispensable, and various nonlinear distortion compensation methods (linearizers) are used. Among them, the Cartesian loop negative feedback linearizer has been used for a long time. A conventional linear feedback amplifier will be described with reference to FIG. FIG. 2 is a block diagram showing a configuration of a transmission unit of a digital wireless device using the Cartesian negative feedback linearizer system.

A baseband signal generator 1 outputs an in-phase component (hereinafter, referred to as an I component) and a quadrature component (hereinafter, referred to as a Q component) of a baseband signal. And I component,
The feedback signal is added to the feedback signal by the adder 2-1 and output, and input to the loop filter 3-1. Similarly, the Q component is added to the feedback signal by the adder 2-2, output, and input to the loop filter 3-2. Loop filters 3-1 and 3-2 determine the input I component and Q
The components are band-limited, and input to quadrature modulator 4.

[0004] A reference signal generator 11 generates a reference frequency signal, and a PLL frequency synthesizer 12 and a PLL frequency synthesizer are provided.
13 Input the reference signal. The PLL frequency synthesizer 12 generates a first local oscillation signal (hereinafter, referred to as an LO signal) based on the reference signal, and outputs the first LO signal to the quadrature modulator 4 and the phase shifter 18.
Give a signal. Further, the PLL frequency synthesizer 13 generates a second LO signal based on the reference signal, and the mixer 6 and the mixer 1
Give 5 the second LO signal. The phase shifter 18 controls the phase of the first LO signal according to a control signal input from the phase controller 19, and supplies the phase-controlled first LO signal to the quadrature demodulator 16.

[0005] The quadrature modulator 4 modulates the I component I 'and the Q component Q' of the input baseband signal into a signal of an intermediate frequency band (hereinafter referred to as an IF frequency band) by the first LO signal. The modulated modulated signal is converted to a bandpass filter (BP
F) A signal obtained by removing unnecessary components from the modulated wave signal input to 5 and input to the mixer 6 is input to the mixer 6. The mixer 6 converts the input modulated wave signal into a desired frequency by a second LO signal output from the PLL frequency synthesizer 13 and supplies the converted signal to a band-pass filter (BPF) 7. The bandpass filter 7 removes unnecessary spurious components from the input signal and supplies the signal to a power amplifier (PA) 8. The power amplifier 8 amplifies the input signal to a specified output level and transmits the signal via the antenna 9.

Since this negative feedback amplifier has a negative feedback linearizer configuration using a Cartesian loop, a part of the output signal of the power amplifier 8 is fed back by the directional coupler 10 and given to the attenuator (ATT) 14. Can be The attenuator 14 adjusts the power level of the input signal to an appropriate value and supplies the adjusted value to the mixer 15. The mixer 15 frequency-converts the signal input from the attenuator 14 into an IF frequency using the second LO signal, and supplies the IF frequency to the quadrature demodulator 16.

[0007] The quadrature demodulator 16 receives the first LO signal input from the phase shifter 18 and outputs the baseband signals of the I component and the Q component.
Outputs q and i. I component i is added through switch 20-1
The I component of the feedback signal is input to the subtraction input side of 2-1 and the Q component q is input to the subtraction input side of the adder 2-2 via the switch 20-2 as the Q component of the feedback signal. At this time, switch 20-
The outputs of 1 and 20-2 are connected to the adders 2-1 and 2-2, respectively.

In such a negative feedback, in order to stabilize the system, the input signals I and Q are input on the input side of the adders 2-1 and 2-2.
And the feedback signals i and q need to be in phase (phase difference 0). That is, when a phase difference occurs between the input signal and the feedback signal, it is necessary for the adders 2-1 and 2-2 to control the phase shifted by a maximum of π radians to eliminate the phase difference.

Next, a phase control method will be described. First, the output sides of the switches 20-1 and 20-2 in FIG. 2 are switched and connected to the phase controller 19 side, and the feedback loop is set to an open loop state. A predetermined DC voltage for phase adjustment is given only to the I component from the baseband signal generator 1, the Q component is set to 0 (Q = 0), quadrature modulation is performed according to the above-described operation, and output from the antenna 9 is performed. I do. The output waveform of the power amplifier 8 at this time is a non-modulated carrier. The output of the power amplifier 8 is partially fed back by the directional coupler 10 and the output of the feedback signal of the quadrature demodulator 16 is observed according to the above-described operation. Voltage appears, and no signal (DC) is output to the Q component side. However, when the phases do not match, a DC voltage corresponding to the phase shift appears in the output on the Q component side. Therefore, the rotation angle of the phase can be obtained from the DC voltage of the I component and the Q component. In the phase controller 19,
The phase of the obtained rotation angle is controlled and rotated in the opposite direction by controlling the phase shifter 18 to adjust the phase of the first LO signal, thereby adjusting the output of the feedback signal of the quadrature demodulator 16 to a negative value. Stabilize the feedback. If the phase of the input signal matches the phase of the feedback signal, the output on the Q component side becomes 0.
Switch 20-1 and 20-2 to adders 2-1 and 2-2, and operate in a closed loop.

[0010]

In the above-mentioned prior art,
It is necessary to open and close the feedback loop each time the phase adjustment operation is performed. Therefore, since the open loop is performed during the phase adjustment, the phase cannot be adjusted with respect to the phase change of the closed loop during the continuous operation. In addition, since switching is performed via switching means to open and close the loop, and the phase is controlled by the DC voltage of the feedback signal on the input side of the switching means, the voltage effect at the switching means differs between the open loop and the closed loop. In addition, when the compensation setting of the offset voltage of the system set in the closed loop state becomes an open loop, it does not match, and accurate phase control cannot be performed. Furthermore, for example, even when the phase changes due to a change in phase characteristics due to a temperature change or a gain change, it is not possible to detect the amount of the phase change.

If the operation is performed with the phase shifted, the phase margin of the entire system is lost, and in the worst case, an oscillation phenomenon may occur. Further, in the process, spurious is generated, and the output operation characteristic is deteriorated. Therefore, the transmission operation must be stopped and readjusted. An object of the present invention is to provide a negative feedback amplifier which eliminates these drawbacks and does not deteriorate output operation characteristics.

[0012]

In order to achieve the above object, the present invention provides a phase of an input baseband signal before adding a feedback signal and a signal after adding the feedback signal and before quadrature modulation. By controlling the phase of the second LO signal input to the quadrature demodulator that demodulates the feedback signal based on the compared information, the phase of the second LO signal is always The control is performed. As a result, a negative feedback amplifier that stably operates its output operation characteristic during normal transmission operation without using a test signal for determining the state of the phase, without leaving an open loop, is always realized. Things.

[0013]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS One embodiment of the present invention will be described with reference to FIG. FIG. 1 is a block diagram showing a configuration of a negative feedback circuit of a quadrature modulation type transmitter using the present invention. The baseband signal generator 1 outputs an I component and a Q component of a baseband signal. Then, the I component is added to the feedback signal by the adder 2-1 and output, and input to the loop filter 3-1. Similarly, the Q component is added to the feedback signal by the adder 2-2, output, and input to the loop filter 3-2. Loop filter 3-1
And 3-2 band-limit the input I component and Q component, respectively, and input them to quadrature modulator 4.

A reference signal generator 11 generates a reference frequency signal, and outputs the reference frequency signal to a PLL frequency synthesizer 12 and a PLL frequency synthesizer.
13 Input the reference signal. The PLL frequency synthesizer 12 generates a first LO signal based on the reference signal, and outputs the first LO signal to the quadrature modulator 4.
And the first LO signal to the phase shifter 18 '. Further, the PLL frequency synthesizer 13 generates a second LO signal based on the reference signal, and supplies the mixer 6 and the mixer 15 with the second LO signal. The phase shifter 18 ′ changes the phase of the first LO signal input from the PLL frequency synthesizer 12 according to a control signal (phase difference information and phase delay / advance information) input from the phase controller 17. The first LO signal, which is controlled in phase, is supplied to the quadrature demodulator 16.

The quadrature modulator 4 uses the I component I 'and Q component Q' of the input baseband signal as a first LO signal (carrier signal).
Is modulated into a signal in the IF frequency band. The modulated modulated wave signal is supplied to the band pass filter 5. The bandpass filter 5 removes unnecessary components of the input modulated wave signal,
Give to mixer 6. The mixer 6 converts the input modulated wave signal
The frequency is converted to a desired frequency by the second LO signal output from the PLL frequency synthesizer 13 and is provided to the bandpass filter 7. The bandpass filter 7 removes unnecessary spurious components from the input signal and supplies the signal to the power amplifier 8.
The power amplifier 8 amplifies the input signal to a specified output level and transmits the signal via the antenna 9.

Since this negative feedback amplifier has a configuration of a negative feedback linearizer using a Cartesian loop, a part of the output signal of the power amplifier 8 is fed back by the directional coupler 10 and supplied to the attenuator 14. Attenuator 14 is
Adjust the power level of the input signal to an appropriate value and
Give to 15. The mixer 15 frequency-converts the signal input from the attenuator 14 into an IF frequency using the second LO signal, and supplies the IF frequency to the quadrature demodulator 16.

The quadrature demodulator 16 uses the first LO signal input from the phase shifter 18 to generate I- and Q-component baseband signals.
q and i are output, the I component i is input to the subtraction input side of the adder 2-1 as the I component of the feedback signal, and the Q component q is input to the subtraction input side of the adder 2-2 as the Q component of the feedback signal. Input and negative feedback is applied to each of I and Q.

Next, a method of detecting a phase error will be described with reference to FIGS. 1, 3, 4 and 5. FIG. FIG. 3 is a time chart for explaining the operation of the embodiment of the phase control method of the present invention. FIG. 4 is a diagram for explaining an embodiment of the exclusive OR operation of the phase controller 17 (FIG. 1) of the present invention. FIG.
2 is a detailed block diagram of the phase controller 17 of FIG. First,
When the Q component output from the baseband signal generator 1 in FIG. 1 is a signal as shown in FIG. 3A, when this signal and a reference voltage are input to the comparator 26 (FIG. 5), the signal shown in FIG. Is obtained.

Next, looking at the Q component input Q 'of the quadrature modulator 4, if the phase is delayed, it will be as shown in FIG. 3 (c). When this signal Q 'and the reference voltage are input to the comparator 21 (FIG. 5), the signal of FIG. 3D is obtained. The exclusive OR gate 23 (FIG. 5) takes the exclusive OR of FIG. 3 (b) and FIG. 3 (d) to obtain the signal of FIG. 3 (e).

Next, when the phase advances on the input side of the quadrature modulator 4, the state becomes as shown in FIG. When this signal and the reference voltage are input to the comparator 21, the signal shown in FIG. The exclusive OR of FIG. 3 (b) and FIG. 3 (g) yields the signal of FIG. 3 (h). Here, FIG. 3 (e) and FIG.
Looking at the signal at (h), the delay time (or advance time)
It can be seen that the following data is obtained. The pulse width representing the time lag is counted by the counter 24 as a clock fclk (fclk =
By counting at 2 MHz, the amount of delay can be measured and the phase shift can be corrected.

For example, in the case of a 4 kHz sine wave, if the 4 kHz sine wave deviates by 1 degree (π / 180 radians), the output of the exclusive OR becomes “H” (FIG. 4). Yes)
It is obtained as a pulse with a level of 694 nsec width. The clock that can count 694 nsec is 1.44 MHz (≒ 1 / 0.00000069
4). Therefore, the clock is 2 MHz. 2 at 2 MHz
When counting 50 μsec (4 kHz ≒ 1 / 0.000250), 5
It is necessary to count 00 times (= 200000/4000). Further
Counting two 694 nsec at 2 MHz requires 2.8 counts. Therefore, if the "H" level of the exclusive OR is 2.8 counts or more at 500 counts, the phase is changed to, for example, 1 degree (π / 1
80 radians).

Next, the determination of the phase control direction will be described with reference to FIGS. It is necessary to determine the direction for controlling the phase depending on whether the signal phase is clockwise or counterclockwise on the IQ coordinate. FIG. 6 shows a case where the signal phase is counterclockwise, and FIG. 7 shows a case where the signal phase is clockwise. here,
For example, when the Q component signal changes from-to +, the I component signal
Determines the direction of phase control by determining whether the I component is positive or negative at the zero crossing point of the Q component signal, such as clockwise phase rotation if +, phase rotation counterclockwise if the I component is- I do.

FIG. 6 shows a baseband phase transition of a specific pattern of the π / 4 shift QPSK modulation method.
Shows a constellation when the signal trajectory transitions by + π / 4 radians as shown in FIG. FIG. 6A shows the time waveform of the Q component signal, FIG. 6B shows the time waveform of the Q component at the loop filter output when the phase of the loop is delayed, and FIG. 6 shows the time waveform of the Q component at the output of the loop filter when the phase of the inside advances. Similarly, FIG.
Shows a constellation when the signal locus transitions by -π / 4 radians as shown in FIG. 7D. Fig. 7 (a)
FIG. 7B shows the time waveform of the Q component signal. FIG. 7B shows the time waveform of the Q component at the output of the loop filter when the phase of the loop is delayed, and FIG. 5 shows a time waveform of a Q component at a loop filter output in the case where

When the signal phase is counterclockwise as shown in FIG. 6, the signal at the peak point ta at the reference time t0 as shown in FIG. When the phase in the loop is delayed from the reference time t0 as at the peak point tb, the output of the loop filter has a phase delay because the signal phase is counterclockwise. Also, as shown in FIG. 6 (c), when the phase in the loop advances from the reference time t0 as at the peak point tc,
The output of the loop filter is advanced in phase.

In contrast to the case of FIG. 6, in the case of FIG. 7, the signal phase is clockwise, and therefore, as shown in FIG.
7B, when the phase in the loop advances from the reference time t0 as shown by the peak point tb 'as shown in FIG. 7B, the loop filter output becomes a phase delay. Also, as shown in FIG. 7 (c), the phase in the loop is at the peak point t.
When the time is delayed from the reference time t0 as in c ', the output of the loop filter becomes advanced in phase.

This will be described with reference to FIG. FIG.
FIG. 3 is a block diagram showing an embodiment of the phase controller 17 of the present invention. Reference numerals 31-1, 31-2, and 31-3 denote input terminals, reference numeral 32 denotes a reference voltage input terminal, and reference numerals 21, 22, and 26 denote comparators that use a zero-cross point input from the reference voltage input terminal 32 as a reference voltage. The comparator 21 inputs the Q component signal (Q ′) output from the loop filter 3-2 from the input terminal 31-1, and the comparator 22 receives the I component signal (I) output from the baseband signal generator 1 at the input terminal 31. 2, the comparator 26 inputs the Q component signal (Q) output from the baseband signal generator 1 from the input terminal 31-3, and outputs the respective results. 23 is an exclusive OR circuit, 24 is a counter, 25 is an up / down counter, 27 and 28 are flip-flops, 29 is an inverter, 30 is a switch, 33 is a control unit, 34 is an output terminal, 35
Is a clock input terminal.

First, the operation for obtaining the phase difference will be described. In FIG. 5, the output of the comparator 21 is input to one input terminal of an exclusive OR circuit 23. Comparator 26
The output from the flip-flop 28 and the exclusive OR circuit 23
To the other input terminal. The output of the exclusive OR circuit 23 outputs a logical result as in the operation example described with reference to FIG. 3, and supplies a pulse signal as shown in FIG. The counter 24 counts “H” level pulses with a clock signal fclk (fclk = 2 MHz) that is separately input, and sends the count to the control unit 33 as a phase difference count. As described above, the presence or absence of a phase shift from the reference value is detected instead of measuring the absolute value of the phase difference. And
If the number of counters exceeds a certain value, it is determined that the phase has shifted,
The control unit 33 outputs a control signal for shifting the phase angle of the carrier signal by 1 degree (π / 180 radians). The amount of phase shift (phase adjustment amount) at one time is not limited to 1 degree (π / 180 radians) and may be determined according to the required response characteristics of the system.

Next, the operation for determining the direction in which the phase is shifted will be described. In FIG. 5, the output of the comparator 26 is input not only to the flip-flop 28 and the exclusive OR circuit 23 but also to the flip-flop 27 and the up / down counter 25. The output of the comparator 21 is input to the exclusive-OR circuit 23 and also to the flip-flop 28.
Further, the output of the comparator 22 is input to the flip-flop 28. In the direction of shifting the phase, the flip-flop 27 latches the output of the comparator 21 with the output signal of the comparator 26, and the phase advances according to the state of the I component signal at that time,
Determine the phase lag. The determination information of the phase advance or the phase delay is output from the inverter 29 and the switch 30. When the switch 30 is switched, the flip-flop 28 latches the output of the comparator 22 with the output signal of the comparator 26, thereby determining whether the I component is positive or negative when the Q component is zero-cross, and switches the switch 30. The output of the switch 30 is input to the up / down counter 25, and the input signal is positive (+)
If it is, it is determined that it is late, and if the signal is negative (-), it is determined that it is advanced and sent to the controller 33. The control unit 33 generates a control signal for controlling the phase shift of the phase shifter 18 'based on the two pieces of information "phase difference" and "phase delay or advance".
Output two DC voltages.

Next, the phase shifter 18 'will be described with reference to FIGS.
1 will be described. FIG. 10 is a block diagram showing the configuration of one embodiment of the phase shifter 18 '. The two DC voltages supplied from the phase controller 17 are supplied to mixers 40-1 and 40-2 via input terminals 17-1 and 17-2, respectively. Also, PL
The first LO signal output from the L frequency synthesizer 12 is provided to the mixer 40-2 and the 90 ° phase shifter. The 90 ° phase shifter 42 shifts the phase of the first LO signal by 90 ° (π / 2 radians)
The phase is shifted and given to mixer 40-1. The outputs of the mixers 40-1 and 40-2 are respectively supplied to the adder 41, where the signals are added, and the signals are added to the quadrature demodulator 16.

FIG. 11 is a diagram for explaining the operation principle of the phase shifter 18 '. The horizontal axis represents the I component, and the vertical axis represents the Q component. As shown in FIG. 11, the operation of FIG.
A DC voltage is applied to the component to perform quadrature modulation, and a signal equivalent to an unmodulated carrier signal is output. At this time,
The initial phase angle θ can be varied by the voltages of the I component and the Q component, and an unmodulated carrier can be output at an arbitrary initial phase. That is, the phase shifter 18 'can be replaced by a general quadrature modulator.

In the above-described embodiment, the comparison of the phase difference between the input signal and the feedback signal is performed for the Q component signal. However, it is obvious that the comparison may be performed for the I component signal. While the phase controller 17 of FIG. 5 can be realized by combining a logic gate circuit with a counter and a memory, it can also be implemented by software using a DSP or a microcomputer (not shown). FIGS. 8 and 9 are flowcharts in the case where the operations of the parts other than the comparators 20, 22, and 26 in FIG. 5 are implemented by software. Less than,
This flowchart will be described.

In step 101, the count value K of the counter 24, the initial value φ of the phase angle on the IQ coordinate, and the flag value M indicating whether the baseband Q component has changed from low level to high level. And are set to zero. Next, in step 101, the I component baseband input signal and the output of the comparator 22, the Q component baseband input signal and the output of the comparator 26, the Q component baseband addition signal Q 'and the output of the comparator 21 are fetched.

Next, at step 102, the comparators 21 and 26
Calculate the value of XOR of the output of. Step 103
It is determined whether X is 1 or not. If X is 1, step
Proceed to 104, and return to step 101 if X is not 1. At step 104, the value K of the phase difference counter (corresponding to the counter 24) is calculated.
Increment by one. In step 105, it is determined whether or not the count value K has exceeded the reference value L set for the phase shift difference counter. If the count value K has exceeded the reference value L, the process proceeds to step 106; otherwise, the process returns to step 101.

In step 106, the DC voltage values of the I component and the Q component corresponding to the phase shift set value φ determined in steps 110 to 126 described later are read from a memory (not shown) and output as analog signals. (Equivalent to the operation of the control unit 33). That is, for example, for a ROM
The values of the I component voltage and the Q component voltage corresponding to the phase value for each range of the φ value from 1 to 360 ° are stored. Then, the phase shifter 18 'is controlled by the voltage value determined in step 106, and the phase shift is reset in step 107.

On the other hand, the detection flow of the phase shift, the advance and the delay will be further described. In step 110, it is determined whether or not the flag value M is at a high level (1). If the flag M is at high level (indicating that the previous Q component is at low level), then at step 111, the Q component baseband input signal
Determine if Q is high level. If the Q value is high, the flag M is reset to 0 in step 112. If the flag M is low at step 110 (indicating that the previous Q component is at high level), then step 113
To determine if the Q value is low. If the Q value is low, the flag M is set to 1 in step 114. If the Q value is not low, the process returns to step 101 and repeats the above steps. The operation so far corresponds to the clock operation of the flip-flops 27 and 28.

Next, at step 115, it is determined whether the value of the I component baseband input signal I is high. If the I value is high, it is determined in step 116 whether the Q component baseband addition signal Q 'is high. If it is determined in step 115 that the I value is not at the high level, it is determined in step 117 whether the Q component baseband addition signal Q ′ is high. Step 110
117 correspond to the operations of the flip-flops 27 and 28, the inverter 29 and the switch 30.

Next, at step 120, it is determined whether or not the count value N is larger than zero. If the count value N is larger than 0, it is a phase shift delay, so that a predetermined phase shift amount △ (delta) is added to the current phase shift set value φ in step 121. If the count value N is not larger than 0, the phase is advanced, so that in step 122, a predetermined phase shift amount △ is subtracted from the current phase set value φ.

Next, in step 123, it is determined whether the updated phase shift set value φ is equal to or greater than 360 °. If φ is equal to or greater than 360 °, φ-360 is calculated in step 124 and the result is set as a new φ. If it is determined in step 123 that φ is not 360 ° or more, it is determined in step 125 whether φ is smaller than 0. If φ is smaller than 0, φ-360 is calculated in step 126, and the result is set as a new φ. The determined φ value is used in step 106.

In the embodiment described above, the Q component Q of the input signal and
The phase difference was compared using the Q component Q ′ of the feedback signal, and the rotation direction was obtained using the I component I and Q component Q of the input signal and the Q component Q ′ of the feedback signal. Because of the feedback signal
The I component I 'of the feedback signal may be used in place of the Q component Q', and at that time, the I component I and the Q component Q of the input signal and the I component I 'of the feedback signal are obtained in order to determine the rotation direction. It is obvious that may be used.

[0040]

As described above, according to the present invention, the phase control can be performed by observing the Q component and the I component of the baseband signal. Further, since the phase control can be automatically performed during the normal operation, no special test signal is required for the phase adjustment. Furthermore, accurate phase control can be performed by eliminating the need for switching means between the open loop and the closed loop. Therefore, it is possible to follow the phase change even if the phase in the loop changes due to temperature fluctuation, temporal change, etc. without opening the Cartesian loop and to stably operate the phase characteristic and stabilize the output operation characteristic. And the spurious characteristics can be prevented from deteriorating.

[Brief description of the drawings]

FIG. 1 is a block diagram showing a configuration of an embodiment of a negative feedback circuit of a transmitter according to the present invention.

FIG. 2 is a block diagram showing a configuration of a conventional negative feedback circuit.

FIG. 3 is a time chart for explaining the operation of an embodiment of the phase control method of the present invention.

FIG. 4 is a view for explaining an embodiment of an exclusive OR operation of the phase controller of the present invention.

FIG. 5 is a block diagram showing an embodiment of a phase controller according to the present invention.

FIG. 6 is an explanatory diagram of an operation of the phase controller according to one embodiment of the present invention.

FIG. 7 is an explanatory diagram of the operation of the phase controller according to one embodiment of the present invention.

FIG. 8 is a part of a flowchart in a case where the phase controller of the present invention is implemented by software.

FIG. 9 is a part of a flowchart when the phase controller of the present invention is implemented by software.

FIG. 10 is a block diagram showing a configuration of a specific example of the phase shifter in FIG. 1;

11 is a diagram illustrating I for explaining the operation of the phase shifter in FIG. 10;
-Q coordinate diagram.

[Explanation of symbols]

1: Baseband signal generator, 2: Adder, 3: Loop filter, 4: Quadrature modulator, 5, 7: Bandpass filter, 6, 15: Mixer, 8: Power amplifier, 9: Antenna, 10: Direction Sex coupler, 11: reference signal generator,
12, 13: PLL frequency synthesizer, 14: attenuator, 16: quadrature demodulator, 17, 19: phase controller, 18,
18 ': phase shifter, 20, 30: switch, 21, 22, 26: comparator, 23: exclusive OR circuit, 24: counter, 25: up / down counter, 27, 28: flip-flop, 29: inverter, 30: switch, 31-1,
31-2, 31-3: Input terminal, 32: Reference voltage input terminal, 33:
Control unit, 34: output terminal, 35: clock input terminal.

Claims (4)

[Claims] 1. An input in-phase component and an input quadrature component of a baseband signal are input, the baseband signal is quadrature-modulated by a local oscillation signal frequency, and the quadrature-modulated signal is amplified to a predetermined power level and transmitted. A transmitter, comprising: a branching unit that branches a part of an output signal of the transmitter, and a part of the output signal that is branched by the branching unit, by converting a feedback in-phase component and a feedback quadrature component according to the local oscillation signal frequency. By providing quadrature demodulation means for performing quadrature demodulation, and adding means for adding the feedback in-phase component, the input in-phase component, and the feedback quadrature component and the input quadrature component, respectively,
A nonlinear distortion compensating circuit based on a negative feedback method for compensating for nonlinear distortion of the Q component Q ′, wherein a phase comparing unit that compares a phase of the input in-phase component and a phase of the input quadrature component with a phase of a signal added by the adding unit; And control means for controlling the phase of the local oscillation signal input to the quadrature demodulator based on the information compared by the phase comparison means. A transmitter that stably operates output operation characteristics by performing the following. 2. An input in-phase component and an input quadrature component of a baseband signal are input, the baseband signal is quadrature-modulated by a local oscillation signal frequency, and the quadrature-modulated signal is amplified to a predetermined power level and transmitted. A transmitter, comprising: a branching unit that branches a part of an output signal of the transmitter, and a part of the output signal that is branched by the branching unit, by converting a feedback in-phase component and a feedback quadrature component according to the local oscillation signal frequency. By providing quadrature demodulation means for performing quadrature demodulation, and adding means for adding the feedback in-phase component, the input in-phase component, and the feedback quadrature component and the input quadrature component, respectively,
In a nonlinear distortion compensation circuit using a negative feedback method for compensating for nonlinear distortion of the transmitter, a phase shifter that shifts a phase of the local oscillation signal frequency input to the quadrature demodulator, and the input quadrature component or the in-phase component. A phase control unit that compares the quadrature component or the in-phase component added by the adding unit and outputs phase control information for phase shifting by the phase shifting unit. The input in-phase component and the feedback in-phase A transmitter for adjusting a phase difference between the input quadrature component and a phase difference between the input quadrature component and the feedback quadrature component. 3. The transmitter according to claim 1, wherein said phase control means and said phase shift means are performed in a closed loop. 4. The transmitter according to claim 1, wherein phase control is performed by detecting a phase rotation amount from the input baseband signal and a signal input to a quadrature modulator.