JP2001168838A - Spread spectrum communication method - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a spread spectrum communication method, where a transmitter side uses carrier frequencies apart from each other by an integer multiple of a frequency specified at a data symbol speed to multiplex a plurality of spread spectrum signals and a receiver side avoids and suppresses the effects of an interference signal caused by an adjacent spread spectrum signal and the effect of jitter or the like caused by distortion produced in a transmission line, so as to enhance timing acquisition and tracking accuracy of the receiver side and decrease a timing acquisition time, thereby enhancing a transmission efficiency. SOLUTION: This method includes a step where symbol timing for data discrimination is acquired and traced in the unit of time width, consisting of a data symbol time length by N symbols (N is an arbitrary natural number), with respect to a spread multiplexed signal at the receiver side.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a direct sequence (DS) -spread spectrum (SS).
m) All wireless communication systems such as a fixed satellite communication system, a mobile satellite communication system, a fixed terrestrial radio communication system, a land mobile communication system, a wireless LAN system, a private radio communication system, etc., to which a communication system is applied, or an optical fiber The present invention is applied to all wired communication systems that perform information transmission using a wired cable such as a coaxial cable. It is also used in a system that applies a four-phase phase modulation (QPSK) method as a modulation method for a data signal to be subjected to spread spectrum processing and a differential detection method as a demodulation method.

[0002]

2. Description of the Related Art As a method of increasing the speed of a spread spectrum communication system within a limited communication bandwidth, a method of performing multiple access using only one pseudo random code commonly provided to all simultaneous communication stations. (CFO-SSMA method: Carrier Frequency Offse
t-Spread Spectrum Multiplil
eAccess Method).
68189, Sumitani, Shinonaga, "Spread Spectrum Communication System"). In this method, a plurality of transmitting stations perform direct spread spectrum modulation of independent digital information using the same pseudo-random code to make the center frequencies of a plurality of carriers different from each other as shown in FIG. It is characterized in that the bands are set to overlap each other and transmitted to the receiving side. At this time, if each transmitting station uses a carrier wave frequency separated from each other by an integer multiple of the information transmission rate, the receiving side extracts a desired wave having a known center frequency of the carrier with a band-pass filter. It is ideally possible to receive and demodulate desired information without being affected by signal waves.

FIG. 15 shows an example of the configuration of a transceiver in a radio system to which the CFO-SSMA system is applied, and shows a case where the number of simultaneous transmission stations is n. In the figure, 301-1-30
1-n is a binary coded information signal of each communication station, 302
-1-302-n is a modulator for performing binary multiplication of an information signal and a pseudo-random code, and 303-1-303-n is a pseudo-random (PN: P) for spreading the spectrum of the information signal.
pseudo Noise) code sequence, 304-1-30
4-n is a modulator that modulates a carrier wave given by 305-1-305-n with respect to the baseband signal obtained at 302-1-302-n, and 305-1-305-n is a local unit that gives a carrier wave. The oscillators 306-1 to 306-n respectively represent band-pass filters for extracting frequency components necessary for transmission from the modulated signal. The spread spectrum communication method can be used for both a wired communication system and a wireless communication system. Reference numeral 307 denotes a state in which individual signals are multiplexed in these transmission media. Reference numeral 308 denotes a band-pass filter for extracting a frequency component necessary for demodulation from the received signal, 309 denotes a frequency converter, 310 denotes a local oscillator for frequency conversion, and 311 denotes information transmitted from the band-limited signal. , 312 respectively represent information signals obtained by the demodulator.

As described above, in the CFO-SSMA system, communication is performed with the center frequencies of signal waves separated from each other by an integral multiple of the information transmission rate. Therefore, even when the same pseudo random code is used in a spread spectrum communication system, Each communication station can perform bidirectional communication without causing interference with each other. Therefore, even in a system in which the number of pseudo-random codes is limited, the pseudo-random codes can be reused by shifting the center frequency of the signal wave from each transmitting station by an integral multiple of the information transmission rate. The number of stations capable of performing simultaneous communication within the given frequency bandwidth can be greatly increased. Further, since each communication station uses the same pseudo-random code, the size of the communication device can be reduced. However, in the CFO-SSMA system, ideally interference-free communication cannot be performed unless the timings of the directly spread spectrum modulated wave groups simultaneously transmitted from the respective transmitting stations coincide with each other on the receiving side. That is, if the direct spread spectrum modulated waves from each transmitting station are not received in a synchronized state on the receiving side, the signals from each transmitting station cause interference with each other and greatly degrade the line quality of the transmission path. Exists.

As a method of overcoming the above-mentioned problem, a plurality of communication channels in which a carrier frequency interval is an integral multiple of a frequency amount in units of an information transmission rate are regarded as one composite communication channel, and each communication station issues a composite communication channel. By performing information transmission using a communication channel, without generating a timing control error between communication channels in which the carrier frequency interval is an integral multiple of the frequency amount in units of the information transmission speed,
A method (CFO-SS method: Carr) using a communication channel multiplexing method capable of increasing the information transmission speed per communication station and providing a high quality communication line.
ier Frequency Offset-Spre
ad Spectrum Multiple Access
ss Method) is disclosed (JP-A-10-10777).
1. Ishikawa, Shinonaga, Kobayashi "Spread spectrum communication method")
Have been.

FIG. 16 shows an example of a transmitter configuration of a radio system to which the CFO-SS system is applied. In the case where the number of communication channels forming a composite communication channel is n, that is, the number of channels performing simultaneous transmission is n. Is shown. In the figure, reference numeral 401 denotes an information data sequence transmitted from a communication station;
Reference numeral 2 denotes a serial-parallel converter for sequentially and repeatedly distributing a serial data sequence to n output terminals.
403-n is an information data sequence modulated for each communication channel, and 404-1-404-n is an information signal and 406-1-406-n output from a 405 PN code generator.
Modulator that performs a binary multiplication with a pseudo-random code of
1-406-n is a pseudo-random code sequence for spreading the spectrum of an information signal, 407-1-407-n is a spectrum-spread baseband signal transmitted for each communication channel, and 408-1-408- n is 404
-1-404-n the baseband signal 407-
1-407-n is set to the carrier frequency f1-
modulator for modulating according to fn, 409-1-409
-N is a local oscillator that gives a carrier frequency f1-fn of each communication channel, 410-1-410-n is a spread spectrum signal frequency-converted to a high frequency region by a modulation operation, and 411-1-411-n is a modulation signal. Bandpass filter for extracting frequency components necessary for transmission, 412-1
-412-n is a spread spectrum signal output from each band-pass filter, 413 is a signal combiner that combines the spread spectrum signals 412-1-412-n of each communication channel, and 414 is an output signal of the signal combiner 413. A certain spread spectrum combined signal, 415 is a common amplifier for amplifying the transmission power of the spread spectrum combined signal, 416 is a spread spectrum combined signal power-amplified by the common amplifier 415, 417 is a band limit for eliminating radiation outside the communication bandwidth. Filters 418 represent output signals of the band limiting filter 417, respectively. The spread spectrum communication method can be used in both a wired communication system and a wireless communication system, and the spread spectrum combined signal 418 is transmitted to another communication station via those transmission media.

FIG. 17 shows an example of the configuration of a receiver of a radio system to which the CFO-SS system is applied. The case where the number of communication channels forming a composite communication channel is n, that is, the number of channels performing simultaneous transmission is n is shown. Is shown. In the figure, 421 is a composite communication channel reception signal transmitted from another communication station, 422 is a reception filter for noise removal, 42
3 is an output signal of the reception filter 422, and 424 is an automatic gain controller (AG) for operating the demodulator in a stable state.
C), 425 are AGC output signals, 426-1-426
-N indicates the carrier frequency f0-fn of each communication channel constituting the composite communication channel by the matched filter 431-
A frequency converter 427-1-427-n for converting the frequency to a center frequency f0 of 1-431-n is a local oscillator for generating a frequency corresponding to the center frequency f1-fn of each communication channel. Reference numeral 428-n denotes an intermediate frequency reception signal of each frequency-converted communication channel.
429-n is a band-pass filter for removing a noise component existing outside the frequency spreading bandwidth.
-N is the output signal of the bandpass filter, 431-1-43
1-n is a matched filter that extracts only the information signal component from the spread spectrum signal, 432-1-432-n is a SAW matched filter output signal for each communication channel, 433
-1-433-n is a delay detection circuit for converting a SAW matched filter output signal into a baseband signal;
1-234-n is the output signal of the differential detection circuit of 433-1-433-n, 435-1-435-n removes the harmonic signal component contained in the differential detection output signal, and only the information signal component 436-a low-pass filter for extracting
1-436-n is a baseband signal including an information signal component, 437-1-437-n is a determiner for determining a data signal at a peak point (determination point) of the differential detection output signal, and 438-1-438. -N is a judgment data sequence obtained for each communication channel, 439 is 438
A parallel-serial converter 440 for converting the judgment data sequence of -1-438-n into the original continuous data sequence
Represents information data sequences output from the parallel-serial converter 439.

In general, a matched filter weighted by a spread code sequence is used as a method for extracting an information signal component from a spread spectrum signal, and its output waveform is a sharp triangular pulse-like intermediate frequency signal. Here, the envelope signal of the matched filter output signal or the peak point of the delay detection output waveform obtained by performing the delay detection on the matched filter output signal is detected, and the data determination is performed at the same timing as that point. If the timing for performing data determination deviates from the peak voltage level point of the desired reception signal, the determination error rate characteristic of the received signal is significantly deteriorated. Therefore, immediately after data transmission is started from the transmission side, it is necessary for the reception side to capture the optimum determination timing at which the determination error rate characteristic of the received signal is the best, and establish timing synchronization between the reception side and the transmission side. Further, in order to continuously perform data transmission, a tracking function for maintaining timing synchronization between the receiving side and the transmitting side is required separately.

As a timing acquisition / tracking method for a conventional spread spectrum communication system, it is presumed that a desired signal is received at a moment when a voltage level of a reception signal exceeds a predetermined threshold value. A method of defining the time as the determination timing is known as a simple system. However, in such a method, even when the received signal level exceeds the threshold value due to the influence of noise, an interference signal from an adjacent channel, or the like, it is erroneously determined that the desired signal pulse has been received, and the erroneous determination is made. The inserted data is inserted into the true information data series,
There arises a problem that all the data after the erroneous determination is erroneous in a burst.

In order to avoid the above-mentioned problem, a method of setting a time window having the same periodicity as the transmission rate and detecting a received signal pulse exceeding a threshold value only in the time window section (time window method) (Reference: Tanaka, Ogawa, Kinugasa, Coconut palm, Takei, "Power-saving SS
Development of Communication LSI ", Institute of Electronics, Information and Communication Engineers, Spread Spectrum Research Group, SST95-77, etc.). In this method, as shown in FIG. 13, the sampling point at which the desired wave pulse signal exceeds the threshold value set in advance in the time window section and becomes the maximum is determined as the optimum determination timing, and the time is set as a new time. It is configured to be set as the center point of the window. By setting the time window in this manner, the probability of erroneous detection of a signal pulse generated due to the influence of noise, an interference signal from an adjacent channel, or the like can be significantly reduced. However, in the above-mentioned time window method, although it becomes effective only after the symbol timing of the desired wave pulse signal is captured, the timing detection is usually performed by trial and error in the timing capture preamble data section added to the head of the transmission data signal. By doing
Initial timing capture is easily possible.

[0011]

The above-mentioned CFO-SS
In the MA system and the CFO-SS system, the output waveform of the matched filter and the output waveform of the delay detector are used to multiplex a plurality of spread spectrum signals and transmit them simultaneously, as shown in FIG. Large waveform fluctuations occur in addition to the peak points. That is, CFO-SSMA
In the system and the CFO-SS system, the orthogonal relationship is established between a plurality of spread spectrum signals to be multiplexed with respect to a peak point of a desired wave serving as a data determination timing.
Although they do not interfere with each other, the orthogonality relationship does not hold except at the peak point of the desired wave, so that an interference signal component from another spread spectrum signal is specifically generated as a waveform level. Therefore, when the conventional timing acquisition and tracking method is applied to the CFO-SSMA method and the CFO-SS method, it is impossible to determine which of the desired wave peak and the interference peak is the true determination timing, and the interference signal There is a high possibility that the peak level due to is erroneously captured as the determination timing. In this case, since the time window itself is set around the erroneous determination timing, a continuous error occurs.

Further, in the case of a timing tracking method in which a peak level which becomes maximum within a time window is detected and the time window is reset for each symbol centering on the point, in the case of a band limitation of a transmission system filter or a transmission line distortion ( The peak point of the desired wave pulse signal moves back and forth on the time axis due to the influence of noise (multiple reflected waves) and noise (jitter is generated). Therefore, between the center point of the time window and the peak point of the desired wave pulse signal. Are accumulated over time. for that reason,
Depending on the state of the distortion of the transmission line, there are many cases where the desired wave peak pulse signal easily deviates from the time window, so that the timing tracking state cannot be maintained, and the determination error of the received data occurs in a burst.

[0013] The present invention relates to a spread spectrum communication system for multiplexing a plurality of spread spectrum signals using carrier frequencies separated from each other by an integral multiple of a frequency amount defined by a data symbol rate, and acquires and tracks timing on a receiving side. The present invention relates to a timing acquisition / tracking method for improving accuracy and avoiding the influence of jitter generated due to transmission path distortion.

[0014]

In order to achieve this object, a spectrum communication method according to the present invention uses a time width consisting of a data symbol time length of N symbols for a spread multiplexed signal on the receiving side. As a unit, there is a step of capturing and tracking a symbol timing for data determination.

According to another embodiment of the present invention, on the transmitting side, the step of modulating uses a phase modulation method, and on the receiving side, before the step of capturing and tracking the symbol timing, spread multiplexing is performed. It is preferable that the method further includes a step of demodulating the signal by a delay detection method.

According to another embodiment of the present invention, the average value of the absolute values of the differential detection output signal levels for the received data for N symbols is set, using the data symbol length and the same time length of the information data sequence as the calculation cycle. Alternatively, the addition value is calculated continuously over one symbol time length, and is significant when the average value or the maximum value of the addition value exceeds the threshold value Th1 set in advance consecutively A times in blocks of N symbol time lengths. It is determined that a proper received signal exists. Next, starting from the time when it is determined that the received signal exists, the time position indicating the latest maximum value at that time is assumed to be the determination timing, and the calculation result at the determination timing position is set in advance. Set threshold level Th2 to N
When the number of consecutive blocks exceeds the symbol time length in B units, it is determined that the determination timing has been correctly captured. Next, after the time at which it is determined that the determination timing has been correctly captured, a time window centering on the latest determination timing is set for each block of N symbol time lengths. The average value of the absolute values of the differential detection output signal levels obtained within the time window or the maximum value of the added value is:
When the threshold value exceeds a preset threshold level Th3, the time position indicating the maximum value is determined as the optimal determination timing, and the determination timing is changed. When the maximum value falls below the threshold level Th3, When the same time position as the previous time is continuously used as the determination timing, and the maximum value falls below the threshold level Th3 successively C times in blocks of N symbol time lengths, the reception of the spread multiplexed signal ends. And judge. Thereafter, it is configured to return to the first stage.

According to another embodiment of the present invention, the received signal is sampled at a sampling rate S (S = R × M) M times the data symbol rate R (M is an arbitrary natural number), and M sampling numbers are sequentially assigned to the sampling points within the data symbol time length, and all of the arithmetic processing and determination processing performed in units of N symbol time length blocks are performed at the sampling points assigned the sampling numbers. Is also preferable.

According to another embodiment of the present invention, the sampling rate R is set to M / L (L is an arbitrary natural number satisfying M> L).
The sampling rate of the received signal is sampled at a sampling rate T (T = R × M / L) that is M / L times the data symbol rate R, and the sampling output signal of the received signal is L-branched. , And the sampling interval 1 / T is divided into L, and the time length 1 / T
A delay circuit having (T × L) as a minimum unit of the delay amount is arranged for each of the parallel reception sampling signals, and the parallel reception sampling signals are mutually shifted by a time length of 1 / (T × L) on the time axis. Form a relationship that offsets each other. Next, M sampling numbers are sequentially assigned to the sampling points within an arbitrary data symbol time length to the L parallel reception sampling signals, and the calculation is performed in N symbol time length block units. It is also preferable that all of the processing and the determination processing be performed on the sampling points to which the sampling numbers are assigned.

According to the spread spectrum communication method of the present invention, a receiving side receives a spread signal having a center frequency at a specific carrier frequency among a plurality of spread signals of a spread multiplexed signal as preamble data for timing acquisition. Setting a preamble data to be assigned to each spread signal in advance so that the signal waveform becomes a demodulated waveform suitable for timing reproduction.

According to another embodiment of the present invention, the step of modulating the transmitting side uses a phase modulation method, and the step of setting the preamble data on the receiving side delays the spread multiplexed signal before the step of setting the preamble data. It is preferable that the method further includes a step of demodulating by a detection method.

According to another embodiment of the present invention, the phase modulation method is such that the phase difference change amounts of 0, π / 2, π, and 3π / 2 are respectively assigned codes of 00, 01, 11, and 10. This is a QPSK system to which dynamic encoding is applied. For H spread signals (H = 2m + 1, m is an arbitrary positive integer) transmitted simultaneously, preambles of spread signals from the lowest carrier frequency to the m spread signals are used. A repetition pattern of 11 codes is set as data, and a repetition pattern of 01 is set as preamble data of the remaining (m + 1) spread signals, so that the (m + 1) -th carrier wave frequency starts from the lowest frequency. It is also preferable that the absolute value of the differential detection output signal of the spread signal becomes a waveform suitable for timing reproduction.

In a communication system to which the CFO-SSMA system and the CFO-SS system are applied as in the system of the present invention, the absolute value of the differential detection output signal level is determined in units of a time width consisting of a data symbol time length of N symbols. By performing a value averaging process (or an addition process), it is possible to suppress a severe waveform fluctuation in a side lobe section generated by an interference signal from an adjacent spread spectrum signal, and to extract only a desired wave peak level as an effective waveform. Can be. That is, since the eye is always open at the desired wave peak point regardless of the information data sequence pattern to be transmitted, the calculation result at the desired wave peak point is ideally constant irrespective of the value of the number of symbols N to be averaged. However, the waveform fluctuates significantly in other time sections, so that as the value of the number N of symbols subjected to the averaging process increases, the waveform fluctuation is suppressed and the average level becomes smaller than the peak level of the desired wave. Become. Therefore, when the maximum value of the output waveform after the averaging process is detected, the desired wave peak point can be easily and accurately detected as the data determination timing.

In addition, a transition is made between four types of states, such as a reception signal detection mode, a timing acquisition mode, a timing tracking mode, and a reception standby mode, by setting an independent protection stage number and a reception level threshold value in advance, respectively, and In the tracking mode, a time window is set, the maximum value is detected, and the center position of the time window is updated every N symbol blocks, so that the band limitation of the transmission system filter, transmission line distortion (multipath fading), noise Thus, it is possible to absorb the influence of the time variation (jitter) of the desired wave pulse signal generated by the influence of the above, and to always maintain the accurate timing tracking state. As a result, it is possible to avoid the influence of the burst error, which has been a problem in the conventional timing acquisition and tracking method, and to provide a stable communication line at all times.

Further, when there is a limit to the processing capacity and power consumption of the digital circuit, the sampling frequency is reduced, a delay circuit having a time width smaller than the minimum sampling interval is separately prepared, and the same parallel signal is supplied to the branch circuit. By inserting an independent delay amount into the sampling signal group converted to the above, and applying the above-described timing acquisition and tracking method, it is possible to perform timing acquisition and timing tracking with the same accuracy as at the time of high-speed sampling. Can be.

Furthermore, a specific code pattern is set in advance for each spread spectrum signal as a preamble pattern for timing acquisition, and is simultaneously transmitted.
In a differential detection output waveform of a spread spectrum signal having a certain carrier frequency as a center frequency, it is possible to suppress an interference signal component from another spread spectrum signal group generated in a time section (side lobe section) other than a desired wave peak point. it can. When the timing acquisition / tracking method of the present invention is applied, the timing acquisition time can be significantly reduced by the suppression effect of the side lobe section, that is, the number of preamble data symbols can be reduced. The time width to be allocated can be shortened, and the throughput characteristics of the communication line can be improved. Further, since the detection accuracy of the desired wave peak point can be improved, the probability of tracking an erroneous timing can be significantly reduced, and the frequency of occurrence of burst errors can be reduced.

[0026]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS First, an embodiment of a timing acquisition / following method and a preamble data setting method of a spread spectrum communication method according to the present invention will be described below.

As shown in FIG. 14, in a radio system to which the CFO-SSMA system and the CFO-SS system are applied, a plurality of spread spectrum signals are simultaneously transmitted.
In the matched filter output of the receiver, a correlation interference pulse of the adjacent spread spectrum signal other than the level peak point 221 of the desired wave pulse signal is randomly generated in the side lobe section 222. For the main purpose of avoiding the influence of the interference wave pulse signal, in the method of the present invention, as shown in FIG. 1, all the calculation / determination processes are performed in units of 5 N symbol blocks (if N = 1, every symbol). I do. This N
The operation on a symbol block basis is performed over the entire one packet frame section 4 including the preamble section 1, the unique word symbol section 2, and the data symbol section 3. By this operation, the interference wave pulse signal generated in the side lobe section 222 other than the level peak point 221 of the desired wave signal shown in FIG. 14 is suppressed, and the probability that the interference wave pulse is erroneously captured as the desired wave pulse at the time of timing acquisition is reduced. Can be reduced. Furthermore, it is also possible to equalize the peak level fluctuation of the desired wave pulse that fluctuates due to the influence of noise, the band limitation of the transmission signal, and multipath,
It is possible to improve the detection accuracy of the optimum sampling point (determination timing). Hereinafter, the invention system will be specifically described. In the embodiment, the data symbol rate is 1M symbol / sec, and the sampling frequency is 44M.
Hz. At this time, 44 samplings are performed per symbol.

FIG. 2 shows the relationship between the sampling number and the received signal on the time axis, where 11 is the length of one symbol section, 12 is the correlation peak level of the desired signal, and 15 is an arbitrary time 13 at the beginning. Each of the symbol periods indicates a sampling number that is repeatedly assigned. Here, the beginning of the sampling number is set to an arbitrary time 13, and thereafter, sampling is performed at a speed of 44 MHz, and 14 sampling numbers 0 to 43 are sequentially assigned to each sampling value. Next, the reception level value at each sampling number is calculated over N symbol blocks 5, and the timing is determined by judging whether or not the maximum value of the average value (any one of the numbers 0 to 43) exceeds a preset threshold level. Acquisition / following is performed.

Next, FIGS. 3 and 4 show a sampling frequency reduction method and an optimum sampling point selection method according to the present invention. FIG. 3 shows the difference between the sampling position and the sampling number between the 44 MHz sampling and the 22 MHz sampling, where 21 is one symbol section length, 22 is the correlation peak level of the desired signal,
5, a symbol period starting at an arbitrary time, 23, a sampling number when the sampling frequency is 44 MHz, and 24, a sampling number when the sampling frequency is 22 MHz. Here, the starting point of the sampling number is the same time, and the even-numbered sampling number at the time of 44 MHz sampling is 22 MHz.
It is the same as the sampling number at the time of sampling.

FIG. 4 is a conceptual diagram of an optimum sampling point selecting method according to the present invention, and shows an embodiment in which a received signal is split into two. In the figure,
The reception signal 31 is a reception signal 32 that is not subjected to delay processing,
The signal is branched into a reception signal 33 that has passed through a delay circuit of Ts / 2 (Ts is a sampling period). For each of the received signals, the absolute value average over N symbol blocks is calculated at each sampling point of the sampling numbers 34 and 35 independently, and the maximum values 36 and 37 are detected. Next, the levels of the maximum values 36 and 37 are compared, and the sampling number indicating the higher level is determined as the optimum sampling point. As described above, the reception signal is branched into L, L delay circuits are set with 1 / L of the sampling period as the minimum time amount, and the maximum value of the average of the absolute values is detected. The sampling frequency can be reduced while maintaining the sampling accuracy of.

FIG. 5-9 shows an embodiment of the timing acquisition / tracking method according to the present invention, which includes a carrier detection mode for judging the presence / absence of a received signal, a timing acquisition mode for detecting a judgment timing from the time of carrier detection, and a detection mode. It consists of a timing tracking mode for tracking the temporal variation of the determined timing. First, a transition procedure from the carrier detection mode to the timing acquisition mode is shown in FIG. 5 as a state transition diagram. In the figure, starting from a carrier detection mode initial state 41, the maximum absolute value average over N symbols at an arbitrary sampling number exceeds a preset threshold level Th1 and carrier detection is determined, and the number of backward protection stages is determined. 45-1-45-
A consecutive carrier detections 43-1-43 corresponding to A
When −A continues, the mode shifts to the timing acquisition mode 62. On the other hand, during the state transition, carrier non-detection 44-0-
When 44- (A-1) is confirmed, the process returns to the carrier detection mode initial state 41, and the counter value is reset. By adopting this method, it is possible to avoid the influence of the level fluctuation due to AGC occurring at the head of the preamble section. Here, for the A time, a sampling number indicating the maximum value of the average of the absolute values of the sampling numbers calculated over the N symbol blocks is detected, and that point is set as the determination timing.

FIG. 6 is a state transition diagram showing a procedure for shifting from the timing acquisition mode to the timing tracking mode. In the figure, starting from the timing acquisition initial state 51, the sampling number at which the maximum value of the absolute value average determined at the A-th time in the carrier detection mode is obtained corresponds to the backward protection stage number 55-1-55-B. If the average of the absolute values exceeds the threshold level Th2 consecutively (53-1-53-B), it is determined that the symbol timing has been captured, and the mode shifts to the timing following mode 52. When the maximum value of the average of the absolute values falls below the threshold level Th2 during the state transition (54-0-54).
-(B-1)), the process returns to the initial state 51 of the timing capture mode, and resets the counter value. By employing this method, the accuracy of the determination timing can be improved.

If the decision timing is determined only in the preamble section, it is not possible to cope with fluctuations in the propagation path characteristics due to multipath and noise, and fluctuations in the optimum decision timing due to the fluctuations in the frequency of the oscillator. Therefore, the timing tracking function according to the present invention is indispensable. Becomes FIG. 7 shows a method of setting the time window 72 in the timing tracking mode. The time window 72 is centered on the determination timing 77 and is a front window sampling number FF (74) and a rear window sampling number BB (7).
3). Among the sampling numbers 4-6 included in the time window 72, the absolute value average is equal to the maximum level 71.
Is selected as the determination timing. Note that the center point of the time window is updated so as to coincide with the optimum determination timing obtained every N symbols. Further, in the timing tracking mode, by performing the arithmetic processing only in the section of the time window 72, it is possible to reduce the amount of arithmetic processing, power consumption, and the like.

Next, FIG. 8 shows a state transition in the timing tracking mode. As shown in the figure, in the timing tracking mode 83, the number of C front protection steps from 87-1 to 87-C is prepared, and each time the maximum value of the average of the absolute values falls below a preset threshold level Th3. To
85-1-85-C, the front protection stage number counter is 1
It is counted up one by one. On the other hand, when the maximum value of the average of the absolute values exceeds the threshold level Th3, the forward protection stage number counter becomes 1 as shown in 84-1-84- (C-1).
The counter is counted down one by one and returns to the stable state 84-0, but when the value falls again, the counter counts up by one. Thereafter, when this operation is repeated and the counter reaches the protection stage number C, it is determined that the line state is bad, and the state shifts to the carrier detection mode initial state 81.

Finally, FIG. 9 is a state transition diagram showing a method of detecting a frame end by recognizing the moment when the state changes from the carrier detection state to the carrier non-detection state. If the line quality is extremely deteriorated due to the influence of shadowing or the like, the mode shifts to the carrier detection mode initial state 91, and shifts to the timing tracking mode 94 via the timing acquisition mode 93. It is necessary to always operate the carrier detection mode. As shown in FIG. 9, 97-
Considering the number of forward protection stages of D stages up to 1-97-D, when no carrier is detected, the forward protection stage number counter is set to 96-1-9.
The count is incremented by one as shown in 6-D. Here, every time a carrier is detected, the front protection stage number counter is counted down one by one and returns to a stable state. However, when no carrier is detected, the counter is further incremented by one. Thereafter, when this operation is repeated and the counter reaches the protection stage number D, it is determined that the frame is end (no carrier input state), and the operation shifts to the initial state of the carrier detection mode.

Next, an embodiment of a method for setting preamble data according to the present invention will be described. A case where five spread spectrum signal waves are multiplexed will be described as an example. In FIG. 14, since the data of the spread spectrum signal wave to be multiplexed is independent and random, a large interference wave level occurs in the side lobe section 222. However, the data pattern transmitted by each spread spectrum signal wave is By using a certain combination of sequences, the interference level in the side lobe section can be suppressed.

FIG. 10 shows preamble data transmitted by each spread spectrum signal, spread spectrum signal 1: 11111111... Spread spectrum signal 2: 11111111... Spread spectrum signal 3: 01010101. 01010101 ... Spread spectrum signal 5: This represents the eye diagram characteristics when set as 0101101 .... Here, the spread spectrum signal 1-5 is
They are arranged in ascending order of carrier frequency, and are 11-bit sequence of Barker code as a spread code sequence, and 00, 01, 11, and 10 for phase difference change amounts 0, π / 2, π, and 3π / 2 as a phase modulation method. QPSK system with differential encoding to assign codes, transmission speed of 2 Mbit / s,
FIG. 9 shows the in-phase components of the differential detection output waveform when the carrier frequency interval is 2 MHz. Also,
The abscissa in the figure indicates the normalized symbol time 103, and the ordinate indicates the absolute value level 104 of the differential detection output signal. In the figure, it can be confirmed that in the side lobe section 102 other than the peak level 101 of the desired signal, the interference wave level is suppressed and the level fluctuation is reduced. As a result, a quick timing acquisition time in the preamble section becomes possible, and the preamble section can be shortened to improve the throughput characteristics.

Finally, FIG. 11 shows a circuit configuration example of the timing acquisition and tracking method according to the present invention. In the figure, 11
1 is a reception signal, 112 is a reception filter for band limitation, 1
13 is an output signal of the band limiting filter 112, 114 is a local frequency oscillator for frequency conversion, 115 is a multiplier,
116 is an output signal of the multiplier 115, 117 is a low-pass filter for removing high-frequency components, 118 is an output signal of the low-pass filter 117, 119 is a matched filter for extracting a phase modulation signal component from a spread spectrum signal, 12
0 is an output signal of the matched filter 119, 121 is a delay detection circuit for demodulating a phase modulation signal, 122 and 125 are output signals of the delay detection circuit 121, 123 is a judgment circuit for data judgment, 124 is judgment output data, and 126 is a judgment output data. An absolute value detection circuit for detecting the absolute value level of the differential detection output signal 125, 127 is an output signal of the absolute value detection circuit 126, 12
8, a sampling circuit for sampling the absolute value level signal 128; 129, a sampling signal output from the sampling circuit 128; 130, a sampling frequency oscillator for generating a sampling frequency; 131, an output signal of the sampling frequency oscillator 130; An N-symbol block averaging circuit 133 for detecting the average value of the absolute value level of the received signal over the symbol block is an N-symbol block averaging circuit 132
134, a maximum value sampling point detection circuit for detecting a sampling number at which the average value of the absolute value level of the reception signal is maximum, and 135, a maximum value level signal output from the maximum value sampling point detection circuit and its sampling The number information 136 is a comparison circuit that compares the threshold value 138 with a preset maximum value level signal 135, and the number information 136 switches between a carrier detection mode, a timing acquisition mode, and a timing tracking mode. 138 is a threshold value set in the mode switching circuit 137, 139 is a protection stage number counter for counting the number of protection stages set in the mode switching circuit, 140
Is a threshold signal output from the mode switching circuit 137;
1 is a comparison result output signal output from the comparison circuit, 142
Is the sampling number of the maximum level signal output when the value exceeds the threshold value 138 in the comparison circuit 136;
Numeral 3 denotes a judgment timing clock reproducing circuit for reproducing a clock signal for judgment timing from the sampling number of the maximum value level signal, 144 a reset signal for resetting the judgment timing clock, and 145 a reset signal for resetting the set value of the time window. The determination signal 146 is a determination timing signal output from the determination timing clock recovery circuit, 147 is sampling number information for setting a time window,
148 is a time window setting circuit for setting a time window from the sampling number information 147, 149 is an information signal relating to a time window set in the timing tracking mode, and 150 is a timing acquisition / tracking circuit. By applying the timing acquisition and tracking method according to the present embodiment, the method of the present invention can be realized as a practical circuit.

[0039]

(1) CFO-SSMA system and CF
In a communication system to which the O-SS system is applied, it is possible to suppress a sharp waveform fluctuation in a side lobe section generated by an interference signal from an adjacent spread spectrum signal and extract only a desired wave peak level as an effective waveform. (2) The desired wave peak point can be easily and accurately detected as the data determination timing. (3) Absorbs the influence of the time variation (jitter) of the desired wave pulse signal generated by the influence of the band limitation of the transmission system filter, transmission line distortion (multipath fading), noise, etc., and always maintains the accurate timing tracking state. It is possible to do. (4) The influence of a burst error, which has been a problem in the conventional timing acquisition and tracking method, can be avoided, and a stable communication line can be provided at all times. (5) The sampling frequency can be reduced while maintaining the same accuracy as that at the time of high-speed sampling, so that the amount of arithmetic processing of the digital circuit can be suppressed and power consumption can be reduced. (6) In a delay detection output waveform of a spread spectrum signal having a carrier frequency as a center frequency, an interference signal component from another spread spectrum signal group generated in a time section (side lobe section) other than a desired wave peak point is suppressed. And the timing acquisition time can be greatly reduced. (7) Since the number of preamble data symbols can be reduced, the time width allocated to the header section of the packet signal can be shortened, and the throughput characteristics of the communication line can be improved. (8) Since the detection accuracy of the desired wave peak point can be improved, the probability of tracking an erroneous timing can be greatly reduced, and the frequency of occurrence of burst errors can be suppressed.

[Brief description of the drawings]

FIG. 1 is a diagram showing an embodiment of a symbol block serving as an operation processing unit in a timing acquisition and tracking method according to the present invention.

FIG. 2 is a diagram illustrating a relationship between a sampling number defined in a timing acquisition and tracking method according to the present invention and a demodulated signal.

FIG. 3 is a diagram showing a relationship between a sampling number defined in a sampling frequency reduction method according to the present invention and a demodulated signal.

FIG. 4 is a diagram showing an embodiment of a sampling frequency reduction method according to the present invention.

FIG. 5 is a diagram showing an embodiment of a state transition from the carrier detection mode to the timing acquisition mode in the timing acquisition / tracking method according to the present invention.

FIG. 6 is a diagram showing an embodiment of a state transition from the timing acquisition mode to the timing tracking mode in the timing acquisition / tracking method according to the present invention.

FIG. 7 is a diagram showing an embodiment of a time window setting method in the timing acquisition / tracking method according to the present invention.

FIG. 8 is a diagram showing an example of a state transition of the timing tracking mode in the timing acquisition / tracking method according to the present invention.

FIG. 9 is a diagram showing an embodiment of a state transition of the carrier detection mode in the timing acquisition and tracking method according to the present invention.

FIG. 10 is a diagram showing an embodiment of a demodulated waveform improved by the preamble data setting method according to the present invention.

FIG. 11 is a diagram showing an embodiment of a circuit configuration for realizing the timing acquisition / tracking method according to the present invention.

FIG. 12 is a diagram illustrating a channel multiplexing method of a spread spectrum signal wave in a carrier frequency offset-spread spectrum multiple access system and a carrier frequency offset-spread spectrum communication system targeted by the present invention.

FIG. 13 is a diagram illustrating a time window setting method in a conventional timing acquisition / tracking method.

FIG. 14 is a diagram illustrating an embodiment of a delay detection output signal in a carrier frequency offset-spread spectrum multiple access system and a carrier frequency offset-spread spectrum communication system to which the present invention is applied.

FIG. 15 is a diagram showing an embodiment of a transceiver configuration method in a carrier frequency offset-spread spectrum multiple access system targeted by the present invention.

FIG. 16 is a diagram illustrating an embodiment of a transmitter configuration method in a carrier frequency offset-spread spectrum communication system targeted by the present invention.

FIG. 17 is a diagram illustrating an embodiment of a receiver configuration method in a carrier frequency offset-spread spectrum communication system targeted by the present invention.

[Explanation of symbols]

1 Preamble section 2 Unique word symbol section 3 Data symbol section 4 1 packet frame section 5 N symbol block 11 1 symbol section length 12 Correlation peak level of desired signal 13 Sampling start time 14 Sampling number 15 Symbol period of sampling number 21 1 symbol Section length 22 Correlation peak level of desired signal 23 Sampling number 24 Sampling number 25 Symbol period of sampling number 31 Received signal 32 Received signal not subjected to delay processing 33 Received signal passed through delay circuit 34 Sampling number 35 Sampling number 36 N Maximum value of absolute value average over symbols 37 Maximum value of absolute value average over N symbols 41 Initial state of carrier detection mode 42 Timing acquisition mode 43-1 to 43-A State transitions due to carrier detection 44-0 to 44- (A-1) State transitions due to carrier non-detection 45-1 to 45-A States at each stage of back protection 51 Timing capture mode initial state 52 Timing tracking mode 53- 1-53-B State transition when the maximum value exceeds the threshold value 54-0 to 54- (B-1) State transition when the maximum value falls below the threshold value 55-1 to 55-B Number of stages of back protection 60 Data determination circuit 61 Preamble section 62 Unique word symbol section 63 Data symbol section 64 1 packet frame section 65 N symbol block 66 Correlation peak level of desired signal 68 1 Symbol section length 69 Sampling number 70 1 Symbol time length 71 Absolute Maximum value level of average value level 72 Time window 73 Back window sampling Number 74 Forward window sampling number 81 Carrier detection mode initial state 82 Timing acquisition mode 83 Timing tracking mode 84-0 to 84- (C-1) State transition when maximum value exceeds threshold value 85-1 to 85-C maximum State transition when value falls below threshold 86 Transition path when counter value reaches forward protection stage number C 87-1 to 87-C State in each stage of forward protection 91 Initial state of carrier detection mode 92 Carrier detection mode 93 Timing acquisition mode 94 Timing tracking mode 95-0 to 95- (D-1) State transition due to carrier detection 96-1 to 96-D State transition due to carrier non-detection 97-1 to 97-D Number of stages of forward protection 98 Transition path 101 when the counter value reaches the number of forward protection stages D 101 Peak level of the desired signal Bell 102 Side lobe section 103 Normalized symbol section 104 Amplitude level 111 Received signal 112 Band limiting filter 113 Band limiting filter output signal 114 Local frequency oscillator 115 Multiplier 116 Multiplier output signal 117 Low pass filter 118 Low pass filter output signal 119 Matching filter 120 Matching filter output signal 121 Delay detection circuit 122 Delay detection circuit output signal 123 Judgment circuit 124 Judgment output data 125 Delay detection circuit output signal 126 Absolute value detection circuit 127 Absolute value detection circuit output signal 128 Sampling circuit 129 Sampling signal 130 Sampling frequency oscillator 131 Sampling frequency signal 132 N symbol block averaging processing circuit 133 N symbol block averaging processing circuit output signal 134 Large value sampling point detection circuit 135 Maximum value level signal and sampling number information 136 Comparison circuit 137 Mode switching circuit 138 Threshold 139 Protection stage number counter 140 Threshold signal 141 Comparison result output signal 142 Sampling number information of maximum value level signal 143 Judgment timing clock reproduction Circuit 144 Judgment timing reset signal 145 Time window set value reset signal 146 Judgment timing signal 147 Time window setting sampling number information 148 Time window setting circuit 149 Time window information signal 150 Timing acquisition / tracking circuit 201-1 to 201-n Spread spectrum Signals 202-1 to 202-n Carrier frequency 203 Carrier frequency interval 204 Frequency axis 205 Power density 211 Matched filter output waveform 212 Maximum value of demodulated signal 213 Sampling point 214 Sampling interval 215 Window width 221 Level peak point of desired signal 222 Side lobe section 223 1 symbol section length 224 Relative voltage level of differential detection output signal 300-1 to 300-n Information data sequence 301-1 to 301- n modulators 302-1 to 302-n pseudo-random codes 303-1 to 303-n modulators 304-1 to 304-n local oscillators 305-1 to 305-n band-pass filters 306 combiners 307 representing propagation path models Bandpass filter 308 Local oscillator 309 Frequency converter 310 Demodulator 311 Judgment data sequence 312 Automatic gain control circuit 401 Information data sequence 402 Serial-parallel converter 403-1-3 -n Information data sequence 404-1-4-n Modulation 405 PN code Generators 406-1 to 406-n PN (pseudo-random) code sequences 407-1 to 407-n Spread spectrum signals 408-1 to 408-n Modulators 409-1 to 409-n Local oscillators 410-1 to 410- n Spread spectrum high frequency signal 411-1 to 411-n Band pass filter 412-1 to 412-n Spread spectrum high frequency signal after band pass 413 Signal combiner 414 Spread spectrum combined signal 415 Common amplifier 416 Spread spectrum combined signal 417 For band limitation Filter 418 Spread spectrum combined transmission signal 421 Complex communication channel received signal 422 Receive filter 423 Receive filter output signal 424 Automatic gain controller (AGC) 425 AGC output signal 426-1 to 426-n Frequency converter 427-1 to 427-n Local oscillation 428-1 to 428-n Intermediate frequency reception signal 429-1 to 429-n Bandpass filter 430-1 to 430-n Bandpass filter output signal 431-1 to 431-n Matching filter 432-1 to 432-n Matched filter output signal 433-1 to 433-n Delay detection circuit 434-1 to 434-n Delay detection output signal 435-1 to 435-n Low-pass filter 436-1 to 436-n Low-pass filter output signal 437 -1 to 437-n Judgment device 438-1 to 438-n Judgment data sequence 439 Parallel-serial converter 440 Information data sequence

Claims (8)

[Claims] The transmitting side modulates a plurality of data signals each having the same data symbol rate and symbol timing and composed of independent data sequences by the same pseudo-random code sequence in a direct spreading system. A spread spectrum communication method comprising the steps of: multiplexing carrier frequencies separated by an integral multiple of a frequency amount defined by a data symbol rate as the center frequency of each spread signal; and simultaneously transmitting a spread multiplexed signal. For the receiving side, the symbol timing for data determination is captured for the spread multiplexed signal in units of a time width consisting of a data symbol time length of N symbols (N is an arbitrary natural number); A spread spectrum communication method, comprising a step of tracking. 2. The method according to claim 1, wherein the modulating step uses a phase modulation method for the transmitting side, and the receiving side performs a differential detection of the spread multiplexed signal before the step of capturing and tracking the symbol timing. 2. The spread spectrum communication method according to claim 1, further comprising the step of demodulating in a system. 3. An average value or an added value of the absolute values of the differential detection output signal levels for N symbols of received data for one symbol time, with the data symbol length of the information data sequence and the same time length as an operation period. If the average value or the maximum value of the addition value continuously exceeds the threshold value Th1 set in advance in blocks of N symbol time lengths A consecutive times, there is a significant received signal. A first position for determining that the received signal is present, and a time position indicating the latest maximum value at that time is assumed to be a determination timing, starting from a time when it is determined that the received signal is present, and the determination timing position Is calculated by setting the threshold level Th2 set in advance to N
In the second stage of determining that the determination timing has been correctly captured when the number of consecutive blocks exceeds the symbol time length in B units, and after the time at which the determination timing is determined to be correctly captured, N A time window centering on the latest decision timing is set for each block of the symbol time length, and the average value of the absolute values of the differential detection output signal levels obtained in the time window or the maximum value of the added value is determined in advance. When the threshold value Th3 is exceeded, the time position indicating the maximum value is determined as the optimal determination timing, and the determination timing is changed. When the maximum value falls below the threshold level Th3, The same time position is continuously used as the determination timing, and the maximum value is equal to the threshold level Th3.
, A third step of judging that the reception of the spread multiplexed signal has been completed when the number of signals falls below C times in blocks of N symbol time lengths, and thereafter returning to the first step. The spread spectrum communication method according to claim 2, wherein: 4. A received signal is sampled at a sampling rate S (S = R × M) M times the data symbol rate R (M is an arbitrary natural number), and the sampling point is set at a sampling point within an arbitrary data symbol time length. On the other hand, M sampling numbers are sequentially assigned, and all of the arithmetic processing and determination processing performed in units of the block having the N symbol time length are performed.
The spread spectrum communication method according to claim 2, wherein the method is performed on a sampling point to which the sampling number is assigned. 5. The sampling rate R is set to M / L (L is M>
L, and a sampling rate T (T = R × M / M) which is M / L times the data symbol rate R.
L), the received signal is sampled, and the sampling output signal of the received signal is converted into L parallel received sampling signals by branching into L, and the sampling interval is 1 / L.
Delay circuits each having a time length 1 / (T × L) obtained by dividing T by L as a minimum unit of the delay amount are arranged for the parallel reception sampling signals, and the parallel reception sampling signals are 1
/ (T × L) are formed such that they are offset from each other on the time axis by a time length, and M sampling numbers are assigned to the sampling points within an arbitrary data symbol time length. The parallel reception sampling signals are sequentially given, and the N
5. The method according to claim 4, wherein all of the arithmetic processing and determination processing performed in units of a symbol time length block are performed on the sampling points to which the sampling numbers are assigned.
2. The spread spectrum communication method according to 1. 6. On the transmitting side, modulating a plurality of data signals having the same data symbol rate and symbol timing and comprising independent data sequences, respectively, by the same pseudo-random code sequence in a direct spreading manner. A spread spectrum communication method comprising the steps of: multiplexing carrier frequencies separated by an integral multiple of a frequency amount defined by a data symbol rate as the center frequency of each spread signal; and simultaneously transmitting a spread multiplexed signal. On the reception side, as preamble data for timing acquisition, a reception signal waveform of a spread signal having a center frequency at a specific carrier frequency among a plurality of spread signals of the spread multiplexed signal is suitable for timing reproduction. Set preamble data to be assigned to each spread signal in advance so that it becomes a demodulated waveform Spread spectrum communication method characterized by having a floor. 7. A phase modulation method is used in the step of modulating the transmitting side, and the spread multiplexed signal is demodulated by a differential detection method before the step of setting the preamble data in the receiving side. The spread spectrum communication method according to claim 6, further comprising the step of: 8. The phase modulation method according to claim 1, wherein a phase difference change amount is 0,
00, 01, 11, for π / 2, π, 3π / 2 respectively
This is a QPSK system to which differential coding is performed by allocating 10 codes. For the H (H = 2m + 1, m is an arbitrary positive integer) spread signals transmitted simultaneously, m As preamble data of up to the number of spread signals, 11
Is set, and the remaining (m + 1)
By setting a repetition pattern of 01 as the preamble data of the spread signals, the absolute value of the delay detection output signal of the (m + 1) th spread signal from the lowest carrier wave frequency is suitable for timing reproduction. 8. The spread spectrum communication method according to claim 7, wherein the waveform is changed.