JP2001160798A - Time division duplex synchronization - Google Patents

Abstract

PROBLEM TO BE SOLVED: To improve a signal-to-noise ratio by supplying a code without a comma from the alphabet of the sum of modulated secondary synchronizing codes to a TDD mode cell search and removing redundancy. SOLUTION: A component generated by the QPSK-modulated secondary synchronizing codes is included and the code without the comma is used in a time division duplex mode spread spectrum communication system. Synchronization in the system is provided.

Description

DETAILED DESCRIPTION OF THE INVENTION TECHNICAL FIELD OF THE INVENTION

[0001] The present invention relates to communications, and more particularly to
It relates to spread spectrum digital communication and related systems and methods.

[0002]

Although spread spectrum wireless communication uses a radio frequency bandwidth that is greater than the minimum bandwidth required for the transmitted data rate, many users simultaneously use that bandwidth. Occupy. Each of the users "spreads" the information to encode it,
And a pseudo-random code to "despread" (by correlation) the spread spectrum signal for recovery of the corresponding information. FIG. 2 shows a system block diagram, and FIGS. 3a and 3b show a pseudo-random code plus QPSK (quadrature shift keying modulation (qu)
addrture phase-shift keyi
ng)) An encoder is exemplified. This multiple access is typically performed using code division multiplexing (code division multiplexing).
It is called multiple access (CDMA). The pseudo-random code is orthogonal (Walsh (Wa)
lsh)) sign, pseudo-nois
e) (PN) code, Gold code or a combination of those codes (modulo 2 addition (modulo 2 addition)
-2 additions): exclusive OR operation). After despreading the signal received at regular time instants, the user recovers the corresponding information, while the remaining interfering signals appear as noise. For example, an interim standard IS-95 for such CDMA communication is:
0.81 with 1.25 MHz bandwidth channel and symbols (bits) lasting 64 chips to be transmitted
38 microsecond code pulse interval (chip (chi
p)) to use the T _C. Recent wideband CDMA (WCD
MA) uses a 3.84 MHz bandwidth and the CDMA code length applied to each information symbol varies from 4 chips to 256 chips. C for each user
DMA codes are typically pseudo-random codes (QPSs) that increase the noise-like properties of the resulting signal.
It is generated as a modulo-2 addition of a Walsh code with two pseudo-random codes for K modulation). FIG.
The cellular system illustrated in FIG. 1 has an air interface (air in) between a base station and a mobile user station.
IS-95 or WCDMA for terface)
Can be used.

[0003] A spread spectrum receiver synchronizes with a transmitter by code acquisition, which is followed by code tracking. Code acquisition performs an initial search that brings the phase of the receiver's local code generator, typically within a half chip of the transmitter, and code tracking occurs on the incoming and locally generated Maintain fine alignment of code chip boundaries. Normal code tracking involves a delay locked loop (delay
-Lock loop (DLL) or tau-dither loop (T
DL), both of which use the well-known Early-L
The ate Gate method is based on the principle.

In a multipath condition, a rake (RAK)
E) The receiver has individual demodulators (fingers) that track separate paths, and the results are combined to improve the signal-to-noise ratio (SNR), typically with individual detected Maximum radio coupling (maxima) in which the synchronized signals are synchronized and weighted according to their strength.
Follow a method such as l ratio combining (MRC). Rake receivers typically have a DLL or TDL code tracking loop for each finger, along with a control circuit that specifies a tracking unit in the received signal path. FIG. 5 illustrates a receiver with N fingers.

UMTS (Universal Mobile Communication System) Approach UTRA (UMTS Terrestrial Radio Access)
Provides both FDD (Frequency Division Duplex) and TDD (Time Division Duplex) modes of operation for the spread spectrum cellular air interface. UT
RA currently uses a 10 ms period frame, where each time slot is divided into 15 time slots of 2560 chips. In the FDD mode, the base station and the mobile user transmit on different frequencies, while in the TDD mode, the time slot is the base station (downlink)
Or transmission by mobile user (uplink). Furthermore, TDD systems are different from FDD systems in that there is interference cancellation at the receiver. The small spread gain of the TDD system (8-16) and the absence of long spreading codes mean that multi-user multipath interference is not Gaussian shaped and needs to be canceled at the receiver. .

In the currently proposed UTRA,
The mobile user performs an initial cell search when first launched or enters a new cell, which search detects base station transmissions on the Physical Synchronization Channel (PSCH) without scrambling. The initial cell search by the mobile user must measure timing (time slots and frames) and identify the appropriate parameters of the found cell, such as scrambling codes.

[0007] For the FDD mode, a physical synchronization channel appears in each of the 16 time slots of the frame, and 25 out of the 2560 chips of the time slot.
Occupies 6 chips. Thus, in the synchronous channel, a comma-free code of length 16 (comma-fr
ee code (CFC) modulated length 25
The base station transmitting the 6-chip pseudo-noise repeated primary synchronization code allows the mobile user to first set the slot timing in synchronization with the 256-chip pseudo-random code, and then use the unique CFC cyclic shift By setting the frame timing using the characteristics, it is possible to synchronize. In addition, C
Decoding the FC reveals the scrambling code used by the base station.

[0008] In contrast, for the TDD mode, only the physical synchronization channel appears in one or two time slots per frame, so a length 16 CFC
Modulated primary synchronization codes are not easily applied. Alternatively, the proposed initial cell search in TDD mode consists of a primary synchronization code (PSC) plus six secondary synchronization codes (SS).
Using the sum of Cs), each code is a 256 chip pseudo-noise sequence and the codes are orthogonal.
In this proposal, the initial cell search consists of code group identification with slot synchronization, frame synchronization and scrambling code determination. In particular, during slot synchronization, the mobile user uses the PSC to obtain slot synchronization to the strongest cell (the strongest received base station transmission). Here, the PSC is common to all cells. Only one matched filter (or any similar device that matched the PSC) is used for detection.

[0009] The mobile user then uses the six SSCs to find frame synchronization and identify one of the 32 code groups used by the base station. Each of the six SSCs is modulated by +1 or -1, which means that 5 bits of information identify which of the 32 possible code groups is used by the found base station. (Scrambling code and midambles) and 6
The th SSC is modulated by +1 or -1;
Identify whether the time slot is the first physical synchronization channel slot in the frame or the second (frame synchronization). Each of the six SSCs is reduced to 1 / √6 and the power of the sum of the six modulated SSCs
Is equal to the radix of PSC. Finally, the mobile user measures which of the four scrambling codes of the code group of the cell is being used, for example, by correlation on the common control physical channel.

However, there are problems with this TDD mode proposal, including the low signal-to-noise ratio in the sum of the six modulated SSCs.

[0011]

The present invention provides a TDD mode cell search with a comma free code from the alphabet of the sum of the modulated secondary synchronization codes. The preferred embodiment is a comma free code of length 2 or 4 for frames with two time slots for the synchronization channel, each of which is not interleaved (n
on-interleaved or interleaved to two levels (two-level int)
used.

This includes the advantage of improving the signal-to-noise ratio by removing redundancy.

[0013]

BEST MODE FOR CARRYING OUT THE INVENTION Overview In the preferred embodiment, the synchronization method for UTRA-type TDD mode initial cell search is of length 2 or 4
(Or larger) comma-free code (CFC
s) is used to encode both the frame timing and base station code group information plus the frame position for the interleaved frame. CF
The use of C effectively removes the redundancy of frame timing and direct coding of code groups in both time slots occupied by the physical synchronization channel. Some preferred embodiments use a CFC with an alphabet generated from a reduced sum of two or three QPSK modulated secondary synchronization codes added to the primary synchronization code. The preferred embodiment spread spectrum communication system incorporates the preferred embodiment synchronization method.

In a preferred embodiment communication system,
The base station and the mobile user each include one or more digital signal processors (DSP's) and / or other programmable devices on which programs are stored to perform the signal processing of the preferred embodiment synchronization method. Can be. Base stations and mobile users also house analog integrated circuits for amplification and conversion between analog and digital inputs to or outputs from antennas, and these analog and processor circuits are Integrated on die. The stored program is, for example, in a ROM mounted on the processor or in an external flash EEPROM. The antenna is part of a rake detector with multiple fingers for each user's signal. The DSP core is a TMS320C6x or TMS32 manufactured by Texas Instruments.
0C5x.

Preferred with 2.12 secondary synchronization code
Embodiment 1 First Preferred Embodiment UTRA in TDD Mode
Initial cell search by mobile users in the system
The method involves two levels of frame interleaving and one
Using the physical synchronization channel illustrated in FIG.
You. In particular, FIG. 1a shows a 10 ms frame, slot
15 hours per frame with 2560 chips per frame
Slot, and the physical synchronization channel
Appear in two time slots per frame (ie,
0 and 8), the time offset from the beginning of the slot
Occupies 256 chips in each slot with
You. The base station has a 256-chip primary synchronization code C_Pplus
QPSK symbol b in the synchronization channel₁, ...
・, B_N(Each b_iIs modulated by one of ± 1, ± j)
N (N is equal to 3 in the example of FIG. 1b)
56-chip secondary synchronization code c₁, ..., c_NSend the sum of
I do. This code set of N secondary synchronization codes 二 c ₁,
.., c_N｝ Is ｛C₀, C₁, ..., C₁₁部分 partial collection
, Ie, ｛C_P, C₀, C₁, ..., C₁₁｝ Is ± 1
C, a 256-chip pseudo-noise sequence of components
_P, C₀, C₁, ..., C_NForm a set orthogonal to each of
Set of twelve secondary synchronization codes selected to be
You. For example, C_iIs derived from the 256 component Gold code
be able to. Each transmitted secondary modulated synchronization code
The radix is C_PIs reduced to 1 / N of the radix of
Keyed c₁, ..., c_NThe base of the sum of C is_PTo the radix of
New Of course, the sum c_kThe correlation with
｛C_P, C₀, C₁, ..., C₁₁｝ That is, b_k= √N
<C_k, C_P+ Σb_ic_i/ √N>, b_k
To recover.

The time offset varies between cells and helps to avoid interference from adjacent cell synchronization channel transmissions by all cells broadcasting the same primary synchronization code.

Frames are interleaved (typically to a depth of 2 or 4) to mitigate burst noise. The CFC of the preferred embodiment is multiplied by the frame interleaving depth (multipl
ied) has a codeword length equal to the number of physical synchronization channel time slots per frame.

Each cell (base station) belongs to one of 32 code groups (code groups are four 16-chip
Measure which set of scrambling codes contains the scrambling codes used by the base station and measure the time offset of the synchronization channel from the beginning of the time slot). The mobile user must measure the code group during the initial cell search, and the preferred embodiment implements the sum b ₁ described in the previous paragraph.
using _{c 1 + ··· + b N c} N as their alphabet, the comma-free code with a codeword of length 4, encodes the code group information. Length 4 specifies which two time slots per frame and 2
It is sufficient to determine which two frame positions in one level of interleaving are detected. Codes without commas with codewords of length m are codewords x = <x (1), x (2),..., X (m)> and y = <y (1), y (2 ),..., Y (m)>, if i> 1, the sequence <x (i), x (i +
, X (m), y (1), ..., y (i-
1) Recall that> has the property of not being a codeword, which includes the case where y = x, which is a cyclic shift of x. Thus, frame timing and frame interleave position information are provided by the comma-free property, and code group information is provided by a modulation code set.

The table of FIG. 1b lists 32 code groups and the corresponding preferred embodiment, length 4 comma-free codewords. First CFC codeword components frame 1, a filling of the slot k rows of three b _i c _i s to be the sum, the second component, the frame 1, slot k + 8
Column entries, the third component is an entry in frame 2, slot k column, and the fourth component is an entry in frame 2, slot k + 8 column. For example, reading the third row of code group 2, the 3 to be added to the PSC
The sum of the two modulated SSCs is (jC ₀ + jC ₁ + C ₂ ) / √3 in the first time slot of frame 1, and in the second time slot of frame 1,
(JC ₀ + jC ₁ −C ₂ ) / √3, and in the first time slot of frame 2, (−jC ₀ −jC ₁ +
C ₂ ) / √3, and (−C ₀ −jC ₁ −C ₂ ) / √3 in the second time slot of frame 2.

The codeword of FIG. 1b is generated as follows.
be able to. A collection of three SSCs for a given code
If, for example, C₀, C₁And C_TwoBut first, two of SSC
And difference of C₀+ C₁And C₀-C₁To form Next
And sum C₀+ C₁And the third SSC C_TwoIn phase (actual)
Modulation <(C₀+ C₁) + C_Two, (C₀+ C₁) -C_Two,-(C
₀+ C₁) + C_Two,-(C₀+ C₁) -C_Two> Used in length
4 form the first codeword. Sum (C₀+ C₁) Polarity
(Sign) indicates the frame number, and C _TwoPole of
Note that gender indicates the time slot number
No. Next, by analogy, instead of the sum, the difference C
₀-C₁, That is, <(C₀-C₁) + C_Two,
(C₀-C₁) -C_Two,-(C₀-C₁) + C_Two,-(C₀−
C₁) -C_Two> Forms the second codeword. even here,
(C₀-C₁) Indicates the frame number.
_TwoIndicates the slot number. These two
The codeword is a code set C₀, C₁And C_TwoThe eight possible
The combination of the real coefficients, ie the first codeword used ±
(C₀+ C₁) ± C_TwoAnd the second used ± (C₀
-C₁) ± C_TwoExhaust. Thus, the third and the third
4 also have the first and second codes, respectively.
C in word₀+ C₁And C₀-C₁Instead of
Phase (quadrature) modulation j (C₀+ C₁)
And j (C₀-C₁) Is used. That is, the third code
The words are <j (C₀+ C₁) + C_Two, J (C₀+ C₁) -C_Two,
−j (C₀+ C₁) + C_Two, -J (C₀+ C₁) -C_Two>
Similarly, for the fourth codeword, j (C₀-C₁)
Used. Again, the sum (C₀+ C₁) Or difference (C₀−
C₁) Indicates the frame number._TwoPolarity
Indicates a time slot number. Next, the fifth and
For the 6th codeword, the third and fourth codewords
C₁⇔C_TwoExchange. That is, the fifth codeword is <
j (C₀+ C_Two) + C₁, J (C₀+ C_Two) -C₁, -J (C
₀+ C_Two) + C₁, -J (C₀+ C_Two) -C₁>
For the sixth codeword, j (C₀-C_Two) And C₁
Is used. Finally, for the seventh and eighth codewords,
In the fifth and sixth codewords, C₀⇔C₁Exchange
Thus, for the seventh codeword, <j (C₁+ C_Two) + C₀, J
(C₁+ C_Two) -C₀, -J (C₁+ C_Two) + C₀, -J (C
₁+ C_Two) -C₀> And similarly for the eighth codeword
And j (C₁-C_Two) And C₀Is used.

The other 24 code words of FIG. 1b also use the code sets {C ₃ , C ₄ , C ₅ }, {C ₆ , C ₇ , C ₈ } and {C ₉ , C ₁₀ , C ₁₁ }. Generated in the same way.

Time offset t₀-T₃₁Is the channel
Starts at 0, 2208 in the possible slot
Chip (= 2560-256-96 guard (guar
d) 32 equally spaced possibilities in part). sand
That, t₀= 0, t₁= 71, t _Two= 142, t_Three= 21
3, ..., t₃₁= 2201 chips.

The cell search of the first preferred embodiment is:
The process proceeds in the following three steps. Step 1: Time Slot Synchronization During the first step, the mobile user selects a primary synchronization code (primary synchronization code).
code) (PSC) to obtain time slot synchronization to the strongest cell (strongest received base station). The PSC is common to all cells and consists of a short pseudo-noise sequence (256 chips). A single matched filter (or any similar device that matches the PSC) is used. Since the physical synchronization channel occupies two slots every 10 ms frame, slot synchronization should be achieved within about 5 ms. The time offset of the channel within the slot is measured in step 2.

Step 2: Frame synchronization and code group identification plus frame position During the second step, the mobile user finds frame synchronization using the secondary synchronization code (SSC) and 3
Identify one of the two code groups. Each code group has a time offset of the synchronization channel in the slot,
So it is linked to a specific frame timing,
It is also linked to a set of four scrambling codes (and a basic midamble). To detect the position of the next synchronization slot, the PSC
Is correlated with the signal received in both 7 and 8 time slots after the slot detected in step 1 (7 and 8 occur because the frame has 15 time slots). The signal received at the position of the synchronization slot is correlated with all of the PSC and SSC. These correlations are phase-corrected by the correlation with the PSC and coherent over one or many time slots.
ly) is performed. The correlation set of three b _i modulation of a time slot is recovered. By looking up in the table of FIG. 1b, the code group, the time offset of the synchronization channel in the time slot and
Frame timing and interleave positions are provided.

Step 3: Scrambling code identification During the third step, the mobile user measures which of the four scrambling codes (and the basic midamble) in the code group the cell actually uses. . This is due, for example, to the correlation with all four scrambling codes in a common control channel transmission by the base station.

The mobile user of the preferred embodiment will have PSCs and SSCs (and scrambling codes, etc.) stored in memory and programmed to perform the cell search described above.

3. Code Minimum Distance The correlation in step 2 to find the QPSK modulation symbol {b _i } is limited by the signal to noise component related to the minimum distance between the detected possible codeword components. Typical minimum distances between the two sums of the three SSCs with the QPSK modulation are (eg, Σb _i c _i / √3 and Σb
_{i 'c i / √3),} 2 two but equal and different symbols of three modulation symbols occurs when the one of the real part and the others have the imaginary part. Therefore, the minimum distance is √
2 / $ 3. Similarly, if the modulation symbols are restricted to being the real part (eg, BPSK modulation), six SSCs are required for the same modulation information, and the minimum distance is three QPSK modulated SSCs. Same as 2 /
√6 = √2 / √3. In contrast, as illustrated in FIG. 1b, the preferred embodiment provides a QP for each code set.
Using only half of the SK modulation possibilities, either 0 or 2,
An imaginary modulation that is never 1 or 3 is used. Thereby, the minimum distance between CFC codeword components increases by ２2 to 2/２3. Further, FIG. 1b also shows that each component (b
₁ , b ₂ and b ₃ ) appear only once in the table, so by decoding only one of the four components of the CFC codeword, the code group, frame timing and frame interleave position Is measured.

However, the preferred embodiment using CFCs is in the ordering (in time) of the sum of the SSCs in which some of the information is modulated (ie, in ordering the components of the CFC codeword). This is advantageous over simply using the sum of the modulated SSCs in the time slot. In fact, one SSC modulation to indicate which time slot and the other five modulated SSs to indicate the code group
C in both time slots of the frame comprising
Using six SSCs modulated by ± 1 has the redundancy that five modulated SSCs are repeated.

4. Preferred Embodiment with Six Secondary Synchronization Codes The second preferred embodiment applies to a physical synchronization channel with only one time slot per frame and frame interleaving at a depth of two. FIG. 6 shows a codeword using two code sets of three SSCs encoding 32 code groups. Basically, first the 16 length 4 codeword of FIG. 1b is cut in half to form the 32 length 2 codeword of FIG. This means that only two code sets are needed.

Of course, the same codeword can be used for two time slots per frame without frame interleaving.

5. Preferred embodiment with a secondary synchronization code of 5.16 A third preferred embodiment consists of two time slots per frame, two frame interleaving depths plus
Applied to three additional bits of information for the transport channel identified. As shown in FIG. 7a, this increases the number of codewords required compared to FIG. 1b by a factor of eight. The codeword consists of three SS
Note that the C code set is generated in the same manner as in FIG. 1b. Therefore, the number of code sets is 1
By repeatedly combining the 6 SSCs, 3
It is increased to two. Notably, the code set can be as in FIG. 7b.

A preferred embodiment for selecting a code set is as follows. All code sets are set {C ₀ ,.
, C ₁₅ } are broken up into subsets. The number of code sets depends on the number of groups to be represented. Each code set can provide eight length 4 comma-free code words. Thus, when there are only thirty-two groups (and hence thirty-two codewords), only four code sets are needed, and four disjoint subsets of {C ₀ ,..., C ₁₅ } It is easy to choose. 256
For an embodiment with codewords, 32 code sets are required. Since there are only 16 SSCs, not all code sets can be separated. In order to keep the minimum distance, the maximum value of one SSC can be the same as between any of the two code sets that are brought together. For example, {C ₀ , C ₁ , C ₂ }, ΔD ₃ ,
C ₄ , C ₅ }, {C ₀ , C ₄ , C ₆ } are among the valid code sets that can be used in this case, but {C ₀ ,
C ₄ , C ₅ } are not so.

6. Preferred Embodiment without Frame Interleaving A fourth preferred embodiment applies to synchronization with non-interleaved frames, but with two (or more) time slots per frame occupied by the physical synchronization channel. You. In particular, the six SSCs
C _1, C ₂ is, ..., at C _6, it is possible to form a CFC codewords of length 2 using QPSK modulation. <
(C _i + C _k ) / √2, (C _i −C _k ) / √2>, <− (C
_{i + C k) / √2,} - (C i -C k) / √2>, <j (C i
+ C _k ) / √2, j (C _i −C _k ) / √2>, and <−
_{j (C i + C k)} / √2, where _{-j (C i -C k) /} √2>, C i and C _k is a set C _1, C _2, replacing ..., C ₆ to It is a set selected without. 6 SS
In C, there are 15 pairs {C _i , C _k } and therefore 60 codewords of the type described above.

For the case of two time slots per frame (not interleaved) and 32 code groups to identify upon detection in a single time slot, BPSK modulated and $ 6 as described above.
With the six SSCs reduced to?, The minimum code distance between the two possible detections is 2 / √6 = 0.816. In contrast, in the length-2 CFC codeword described above, the frame timing is unique in the codeword component (C
_(i + C _k is the first time slot and C _i -C _k is the second time slot) and the minimum code distance between the two possible detections is equal to one. 30 of the 32 codewords required are
Note that real (in-phase) modulation can be applied and only two codewords require imaginary (fourth) modulation. That is, the 30 code words are:
<(C _i + C _k ) / √2, (C _i −C _k ) / √2> and <
− (C _i + C _k ) / √2, − (C _i −C _k ) / √2> and two imaginary modulated codewords, for example, <j (C ₁ + C ₂ ) / √2, j (C ₁ -C ₂ ) / √2>
And <−j (C ₁ + C ₂ ) / √2, −j (C ₁ −C ₂ ) /
It is necessary to use only <2>. In addition, reflecting the larger minimum code distance, FIGS.
CF of length 2 for 6 PSK modulated SSCs
It illustrates the excellent performance of the C code.

Furthermore, the computational complexity of a length-2 CFC is comparable to that of six BSPK modulated SSCs. In particular, for six SSCs, thirty-two correlations with a length eight correlation are obtained, and the fast Hadamard transform (Fast Hadam transform) is added to these correlation values.
ard Transform) is applied and 8x32
+ 2 × 16 × log ₂ 16 + 7 = 391 Obtain a correlation with 6 SSCs and PSCs requiring complex addition. Again, the phase of the correlation with the PSC is used as a reference for the correlation with the six SSCs.
This requires 28 real multiplications and 13 real additions. There are 64 possible combinations (codewords) of the 6 SSCs. Note that some of these combinations are simply negative of the others, and with other redundancy, 64 combinations require 564 real additions. Averaging 64 decision variables over K slots requires approximately 64 actual additions per slot for large K. Selecting the maximum value after averaging over K slots requires a comparison of (log + 2 + 64) / K for each slot. This number is the CFC and the six SSCs
The same is true for both methods, and very small for large K. So it is ignored in the analysis. Thus, the six modulated SSC methods require 118 actual additions per slot to calculate and average the decision variables and to select the maximum.

Multiplication by j for CFC means flipping the imaginary and real parts. Also, using ± <(C _i + C _k ) / √2, (C _i −C _k ) / √2> as described above, 30 code words are generated, and therefore ± j <
(C ₁ + C ₂ ) / {2, (C ₁ −C ₂ ) / √2> only 3
2 needed to complete the required codeword.
Again, note that some of these combinations are simply negative of the others, and the method requires 32 additions per slot to obtain 64 decision variables. Averaging 64 decision variables over K slots requires approximately 64 actual additions per slot for large K. Again, ignoring the comparison required to select the maximum, which is the same for both methods, the length 2 CFC method calculates the decision variables, averages, and selects the maximum. Requires 96 real additions per slot.

7. Modification The preferred embodiment uses a comma free code (CFC) for synchronization in TDD systems to remove redundancy.
It can be changed in various ways while retaining the features of. For example, code sets in four or more SSCs can be used. 1 + j, 1-j, -1 + j, -1-j
Rotation to modulation by. Using comma-free codes for TDD ensures that all time slots (in the synchronization channel) that contribute to the distance between codewords avoid the problem of loss of diversity, which leads to loss of codeword separation distance. Note that it is possible.

[Brief description of the drawings]

The drawings are heuristic for clarity.
tic).

FIG. 1a shows a code without synchronization channels and commas.

FIG. 1b shows a code without synchronization channels and commas.

FIG. 2 shows a spread spectrum system.

FIG. 3 illustrates pseudo-random codes and symbols.

FIG. 4 shows a cellular system and block diagram of a receiver.

FIG. 5 shows a cellular system and block diagram of a receiver.

FIG. 6 illustrates a code without a comma.

FIG. 7a illustrates a code without commas.

FIG. 7b illustrates a code without commas.

FIG. 8 is a simulation result.

FIG. 9 is a simulation result.

FIG. 10 is a simulation result.

FIG. 11 is a simulation result.

FIG. 12 is a simulation result.

FIG. 13 is a simulation result.

FIG. 14 is a simulation result.

FIG. 15 is a simulation result.

──────────────────────────────────────────────────の Continued on the front page (51) Int.Cl. ⁷ Identification FI FI Theme Court II (Reference) H04L 27/36 H04L 27/00 F 27/38 G East Lenner 2600 Number 165 (72) Inventor Sundararajan Suriram United States Texas, Dallas, Lauder 16500 Number 18208

Claims (10)

[Claims] 1. A method for time division duplex communication synchronization comprising: (a) detecting a codeword component without commas in a synchronization channel time slot; and (b) decoding the codeword component. 2. The method of claim 1, wherein (a) the codeword component is a combination of QPSK modulated synchronization codes. 3. (a) said combination is selected from one of FIGS. 1b, 6 and 7a, wherein C _n represents a synchronization code;
The method according to claim 2. 4. A synchronization channel detector operable to detect a comma free codeword component in the received signal, and (b) for the codeword component, coupled to the detector. A mobile receiver in a time division duplex communication system comprising a decoder. 5. The mobile station of claim 4, wherein: (a) the detector includes a programmable processor with programming to detect the codeword component as a combination of QPSK modulated synchronization codes. 6. The method according to claim 6, wherein said combination is selected from one of FIGS. 1b, 6 and 7a, wherein C _n represents a synchronization code.
The mobile device according to claim 5. 7. A method comprising: (a) transmitting a comma-free codeword component in a synchronization channel time slot, wherein (b) the codeword component is at least partially transmitted to a transmitter. Identifying time division duplex synchronization method. 8. The method of claim 7, wherein (a) the codeword component is a combination of QPSK modulated synchronization codes. 9. The method of claim 8, wherein (a) the combination is selected from one of the rows of FIGS. 1b, 6 and 7a, where C _n represents a synchronization code. 10. A transmitter for a codeword component without comma in a synchronization channel time slot, wherein (b) said codeword component is at least partially associated with a base station. Said base station in a time division duplex communication system identifying.