JP2000315966A - Space time transmission diversity for tdd/wcdma - Google Patents

Abstract

PROBLEM TO BE SOLVED: To improve reception by providing diversity of at least 2L via time and a space by receiving a data symbol by a circuit using a matching filter circuit including plural fingers, generating respective output signals by respective fingers, receiving the output signals by plural decoder circuits and generating respective output signals by respective decoder circuits. SOLUTION: A finger L is provided to inversely spread the received signal from a user K. Therefore, L signals corresponding to respective multipaths of the user K selectively pass matching filters 700 to 704. Output signals of the matching filters are supplied to each of STTD(space time transit diversity) decoders 706 to 710 and then supplied to a rake synthesizer 712. Respective multipath signals of the user K are synthesized by the rake synthesizer 712. The synthesized signal of the user K is supplied to a symbol decision 714.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

This application is a provisional application No. 60 / 121,541,199 filed on Feb. 25, 1999.
Provisional application number 60 / 121,65 filed on February 25, 9
7, Provisional application number 60/13 filed on May 21, 1999
5,263 priority of 35U. S. C. §119 (e)
(L).

[0002] The present invention relates to wideband code division multiple access (WCDMA) for communication systems and, in particular, to WCDM.
A space-time block coded transmission diversity for A (sp
ace time block coded tran
(Smith antenna diversity).

[0003]

BACKGROUND OF THE INVENTION Code division multiple access (CDMA) systems are characterized by assigning unique codes to different data signals and transmitting them simultaneously on a common channel. This unique code is matched with the code of the selected receiver to determine the proper data signal recipient.
The different data signals reach the receiver through multiple paths due to ground clutter and unexpected signal reflections. Due to the additive effects of these multiplexed data signals at the receiver, large fading or fluctuations in the received signal strength can occur. Generally, this fading due to multiple data paths is
It can be mitigated by spreading the transmitted energy over a wide bandwidth. The use of such a wide band significantly reduces fading as compared to narrow band transmission modes such as frequency division multiple access (FDMA) and time division multiple access (TDMA).

The next generation wideband code division multiple access (WCDM)
A) A series of new standards for communication systems are emerging, one of which is U.S. Pat. 90 / 205,029, which is incorporated herein by reference. The U.S. patent application includes Da
frequency division duplex (FDD) by bak et al.
A method of space-time transmit diversity (STTD) for a WCDMA system is described. These FDD
The system is a coherent communication system using a pilot symbol based channel estimation scheme. These pilot symbols are QPSK (quadratu)
The data is transmitted to all receivers within the range in a predetermined time frame as known data of a re-shift keyed. Frames can be transmitted in discontinuous transmission (DTX) mode. In voice traffic, user data is transmitted when the user is talking, and no data symbols are transmitted when the user is silent. The same applies to packet data, in which user data is transmitted only when preparation for packet transmission is completed. The frame includes control symbols such as transmission power control (TPC) symbols and rate information (RI) symbols in addition to pilot symbols. These control symbols include multiple bits called chips to distinguish them from data bits. Therefore, the chip transmission time (Tc) is equal to a value obtained by dividing the symbol rate (T) of the symbol (G) by the number of chips.

The time division duplex (TDD) is WCD
MA and FDD systems provide another communication standard.
The TDD data is transmitted as a QPSK symbol in a data packet of a predetermined period, that is, a time slot. Each data packet has a predetermined training sequence or midamble during a time slot.
e). Data packets are exchanged in cells formed by base stations communicating with nearby mobile stations. Data in neighboring cells is modulated by different periodic codes. The midamble is formed by adding sequences shifted in time from the base sequence, each time shift corresponding to a mobile station in the cell. The spreading factor (SF), that is, the number of chips per modulation symbol is desirably 16 or less. The basic period code that modulates the midamble symbol in the cell is shifted so that each mobile station in the cell can be individually identified. However, since the periodic codes in the cell are the same and the spreading factor is small, interference from the base station and other mobile stations in the cell is not received as Gaussian noise. Therefore, F
Typical matched filter circuits used in DD systems are not suitable for removing this intracell interference. A solution to this problem is described by Anja Klein et al.
orcing and Minimum Mean-S
query-Error Equalization
for Multiuser Detection i
n Code-Division Multiple-
Access Channels, IEEE Tra
ns. on Vehicular Technology
gy, 276-287 (1996) "and incorporated herein by reference. There are intersymbol interference (ISI) and multiple access interference (MAI)
Klein et al. Describe zero forcing (ZF) and minimum mean square error (MMSE) equalization with and without decision feedback (DF) to reduce both. Kle
in et al. Jung, J.A. Blanz,
P. W. "Coherent Rec by Baicr
eever Antenna Diversity f
or CDMA Mobile Radio System
emsUsing Joint Detection,
Proc. IEEE Int. Symp. Pe
rs. Indoor and Mobile Rad
io Communications, 488-49
2 (1993), and proposes the possibility of combining these techniques with antenna diversity.
A. Naguib, N .; Seshadri, A.
R. "Applicati" by Calderbank
ons of Space-Time Block C
odes and Interference Sup
Pression for High Capacit
y and High Data Rate Wire
less Systems, Proc. of th
e Asilomar Conference, 18
03-1810 (1998) "further extends the work of Klein et al. However, at that time, spatio-temporal transmit diversity was not known. Therefore, Klein et al. And Jung et al. Have joint detection (joint detection) with STTD in a TDD system.
Ction) is not proposed nor disclosed. Also, Klein et al., Jung et al.
A communication system with the advantages of STTD and joint detection in a TDD system is not described.

[0006]

SUMMARY OF THE INVENTION These problems are solved by a circuit designed with a matched filter circuit including a plurality of fingers connected to receive data symbols. Each finger corresponds to a respective path of the data symbol. Each finger produces a respective output signal. Output signals generated from each of the plurality of fingers are received by a plurality of decoder circuits. Each decoder circuit produces a respective output signal. A joint detector circuit is connected to receive output signals from the plurality of decoder circuits, respectively. The joint detector circuit generates an output signal corresponding to a predetermined code.

[0007] The present invention improves reception by providing at least 2L diversity over time and space for TDD systems. No transmission power enhancement or bandwidth extension is required. Balance the power between multiple antennas.

[0008]

FIG. 1 shows a STTD (Space-Time).
FIG. 2 is a simplified block diagram illustrating a transmitter of the present invention that utilizes Transit Diversity. The transmitter circuit receives the Doppler control signal on lead 102 and
4 includes a diversity control circuit 100 connected to receive the handoff control signal. The Doppler control signal is determined by comparing consecutive midamble symbols from the mobile station in the same cell as the transmitter. An increase in the difference between the received midamble symbols indicates that the Doppler rate based on mobile station speed relative to the transmitter is large. The handoff signal is determined by a mobile station report indicating the strength of a signal received from a surrounding base station. When the base station handoff is not required and the Doppler rate is low, the first value of the control signal is output to the lead 108 from the diversity control circuit. When this first value is provided to the STTD encoder circuit 110, the encoder circuit switches the transmit antennas 112, 114 to switch transmit diversity (S
TD) applies. Therefore, the symbols received on lead 106 are transmitted alternately from antennas 112 and 114.

On the other hand, when base station handoff is required and the Doppler rate is increased, the second value of the control signal is output from the diversity control circuit to the lead 108.
When this second value appears, STTD encoder circuit 110 applies STTD to transmit antennas 112,114. Thus, the encoder circuit receives the symbol S ₁ from antenna 112 and the transformed symbol from antenna 114

[Outside 1] Are output simultaneously. These symbols are multipath 11
It is transmitted to the remote mobile antenna 120 via 6, 118. This design uses STTD for high Doppler rates.
It is very advantageous when coding improves communication, and also during weak signal periods, such as during base station handoff. For example, PCCPCH (primary commo)
n broadcast channel such as n control channel), STT is used for all transmissions.
It is desirable to use D coding. This is advantageous because the broadcast channel is transmitted to all mobile receivers regardless of specific diversity conditions.

FIG. 2 is a block diagram of a communication system according to the present invention, showing diversity communication with a mobile station and communication without diversity. In this typical configuration, STTD is applied to users 1 to Z and diversity is not applied to Z + 1 to K. Therefore, in this communication system, STTD is added to the data symbol in lead 202.
And no diversity is applied to the data symbols on lead 218. The data symbol D ^{1 on the} lead 202 is STTD encoded by the encoder circuit 200 and the encoded data symbol

[Outside 2] Is the encoded data symbol from lead 204

[Outside 3] Are output from the leads 206, respectively. Lead 204
Coded data symbol in

[Outside 4] Is multiplied by a predetermined user-specific code, that is, a sequence C ^1, by a circuit 208 and supplied to an addition circuit 212. These encoded data symbols are added together with other user-specific data symbols by an adding circuit 212,
It is supplied from the lead 230 to the antenna 1. Similarly, the data symbol in lead 206

[Outside 5] Is multiplied by the same user unique code C ¹ by the circuit 214 and supplied to the adding circuit 216. These added symbols are transmitted from lead 250 on radio channel 261 to the mobile receiver antenna. The transmitted symbols are effectively multiplied by H ₁ 232 and H ₂ 238 of the channel impulse response matrix of respective paths 234, 240 and then added by path 242. Noise N is added by the path 246, and a reception signal is obtained at the antenna 250. Joint STTD decoder circuit 26
0 receives the composite signal and the user specific symbol sequence from lead 252 for each of the K users.

[Outside 6] , From lead 254

[Outside 7] From lead 256

[Outside 8] Is output.

If no transmit diversity is applied, or if another type of diversity is applied, such as switched transmit diversity (STD) or adaptive transmit array diversity (TxAA), the transmitter transmits the symbol sequence from lead 218.

[Outside 9] Is output. This sequence is multiplied by the user unique code C ^K in the circuit 220 and supplied to the addition circuit 212. Symbol sequence

[Outside 10] Is added together with other user-specific signals in the circuit 212 as described above, and is transmitted on the wireless channel 261. Therefore, the communication circuit of the present invention is STTD compatible and has compatibility with transmission to which diversity is not applied.

FIG. 3 shows a TDD radio frame that can be transmitted by the communication system of FIG. The duration of a radio frame, eg, radio frame 300, is 10 ms.
The radio frame has the same 15 time slots 302-3.
It is divided into ten. Each of these time slots is further divided into 2560 chip times (Tc). FIG. 4 illustrates the structure of a TDD time slot. The time slot includes a first group of data symbols 420 including 1104 chips. This first group is 6 for a typical spreading factor of 16
This corresponds to 9 data symbols. The first group is followed by a midamble 422 containing 16 symbols for a typical spreading factor of 16. These midamble symbols are a predetermined training sequence similar to the pilot symbols of the FDD system. As mentioned above, the midamble symbols are time-shifted cyclically for different users in the cell. The midamble is followed by a second group of data symbols 424 that includes another 1104 chips. Finally, the data symbol of the second group is followed by a guard period 426 of 96 chips.

FIG. 5A shows an embodiment of a symbol transmission sequence for TDD using STTD coding. Typical symbol sequences S ₁ -S ₈ show a partial sequence of data symbols supplied to the transmitting circuit from leads 106 (FIG. 1). This symbol sequence is midamble 4
22 (FIG. 4). These symbols are the antennas ANT1, ANT2
, Symbol transmission time 0, T, 2T ...
Reconstructed and transformed according to (N + 3) T. The symbol transmission time of 2NT corresponds to the data symbol 420 of the first group. Therefore, the symbol transmission time NT is located substantially at the center of the transmission sequence of the data symbol 420. For example, symbols S ₁ and S ₂ are transmitted at antennas ANT at transmission times T and 2T, respectively.
Sent from 1. Conversion symbol

[Outside 11] Are antennas AN at transmission times T and 2T, respectively.
Transmitted simultaneously from T2. These transform symbols are the complements of the complex conjugate of the respective symbols S ₃ and S ₄ . These sequences continue up to symbols 420, 424 (FIG. 4). In this transmission sequence, N out of 2NT symbols
For the T symbols, the receiver sets the data symbols 0 to (N
-1) Zero forcing (ZF-STTD) by ignoring intersymbol interference (ISI) of T blocks
Is reduced and the minimum mean square error (MMSE-S
(TTD) STTD decoder is obtained.

FIG. 5B shows an embodiment of a midamble pattern used for channel estimation. The basic sequence extends over the entire length of the midamble excluding the cyclic prefix. This basic sequence corresponds to B.I. Ste
inner and P.S. W. "Low C by Baier
ost Channel Estimation in
the Uplink Receiver of CD
MA MobileRadio Systems, F
requenz. , Vol. 47, 292-298 (1
993)) to cyclically shift to provide channel estimation to different users. The cyclic prefix 516 is obtained by copying to the end 514 of the cyclically shifted basic sequence. The hatched area 510 is the first 64 bits of the basic midamble sequence. In the present invention, the first two time shifts 511-512 are respectively assigned to the channel estimates for antenna 1 and antenna 2 of the broadcast channel. Thus, the broadcast channel is such that midamble shift 511 is transmitted from antenna 1 and midamble shift 512 is transmitted from antenna 2. As with broadcast channels, receivers employing STTD preferably use two midamble shifts for channel estimation. On the other hand, non-STTD receivers preferably use the same midamble shift from both antennas with the appropriate weight corresponding to transmit beamforming for that user.

FIG. 6A is a block diagram showing a signal flow in a TDD receiver for a single user according to the present invention to which STUD coding is applied. The receiver includes matched filters 600 to 604. Each matched filter circuit is a respective STTD decoder circuit 606-61.
Connected to 0. STTD decoder circuits 606-610 are connected to rake combiner circuit 612. Each matched filter and STTD decoder corresponds to a finger of a RAKE combiner circuit 612. These fingers correspond to paths 1 (116) to j (118) in FIG.
Selectively connected to different multipath signals. The selected multipath signals are then combined by a rake combiner 612 and sent to a channel decoder, such as a turbo decoder or a Viterbi decoder, for new processing.

A typical STTD decoder 60 shown in FIG. 6B
6 can be used for the STTD decoders 606-610 of FIG. 6A. The Rayleigh fading parameter is determined from the channel estimates of the midamble symbols transmitted from the respective antennas on leads 112 and 114. Rayleigh fading parameters to simplify analysis

[Outside 12] Is a parameter of a signal transmitted from the first antenna 112 along the j- ^th (j ^th ) path. Similarly, the Rayleigh fading parameter

[Outside 13] Is a parameter of a signal transmitted from the second antenna 114 along the j- ^th (j ^th ) path. When the transmission time τ _j corresponding to the j- ^th (j ^th ) path elapses, the respective i- ^th (i ^th ) chip or bit signal r _j (i +
τ _j ) are sequentially received by the remote mobile station antenna 120. The chip signal on lead 620 and the channel orthogonal code on lead 622 are multiplied by circuit 624 and the user-specific signal is output on lead 626. Lead 62
6 is supplied to a despreader input circuit 628, where it is added every symbol time and placed on leads 632 and 634 corresponding to the jth of the L-duplex signal path.

[Outside 14] Are respectively output.

[Outside 15] Is delayed by one symbol time by the circuit 630,

[Outside 16] Sync with Represented by the formula [1-2]

[Outside 17] Is supplied from the leads 632 and 634 as an input signal of the phase correction circuit.

(Equation 1)

The first phase correction circuit has a first
Rayleigh fading parameter corresponding to antenna

[Outside 18] Channel estimate and another Rayleigh fading parameter corresponding to the second antenna on lead 646

[Outside 19] The complex conjugate with the channel estimate of is provided. The complex conjugate of the input signal is derived from leads 648,
650 respectively. These input signals and their complex conjugates are multiplied by the Rayleigh fading parameter estimate signal and added as described above to obtain the equation [3-
4] are supplied from output leads 668 and 670, respectively.

(Equation 2)

These path-specific symbol estimates are then supplied to a rake combiner circuit 612, where the individual path-specific symbol estimates are added, resulting in pure software as shown in equation [5-6]. The symbol is output on lead 616.

(Equation 3) These pure symbols or estimates provide path diversity L and transmit diversity 2. Therefore, the total diversity of the STTD system is 2L.
This increase in diversity is very effective for reducing bit errors.

FIG. 7 is a block diagram showing a signal flow when a plurality of users use the TDD receiver of the present invention employing STTD coding. As will be described in greater detail below, this figure is an extension of the circuits of FIGS. 6A and 6B to perform parallel interference cancellation for multiple users. L fingers are provided to despread the received signal from K users. Therefore, the L signals corresponding to the respective multipaths of the K users are selectively applied to the matched filter circuits 700 to 700.
Go through 704. These matched filter output signals are supplied to respective STTD decoder circuits 706 to 710 and subsequently to a rake combiner circuit 712. A rake combiner circuit 712 combines the K respective L multipath signals. The combined K user signal is supplied to a symbol determination circuit 714. Each of the K symbols is determined and provided on bus 716 as an output signal.

As described above, for these TDD data symbols, it is desirable that the spreading factor (SF), that is, the number of chips per modulation symbol is 16 or less. Also, the basic period code that modulates the midamble symbol in the cell is shifted to identify each mobile station in the cell. Therefore, since the periodic code in the cell is the same and the spreading factor is small, interference from a base station or another mobile device in the cell is not received as Gaussian noise. Typical matched filter circuits used in FDD systems are not suitable for removing this intra-cell interference. The circuit of FIG. 8 is a block diagram of a first embodiment of the present invention, illustrating parallel interference cancellation for TDD using STTD coding. Data symbols from the matched filter circuits 700 to 704 as shown in Expression [7] are stored in the memory circuit 800.

(Equation 4)

A cross-correlation matrix is calculated and stored in memory circuit 802 to determine the interference effect each path of each finger of each user has on all other paths. First, a cross-correlation matrix R is calculated by calculating all cross-correlations between symbols on each finger of each user. Since this step is performed for the previous symbol, the current symbol, and the next symbol, three LK matrices are obtained. Then, to eliminate the autocorrelation, the diagonal of the intermediate LK matrix is set to zero. Therefore, the cross-correlation matrix R
Is LK × 3LK. The initial channel estimates for each antenna given by equations [8-9] are stored in memory circuit 80
4 is stored.

(Equation 5)

The initial data symbol estimates include two symbols for each user, which are obtained from equation [10] and stored in memory circuit 818.

(Equation 6)

These initial data symbols are STTD coded, multiplied by the initial channel estimate stored in memory circuit 804, and stored in circuit 814, as shown in equation [11-12] for path p. .

(Equation 7)

The STTD encoded data symbol obtained from the equation [11-12] is supplied to the circuit 814 by the circuit 802.
Is multiplied by the cross-correlation matrix R from, and the signal estimate given by equation [13] is output.

(Equation 8)

Next, this signal estimate is multiplied by the cross-correlation matrix R, and the intersymbol interference (ISI) estimate is output on lead 812. The circuit 820 includes the lead 806
The ISI estimate on lead 812 is subtracted from the stored matched filter symbol Y appearing at, and the first iteration of the modified data symbol is output on lead 822. The first iteration of the new data symbol is decoded and rake combined in circuit 824 to output the new symbol decision on lead 826. These new symbols on lead 826 then replace the initial symbols stored in memory circuit 818. The above procedure is repeated to output the second and subsequent iterations of the modified data symbol on lead 822. Until the ISI is effectively removed, a new symbol decision Y _i is generated according to equation [14] until a predetermined number of iterations is reached. Therefore, the parallel interference elimination circuit of FIG. 8 performs the previous symbol decision y _i−1 , correlation matrix R, and previous signal estimated value matrix E _{i −1}
Is output as a new symbol decision Y _i .

(Equation 9)

FIG. 2 details a system model according to an alternative embodiment of the interference cancellation of the present invention. The circuit of FIG. 2 includes a base station on the left side of the radio channel 261. The base station
The STTD encoded data symbol given by equation [15] is transmitted from antenna 1 at 230 to L of K users. The base station transmits a similar data symbol given by equation [16] from antenna 2 (236) to the same user.

(Equation 10)

The denominator of the equation [15-16]

[Outside 20] Is due to the balanced transmit power at each antenna for SITU coding. For the remaining KL users, equation [17]
The non-STTD encoded data symbol given by
Sent only from antenna 1 at 0.

[Equation 11]

The transmission data rate is the same for all users. Each data symbol is repeated G times, and in circuits 208, 220 and 214, equation [18]
Is multiplied by the orthogonal code unique to each user.

(Equation 12)

The chip period of each data symbol is represented by T _c = T _S / G. After user-specific spreading is performed, the signal data symbols of all K users are
It is added at 12 and supplied to the antenna 1 at 230. As shown in equation [19], the wireless channel adds 232 an impulse response of length W to the transmission data symbol from antenna 1 sampled at the chip rate. The corresponding impulse response of the transmission data symbol from antenna 2 is given by equation [20].

(Equation 13)

If the W value for any user is greater than 1, loss or orthogonality causes inter-symbol interference (ISI) for user symbols and multiple access interference (MAI) for other user symbols. For the purpose of illustration, a typical chip rate sampling was assumed, but as will be apparent to those skilled in the art, a mobile device may have a fractional, space,
Equalizer (fractionally spaced)
In order to provide equalizers, it may be necessary to sample channels at twice the chip rate. However, Fractional Spatial
Equalizer (fractionally spaced)
The analysis of the STTD decoder for equalizers and multi-user detectors is the same as typical chip rate sampling. The combined channel response of antennas 1 and 2 is given by equations [21] and [22], respectively.

[Equation 14] A combined data symbol vector of one block composed of M symbols transmitted from both antennas is represented by Expression [23], and is output from 242 on transmission path 244.

(Equation 15)

Additive Gaussian noise for sampling at chip rate

[Outside 21] Is represented by equation [24] and added at 246 to generate a composite signal at the receiving antenna 250 of the mobile device.

(Equation 16) This received sequence sampled at the chip rate at 250

[Outside 22] Is MG + W−1, which is the sum of the signals from the two antennas and the additive Gaussian noise given by equation [25].

[Equation 17]

The matrices A = (A _ij ) and B =
The elements of (B _ij ) are given by equations [26 to 30]. Where i = 1, 2,..., M * G + W−1, j = 1,
2, ..., K * M. Regarding the elements of the matrix B, even elements are given by equations [28] and [30], and odd elements are given by equations [29 to 30].

(Equation 18)

The structure of the matrix B is generated by STTD coding. This structure is substantially different from the prior art. For example, the structures of Klein et al. And Naguib et al. Correspond to equation [25] where matrix B is zero. These prior art formulas are valid in the absence of multipath channels. However, if there is a multipath channel, the structure of Equation [25] cannot remove either intersymbol interference (ISI) or multiple access interference (MAI). Received signal matrix

[Outside 23] And the complex conjugate matrix

[Outside 24] Only when is combined, both ISI and MAI can be removed. This structure is very advantageous for the design of the joint detector of the present invention. Therefore, equation [25]
The structure of the matrix B represented by
Can be rewritten.

[Equation 19]

[0034]

[Outside 25] Is the equation of equation [25]

[Outside 26] Equation [31] is linearly independent if the original equation [25] is linearly independent. Therefore, the conjugate matrix can be rewritten into equation [32] and equation [3]
1] can be rewritten into equation [33].

(Equation 20)

FIG. 9A is a block diagram showing a new embodiment of the interference cancellation circuit of the present invention using an STTD decoder and a zero-forcing STTD equalizer. Formula [33]
Received signal

[Outside 27] Is supplied to the whitening matched filter 902 through the lead 900. This whitening matched filter includes multiple finger matched filters 700-704 and their corresponding sampling STTD decoders 706-710 of FIG. The product of the received signal and the output of the whitening matched filter is supplied to a zero forcing STTD equalizer circuit 904,
Data symbol matrix

[Outside 28] Is output on the lead 906. Zero Forcing ST
The term in the TD equalizer circuit 904 provides the zero forcing solution of equation [33] that does not cause intersymbol interference (ISI) or multiple access interference (MAI), as shown in equation [34]. However,

[Outside 29] ^H represents the Hermitian operator on the matrix.

(Equation 21)

[0036]

(Equation 22) In the special case of, SF-STTD is given by equation [35].

(Equation 23) The same estimate of the received data symbol is

[Outside 30] , So only one of them needs to be calculated.
But in the middle step,

[Outside 31] Needs to be calculated.

[Outside 32] The Cholesky decomposition of is given by equation [36].

(Equation 24)

Σ is a diagonal matrix, and H is an upper triangular matrix. Cholesk in equation [36]
The y decomposition is

[Outside 33] , The amount of calculation of equation [35] is greatly reduced. Cholesky's formula [36] provides a means for solving equation [35] using the forward equation obtained from the upper triangular matrix H. The detailed block diagram of FIG. 9B shows an iterative solution of equation [34] for a zero-forcing STTD equalizer using decision feedback. For an introduction and use of feedback operator 924, see Anja Klein et al.
0.

FIG. 10A is a block diagram showing a third embodiment of the interference cancellation according to the present invention using an STTD decoder and a minimum mean square error STTD equalizer. Data covariance matrix

[Outside 34] , The minimum mean squared error solution for STTD decoding (MMSE-STTD) is given by equation [37].

(Equation 25)

(Equation 26) In the special case of MMSE-STTD decoder solution, the equation [3
8].

[Equation 27] Again,

[Outside 35] Since the same data estimation value is obtained from, one of the calculations and the calculation of the intermediate step are performed.

[0039]

[Outside 36] The Cholesky decomposition of is given by equation [39].

[Equation 28] The Cholesky decomposition of equation [39] reduces the complexity of equation [38].

[Outside 37] This is very advantageous because the computation of is complicated. C
Holsky's formula [39] is a forward equation (fowa) obtained from the upper triangular matrix H.
rd equation) to provide a means to solve equation [38]. The block diagram of FIG.
Iterative (i) using decision feedback
(Tertive) shows a minimum mean square error STTD equalizer. For an introduction and use of the feedback operator 1024, see Klein et al.
Is described in detail.

FIG. 11 is a simulation diagram showing the bit error rate (BER) as a function of the bit energy-to-noise ratio (Eb / No) with and without the application of vehicle Doppler rate diversity with a spreading factor of 16. is there. With a typical BER of 10 ² , a zero-forcing STTD receiver shows a ^2.5 dB improvement over an equivalent receiver without STTD. The simulation diagram of FIG. 12 shows the bit error rate (BER) with and without pedestrian Doppler rate diversity with a spreading factor of 16 and eight users as the bit energy to noise ratio (Eb / No). As a function of
Both curves show an improvement in the bit energy to noise ratio compared to the simulation of FIG. 11 when the vehicle Doppler rate is relatively high. Furthermore, the STTD curve of the pedestrian Doppler rate shows a 3 dB improvement over the solid curve without STTD. Therefore, compared to the prior art system, the STTD for TDD of the present invention greatly improves the reception state.

FIG. 13A is a block diagram of a receiver according to the present invention, in which an STTD decoder is provided before a rake receiver and a joint detector. This circuit design is similar to that of FIG. This circuit comprises STTD decoder circuits 1302-1304 corresponding to the respective multipath signals. A plurality of output signals from each STTD decoder are supplied to each rake receiver, and a multipath signal of each user is combined. The synthesized signal is supplied to a joint STTD detector circuit 1310. As before, the detected signal is used in a joint detector circuit to remove interference from other users' signals. The circuit of FIG. 13B shows an alternative embodiment of the present invention. This embodiment includes rake receivers 1312 to 1314 configured to combine the multipath signals of each user. These combined signals are applied to a joint detector and STTD decoder circuit 131.
6. Joint detector 1316 decodes the received signal of each user, subtracts an interference signal of an unintended user, and outputs a predetermined output signal Do on lead 1320. The circuit of FIG. 13C is another embodiment according to the present invention. In this embodiment, the rake receiver 1312 as described above is used.
1314 is provided. The combined signal from the rake receiver is provided to a joint detector circuit 1318 for user identification and interference cancellation. And this signal is STTD
This is supplied to a decoder 1319. The decoded output signal Do from the STTD decoder is provided from lead 1320 to the intended user.

Although the details of the invention have been described in accordance with the preferred embodiments, this description is by way of example only and is not meant as limiting. For example, the same 2L diversity can be obtained even if the symbol transmission order is changed. The use of multiple transmit or receive antennas increases the typical diversity of the present invention. Further, the novel concept of the present invention is not limited to the example circuit, and can be realized by digital signal processing as will be apparent to those skilled in the art from the present specification. For example, an alternative embodiment of the present invention with a spreading factor of 1 is equivalent to a time division multiple access (TDMA) system. Therefore, the present invention using the STTD encoded multipath signal for the reception channel equalization is based on IS-136, EDGE (Enhanced)
Data GSM Environment) and other cellular systems.

Various modifications can be made to the details of embodiments of the invention as apparent to those skilled in the art from this description. Such modifications and additions are intended to be within the spirit and scope of the following claims.

[Brief description of the drawings]

FIG. 1 is a block diagram of a transmitter of the present invention using diversity control.

FIG. 2 is a block diagram of the communication system of the present invention,
The figure which shows the communication with the mobile station in the case where diversity is applied and the case where it is not applied.

FIG. 3 is a diagram showing a TDD radio frame.

FIG. 4 is a diagram showing time slots included in the radio frame of FIG. 3;

FIG. 5A is a diagram showing an embodiment of a symbol transmission sequence using STDD encoding for TDD. B is a diagram showing an embodiment of a midamble structure used for channel estimation.

FIG. 6A is a block diagram showing a signal flow of a single user in the TDD receiver of the present invention using STTD encoding. 6B is a schematic diagram of the STTD decoder of FIG. 6A.

FIG. 7 is a block diagram showing a flow of signals of a plurality of users in the TDD receiver of the present invention using STTD encoding.

FIG. 8 is a block diagram illustrating parallel interference cancellation according to the present invention using STTD coding for TDD.

FIG. 9A shows an STTD decoder and zero forcing S
FIG. 3 is a block diagram illustrating interference cancellation by a TTD equalizer. B
FIG. 9B is a detailed block diagram of FIG. 9A, showing a zero-forcing STTD equalizer to which decision feedback has been applied.

FIG. 10A is a block diagram showing interference removal by an STTD decoder and a minimum mean square error STTD equalizer.
FIG. 10B is a detailed block diagram of FIG. 10A, showing a minimum mean square error STTD equalizer to which decision feedback has been applied.

FIG. 11 shows the bit error rate (BER) with and without the application of vehicle Doppler rate diversity with a spreading factor of 16 to the bit energy to noise ratio (Eb).
/ No).

FIG. 12 shows the bit error rate (BER) as a function of bit energy to noise ratio (Eb / No) with and without pedestrian Doppler rate diversity with a spreading factor of 16 and eight users. FIG.

FIG. 13A is a block diagram of a receiver according to the present invention in which an STTD decoder is provided before a rake receiver and a joint detector. B is a block diagram of the receiver of the present invention in which a joint detector and an STTD decoder are combined. C is a block diagram of the receiver of the present invention in which an STTD decoder is provided at a stage subsequent to the joint detector.

[Explanation of symbols]

 REFERENCE SIGNS LIST 100 Diversity control circuit 102 Lead 104 Lead 106 Lead 108 Lead 110 STTD encoder circuit 112 Transmit antenna 114 Transmit antenna 116 Multipath 118 Multipath 120 Mobile station antenna 200 STTD encoder 212 Adder circuit 216 Adder circuit 260 STTD decoder circuit 600 Matching Filter 602 Matched filter 604 Matched filter 606 STTD decoder 608 STTD decoder 610 STTD decoder 612 Rake combiner 700 Matched filter 702 Matched filter 704 Matched filter 706 STTD decoder 708 STTD decoder 710 STTD decoder 712 Rayclay Decision circuit 716 Bus 902 Whitening matched filter 904 Zero forcing STTD equalizer times 1002 Whitening matched filter 1004 Minimum mean square error STTD equalizer circuit 1302 STTD decoder 1304 STTD decoder 1306 Rake receiver 1308 Rake receiver 1310 Joint STTD detector 1312 Rake receiver 1314 Rake receiver 1316 Joint detector + STD 1318 Joint detector 1319 Joint decoder

Claims (28)

[Claims] 1. A matching circuit coupled to receive during a time slot a plurality of signals including a predetermined signal sequence inserted into the plurality of data signals, and generating an output signal in response to the data signals and the predetermined code. A circuit having a filter circuit and a decoding circuit connected to receive an output signal including a first data symbol and a conversion of the second data symbol, the decoding circuit generating a first decoded data symbol and a second decoded data symbol. . 2. The circuit according to claim 1, wherein the first decoded data symbol and the second decoded data symbol are output from the decoding circuit in response to the conversion of the first data symbol and the second data symbol, respectively. Circuit to do. 3. The circuit according to claim 2, wherein the conversion of the second data symbol is a complex conjugate of the second data symbol. 4. The circuit according to claim 1, wherein the predetermined signal includes a midamble. 5. The circuit of claim 1, wherein the matched filter circuit further includes a plurality of fingers connected to receive the plurality of signals, each finger corresponding to each path of the plurality of signals. A circuit for generating an output signal from each finger. 6. The circuit according to claim 5, wherein an output signal at each finger of the plurality of fingers includes a plurality of output signals, each of the plurality of output signals corresponding to each shift of a code of a predetermined signal. circuit. 7. The circuit of claim 6, wherein each shift of the code of the predetermined signal is a shifted sample of the code sequence. 8. The circuit of claim 6, further connected to receive a plurality of output signals from each finger, each outputting a first decoded symbol and a second decoded symbol, respectively. A plurality of decoding circuits; a first decoding symbol and a second decoding symbol from each decoding circuit;
A joint detector circuit connected to receive the decoded symbols, and receiving and mixing the first decoded symbol and the second decoded symbol from the respective fingers corresponding to the respective codes. circuit. 9. The circuit of claim 8, wherein said joint detector attenuates interference by removing parallel interference. 10. The circuit according to claim 8, wherein the joint detector has zero forcing interference cancellation (zero).
forcing interference can
circuit that attenuates interference by means of a cell. 11. The circuit according to claim 8, wherein said joint detector includes a minimum mean square error interference canceller (mini).
mum mean squared interfer
circuit that attenuates the interference by means of an interference cancellation. 12. The circuit according to claim 1, wherein a predetermined signal in each time slot corresponds to each user, and the predetermined signal corresponding to each user is encoded by shifting a code sequence. 13. A matched filter circuit, each corresponding to a respective path of the data symbol, connected to receive the data symbol, and including a plurality of fingers for generating respective output signals; and from each of the plurality of fingers. A plurality of decoder circuits respectively connected to receive the output signals of the plurality of decoder circuits respectively outputting the output signals; and a plurality of decoder circuits connected to receive the respective output signals from the plurality of decoder circuits. And a joint detector circuit for generating an output signal to be output. 14. The circuit according to claim 13, wherein said predetermined code corresponds to a mobile receiver. 15. The circuit according to claim 13, wherein said predetermined code is a subset of a code sequence corresponding to a plurality of mobile receivers. 16. The circuit of claim 13, further comprising a plurality of RAKE receiver circuits each connected between each decoder circuit and the joint detector circuit. 17. A matched filter circuit, each corresponding to a respective path of the data symbol, connected to receive the data symbol and including a plurality of fingers for producing respective output signals, and a plurality of output signals from each of the plurality of fingers. A plurality of rake receiver circuits, each connected to receive an output signal, each of which outputs an output signal; and a plurality of rake receiver circuits connected to receive each output signal from the plurality of rake receiver circuits. And a joint detector circuit for generating an output signal corresponding to the following. 18. The circuit according to claim 17, wherein said predetermined code corresponds to a mobile receiver. 19. The circuit according to claim 17, wherein said predetermined code is a subset of a code sequence corresponding to a plurality of mobile receivers. 20. The circuit according to claim 17, wherein said joint detector circuit further comprises a decoding circuit. 21. The circuit of claim 17, further comprising a decoder circuit connected to receive the signal. 22. A circuit having an encoder circuit connected to receive a plurality of first and second symbols, the encoder circuit receiving a plurality of first symbols from a first output terminal during a time slot. And outputs a conversion of a plurality of second symbols from a second output terminal, and outputs a sequence of predetermined signals inserted into the plurality of first symbols during the time slot from an encoder circuit. circuit. 23. The circuit of claim 22, wherein the encoder circuit is connected to receive a control signal, and wherein when the control signal is a first value, a plurality of first symbols are output from a first output terminal. Outputting a plurality of conversions of a plurality of second symbols from a second output terminal, and when the control signal has a second value,
A circuit that outputs a plurality of first symbols from a first output terminal but does not output conversions of a plurality of second symbols from a second output terminal. 24. The circuit of claim 22, further comprising a first diversity control circuit connected to receive the first input signal and generating a control signal corresponding to the input signal. 25. The circuit according to claim 22, wherein said input signal corresponds to a Doppler frequency. 26. The circuit according to claim 22, further comprising a diversity control circuit connected to receive the second input signal corresponding to the handoff signal. 27. The circuit according to claim 22, wherein said first input signal corresponds to a handoff signal. 28. The circuit according to claim 22, wherein said decoder circuit generates a midamble after a first symbol and before a second symbol.