JP2000286643A - Frequency converter circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To obtain a frequency converter circuit which can effectively suppress odd order harmonic distortion, such as the third harmonic distortion, etc., and can obtain a sufficient conversion gain. SOLUTION: A frequency converter circuit is composed of three multipliers 11, 12, and 13 which multiply high-frequency signals by local oscillation signals, an adder 14, and a three-phase local signal generator 20. The circuit inputs the high-frequency signals, the phases of which are shifted 60 deg. by 60 deg. and the local oscillation signals to the multipliers 11, 12, and 13 and generates output signals by adding up the output signals of the multipliers 11, 12, and 13 by means of the adder 14.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to a frequency conversion circuit suitable for a modulator or the like of a radio communication device.

[0002]

2. Description of the Related Art A double balance mixer is known as a modulator used in a radio communication device or the like. The double balance mixer is a frequency conversion circuit composed of two multipliers for multiplying a high-frequency signal and a local oscillation signal (local signal), and is generally configured as shown in FIG.

That is, transistors Q1, Q2, Q5
To Q8, current sources I1 and I2, and degeneration resistor R3 to form a first multiplier, and transistors Q3, Q4, Q9 to Q
12, a current multiplier I3, I4 and a degeneration resistor R4 constitute a second multiplier. The transistors Q1 and Q2 and the transistors Q5 to Q8 and the transistors Q3 and Q4 and the transistors Q9 to Q12 are connected in a vertically stacked manner between the power supply Vcc and the ground via common load resistors R1 and R2.

A phase of 0 ° is applied between base electrodes of transistors Q1 and Q2, which are RF input terminals D1 and D2 of a first multiplier.
Are input to the LO input terminals L1, L2
A differential local signal having a phase of 0 ° is input between the base electrodes of the transistors Q5 and Q8 and the base electrodes of the transistors Q6 and Q7. Further, a differential high-frequency signal having a phase of 90 ° is input between base electrodes of transistors Q3 and Q4, which are RF input terminals D3 and D4 of the second multiplier.
Transistors Q9 and Q12 which are O input terminals L3 and L4
, And a differential local signal having a phase of 90 ° is input between the base electrodes of the transistors Q10 and Q11.

Each output signal from the first and second multipliers is a modulated signal. By adding the two, the signals are extracted through an emitter follower circuit including transistors Q13 and Q14 and resistors R5 and R6. , The desired modulated output signal in which the image signal is suppressed is output to output terminals O1 and O2.
2 are obtained.

In such a double-balanced mixer, transistors having completely matched characteristics are used, and an appropriate bias level is applied to each transistor. In principle, leakage of an image signal or a local signal to a modulation output signal is prevented. Does not occur. However, a distortion component is superimposed on the modulation output signal due to the nonlinearity of the transistor. When the distortion component is superimposed on the output signal, in the case of the modulator, the modulation accuracy deteriorates, and the output signal quality deteriorates.

Generally, in the case of a double balance mixer, the current gain ΔI is expressed by the following equation.

[0008]

(Equation 1)

Here, C is a constant, VLO is a local signal voltage, Vin is a high-frequency signal voltage, and Ve is degeneration resistors R3 and R4.
Is a voltage generated at both ends of tanhx @ xx- ³ /
3, so each signal

[0010]

(Equation 2)

Substituting into equation (1) as

[0012]

(Equation 3)

## EQU1 ## In Equation (3), the first term is a desired wave term, and the second term is a term representing third-order distortion.

Among modulators, a modulator used in a direct conversion type high frequency receiver for directly converting an input high frequency signal into a baseband signal generates a third harmonic of a baseband signal near a desired frequency. Therefore, the third-order distortion is a problem because the modulation accuracy is deteriorated and the adjacent channel is spurious.

Generally, the desired output signal increases in proportion to the input signal power, whereas the third-order distortion is 3 times the input signal power.
Conventionally, in order to suppress the third-order distortion, a method of suppressing the input signal power to some extent and lowering the output signal power to reduce the distortion level has been adopted. However, in this method, it is necessary to increase the gain of the subsequent amplifier in order to secure the required gain, and there arises a problem that the power consumption increases and the number of elements increases the cost.

Further, by increasing the resistance values of the degeneration resistors R3 and R4 and increasing the terminal voltage Ve, it is possible to suppress the third-order distortion even when the power level of the input signal is high. The noise level increases and N
F (noise figure) is degraded.

[0017]

As described above, the third-order distortion, which is a main factor of the distortion of the modulator, increases when the output power is increased. Therefore, conventionally, the output power of the modulator is suppressed and the amplifier at the subsequent stage is suppressed. The method of earning a gain is used,
There has been a problem that power consumption is increased and cost is increased due to an increase in the number of elements. On the other hand, a method of suppressing the third-order distortion by increasing the terminal voltage of the degenerate resistor used in the multiplier is NF
There is a problem that it deteriorates.

SUMMARY OF THE INVENTION The present invention has been made to solve such a conventional problem, and it is possible to effectively suppress odd-order distortion such as third-order distortion and obtain a sufficient conversion gain. An object of the present invention is to provide a frequency conversion circuit suitable for a modulator that can be used.

[0019]

In order to solve the above problems, a frequency conversion circuit according to the present invention comprises n multipliers for multiplying a high frequency signal and a local oscillation signal (where n is an integer of 3 or more). ), A high-frequency signal having a phase shift of 180 ° / n and a local oscillation signal having a phase shift of 180 ° / n are input to each multiplier, and the output signals of the multipliers are added and synthesized. Is obtained.

Here, the number n of multipliers is typically n =
3, in which case the high-frequency signal and the local oscillation signal, which are shifted in phase by 60 ° from each other,
° phase high frequency signal and local oscillation signal
A high-frequency signal and a local oscillation signal shifted by 60 ° and a high-frequency signal and a local oscillation signal similarly shifted by −60 ° are input.

The case where n = 3 will be described as an example. Assuming that the local signal is VLO and the high frequency signal is Vin, each signal wave is

[0022]

(Equation 4)

## EQU2 ## Further, the local signal LO,
The signal waves of the high-frequency signal Vin shifted by + 60 ° and -60 ° are

[0024]

(Equation 5)

## EQU2 ##

Examining the effect of the present invention with respect to the third-order distortion of the second term in equation (3), equation (4) is

[0027]

(Equation 6)

Is as follows. Similarly, for equations (5) and (6),

[0029]

(Equation 7)

## EQU2 ##

Therefore, when these three multipliers are connected in parallel to form a frequency conversion circuit (modulator), the third-order distortion contained in the output signal is expressed by the following equations (7), (8) and (9). This is the sum and becomes 0.

In the frequency conversion circuit according to the present invention, which is configured by connecting three multipliers in parallel, no third-order distortion is generated in principle. Similarly, image components included in the output signal do not occur in principle. Therefore, the frequency conversion circuit of the present invention can obtain an ideal frequency conversion output signal (modulation output signal) from which distortion has been completely removed.

[0033]

Embodiments of the present invention will be described below with reference to the drawings. FIG. 1 is a block diagram illustrating an example in which a frequency conversion circuit according to an embodiment of the present invention is configured as a modulator. This modulator comprises three multipliers 11, 12, 1
3 and the adder 14.

One of the input terminals of the multipliers 11, 12, and 13 has a phase of 0 ° from the input terminals 1, 2, and 3, respectively.
High frequency signals (RF signals) of 60 ° and 120 ° are input. The other input terminal of each of the multipliers 11, 12, and 13 has a phase of 0 ° from the three-phase local signal generator 20,
Local signals of 60 ° and 120 ° are input, respectively. The output signals of the multipliers 11, 12, and 13 are added and synthesized by the adder 14, and output from the output terminal 4 as a modulated output signal.

The three-phase local signal generator 20 includes two 60 ° phase shifters 22 cascaded with a local oscillator 21.
And a local signal having a phase of 0 ° from the output of the local oscillator 21, a local signal having a phase of 60 ° from the output of the first-stage 60 ° phase shifter 22, and an output of the second-stage 60 ° phase shifter 23 , Local signals having a phase of 120 ° are respectively extracted.

Next, the operation of the modulator of this embodiment will be described. Now, it is assumed that a signal wave of a 10 MHz band is input to the multipliers 11, 12, and 13 as an RF signal, and a signal wave of 2 GHz is input as a local signal. in this case,
As is apparent from equation (3), the modulated output signal is 1.99.
GHz, and the third-order distortion at each output terminal of the multipliers 11, 12, and 13 occurs at 2.03 GHz.

Here, the third-order distortions generated at the output terminals of the multipliers 11, 12, and 13 are 0 °, 60 °, and 120 °, respectively.
Is maintained, the composite wave of the two signal waves has an opposite-phase relationship with the remaining one signal wave. Therefore, when the output signals of the multipliers 11, 12, and 13 are added by the adder 14, the tertiary distortion is canceled in the output signal of the adder 14, and no signal is generated in principle.

FIG. 2 shows a specific circuit configuration of the multipliers 11, 12, and 13 and the adder 14 in FIG. This circuit has three balanced multipliers in parallel, and can be said to be a triple balanced mixer as compared with a conventional double balanced mixer.

Transistors Q1, Q2, Q11-Q14
, Current sources I1 and I2 and degeneration resistor R3 constitute a first multiplier, and transistors Q3, Q4, Q15-Q18
, Current sources I3, I4 and degeneration resistor R4 constitute a second multiplier, and transistors Q5, Q6, Q19-Q22
, The current sources I5 and I6 and the degeneration resistor R5 constitute a third multiplier.

Transistors Q1, Q2 and transistor Q
The transistors Q11 to Q14, the transistors Q3 and Q4 and the transistors Q15 to Q18, and the transistors Q5 and Q6 and the transistors Q19 to Q22 are connected in a vertically stacked manner between the power supply Vcc and the ground via common load resistors R1 and R2. ing.

The RF input terminals D1, D of the first multiplier 11
2, a differential RF signal having a phase of 0 ° is input between the base electrodes of the transistors Q1 and Q2, and the LO input terminals L1 and L
2 between the base electrodes of the transistors Q11 and Q14 and the base electrodes of the transistors Q12 and Q13.
° differential local signal is input.

The RF input terminal D of the second multiplier 12
A differential RF signal having a phase of 60 ° is input between the base electrodes of the transistors Q3 and Q4, which are D3 and D4, and the base electrodes of the transistors Q15 and Q18 and the base electrodes of the transistors Q16 and Q17, which are LO input terminals L3 and L4. During this period, a differential local signal having a phase of 60 ° is input.

The RF input terminal D of the third multiplier 13
5, a differential RF signal having a phase of 120 ° is input between the base electrodes of the transistors Q5 and Q6, and the LO input terminals L3 and L4 of the base electrodes of the transistors Q19 and Q22 and the base electrodes of the transistors Q20 and Q21. During this period, a differential local signal having a phase of 120 ° is input.

As described above, in the present embodiment, each multiplier is R
Transistors Q1 to Q6 to which an F signal is input and transistors Q11 to Q22 to which a local signal is input are stacked in cascade, and further, transistors Q1 and Q2, transistors Q3 and Q4, transistors Q5 and Q6, transistors Q11 and Q12, and transistors Q13 and Q14, transistors Q15 and Q16, transistors Q17 and Q18,
Transistors Q19 and Q20, Transistors Q21 and Q
Numerals 22 each constitute a differential circuit.

First, second and third multipliers 11, 12, 1
Each output signal from 3 is a modulated signal;
After these are added by common load resistors R1 and R2, they are taken out via an emitter follower circuit composed of transistors Q31 and Q32 and resistors R6 and R7 connected to their respective emitter electrodes, whereby the image signal is suppressed. The desired modulated output signal is obtained from output terminals O1 and O2.

Next, the effect of the modulator according to the present invention will be described with reference to specific data examples.

FIG. 3 shows a 10 MHz signal wave as an RF signal and a 2 GHz signal as a local signal in the modulator of this embodiment.
3 shows a modulated output signal waveform when each signal wave is input. It can be seen that the modulated output signal has a large power output with an amplitude of about 150 mV, but a good waveform with almost no distortion is obtained.

FIG. 6 shows a modulated output signal waveform of the conventional double balance mixer shown in FIG. 5 for comparison. The amplitudes of the RF signal and the local signal are the same as in FIG. However, FIGS. 3 and 6 compare the current consumption per multiplier as the same. As shown in FIG. 6, the conventional double-balanced mixer has a waveform with extremely large distortion even though the output signal amplitude is 100 mV or less.

FIG. 4 shows the result of Fourier transform of the modulated output signal waveform of the modulator of this embodiment. The spectrum at 2.03 GHz, which is the frequency at which the third-order distortion occurs, is below the noise level and is sufficiently suppressed. Further, it can be seen that the image spectrum and the local leak did not occur.

On the other hand, according to the Fourier transform result of the modulation output signal waveform of the conventional double balance mixer, as shown in FIG. 7, a spectrum of 2.03 GHz, which is the frequency of the third-order distortion, is generated at about 0.1 mV. It can be seen that distortion is largely included.

In the above embodiment, an example in which three multipliers are arranged in parallel has been described. However, four or more multipliers may be arranged in parallel. For example, using 5 multipliers, 36 °
By inputting the RF signal and the local signal, each of which is out of phase, an output signal in which the fifth-order distortion is suppressed can be obtained.

Further, the present invention is not limited to a modulator, but can be applied to, for example, a demodulator and can be widely applied to a frequency conversion circuit.

[0053]

As described above, the frequency conversion circuit of the present invention uses three or more multipliers, and each multiplier has 180
By inputting an RF signal and a local signal having a phase difference obtained by dividing ° by the number of multipliers, and synthesizing their outputs to obtain a frequency-converted output signal, 3
It is possible to completely remove odd-order distortions such as second-order distortions and images.

Further, in the frequency conversion circuit of the present invention, since the conversion gain can be easily increased and a high-power output signal can be obtained, it is not necessary to increase the gain of the subsequent-stage amplifier more than necessary. The problem of an increase in cost due to an increase in the number of elements can be avoided.

Further, in the present invention, the NF is not deteriorated unlike the method of suppressing the third-order distortion by increasing the terminal voltage of the degeneration resistor used in the multiplier.

[Brief description of the drawings]

FIG. 1 is a block diagram showing an example in which a frequency conversion circuit according to one embodiment of the present invention is applied to a modulator.

FIG. 2 is a circuit diagram showing a more specific configuration of the modulator according to the embodiment.

FIG. 3 is a view showing a modulation output signal waveform of the modulator according to the embodiment.

FIG. 4 is a diagram showing a Fourier transform result of a modulated output signal waveform of the modulator according to the embodiment.

FIG. 5 is a circuit diagram showing a configuration of a double balance mixer based on a conventional technique.

FIG. 6 is a diagram showing a modulation output signal waveform of a double balance mixer based on the prior art.

FIG. 7 is a diagram showing a Fourier transform result of a modulation output signal waveform of a double balance mixer based on the prior art.

[Explanation of symbols]

 1, 2, 3, RF signal input terminal 4, modulation signal output terminal 11, 12, 13, multiplier 14, adder 20, three-phase local signal generator 21, local oscillator 22, 23, 60 ° phase shifter Q1 to Q32: Transistors R1 to R7: Resistors L1, L2: 0 ° phase local signal input terminals L3, L4: 60 ° phase local signal input terminals L5, L6: 120 ° phase local signal input terminals D1, D2: 0 ° phase RF signal input terminals D3, D4... 60 ° phase RF signal input terminals D5, D6... 120 ° phase RF signal input terminals O1, O2...

Claims (4)

[Claims] An n-number (where n is an integer of 3 or more) multiplier for multiplying a high-frequency signal and a local oscillation signal, and a phase shift of 180 ° / n with respect to the n-numbered multipliers. High-frequency signal input means for inputting a high-frequency signal; local oscillation signal input means for inputting local oscillation signals 180 ° / n out of phase to each of the n multipliers; and output signals of the n multipliers And a signal output means for adding and combining the signals. 2. Multipliers for multiplying a high-frequency signal by a local oscillation signal; high-frequency signal input means for inputting high-frequency signals having a phase shift of 60 ° to each other to the three multipliers; Local oscillation signal input means for inputting local oscillation signals having a phase shift of 60 ° to n multipliers, and signal output means for adding and synthesizing output signals of the three multipliers and outputting the result. The frequency conversion circuit characterized by the above. 3. The frequency according to claim 1, wherein the multiplier is configured by cascading a transistor to which the high-frequency signal is input and a transistor to which the local oscillation signal is input. Conversion circuit. 4. The frequency converter according to claim 3, wherein said multiplier is configured such that a transistor to which said high-frequency signal is inputted and a transistor to which said local oscillation signal is inputted respectively constitute a differential circuit. Conversion circuit.