JP2000102168A - Active filter control method - Google Patents
Active filter control methodInfo
- Publication number
- JP2000102168A JP2000102168A JP10267778A JP26777898A JP2000102168A JP 2000102168 A JP2000102168 A JP 2000102168A JP 10267778 A JP10267778 A JP 10267778A JP 26777898 A JP26777898 A JP 26777898A JP 2000102168 A JP2000102168 A JP 2000102168A
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- JP
- Japan
- Prior art keywords
- current
- load
- component
- phase
- receiving end
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
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Classifications
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
- Y02E40/00—Technologies for an efficient electrical power generation, transmission or distribution
- Y02E40/20—Active power filtering [APF]
Landscapes
- Networks Using Active Elements (AREA)
- Supply And Distribution Of Alternating Current (AREA)
- Control Of Electrical Variables (AREA)
- Power Conversion In General (AREA)
Abstract
Description
【０００１】[0001]
【発明の属する技術分野】本発明は、電力系統に接続さ
れた負荷側で発生する高調波無効電力及び不平衡を打ち
消すように補償するアクティブフィルタ制御方法に関す
る。BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an active filter control method for compensating so as to cancel out harmonic reactive power and imbalance generated on a load side connected to a power system.
【０００２】[0002]
【従来の技術】半導体応用機器の普及により電力系統の
高調波が問題となっている。これに加えて、電力系統の
不平衡が将来問題となる可能性があることが論文などで
指摘され、不平衡と高調波を一括して補償する補償装置
が、重電機メーカの発表する論文などで提案されるとと
もに系統安定用として実用化されつつある。これらの補
償装置は、いわゆる無効電力補償装置ＳＶＣに不平衡と
高調波の補償機能を付加したものと考えることができ
る。従来、この様な目的として、平成７年電気学会全国
大会一般講演Ｎｏ．８１０及び平成５年電気学会全国大
会一般講演Ｎｏ．６０４等で発表されているように、そ
の制御方式は、表現方法は異なるものの、概ね三相ａ−
ｂ−ｃ座標の電流を正相変換し、フィルタを用いて高調
波分と基本波分を分離し、正相分の有効無効電流を検出
する。他方、逆相変換を行って同様のことを行う。これ
らから、高調波電流及び逆相電流を検出し、これを指令
値として不平衡と高調波の補償を行うものである。従っ
て、これまでの方法は、少なくとも正相変換及び逆相変
換の２系統の座標変換を行う必要があった。2. Description of the Related Art Harmonics in power systems have become a problem due to the spread of semiconductor application equipment. In addition, it has been pointed out in papers and others that imbalance in the power system may be a problem in the future, and a compensator that collectively compensates for imbalance and harmonics has been published in a paper published by heavy equipment manufacturers. And it is being put to practical use for system stabilization. These compensating devices can be considered as those obtained by adding a function of compensating for unbalance and harmonics to the so-called reactive power compensating device SVC. Conventionally, for such purpose, a general lecture No. 810 and 1993 IEEJ General Conference No. As disclosed in 604, etc., the control method is generally three-phase a-
The current of the bc coordinate is subjected to positive phase conversion, and a harmonic component and a fundamental component are separated using a filter, and an active / reactive current for the positive phase is detected. On the other hand, the same is performed by performing an inverse phase conversion. From these, a harmonic current and a negative-sequence current are detected, and these are used as command values to compensate for unbalance and harmonics. Therefore, in the conventional methods, it is necessary to perform at least two systems of coordinate conversion, ie, normal phase conversion and reverse phase conversion.
【０００３】[0003]
【発明が解決しようとする課題】本発明は、高調波およ
び不平衡を一括補償する際に、２系統の座標変換が必要
であったものを、１系統の座標変換だけを行って負荷電
流の基本波正相分の有効電流だけを検出し、これと負荷
電流の差を一括補償することにより、個別の補償対象成
分を個々に検出することのない、簡易なアクティブフィ
ルタ制御方法を提供することを目的とする。SUMMARY OF THE INVENTION According to the present invention, when the harmonics and unbalance are collectively compensated, two systems of coordinate conversion are required, but only one system of coordinate conversion is performed to reduce the load current. To provide a simple active filter control method that detects only an active current for a positive phase of a fundamental wave and collectively compensates for a difference between the active current and a load current so that individual components to be compensated are not individually detected. With the goal.
【０００４】[0004]
【課題を解決するための手段】本発明は、電力系統に接
続された負荷側で発生する高調波、無効電力及び不平衡
を打ち消すように補償するアクティブフィルタ制御方法
において、受電端における該負荷に流入する電流の基本
波正相分の有効電流を検出し、該電流の基本波正相分の
有効電流以外の成分を一括補償することにより、電源側
で正弦波平衡三相状態にすることを特徴とするアクティ
ブフィルタ制御方法を提供するものである。SUMMARY OF THE INVENTION The present invention provides an active filter control method for compensating so as to cancel out harmonics, reactive power and unbalance generated on the load side connected to a power system. By detecting the effective current of the in-phase of the fundamental wave of the inflowing current and compensating components other than the effective current of the in-phase of the fundamental wave of the current at a time, it is possible to make the sine wave balanced three-phase state on the power supply side. An object of the present invention is to provide a featured active filter control method.
【０００５】また、本発明は、前記の一括補償する際
に、前記受電端の電圧を実際の電気角を検出することな
く電源の基本周波数と等しい鋸歯状波を用いてｄ−ｑ座
標上へ変換し、ローパスフィルタを用いて極低周波分を
含む直流分を分離し、該直流分のｄ軸成分とｑ軸成分の
比の逆正接を求め、該逆正接と該鋸歯状波の和を求める
ことにより該受電端電圧の基本波の電気角を求め、該電
気角を用いて負荷電流をｄ−ｑ座標上に変換し、別のロ
ーパスフィルタを用いてｄ軸上の極低周波を含む直流分
を分離し、該電気角を用いてａ−ｂ−ｃ座標上に変換す
ることにより負荷電流の基本波正相分の有効電流を検出
し、該負荷に流入する電流の基本波正相分の有効電流と
負荷電流との差を求め、アクティブフィルタの指令値と
することを特徴とするアクティブフィルタ制御方法を提
供するものである。Further, according to the present invention, at the time of the collective compensation, the voltage at the power receiving end is converted to dq coordinates using a sawtooth wave equal to the fundamental frequency of the power supply without detecting the actual electrical angle. The DC component including the extremely low frequency component is separated using a low-pass filter, the arc tangent of the ratio of the d-axis component to the q-axis component of the DC component is obtained, and the sum of the arc tangent and the sawtooth wave is calculated. Calculate the electrical angle of the fundamental wave of the receiving end voltage, convert the load current into dq coordinates using the electrical angle, and use the low-pass filter to include the extremely low frequency on the d axis. The DC component is separated and converted into an abc coordinate using the electrical angle to detect an effective current corresponding to a fundamental wave positive phase of a load current, and a fundamental wave positive phase of a current flowing into the load is detected. The difference between the active current and the load current is calculated as the command value of the active filter. There is provided an active filter control method.
【０００６】[0006]
【発明の実施の形態】以下、図１〜３を参照して、本発
明の実施例について説明する。DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS An embodiment of the present invention will be described below with reference to FIGS.
【０００７】図１は、本実施例の高調波および不平衡の
同時補償法を示すアクティブフィルタの構成図を示す。
図において、１は平衡三相電源、２は電源インピーダン
ス、３は不平衡負荷、４はサイリスタ整流回路、５はサ
イリスタ整流回路の出力電流、６はフィルタリアクト
ル、７はアクティブフィルタを示す。該平衡三相電源１
には、Ｒ−Ｌ回路から構成される該不平衡負荷３と高調
波の発生源である該サイリスタ整流回路４が接続され、
これらと並列に該アクティブフィルタ７が接続された構
成になっている。FIG. 1 is a configuration diagram of an active filter showing a method for simultaneously compensating for harmonics and unbalance according to the present embodiment.
In the figure, 1 is a balanced three-phase power supply, 2 is a power source impedance, 3 is an unbalanced load, 4 is a thyristor rectifier circuit, 5 is an output current of the thyristor rectifier circuit, 6 is a filter reactor, and 7 is an active filter. The balanced three-phase power supply 1
Is connected to the unbalanced load 3 composed of an RL circuit and the thyristor rectifier circuit 4 that is a source of harmonics.
The active filter 7 is connected in parallel with these components.
【０００８】図２は、本実施例のアクティブフィルタ制
御方式のブロック図を示す。以下に、図１、図２を参照
して、制御方法について説明する。図１において、Ｖｓ
は電源電圧、Ｉｓは電源電流、Ｖｔは受電端電圧、Ｉｌ
は負荷側電流、Ｉｃはアクティブフィルタ７の出力電流
（補償電流）、Ｖｄはアクティブフィルタ７の直流電
圧、Ｚｌは負荷インピーダンス、Ｉｄｃはサイリスタ整
流回路の出力電流を示す。図２において、Ｖｔａ、Ｖｔ
ｂ、Ｖｔｃは各相の受電端電圧で、Ｉｌａ、Ｉｌｂ、Ｉ
ｌｃは各相の負荷側電流である。これら受電端電圧を実
際の電気角θｖを検出することなく周波数６０Ｈｚの理
想的な鋸歯状波θｒを用いてｄ−ｑ座標へ変換する。こ
のとき、受電端電圧のｄ軸成分Ｖｔｄおよびｑ軸成分Ｖ
ｔｑは、下記の数１の式で与えられる。また、数２は、
ａ−ｂ−ｃ座標からｄ−ｇ座標への変換行列Ｃ1 の計算
式を示す。FIG. 2 is a block diagram showing an active filter control system according to this embodiment. The control method will be described below with reference to FIGS. In FIG. 1, Vs
Is the power supply voltage, Is is the power supply current, Vt is the receiving end voltage, Il
Is the load side current, Ic is the output current (compensation current) of the active filter 7, Vd is the DC voltage of the active filter 7, Zl is the load impedance, and Idc is the output current of the thyristor rectifier circuit. In FIG. 2, Vta, Vt
b, Vtc are the receiving end voltages of each phase, and Ila, Ilb, I
lc is a load-side current of each phase. These receiving end voltages are converted to dq coordinates using an ideal sawtooth wave θr having a frequency of 60 Hz without detecting the actual electrical angle θv. At this time, the d-axis component Vtd and the q-axis component V
tq is given by the following equation (1). Equation 2 is
A formula for calculating a transformation matrix C1 from abc coordinates to dg coordinates is shown below.
【０００９】 このとき、ｄ−ｑ座標のｄ軸成分Ｖｔｄおよびｑ軸成分
Ｖｔｑは、電源周波数に比較し極低周波数を含む直流分
と交流分に分解することが出来る。この極低周波分を含
む直流分をＬＰＦ１（ローパスフィルタ）を用いて分離
し、逆ｄ−ｑ変換することで、電源周波数が微妙に変動
した場合でも基本波分を検出することが出来る。これと
同様に、負荷電流についてもｄ−ｑ座標上へ変換するこ
とで、正相分を検出できる。しかしながら、受電端電圧
の電気角θｖと鋸歯状波θｒに位相差が存在するため、
受電端電圧の正相分と同相成分を検出することが出来な
い。そこで、ｄ−ｑ座標上でＬＰＦを用いて検出したＶ
ｔｄ及びＶｔｑに着目すると、受電端電圧の電気角θｖ
は下記の数３の式で与えられる。[0009] At this time, the d-axis component Vtd and the q-axis component Vtq of the dq coordinate can be decomposed into a DC component and an AC component including a very low frequency compared to the power supply frequency. The DC component including the extremely low frequency component is separated using an LPF1 (low-pass filter) and subjected to inverse dq conversion, so that the fundamental component can be detected even when the power supply frequency slightly changes. Similarly, by converting the load current into dq coordinates, the positive phase component can be detected. However, since there is a phase difference between the electric angle θv of the receiving end voltage and the sawtooth wave θr,
The in-phase component and the in-phase component of the receiving end voltage cannot be detected. Therefore, V detected on the dq coordinates using an LPF
Focusing on td and Vtq, the electrical angle θv of the receiving end voltage
Is given by the following equation (3).
【００１０】 数３において、Ｖｔｄ1は受電端電圧のｄ軸電圧の直流
分で、Ｖｔｑ1は受電端電圧のｑ軸電圧の直流分を示
す。数３を用いて負荷電流をｄ−ｑ座標上へ変換すると
下記の数４の式となる。[0010] In Equation 3, Vtd1 is a DC component of the d-axis voltage of the receiving end voltage, and Vtq1 is a DC component of the q-axis voltage of the receiving end voltage. When the load current is converted into dq coordinates using Equation 3, the following Equation 4 is obtained.
【００１１】 数４で、負荷電流のｄ軸成分Ｉｌｄ、負荷電流のｑ軸成
分ＩｌｑからＬＰＦ2を用いてｄ軸成分の直流分Ｉｌｄ1
のみを抽出し、これらを再びａ−ｂ−ｃ座標上へ変換す
ることで受電端電圧の正相分Ｖｔｐと同相の負荷電流の
正相分Ｉｌａｐ、Ｉｌｂｐ、Ｉｌｃｐを検出できる。こ
のとき、前記Ｉｌａｐ、Ｉｌｂｐ、Ｉｌｃｐは下記の数
５の式で与えられる。数６はｄ−ｑ座標からａ−ｂ−ｃ
座標への変換行列Ｃ2の計算式を示す。[0011] In Equation 4, the DC component Ild1 of the d-axis component is obtained from the d-axis component Ild of the load current and the q-axis component Ilq of the load current using LPF2.
Only the positive phase components Ilap, Ilbp, and Ilcp of the load current having the same phase as the positive phase component Vtp of the power receiving terminal voltage can be detected by extracting only the positive polarity component and the positive polarity component Vtp of the power receiving end voltage. At this time, Ilap, Ilbp, and Ilcp are given by the following equation (5). Equation 6 is abc from dq coordinates
The calculation formula of the conversion matrix C2 into coordinates is shown.
【００１２】 今、負荷電流Ｉｌと検出した負荷電流の正相分Ｉｌｐと
の差を求め、これをアクティブフィルタ７の指令値とす
る。これにより、高調波、無効電力および不平衡を一括
して補償できる。このとき、サイリスタ整流回路の各相
への指令値ＩＣａ、ＩＣｂ、ＩＣｃは、下記の数７の式
に示す値となる。[0012] Now, a difference between the load current Il and the positive-phase component Ilp of the detected load current is obtained, and is set as a command value of the active filter 7. This makes it possible to collectively compensate for harmonics, reactive power, and imbalance. At this time, the command values ICa, ICb, and ICc for each phase of the thyristor rectifier circuit are values shown in the following equation (7).
【００１３】 本システムは、補償対象とする電気量を検出するのでは
なく電源側でどの様な電流波形が望ましいかということ
に着目している点に特色がある。[0013] The present system is characterized in that it focuses on what kind of current waveform is desirable on the power supply side instead of detecting the amount of electricity to be compensated.
【００１４】次に、本発明の制御方法の有効性を確認す
るために、計算機シミュレーションを行った。このと
き、不平衡負荷の一例であるＲ−Ｌ負荷の定数を下記の
表１に示す。Next, a computer simulation was performed to confirm the effectiveness of the control method of the present invention. At this time, the constant of the RL load, which is an example of the unbalanced load, is shown in Table 1 below.
【００１５】 表１においてＺ１は不平衡負荷インピーダンス、Ｒａ〜
Ｒｃは各相の負荷抵抗、Ｌａ〜Ｌｃは各相の負荷リアク
トル、Ｉｄｃはサイリスタ整流回路の出力電流、αはサ
イリスタ整流回路の位相制御角である。アクティブフィ
ルタ７の主回路は、ヒステリシスコンパレータ方式ＰＷ
Ｍインバータから構成されている。また、ｄ−ｑ座標上
における直流分抽出のＬＰＦには２次バタワース形を使
用し、カットオフ周波数を５Ｈｚとした。図３にシミュ
レーション結果を示す。図３において、添字”ｓ”は電
源側、”ｔ”は受電端、”ｌ”は負荷側を表している。
一番上はａ相、２番目はｂ相、そして３番目はｃ相の電
圧および電流を示し、一番下はａ、ｂ、ｃ各相の負荷側
電流を示し、いずれも横軸が経過時間である。図から解
るように、不平衡Ｒ−Ｌ負荷と高調波発生源であるサイ
リスタ整流回路が接続されているために、線Ｉｌａ、Ｉ
ｌｂおよびＩｌｃは不平衡状態で高調波電流を含んでい
る。一方、電源側において電源電流Ｉｓａ、Ｉｓｂおよ
びＩｓｃは高調波電流が補償されており、且つ、平衡三
相となっている。また、受電端電圧Ｖｔａ、Ｖｔｂ、Ｖ
ｔｃに比較し、電源電流はそれぞれ同位相となってお
り、電源側で無効電流が補償されていることが確認でき
る。[0015] In Table 1, Z1 is an unbalanced load impedance, Ra to
Rc is the load resistance of each phase, La to Lc are the load reactors of each phase, Idc is the output current of the thyristor rectifier circuit, and α is the phase control angle of the thyristor rectifier circuit. The main circuit of the active filter 7 is a hysteresis comparator type PW
It consists of M inverters. A second-order Butterworth type was used as the LPF for DC component extraction on the dq coordinates, and the cutoff frequency was 5 Hz. FIG. 3 shows a simulation result. In FIG. 3, the suffix “s” indicates the power supply side, “t” indicates the power receiving end, and “l” indicates the load side.
The top shows the voltage and current of the a phase, the second shows the b phase, and the third shows the voltage and current of the c phase, and the bottom shows the load side current of each phase of a, b, and c. Time. As can be seen from the figure, since the unbalanced RL load is connected to the thyristor rectifier circuit, which is a harmonic generation source, the lines Ila, Ila
lb and Ilc are unbalanced and contain harmonic currents. On the other hand, on the power supply side, the power supply currents Isa, Isb and Isc are compensated for higher harmonic currents and are balanced three-phase. Also, the receiving end voltages Vta, Vtb, V
Compared to tc, the power supply currents have the same phase, and it can be confirmed that the reactive current is compensated on the power supply side.
【００１６】[0016]
【発明の効果】本発明によれば、これまで必要とされた
正相及び逆相変換のなかで正相変換を用いることのみで
不平衡と高調波の補償の一括補償が可能であり、且つ、
電源側には、正相分の無効電流も補償されることから、
電力会社から見ると理想的な電流波形となる。一方、需
用家から見ると受電設備には常に正弦波で、且つ有効電
流のみが流れることにより受電設備の有効利用が可能と
なるという効果がある。According to the present invention, it is possible to collectively compensate for unbalance and harmonics only by using the positive-phase conversion among the positive-phase and negative-phase conversions required so far, and ,
On the power supply side, the reactive current for the positive phase is also compensated,
From an electric power company's point of view, it has an ideal current waveform. On the other hand, from the viewpoint of a consumer, there is an effect that the power receiving facility can be used effectively because the power receiving facility always has a sine wave and only an effective current flows.
【図１】実施例のアクティブフィルタの構成図。FIG. 1 is a configuration diagram of an active filter according to an embodiment.
【図２】実施例のアクティブフィルタの制御方法のブロ
ック図。FIG. 2 is a block diagram of a control method of the active filter according to the embodiment.
【図３】実施例のシミュレーション結果。FIG. 3 is a simulation result of the embodiment.
１・・・平衡三相電源 ２・・・電源インピーダンス ３・・・不平衡負荷 ４・・・サイリスタ整流回路 ５・・・サイリスタ整流回路の出力電流 ６・・・フィルタリアクトル ７・・・アクティブフィルタ Ｖｓ・・・電源電圧 Ｉｓ・・・電源電流 Ｉｓａ・・・ａ相の電源電流 Ｉｓｂ・・・ｂ相の電源電流 Ｉｓｃ・・・ｃ相の電源電流 Ｖｔ・・・受電端電圧 Ｉｌ・・・負荷側電流 Ｉｃ・・・アクティブフィルタの出力電流（補償電流） Ｖｄ・・・アクティブフィルタの直流電圧 Ｚｌ・・・負荷インピーダンス Ｉｄｃ・・・サイリスタ整流回路の出力電流 Ｖｔａ・・・ａ相の受電端電圧 Ｖｔｂ・・・ｂ相の受電端電圧 Ｖｔｃ・・・ｃ相の受電端電圧 Ｉｌａ・・・ａ相の負荷側電流 Ｉｌｂ・・・ｂ相の負荷側電流 Ｉｌｃ・・・ｃ相の負荷側電流 Ｖｔｄ・・・受電端電圧のｄ軸成分 Ｖｔｑ・・・受電端電圧のｑ軸成分 Ｖｔｄ1・・・受電端電圧のｄ軸電圧の直流分 Ｖｔｑ1・・・受電端電圧のｑ軸電圧の直流分 Ｉｌｄ・・・負荷電流のｄ軸成分 Ｉｌｑ・・・負荷電流のｑ軸成分 Ｉｌｄ1・・・負荷電流のｄ軸成分の直流分 Ｖｔｐ・・・受電端電圧の正相分 Ｉｌｐ・・・負荷電流の正相分 Ｉｌａｐ・・・ａ相の負荷電流の正相分 Ｉｌｂｐ・・・ｂ相の負荷電流の正相分 Ｉｌｃｐ・・・ｃ相の負荷電流の正相分 ＬＰＦ1・・・ローパスフィルタ ＬＰＦ2・・・ローパスフィルタ Ｃ1・・・ａ−ｂ−ｃ座標からｄ−ｑ座標への変換行列 Ｃ2・・・ｄ−ｑ座標からａ−ｂ−ｃ座標への変換行列 Ｒａ・・・ａ相負荷抵抗 Ｒｂ・・・ｂ相負荷抵抗 Ｒｃ・・・ｃ相負荷抵抗 Ｌａ・・・ａ相負荷リアクトル Ｌｂ・・・ｂ相負荷リアクトル Ｌｃ・・・ｃ相負荷リアクトル α・・・サイリスタ整流回路の位相制御角 ＩＣａ・・・サイリスタ整流回路のａ相指令値 ＩＣｂ・・・サイリスタ整流回路のｂ相指令値 ＩＣｃ・・・サイリスタ整流回路のｃ相指令値 θｖ・・・受電端の実際の電気角 θｒ・・・受電端の周波数６０Ｈｚの理想的な鋸歯状波 DESCRIPTION OF SYMBOLS 1 ... Balanced three-phase power supply 2 ... Power supply impedance 3 ... Unbalanced load 4 ... Thyristor rectifier circuit 5 ... Output current of thyristor rectifier circuit 6 ... Filter reactor 7 ... Active filter Vs ... Power supply voltage Is ... Power supply current Isa ... A-phase power supply current Isb ... B-phase power supply current Isc ... C-phase power supply current Vt ... Receiving end voltage Il ... Load-side current Ic: Output current (compensation current) of active filter Vd: DC voltage of active filter Zl: Load impedance Idc: Output current of thyristor rectifier circuit Vta: Receiving end of a-phase Voltage Vtb ... b-phase receiving end voltage Vtc ... c-phase receiving end voltage Ila ... a-phase load-side current Ilb ... b-phase load-side current Ilc ... c-phase load Current Vtd: d-axis component of receiving end voltage Vtq: q-axis component of receiving end voltage Vtd1: DC component of d-axis voltage of receiving end voltage Vtq1: DC of q-axis voltage of receiving end voltage Component Ild: d-axis component of load current Ilq: q-axis component of load current Ild1: DC component of d-axis component of load current Vtp: Positive phase component of receiving end voltage Ilp: load Positive phase component of current Ilap ... Positive phase component of a-phase load current Ilbp ... Positive phase component of b-phase load current Ilcp ... Positive phase component of c-phase load current LPF1 ... Low-pass filter LPF2: low-pass filter C1: conversion matrix from abc coordinates to dq coordinates C2: conversion matrix from dq coordinates to abc coordinates Ra: a phase Load resistance Rb: b-phase load resistance Rc: c-phase load resistance La: a-phase load Reactor Lb: b-phase load reactor Lc: c-phase load reactor α: phase control angle of thyristor rectifier circuit ICa: a-phase command value of thyristor rectifier circuit ICb: b-phase of thyristor rectifier circuit Command value ICc: c-phase command value of thyristor rectifier circuit θv: Actual electrical angle of power receiving end θr: Ideal sawtooth wave of frequency 60 Hz at power receiving end
Claims (2)
高調波、無効電力及び不平衡を打ち消すように補償する
アクティブフィルタ制御方法において、受電端における
該負荷に流入する電流の基本波正相分の有効電流を検出
し、該電流の基本波正相分の有効電流以外の成分を一括
補償することにより、電源側で正弦波平衡三相状態にす
ることを特徴とするアクティブフィルタ制御方法。1. An active filter control method for compensating for cancellation of harmonics, reactive power, and imbalance generated on a load side connected to a power system, wherein a fundamental wave positive phase of a current flowing into the load at a power receiving end is provided. An active filter control method, comprising: detecting an effective current of a current, and compensating components other than the active current of the positive phase of the fundamental wave of the current at a time, thereby establishing a sine-wave balanced three-phase state on the power supply side.
電圧を実際の電気角を検出することなく電源の基本周波
数と等しい鋸歯状波を用いてｄ−ｑ座標上へ変換し、ロ
ーパスフィルタを用いてｄ軸上の極低周波分を含む直流
分を分離し、該直流分のｄ軸成分とｑ軸成分の比の逆正
接を求め、該逆正接と該鋸歯状波の和を求めることによ
り該受電端電圧の基本波の電気角を求め、該電気角を用
いて負荷電流をｄ−ｑ座標上に変換し、別のローパスフ
ィルタを用いて極低周波を含む直流分を分離し、該電気
角を用いてａ−ｂ−ｃ座標上に変換することにより負荷
電流の基本波正相分の有効電流を検出し、該負荷に流入
する電流の基本波正相分の有効電流と負荷電流との差を
求め、アクティブフィルタの指令値とすることを特徴と
する請求項１記載のアクティブフィルタ制御方法。2. The method according to claim 1, further comprising: converting said voltage at said power receiving end to dq coordinates using a sawtooth wave equal to a fundamental frequency of a power supply without detecting an actual electrical angle. A DC component including a very low frequency component on the d-axis is separated using a filter, an arc tangent of the ratio of the d-axis component to the q-axis component of the DC component is obtained, and the sum of the arc tangent and the sawtooth wave is calculated. The electrical angle of the fundamental wave of the voltage at the receiving end is obtained, the load current is converted into dq coordinates using the electrical angle, and the DC component including the extremely low frequency is separated using another low-pass filter. Then, by converting the current into the abc coordinate using the electrical angle, an effective current corresponding to the fundamental wave positive phase of the load current is detected, and an effective current corresponding to the fundamental wave positive phase of the current flowing into the load is detected. 2. The difference between the load current and the load current is obtained and used as a command value of the active filter. Active filter control method.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP26777898A JP3611235B2 (en) | 1998-09-22 | 1998-09-22 | Active filter control method |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP26777898A JP3611235B2 (en) | 1998-09-22 | 1998-09-22 | Active filter control method |
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| Publication Number | Publication Date |
|---|---|
| JP2000102168A true JP2000102168A (en) | 2000-04-07 |
| JP3611235B2 JP3611235B2 (en) | 2005-01-19 |
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|---|---|---|---|
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Cited By (6)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US7176652B2 (en) | 2004-04-15 | 2007-02-13 | Denso Corporation | Motor control apparatus |
| CN100370671C (en) * | 2006-03-24 | 2008-02-20 | 武汉大学 | Active power filtering method and its device based on alpha-beta current component directly injection |
| JP2012143094A (en) * | 2011-01-04 | 2012-07-26 | Mitsubishi Electric Corp | Harmonic current compensation device |
| CN103647282A (en) * | 2013-11-15 | 2014-03-19 | 许继集团有限公司 | Double-feed wind generating set low-subharmonics current inhibition method |
| CN110176770A (en) * | 2019-06-10 | 2019-08-27 | 上海电力学院 | The control method of MMC type Active Power Filter-APF when unbalanced source voltage |
| CN114256846A (en) * | 2021-12-13 | 2022-03-29 | 湖南大学 | Adaptive impedance coupling series injection type active power filter and control method |
-
1998
- 1998-09-22 JP JP26777898A patent/JP3611235B2/en not_active Expired - Fee Related
Cited By (8)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US7176652B2 (en) | 2004-04-15 | 2007-02-13 | Denso Corporation | Motor control apparatus |
| CN100370671C (en) * | 2006-03-24 | 2008-02-20 | 武汉大学 | Active power filtering method and its device based on alpha-beta current component directly injection |
| JP2012143094A (en) * | 2011-01-04 | 2012-07-26 | Mitsubishi Electric Corp | Harmonic current compensation device |
| CN103647282A (en) * | 2013-11-15 | 2014-03-19 | 许继集团有限公司 | Double-feed wind generating set low-subharmonics current inhibition method |
| CN110176770A (en) * | 2019-06-10 | 2019-08-27 | 上海电力学院 | The control method of MMC type Active Power Filter-APF when unbalanced source voltage |
| CN110176770B (en) * | 2019-06-10 | 2022-12-27 | 上海电力学院 | Control method of MMC type active power filter during power grid voltage unbalance |
| CN114256846A (en) * | 2021-12-13 | 2022-03-29 | 湖南大学 | Adaptive impedance coupling series injection type active power filter and control method |
| CN114256846B (en) * | 2021-12-13 | 2024-01-16 | 湖南大学 | Self-adaptive impedance coupling series injection type active power filter and control method |
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| Publication number | Publication date |
|---|---|
| JP3611235B2 (en) | 2005-01-19 |
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